CN101076004A - Wireless communication device - Google Patents

Abstract

A wireless communication device for receiving a packet composed of a signal modulated by OFDM includes the following elements. A bandpass filter extracts OFDM signals of a desired frequency band. A low-pass amplifier with a gain controlled according to received signal strength amplifies the desired OFDM signal. A frequency converter downconverts the amplified FDM signal to baseband. An analog-to-digital converter converts the baseband signal into a digital signal. A first high pass filter removes a DC offset from the baseband signal corresponding to a predetermined preamble portion of the packet. The frequency offset estimator estimates a frequency offset from sampled signals constituting the baseband signal from which the DC offset has been removed by the first high-pass filter. A frequency offset corrector removes the estimated frequency offset from the baseband signal. The demodulator demodulates the subcarrier signal arranged in the frequency domain from the baseband signal compensated for the frequency offset.

Description

wireless communication device

Cross References to Related Applications

The present invention comprises Japanese patent application JP 2006-137047 submitted to the Japan Patent Office on May 16, 2006, JP2007-037719 submitted to the Japan Patent Office on February 19, 2007, and JP2007-037719 submitted to the Japan Patent Office on April 17, 2007 Subject matter of JP2007-108046 filed, the entire contents of which are hereby incorporated by reference.

technical field

The present invention relates to wireless communication devices for receiving radio frequency (RF) signals modulated by Orthogonal Frequency Division Multiplexing (OFDM). In particular, the present invention relates to wireless communication devices for receiving signals using a direct conversion architecture that does not utilize an intermediate frequency (IF) stage.

More particularly, the present invention relates to a wireless communication device for removing a frequency offset and demodulating OFDM symbols using a training sequence added to a packet header. In particular, the present invention relates to a wireless communication apparatus for accurately estimating frequency offset in the presence of time-varying DC offset and in-phase and quadrature-phase (IQ) imbalances in received OFDM symbols.

Background technique

Wireless networks have attracted attention as cable-free systems replacing conventional wired communication systems. IEEE (Institute of Electrical and Electronics Engineers) 802.11 is a common standard for wireless networks.

For example, when a wireless network is set up in an outdoor environment, there arises a problem that the receiving device receives a superimposition of a direct wave and a plurality of reflected waves and delayed waves, that is, multi-path reception occurs. Multiple reception causes delay distortion (or frequency selective fading), which leads to communication errors. Delay distortion causes inter-symbol interference. In wireless local area network (LAN) standards such as IEEE 802.11a/g, the OFDM modulation scheme is adopted as a multicarrier modulation scheme (see, for example, IEEE 802.11a, Part 11: Wireless LAN Media Access Control (MAC) Layer and Physical Layer (PHY) Description: High-Speed Physical Layer in the 5GHZ Band; and IEEE 802.11g, Part 11: Wireless LAN Medium Access Control (MAC) Layer and Layer Physical (PHY) Description: High-Speed Physical Layer in the 2.4GHZ Band).

sheet standard IEEE 802.11a IEEE 802.11g usage frequency 5.2GHz 2.4GHZ Maximum transmission speed 54Mbps 54Mbps modulation mode OFDM OFDM

The OFDM transmitter converts the information transmitted through the serial signal into parallel data every symbol period at a rate lower than the information transmission rate, and then distributes multiple parallel data streams to the subcarriers for the amplitude of each subcarrier and phase modulation. The OFDM transmitter also performs an Inverse Fast Fourier Transform (IFFT) on multiple subcarriers to convert the frequency domain subcarriers to time domain signals and transmits the resulting signals. An OFDM receiver performs the inverse of that of an OFDM transmitter. That is, the OFDM receiver performs Fast Fourier Transform (FFT) to convert a time-domain signal into a frequency-domain signal for demodulation according to a modulation mode corresponding to a subcarrier. The OFDM receiver also performs parallel-to-serial conversion and regenerates the original information transmitted by the serial signal. The frequency of the carrier is determined such that the subcarriers are orthogonal to each other over the symbol period. Subcarriers that are orthogonal to each other mean that the peak points of the spectrum of a given subcarrier constantly match the nulls of the spectrum of other subcarriers, and that no crosstalk occurs between them. Therefore, transmission data is transmitted on a plurality of carriers having orthogonal frequencies, and the advantages of narrow bandwidth of carriers, high frequency use efficiency, and high resistance to frequency selective fading are realized. Therefore, an efficient OFDM modem can be realized by using the FFT algorithm. The OFDM transmission mode is used in wireless LAN systems, such as terrestrial terrestrial digital broadcasting systems (see, for example, J. OLSSON, "WLAN/WCDMA Dual-Mode Receiver Architecture Design Trade-Offs" Proc. of IEEE 6th CAS Symp., vol.2, pp .725-728, May 31 to June 2, 2004), the fourth generation mobile communication system, and various other broadband digital communication systems of the power line carrier communication system.

In the RF front end of a wireless communication device, a variable gain amplifier circuit is used during transmission, usually after up-converting an analog baseband signal to an RF band signal using a frequency converter (sigma modulator) and limiting the frequency band using a bandpass filter Amplify the transmission power. During reception, the signal received by the antenna is amplified by a low-noise amplifier (LNA) and then down-converted to baseband using the local frequency _fLC . An Automatic Gain Control (AGC) circuit is used to maintain the current of the own signal at a suitably constant level.

In recent wireless communication devices, a frequency converter for up-converting or down-converting a transmit/receive signal uses a direct conversion architecture to perform direct frequency conversion using a carrier frequency f _C as a local frequency f _L0 . Direct conversion architectures do not use external intermediate frequency (IF) filters (also known as "RF interstage filters"), reducing size and power consumption and increasing integration compared to superheterodyne architectures. Also, in principle, no aliasing is generated and the direct conversion architecture is superior in terms of transmitter and receiver design. However, in direct conversion receiver architectures, it has been pointed out that the problem of a direct current component, or DC offset, at the output of the downconverter is caused by the self-mixing of the local signal since the received frequency is equal to the local frequency (see, for example, Anuj Batra , "03267r1P802-15_TG3a-Multi-band-OFDM-CFP-Presentation.ppt", pp.17, July 2003). Self-mixing occurs due to the finite isolation between the local signal and the RF port of the low noise amplifier or mixer. The term DC as used herein is defined as 0 Hz (zero IF) in the baseband signal in OFDM modulation mode.

OFDM communication systems have small errors between the oscillator frequencies in the transmitter and receiver (for example, in a wireless LAN, using an oscillator with an accuracy of about 20 ppm) can cause frequency offset problems in the receiver. Although subcarriers do not interfere with each other, frequency orthogonality between subcarriers cannot be maintained in the presence of frequency offset, resulting in degradation of demodulation characteristics, that is, errors in received data.

In a packet-switched wireless communication system such as the IEEE 802.11 communication system, a symbol known to the transmitter and receiver, ie, a training sequence, is placed at the header of each packet. The receiver performs automatic gain control of the low noise amplifier, DC offset estimation and removal, frequency offset estimation and removal, packet detection, and timing detection using the received training sequence. If the frequency shift processing is performed by an analog circuit, the complexity of the circuit structure and power consumption increase. Therefore, the inventors of the present invention considered that the estimation and compensation processing of the frequency offset is preferably performed by digital processing. In response to the observation of the frequency offset, the phase of the data is inverted to correct for the frequency offset.

The following will examine the issue of frequency offset in the context of IEEE 802.11a/g. Figure 15 shows the preamble structure specified in IEEE 802.11a/g (see, for example, M. MAC) and physical layer (PHY) Description: High-speed physical layer in the 5GHZ frequency band). As shown in Figure 15, a short preamble period of 8.0 μs and a long preamble period of 8.0 μs are added to the packet header. The short-preamble period consists of a short training sequence (STS), in which ten short-preamble symbols t ₁ to t ₁₀ are repeatedly transmitted. The long-preamble period consists of a long training sequence (LTS) in which two long-preamble symbols T ₁ to T ₂ are repeatedly transmitted after a guard interval GI2 of 1.6 μs. A short-preamble symbol consists of 12 subcarriers and has a length of 0.8 μs, which corresponds to a quarter of the IFFT/FFT period _TFFT . One long-preamble symbol consists of 52 subcarriers and has a length of 3.2 μs, which corresponds to the IFFT/FFT period T _FFT . As shown in Figure 30, the OFDM signal does not include DC or 0 Hz subcarriers to avoid DC offset interference.

IEEE 802.11a/g does not specify the use of preambles. Typically, the receiver sets the gain of the receiver and corrects the DC offset using four STS symbols of 0.8 μs, and performs estimation and correction of frequency offset, packet detection, and coarse timing detection using the remaining six STS symbols.

An estimate of the frequency offset can be obtained using an STS with a period of 0.8 microseconds according to equation (1) as follows:

$\mathrm{<span\; class="notranslate">\; \&Delta;\; f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}{\mathrm{<span\; class="notranslate">\; T</span>}}_{\mathrm{<span\; class="notranslate">\; STS</span>}}}\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; m</span>}}\underset{\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; k</span>}}{\overset{\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; m</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\mathrm{<span\; class="notranslate">\; arg</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; S</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

where T _STS represents 0.8 microseconds, S(i) represents the STS signal sampled at a frequency of 20 MHz, S ^* (i) represents the complex conjugate of the STS signal, and M represents the sample mean.

Packet detection and coarse timing detection can be performed based on correction values normalized by the STS power level, which is given by the following equation (2):

$\mathrm{<span\; class="notranslate">\; CF</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; N</span>}}\underset{\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; k</span>}}{\overset{\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; m</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\frac{\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; S</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}}{\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; S</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; )</span>$

Among them, N represents the sampling average.

During packet detection, a threshold level is set for the correction value given by equation (2) above, and a packet is detected when the correction value exceeds the threshold level. During coarse timing detection, the characteristic that the correction value changes from increasing to decreasing at the end of the STS is utilized. That is, the currently detected correction value is compared with the previously determined correction value to determine the coarse timing.

Accordingly, the receiver performs automatic gain control of the low noise amplifier, DC offset estimation and removal, frequency offset estimation and removal, packet detection, and timing detection using the preamble portion of each packet.

However, the accuracy of frequency offset estimation, packet detection, and timing detection is sensitive to DC offset, and there arises a problem that it is difficult to accurately estimate frequency offset in the presence of DC offset. Especially in the above-mentioned direct conversion architecture, the problem of DC offset caused by self-mixing is serious, and the quality of the received signal can be impaired by frequency offset and DC offset.

For example, in the case of DC offsets at the I-axis and Q-axis inputs, the correction value will increase even during dead time. That is, the correction value is constantly increased by one, and due to the continuous increase, the correction value exceeds the threshold value which is the basis of packet detection. Therefore, the receiver recognizes that packets are received even during the silent period, resulting in an operation error.

In addition, in the case where there is a DC offset at the I-axis and Q-axis inputs, even in a portion where the received signals are not correlated with each other, the correction value does not change from increase to decrease due to the influence of the DC offset. Therefore, coarse timing detection characteristics are degraded.

In addition, in the presence of a DC offset, the accuracy of frequency offset estimation decreases, and the remaining frequency offset further degrades the characteristics of the received signal. The frequency offset that has not yet been removed causes a phase rotation of all subcarriers of the OFDM symbols following the training sequence and causes an error floor where packet errors occur even if the signal-to-noise (SN) ratio increases. In contrast, it is difficult to accurately estimate the DC offset when estimating the DC offset in the presence of a frequency offset. Therefore, it is desirable to address the presence of DC offset and frequency offset.

As described above, it is desirable to perform high-precision DC offset correction in a stage preceding the frequency offset correction circuit block within a period as short as the first four STS symbols t ₁ to t ₄ . Generally, it is difficult to realize short-term high-precision DC offset correction, and power consumption and circuit size increase significantly.

There already exist methods for removing DC offset using a high-pass filter (HPF), methods for estimating DC offset and frequency offset simultaneously, methods for estimating DC offset and frequency offset in parallel, methods for repeating DC Methods of offset estimation and frequency offset compensation, etc.

Fig. 16 diagrammatically shows the structure of a receiver using HPF to remove DC offset (for example, see W.Namgoong and T.H.Meng, "Direct-Conversion RF Receiver Design)", IEEE Trans.on Commun.Vol.49, No.3 , March 2001). In the receiver shown in FIG. 16, the DC offset component included in the received OFDM symbol is removed using the HPF. Then, signal processing is performed to estimate the frequency offset and remove the frequency offset from the OFDM symbols following the training sequence. However, in this method, HPF attenuates near-DC signals in OFDM symbols, possibly deteriorating demodulation characteristics.

One way to prevent fading of near-DC signals is to sufficiently reduce the cutoff frequency f _c of the HPF relative to the subcarrier spacing (see FIG. 17A ). However, if the gain of the low noise amplifier is changed by automatic gain control, there is a problem of generating a time-varying DC offset (for example, see S. Otaka, T. Yamaji, R. Fujimoto, and H. Tanimoto, "A Low Offset 1. 9GHz Direct Conversion Receiver IC with Spurious Free Dynamic Range of over 67 dB″, IECE Trans. on Fundamentals, vol. E84-A, no.2, pp.513-519, February 2001). A HPF with a low cut-off frequency _fc has low response and can transmit a time-varying DC offset through the HPF.

For example, in the preamble structure shown in Figure 15, the gain of the LNA changes from high to low around the center of the short preamble period. The DC offset greatly changes with time according to the change of the gain, and high-frequency components are included in the DC offset (see part (a) of FIG. 31 ). Since the HPF with low cut-off frequency _fc has low response, the high-frequency component of the time-varying DC offset is transmitted through the HPF and passes through the frequency offset estimator in a subsequent stage. If such residual DC is still present in the subsequent long-preamble period, it will affect the fine frequency offset estimation performed in the long-preamble period (see part (b) of Figure 31), resulting in lower Estimated accuracy.

For example, in IEEE 802.11a/g, the preamble period is significantly shorter, and it is expected to use HPF to quickly converge the residual DC offset. Minimize the convergence time by greatly increasing the cutoff frequency _fc of the HPF (see, for example, T. Yuba and Y. Sanada, "DecisionDirected Scheme for IQ Imbalance Compensation on OFDCM DirectConversion Receiver", IEICE Trans. on Communications, vol. E89- B, no.1, pp.184-190, January 2006).

An HPF with increased cutoff frequency _fc has a high response to changes in DC offset caused by changing the gain of the LNA, and can cut off even valid near-DC signals (see Figure 17B). Therefore, OFDM demodulation characteristics may be degraded.

Considering better transient response and fast convergence speed, it is preferable that the cut-off frequency f _c of the HPF is very high. In the preamble structure shown in FIG. 15 , in the STS with 1.25 MHz interval between the DC component and the subcarrier closest to the DC component, the signal near the DC subcarrier is not cut off even if the cutoff frequency is high. However, in the subsequent LTS with 312.5kHz spacing between the subcarriers closest to DC, the HPF will cut off the signal of the subcarriers close to DC, resulting in degradation of demodulation characteristics.

Figure 18 diagrammatically shows the structure of a receiver that simultaneously estimates DC offset and frequency offset (see, for example, GT Gil, IHSohn, JK Park, and YHLee, "JointML Estimation of Carrier Frequency, Channel, I/Q Mismatch, and DCOffset in Communication Receivers", IEEE Trans. on Vehi. Tech., Vol.54, No.1, January 2005). In the receiver shown in Fig. 18, the DC offset and the frequency offset are simultaneously estimated and compensated using the maximum likelihood estimation method. However, it is difficult to perform maximum likelihood estimation in a system with limited offset compensation time due to the large computation and long computation time of the maximum likelihood estimation method. In the preamble structure shown in Fig. 15, it is expected that the DC offset estimation is completed within about the first four STS symbols _ti to _t4 , ie, about 3.2 microseconds.

Fig. 19 diagrammatically shows the structure of a receiver for parallel estimation of DC offset and frequency offset (see, for example, C.K.Ho, S.Sun and P.He, "Low Complexity Frequency Offset Estimation in the Presence of DC Offset", Proc. of IEEE International Conference on Communications 2003, Vol.3, pp.2051-2055, May 2003; and U.S. Patent Application Publication Nos. 2003/0174790 and 2005/0078509). The DC offset estimator estimates the DC offset by averaging over the entire preamble. A frequency offset estimator in a subsequent stage calculates a correlation function of the preamble signal and subtracts the estimated DC offset to estimate a precise frequency offset from the DC offset removed signal. However, if the level of the DC offset is changed by changing the gain of the low noise amplifier or the like during preamble reception, the DC offset may be erroneously estimated.

Figure 20 diagrammatically shows the structure of a receiver that repeats DC offset estimation and frequency offset compensation (see, eg, US Patent Publication Nos. 2005/0020226, 2003/0133518, 2004/0202102, and 2005/0276358). In the receiver shown in Fig. 20, the DC offset remover removes the DC offset and then estimates the frequency offset. After the frequency offset is compensated, the DC offset is further estimated to remove the residual DC offset. This method takes a long time to converge the feedback loop of DC offset removal, and it is difficult to use the method for short preambles. In addition, if the DC offset is changed by changing the gain of the low-noise amplifier or the like, an error occurs in frequency offset estimation.

It is well known that the DC offset level varies according to the change of the gain set in the low noise amplifier. The receivers shown in Figures 18-20 do not adequately account for the change in DC offset over time according to gain changes in the low noise amplifier.

OFDM direct conversion receivers also have problems with IQ imbalance and DC offset caused by self-mixing of local signals. Direct conversion architectures do not use IF signals in the digital domain and perform IQ quadrature modulation in the analog domain instead of the digital domain. IQ imbalance is caused by an imbalance between the in-phase (I) component and the quadrature-phase (Q) component. Specifically, the IQ phase imbalance is caused by a non-90 degree phase difference between the local signals input to the I-channel and Q-channel mixers, and the IQ gain imbalance is caused by the difference between the signals in the I-channel and Q-channel Gain difference caused by (see, for example, T.Yuba and Y.Sanada, "Decision Directed Scheme for IQ Imbalance Compensation on OFDCM Direct Conversion Receiver", IEICE Trans.onCommunications, vol.E89-B, no.1, pp.184 -190, January 2006). Similar to DC offset, IQ imbalance causes degradation of frequency offset estimation accuracy, and also affects decoding characteristics.

Therefore, in an OFDM direct conversion receiver architecture, the quality of the received signal can be degraded due to frequency offset, DC offset, and IQ imbalance. The receivers shown in Figures 18-20 do not take time-varying DC offset and IQ imbalance into account.

Contents of the invention

Therefore, it is desirable to provide an excellent wireless communication device using a direct conversion architecture in which the frequency offset can be properly removed in the presence of a time-varying DC offset to achieve OFDM modulation with higher characteristics.

In addition, it is also desirable to provide an excellent wireless communication apparatus in which DC offset can be removed as well as accurate estimation of frequency offset.

According to a first embodiment of the present invention, there is provided a wireless communication apparatus for receiving a packet composed of a signal modulated by Orthogonal Frequency Division Multiplexing (OFDM). The wireless communication device includes the following elements. A bandpass filter extracts OFDM signals of a desired frequency band. A low noise amplifier having a gain controlled according to the intensity of a received signal amplifies an OFDM signal of a desired frequency band. A frequency converter downconverts the amplified FDM signal to baseband. An analog-to-digital converter converts the baseband signal into a digital signal. A first high pass filter removes a DC offset from the baseband signal corresponding to a predetermined preamble portion of the packet. The frequency offset estimator estimates the frequency offset from the sampled signals constituting the baseband signal from which the DC offset has been removed by the first high-pass filter. A frequency offset corrector removes the estimated frequency offset from the baseband signal. The demodulator demodulates the subcarrier signals arranged in the frequency domain from the frequency offset-compensated baseband signal.

Embodiments of the present invention relate to a wireless communication device for receiving OFDM signals using a direct conversion architecture. A direct conversion architecture without an IF filter facilitates wideband receiver implementation and allows flexibility in receiver design. However, there is a problem that a DC offset caused by self-mixing of local signals affects frequency offset or timing detection.

In the wireless communication device according to the embodiment of the present invention, the frequency offset is estimated with high precision after removing the DC offset from the baseband signal portion corresponding to the predetermined preamble portion of the packet using the first high-pass filter. The estimated frequency offset is removed from the portion of the received baseband signal subsequent to the portion for which the frequency offset has been estimated.

A differential filter with a simple circuit structure is used as the first high-pass filter to remove DC offset. This differential filter has a sufficiently high response to changes in the DC offset level caused by changing the gain of the low noise amplifier or the like during preamble reception.

If the cutoff frequency _fc for DC offset removal increases, the response to changes in DC offset caused by changing the gain of the low noise amplifier increases. However, there is a problem that even near-DC signals are cut off, resulting in degradation of demodulation characteristics (see FIG. 17B ). However, since the preamble part for frequency offset estimation is divided and DC offset removal is performed, the increased cutoff frequency _fc may not affect the demodulation characteristics in the subsequent data part.

If the DC offset changes rapidly due to a change in the gain of the low noise amplifier or the like, the DC offset may be transmitted through the differential filter. Therefore, the differential filter serving as the first high-pass filter may be configured to input a detection signal to a frequency offset estimator in a subsequent stage upon detection of a rapid change in DC offset. The frequency offset estimator does not perform frequency offset estimation on the sampled signal obtained from the difference filter when the detection signal is input. Thus, high estimation accuracy is achieved.

Assume that the OFDM signal input to the radio frequency communication device does not include a DC subcarrier. The frequency offset estimator may be configured to estimate the frequency offset using a preamble transmitting two OFDM symbols.

Specifically, one OFDM symbol consists of n subcarriers. When the i-th sample of the temporal waveform of two transmitted OFDM symbols is denoted by s(i), the sample of the first transmitted OFDM symbol is represented by {s(0), s(1), ..., s(n -1)}, the sampling of the OFDM symbol of the second transmission is denoted by {s(n), s(n+1), ..., s(2n-1)}, the frequency offset is denoted by Δf, and When the DC offset is represented by D, the first high-pass filter can be configured to perform the operation given by the following equation (4) on the received baseband signal given by the following equation (3), and output the sampled signal d(i):

r(i)=s(i)exp(j2πΔfi)+D ...(3)

d(i)=r(i+1)-r(i)

＝s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi) …(4)

The frequency offset corrector may be configured to perform the operation given by the following equation (5) using the sampled signal d(i) to estimate the frequency offset Δf included in the received baseband signal r(i):

$\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 5</span>}<span\; class="notranslate">\; )</span>$

In IEEE802.11a/g, which is an example of a wireless communication system according to an embodiment of the present invention, both the short preamble and the long preamble with transmission repetition of the same symbol are included at the header of each packet. Therefore, by performing the above operations, the frequency offset can be estimated more accurately from the DC offset-removed signal.

The sample signal d(i) output from the first high-pass filter constituted by a difference filter is determined by the above-mentioned equation (4). More specifically, as given by equation (6) below, the change in DC offset between the i-th and (i+1)-th samples given by D(i+1)-D(i) Added to the sampled signal d(i). If the DC offset varies by a small amount, then there is no problem. However, if the DC offset varies greatly between sampled signals due to gain changes of the low noise amplifier, etc., the DC offset affects the frequency offset estimation:

d(i)=r(i+1)-r(i)

＝{s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi)}+{D(i+1)-D(i)}

...(6)

Therefore, when the absolute value of the sampled output d(i) is large due to the change in DC offset between the i-th and (i+1)-th samples given by D(i+1)-D(i) , the first high-pass filter may be configured to output the detection signal to a frequency offset estimator in a subsequent stage. In response to the detection signal, the frequency offset estimator does not perform frequency offset estimation on the i-th sampled output using the operation given by equation (5) above. Thus, high estimation accuracy is achieved.

As described above, the frequency offset estimator estimates the frequency offset from the DC offset-removed signal. The frequency offset corrector removes the frequency offset estimated by the frequency offset estimator from the portion of the received baseband signal following the portion used for frequency offset estimation (ie, from which the DC offset has not been removed). In other words, the DC offset is only removed for the preamble portion where frequency offset estimation is performed, while the frequency frequency offset is removed but not for the preamble portion and payload portion following the portion used for frequency offset estimation DC offset. In this case, there may arise a problem that the DC offset affects the demodulation circuit in the subsequent stage even if the frequency offset estimation is performed with high precision.

Therefore, preferably, the means for removing the DC offset is also provided in the section following the section in the received baseband signal in which the frequency offset has been estimated.

For example, in the case where the frequency converter uses a direct conversion architecture, the phase of the local frequency oscillated by the local oscillator may be inverted according to the frequency offset estimated by the frequency offset estimator. Therefore, the influence of the DC offset and the frequency offset can be eliminated from the portion of the received baseband signal that includes the DC offset after the portion where the frequency offset has been estimated.

Optionally, the wireless communication device may further include a DC offset estimator that estimates a DC offset in the digital baseband signal converted by the analog-to-digital converter, and may remove the estimated DC offset from the converted digital baseband signal.

The DC offset is usually estimated by averaging the converted digital baseband signal. Therefore, if the DC offset changes rapidly due to a change in the gain of the low noise amplifier, etc., the average value is no longer useful, and the composition of the DC offset estimate will lead to degradation of accuracy. Thus, upon detection of a rapid change in DC offset, the differential filter may be configured to input the detected signal to the DC offset estimator. The DC offset estimator may be configured to exclude estimation data estimated before the detection signal is input and re-estimate the DC offset to prevent a decrease in estimation accuracy.

Optionally, the received baseband signal down-converted using the direct conversion architecture can be filtered by a second high-pass filter, and the DC offset caused by self-mixing of the local signal etc. can be removed, after which the resulting signal can be converted into a digital signal. The second high-pass filter allows all signals to pass through, and the cut-off frequency of the second high-pass filter is preferably set relatively low so that no near-DC signals in OFDM symbols are cut off.

The second high-pass filter with a low cutoff frequency has a low response and causes the effect of a DC offset for a long time. However, the frequency offset is estimated after removing the DC offset using a first high-pass filter with a sufficiently high cutoff frequency, resulting in high estimation accuracy. In addition, filtering with a high cutoff frequency is performed only on the preamble portion where frequency offset estimation is performed, and demodulation characteristics of subsequent signals are not deteriorated.

In general, characteristic degradation is very sensitive to DC offset not only in frequency offset estimation but also in other signal processing such as packet detection or coarse timing detection.

Therefore, the wireless communication apparatus may further include a detector that performs packet detection and rough timing detection using the signal from which the DC offset has been removed by the differential filter.

The wireless communication device may further comprise a switch exclusively connecting the output of the analog-to-digital converter to the path leading to the first high-pass filter or to the path leading to the DC offset corrector. The switch may be configured to switch the output of the analog-to-digital converter from the path leading to the first high-pass filter to leading to the DC offset corrector when the detector detects the end of a predetermined preamble portion for frequency offset estimation path of.

Since the output of the analog-to-digital converter is connected to the path leading to the first high-pass filter in the period before the end of the predetermined preamble portion is detected, the frequency offset estimator can be The frequency offset is estimated with high accuracy over a period of time using the received baseband signal from which the DC offset has been removed using the first high-pass filter.

The DC offset estimator may be configured to estimate the DC offset for a period of time before the end of the predetermined preamble portion. When the end of the predetermined preamble portion is detected, the output of the analog-to-digital converter is switched to a path leading to the DC offset corrector. DC offset correction can be performed on the received baseband signal portion after the end of the predetermined preamble portion using the DC offset estimated with high accuracy over a long period of time by the DC offset estimator. In the received baseband signal portion after the end of the predetermined preamble portion, the DC offset is not removed using the first high-pass filter, and deterioration of the SNR characteristic is not considered.

The frequency offset estimator may be configured to estimate the frequency offset for a period of time before the end of the predetermined preamble portion, and the frequency offset estimator may be configured to correct the frequency offset from the portion of the received baseband signal after the end of the predetermined preamble portion Estimated frequency offset.

According to an embodiment of the present invention, a wireless communication system conforming to the IEEE 802.11a standard is provided. In the IEEE 802.11a standard, a short preamble part consisting of a short training sequence with a relatively large subcarrier spacing and a long preamble part consisting of a long training sequence with a relatively small subcarrier spacing are added to each in the header of the package.

A wireless communication apparatus according to an embodiment of the present invention is configured such that a short preamble part with a relatively large subcarrier spacing is utilized, and the frequency offset is estimated after removing the DC offset using a differential filter with a sufficiently high cutoff frequency, thereby obtaining High estimation accuracy. That is, since frequency offset correction is performed using only the short-preamble portion, the switch can be configured to route the output of the analog-to-digital converter from the beginning of the long-preamble portion following the end of the short-preamble portion. The path of a high pass filter is switched to the path leading to the DC offset corrector.

Then, a frequency offset estimator estimates a frequency offset in the short preamble part, and a frequency offset corrector removes the estimated frequency offset from the long preamble part. A DC offset estimator estimates a DC offset in the short preamble portion, and a DC offset corrector removes the estimated DC offset from the long preamble portion.

The long-preamble portion transmitted after the short-preamble portion may be used by the receiver in any form. Typically, the short preamble part is used to perform coarse frequency offset estimation, and the long preamble part is used to perform fine frequency offset correction and channel estimation. Therefore, an accurate frequency offset is estimated in the short-preamble part for more accurate channel estimation.

For example, the wireless communication apparatus may further include: a second frequency offset estimator for estimating a frequency offset in a long preamble part following the short preamble part; and a second frequency offset corrector for estimating a frequency offset in the long preamble part The frequency offset estimated by the second frequency offset estimator is removed in part. The second frequency offset estimator receives a received baseband signal portion after the long-preamble portion from which the frequency offset and the DC offset estimated in the short-preamble portion have been removed, and estimates a frequency offset. The second frequency offset corrector removes the frequency offset estimated by the second frequency offset estimator from the received baseband signal portion following the long preamble portion

Alternatively, the received baseband signal portion after the long-preamble portion from which the frequency offset and DC offset estimated in the short-preamble portion have been removed may be fed back to the frequency offset estimator to obtain The frequency offset is estimated in the portion of the received baseband signal following the synchronization code portion. A frequency offset corrector removes the estimated frequency offset from the portion of the received baseband signal following the long preamble portion.

In either case, from the received baseband signal from which the estimated frequency offset and DC offset in the lower short-preamble portion have been removed and the residual frequency offset estimated in the portion following the long-preamble portion has been corrected Estimate the channel to obtain channel information with high precision.

If the received baseband signal includes IQ imbalance, desired reception characteristics cannot be obtained even if frequency offset correction is performed. To avoid this disadvantage, the wireless communication device may further include an IQ imbalance estimator and an IQ imbalance corrector. With this structure, IQ imbalance included in the received baseband signal can be canceled, and further improved reception characteristics can be obtained.

An OFDM direct conversion receiver has not only a DC offset but also a problem of IQ imbalance caused by a phase difference between local signals input to I-axis and Q-axis mixers and an amplitude difference between the mixers. Similar to DC offset, IQ imbalance causes degradation of frequency offset estimation accuracy, and also affects decoding characteristics.

When the frequency offset estimator estimates the frequency offset using a received baseband signal from which the DC offset has been removed and in which IQ imbalance still exists, the frequency offset information includes a frequency offset value Δf and a component brought about by the IQ imbalance, That is, the IQ imbalance component.

In a general communication system, multiple preamble symbols are transmitted from a transmitter, and a frequency offset estimator in a receiver can estimate a frequency offset for each preamble symbol. The frequency offset can be represented as a vector in complex space. The direction of the vector representing the IQ imbalance component differs according to the preamble symbol. By sequentially adding frequency offsets of preamble symbol estimates, the frequency offset estimator can relatively reduce an IQ imbalance component included in an estimated frequency offset value, and can finally obtain a more accurate frequency offset.

Receivers typically include a gain controller that adjusts the gain of the low noise amplifier. When the gain of the low noise amplifier is changed during the frequency offset estimation performed by the frequency offset estimator, the IQ imbalance component included in the frequency offset estimated when the large gain is set increases. Therefore, if frequency offsets estimated for a plurality of preamble symbols are simply added, it may be difficult to effectively reduce the proportion of the IQ imbalance component.

In this case, the frequency offset estimator may weight each frequency offset estimated by the preamble symbol according to the gain set in the low noise amplifier when the corresponding preamble symbol is received, and may use The weighted frequency offsets are added to obtain the final frequency offset value. Therefore, the IQ imbalance component included in the estimated frequency offset value can be relatively reduced, and a more accurate frequency offset can finally be obtained.

Generally, in a receiver, a large gain of a low noise amplifier is determined at the beginning of signal detection, and the gain is changed to a lower gain according to the power of a received signal. Therefore, the frequency offset estimator applies small weights to the frequency offset estimated in the first few preamble symbols (during which a large gain is set in the LNA), and large weights to the frequency offset when the gain is changed to a smaller The estimated frequency offset in subsequent preamble symbols with a small gain, and the weighted frequency offsets are summed to obtain the final frequency offset. Specifically, applying a weight to a frequency offset is equivalent to multiplying the frequency offset estimation vector (described below) or the output of the difference filter by a weighting factor.

Specifically, the frequency offset estimator is configured to calculate a weighting factor based on the absolute value of the frequency offset estimated for each preamble symbol, to weight the frequency offset by the weighting factor, and to weight the weighted frequency offset Add up to get the final frequency offset value.

For example, the frequency offset estimator may estimate the frequency of the preamble symbol by applying a weighting factor of 0 to a frequency offset whose absolute value exceeds a predetermined threshold and a weighting factor of 1 to a frequency offset whose absolute value does not exceed a predetermined threshold. The offsets are weighted, and the weighted frequency offsets can be summed to obtain a final frequency offset value. That is, the estimated frequency offset within the preamble period where a large gain is set in the LNA is ignored. For example, the predetermined threshold may be determined based on the strength of the received signal.

Alternatively, the frequency offset estimator may weight the frequency offset of the preamble symbol estimate by applying a weighting factor formed by the reciprocal of the absolute value of the frequency offset to the frequency offset, and may apply the weighted The frequency offsets are added to obtain the final frequency offset value.

According to the embodiments of the present invention, an excellent wireless communication device using a direct conversion architecture can be realized, where frequency offset can be properly removed to achieve OFDM demodulation with higher characteristics in the presence of time-varying DC offset.

According to another embodiment of the present invention, an excellent wireless communication device can be realized, wherein DC offset can be removed, and time-varying DC offset, IQ imbalance, and frequency offset can exist simultaneously in received OFDM symbols Accurately estimate the frequency offset in the case of .

A wireless communication device according to an embodiment of the present invention receives OFDM signals using a direct conversion architecture. Even if an OFDM signal includes a DC offset, the wireless communication device can perform high-speed and high-precision frequency offset estimation by removing the DC offset using a differential filter. Also, if a rapid change in DC offset occurs due to a change in the gain of the low noise amplifier, the DC offset change is detected from the output of the differential filter, and frequency offset estimation is not performed on the output. Therefore, the accuracy of frequency offset estimation can be increased.

Also, according to an embodiment of the present invention, when vector signals including frequency offset information are added, the vector signals are multiplied by a weighting factor according to a level corresponding to a preamble signal (ie, gain of a low noise amplifier). Thus, more accurate frequency offset estimation can be performed using simple signal processing while reducing the effects of IQ imbalance and time-varying DC offset during frequency offset estimation.

Other features and advantages of the present invention will become apparent from the following detailed description of preferred embodiments of the present invention and accompanying drawings.

Description of drawings

FIG. 1 is a diagram illustrating a structure of a receiver in a wireless communication device according to an embodiment of the present invention;

2 is a diagram illustrating a subcarrier structure of an OFDM symbol in a wireless LAN system conforming to the IEEE 802.11a/g standard;

FIG. 3 is a graph showing a squared error of an estimated frequency offset value obtained when a ratio of DC offset power to OFDM signal power is 30 dB and a frequency offset value normalized by subcarrier spacing;

FIG. 4 is a diagram showing another example of the structure of a receiver in a wireless communication device;

FIG. 5 is a diagram showing another example of the configuration of a receiver in a wireless communication device;

FIG. 6 is a diagram showing another example of a configuration of a receiver in a wireless communication device;

FIG. 7 is a diagram showing another example of a configuration of a receiver in a wireless communication device;

8 is a diagram showing a configuration example of a peripheral synchronization circuit that performs frequency offset estimation and correction, packet detection, and coarse timing detection using an output of a high-pass filter;

9 is a diagram showing a configuration example of another peripheral synchronization circuit that performs frequency offset estimation and correction, packet detection, and coarse timing detection using the output of a high-pass filter;

FIG. 10 is a diagram showing an example of a method for adjusting switching timing by the switch controller 28;

FIG. 11A is a diagram showing a structural example of a high-pass filter 21 using a differential filter;

FIG. 11B is a diagram showing another structural example of the high-pass filter 21 using a moving average-based DC offset estimator and corrector;

FIG. 12 is a diagram showing a structural example of the DC offset estimator 25;

13 is a diagram showing a structural example of a peripheral synchronization circuit including a circuit module performing frequency offset correction and channel estimation in the LTS;

14 is a diagram showing a configuration example of a peripheral synchronization circuit configured such that a frequency offset estimator 22 and a frequency offset corrector 24 respectively estimate and remove the frequency offset of the received baseband signal portion after the LTS;

15 is a diagram showing a preamble structure specified in IEEE 802.11a/g;

16 is a schematic diagram showing a receiver structure using HPF to remove DC offset;

17A is a diagram illustrating removal of a DC component of an OFDM signal using an HPF having a sufficiently small frequency with respect to subcarrier spacing;

FIG. 17B is a diagram illustrating the removal of a DC component of an OFDM signal using an HPF having a large frequency with respect to subcarrier spacing to cut off a near-DC signal;

18 is a schematic diagram showing a receiver structure for simultaneously estimating a DC offset and a frequency offset;

FIG. 19 is a schematic diagram showing a receiver structure for estimating DC offset and frequency offset in parallel;

20 is a schematic diagram showing a receiver structure for repeated DC offset estimation and frequency offset compensation;

FIG. 21 is a diagram showing a specific example of the structures of the differential filter 5 and the frequency offset estimator 6;

22 is a diagram showing that changing the gain of the low noise amplifier 2 by automatic gain control affects the output of the differential filter 5;

FIG. 23 is a diagram illustrating the cause of IQ imbalance;

24 is a diagram illustrating a vector representation of frequency offset information in complex space;

FIG. 25 is a diagram showing a specific example of the structure of the differential filter 5 and the frequency offset estimator 6;

FIG. 26 is a diagram showing estimated vectors of frequency offset before and after a gain change in the low noise amplifier 2;

FIG. 27 is a graph showing the 15 sample outputs of multiplier 305 corresponding to short preamble symbols t ₃ and t ₄ and the outputs corresponding to short preamble symbols t ₅ and t ₁₀ by sequentially adding the output of storage element 310 to A diagram of the estimated composite vector obtained from the 79 sample outputs of the multiplier 305;

28 is a diagram showing an estimated composite vector obtained by weighting frequency offsets with weighting factors according to the absolute value of the output signal of the multiplier 305 and adding the weighted frequency offsets;

29 is a graph showing frequency offset estimation accuracy values (mean square error versus normalized frequency offset value) using the related art method and the proposed method;

FIG. 30 is a diagram showing a configuration as a subcarrier specified in IEEE 802.11a/g;

Figure 31 is a diagram illustrating the effect of DC offset on frequency offset;

FIG. 32 is a diagram illustrating the removal of residual DC offset using differential filter 5;

33 is a diagram showing a structure of a receiver in a wireless communication device according to another embodiment of the present invention;

FIG. 34 is a diagram showing a specific example of the structures of differential filter 5, frequency offset estimator 6, and IQ imbalance estimator 1000 in the receiver shown in FIG. 33;

35 is a diagram showing MSE in α estimation in an environment where the gain of the LNA is not changed, where α=0.05 and θ=5°;

36 is a diagram showing MSE in θ estimation in an environment where the gain of the LNA is not changed, where α=0.05 and θ=5°

FIG. 37 is a diagram showing a specific example of the structures of the differential filter 5, the frequency offset estimator 6, and the IQ imbalance estimator 1000;

FIG. 38 is a diagram showing a structural example of another receiver in the wireless communication device;

FIG. 39 is a diagram showing a structural example of another receiver in the wireless communication device;

FIG. 40 is a diagram showing a structural example of another receiver in a wireless communication device;

FIG. 41 is a diagram showing a structural example of another receiver in a wireless communication device;

42 is a diagram showing a configuration example of a peripheral synchronization circuit that performs frequency offset estimation and correction, packet detection, and rough timing detection using the output of a high-pass filter;

43 is a diagram showing a configuration example of another peripheral synchronization circuit that performs frequency offset estimation and correction, packet detection, and coarse timing detection using the output of a high-pass filter;

FIG. 44 is a diagram showing a configuration example of a peripheral synchronization circuit including circuit modules that perform frequency offset correction, IQ imbalance correction, and channel estimation in the LTS; and

Fig. 45 is a diagram showing a configuration example of a peripheral synchronization circuit including circuit modules that perform frequency offset correction, IQ imbalance correction, and channel estimation in the LTS.

Detailed ways

first embodiment

Hereinafter, a first embodiment of the present invention will be described in detail with reference to the drawings.

The present invention contemplates a wireless communication device for receiving OFDM signals using a direct conversion architecture. A direct conversion architecture without the use of an IF filter easily implements a wideband receiver and increases receiver design flexibility.

A problem with OFDM communication systems is that small errors between the frequencies of the oscillators in the transmitter and receiver can cause a frequency offset, which is seen as a phase rotation phenomenon of the received signal in the digital part of the receiver. In a usual procedure, the frequency offset is observed and corrected for using a known training sequence added to the packet header of each packet.

However, direct conversion receivers have the problem of inducing a direct current component, or DC offset, at the output of the downconverter due to self-mixing of the local signal. The accuracy of frequency offset estimation and timing detection is easily affected by DC offset, and it is difficult to accurately estimate frequency offset in the presence of DC offset.

A wireless communication device according to an embodiment of the present invention realizes high-speed and high-precision frequency offset estimation by removing a DC offset using a differential filter.

FIG. 1 shows the configuration of a receiver in a wireless communication device according to a first embodiment of the present invention. The device shown in Fig. 1 has a direct conversion receiver for receiving OFDM signals and means for compensating for frequency offset.

When the antenna receives an OFDM signal, only a signal of a desired frequency band in the OFDM signal is transmitted through a band pass filter (BPF) 1 and amplified by a low noise amplifier (LNA) 2 . The received RF signal has a frequency offset caused by a frequency error between the local oscillators of the transmitter and receiver.

An automatic gain control (AGC) circuit adjusts the gain of the low noise amplifier 2 to maintain the power of the received signal at an appropriate constant level. For example, a gain control range of 50 dB or more is specified in IEEE 802.11a/g. Typically, a large gain is set in LNA 2 at the beginning of signal detection, and then, for example, around the center of the short-preamble period (in the first embodiment, at the end of the fourth short-preamble _t4 ), switch to a lower gain according to the power of the received signal. Gain switching level (gain switching level) is about 20dB. The AGC mechanism is well known and will not be repeated here.

The amplified received signal is multiplied by the local frequency f _L0 generated by the local oscillator 11 using the mixer 3 and frequency converted to a baseband signal using the direct conversion mode. The baseband signal is converted to a digital signal by an analog-to-digital (AD) converter (ADC) 4 .

In a direct conversion architecture of a receiver, since the received frequency and the local frequency are equal, a direct current component, or DC offset, is induced at the output of the downconverter by self-mixing of the local signal. If the gain of LNA 2 is changed by automatic gain control, the DC offset also changes over time (see, for example, IEEE 802.11a, Part 11: Wireless LAN Medium Access Control (MAC) Layer and Physical Layer (PHY) Description: High-speed physical layer in the 2.4GHZ frequency band). The received baseband signal in the previous stage has a time-varying DC offset as well as a frequency offset.

A predetermined cycle of the preamble signal in the digital baseband signal is divided and input to the differential filter 5 . After removing the DC offset component, the resulting signal is input to a frequency offset estimator 6 to estimate a more accurate frequency offset from the signal from which the DC offset has been removed by subtracting the estimated DC offset. FIG. 21 shows a specific example of the structures of the difference filter 5 and the frequency offset estimator 6 . The differential filter 5 includes a delay unit 201 and an adder 202 . The frequency offset estimator 6 includes a delay unit 203 , a complex conjugate calculation circuit 204 , a multiplier 205 , an adder 206 , a storage element 207 , and a phase detection circuit 208 .

The differential filter 5 is a high-pass filter with a simple circuit structure and high response. When the gain of the low noise amplifier 2 is changed to a lower gain (as described above) according to the power of the received signal at the end of the short preamble _t4 , the DC offset level changes. The differential filter 5 has a very high response to changes in the DC offset level, thereby preventing high frequency components from passing therethrough.

If the cutoff frequency f _c for DC offset removal is increased, the response to a change in DC offset produced by changing the gain of the low noise amplifier 2 increases. However, even near-DC signals are cut off, causing a problem of degradation of demodulation characteristics (see FIG. 17B ). In contrast, the receiver shown in FIG. 1 is configured such that the preamble portion used for frequency offset estimation (ie, the STS with relatively large subcarrier spacing) is divided for DC offset removal. In other words, the DC offset is removed only for the preamble portion where frequency offset estimation is performed, while the frequency offset is removed but not the DC offset for the preamble portion and payload portion following the portion used for frequency offset estimation. shift. Therefore, even if the cutoff frequency _fc is increased, there is no adverse effect on the demodulation characteristics of the subsequent data portion (after LTS with short subcarrier spacing).

If the DC offset changes rapidly due to a change in the gain of the low noise amplifier 2 or the like, the DC offset can be transmitted through the differential filter 5 . When such a pulse waveform is input to the frequency offset estimator 6, a mean square error (MSE) may increase. Therefore, differential filter 5 inputs the detection signal to frequency offset estimator 6 upon detection of a rapid change in DC offset. When the input detection signal is a signal including a DC offset, the frequency offset estimator 6 determines a sample output from the difference filter 5 and does not perform frequency offset estimation on the sample output. Therefore, high estimation accuracy can be maintained.

For example, the threshold value for the differential filter 5 to detect a rapid change of the DC offset may be calculated based on the number of gain changes and the level of the received signal.

The estimated frequency offset value is input to the frequency offset corrector 7, and compensates for the frequency offset in the baseband signal of the OFDM symbol section following the section for frequency offset estimation.

The output of the frequency offset corrector 7 is input to a discrete Fourier transform (DFT) unit 8, and demodulates the subcarrier signals arranged in the frequency domain.

It is assumed that the OFDM signal input to the receiver according to the first embodiment of the present invention does not include a DC subcarrier (DC corresponds to 0 Hz in the baseband signal in OFDM demodulation). The frequency offset estimator 6 estimates the frequency offset in the preamble of each packet (ie, in STS with relatively large subcarrier spacing) and in preambles where there are two transmissions of the same OFDM signal symbol.

A wireless communication system according to an embodiment of the present invention is a wireless LAN system conforming to the IEEE 802.11a/g standard. FIG. 2 shows the subcarrier structure of OFDM symbols in this wireless LAN system. As shown in Figure 2, one OFDM symbol consists of 64 subcarriers, 52 of which are modulated as information signals, and 4 subcarriers are used as pilot signals. No signal is transmitted on the remaining subcarriers including the DC component (ie, the remaining subcarriers carry null signals).

When the part of the received baseband signal (which is affected by the DC offset and frequency offset) after the part used for frequency offset estimation (i.e., after the LTS) is input to the frequency offset corrector 7, accurately compensates for And demodulate the frequency offset without bringing about the degradation of the demodulation characteristics caused by removing the DC offset through the high-pass filter.

Fig. 15 shows a preamble structure specified in IEEE 802.11a/g. In the long preamble period, OFDM symbols consisting of 3.2 microsecond long training sequence (LTS) symbols are continuously transmitted twice. The ith sample of the time waveform of the OFDM symbol is denoted by s(i). Samples {s(0), s(1), ..., s(63)} are associated with the first OFDM symbol, and samples {s(64), s(65), ..., s(127)} Associated with the second OFDM symbol (if the order of the discrete Fourier transform is denoted by N, the first OFDM symbol is the set of samples {s(0), s(1), ..., s(N/4-1)} , and the second OFDM symbol is a set of samples {s(N/4), s(N/4+1), ..., s(2N/4-1)}).

If the frequency offset at this time is denoted by Δf and the DC offset is denoted by D, then the received baseband signal associated with the ith short preamble is given by the following equation:

r(i)=s(i)exp(j2πΔfi)+D ...(7)

The differential filter 5 includes a delay unit 201 and an adder 202 . The AD converted signal obtained by the AD converter 4 is input to the input terminal of the delay 201 . The AD converted signal is also input to the first input terminal of the adder 202, and the output of the delay unit 201 is transformed and then input to the second input terminal of the adder 202 for subtraction therebetween. Therefore, the differential filter 5 processes the received baseband signal according to the following equation (8):

d(i)=r(i+1)-r(i)

＝s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi) …(8)

Equation (8) represents the output signal of the differential filter 5 with respect to the i-th short preamble.

The frequency offset estimator 6 in the subsequent stage includes a delay unit 203 , a complex conjugate calculation circuit 204 , a multiplier 205 , an adder 206 , a storage element 207 , and a phase detection circuit 208 . The output of the adder 202 is input to a delay unit 203 and a first input terminal of a multiplier 205 . The delay unit 203 delays the input signal by N/4 (=16) samples corresponding to the short preamble length, and inputs the delayed signal to the complex conjugate calculation circuit 204 in the subsequent stage. The output of the complex conjugate calculation circuit 204 is input to the second input terminal of the multiplier 205 . Therefore, the multiplier 205 performs a cross-correlation operation given by the following equation on each short-preamble t ₁ , t _{2 ,} etc.

$\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 9</span>}<span\; class="notranslate">\; )</span>$

The output of the multiplier 205 is connected to a first terminal of the adder 206 and the output of the storage element 207 is connected to a second input terminal of the adder 206 . The output of the adder 206 is input to a storage element 207 and a phase detection circuit 208 . Then, the cross-correlation results determined by the above equation (9) for all short preambles are added using the adder 206, and the frequency offset Δf is estimated.

The frequency offset of the portion of the received baseband signal (affected by the frequency offset) following the portion used for frequency offset estimation is compensated using the frequency offset corrector 7 . Specifically, the frequency offset is corrected by inverting the data phase according to the frequency offset. A short training sequence (STS) can also be used to perform frequency offset estimation in a similar manner as above (in this case, the number of samples is 16).

Fig. 3 shows the mean square error of the estimated frequency offset value obtained when the ratio of DC offset power to OFDM signal power is 30 dB and the frequency offset value is normalized by the subcarrier spacing. As can be seen from Fig. 3, accurate frequency offset estimation is achieved by removing the DC offset from the received preamble for frequency offset estimation using a differential filter 5. Since the STS has repetitions of the same training sequence symbols (see Figure 15), the frequency offset can be estimated using a similar operation.

In equation (9) above, it is assumed that the DC offset is constant or does not change over time, and the differential filter 5 can remove the DC offset. However, when the gain of the low-noise amplifier 2 is changed, a change in the DC offset level appears as a high-frequency component, and the output of the differential filter 5 represents the amplitude of the DC offset.

Specifically, the automatic gain control circuit sets a large gain for the low-noise amplifier 2 at the start of signal detection, and uses the first to fourth short-preambles _t1 to _t4 to determine the gain for maintaining the power of the received signal at a constant level Proper gain (preambles t ₁ and t ₂ are not used due to multipath effects). The gain is switched to a lower gain at the beginning of the fifth short preamble _t5 . The gain switching level is about 20dB. The DC offset level also changes according to the switching of the gain, which affects the output of the difference filter ₅ at the beginning of the preamble t5 (see Fig. 22).

If the DC offset value of the i-th sample is represented by D(i), when the DC offset in the OFDM sample changes due to a change in the gain of the low-noise amplifier 2, etc., the differential filtering is determined by the following equation (10) The output of device 5:

d(i)=r(i+1)-r(i)

＝{s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi)}+{D(i+1)-D(i)}

...(10)

From the above equations, when the DC offset changes rapidly, the difference between the DC offset of the (i+1)th sample and the DC offset of the i-th sample (D(i+1)-D( i)), and the absolute value of the output d(i) of the differential filter 5 increases. Therefore, when a DC offset change is detected, the DC offset is not removed and passed through the differential filter 5 .

In the receiver shown in FIG. 1, once it is detected that the absolute value of the output d(i) exceeds a predetermined value, the differential filter gives an indication (detection signal) to the frequency offset estimator 6 in the subsequent stage, so as not to The ith sampled output d(i) including the effect of the DC offset performs frequency offset estimation. Thus, the frequency offset estimator 6 estimates the frequency offset without considering the DC offset, thereby improving estimation accuracy. Even in the case where the automatic gain control of the low noise amplifier 2 is frequently performed, frequency offset estimation is not performed on the transmitted DC offset component, thereby maintaining high estimation accuracy.

As described above, since the frequency offset is estimated after the DC offset is removed using the differential filter 5, the structure shown in Fig. 1 can achieve accurate frequency offset estimation while dealing with the occurrence of DC offset and frequency offset Case. In addition, since the estimated frequency offset value obtained in the cycle of changing the gain of the low noise amplifier 2 is excluded, high estimation accuracy can be maintained.

In the receiver structure shown in FIG. 1, the frequency offset corrector 7 removes the pass frequency offset estimator from the portion of the received baseband signal (i.e., from which the DC offset is not removed) following the portion used for frequency offset estimation. 6 Estimated frequency offset. In other words, when the DC offset is removed for the preamble part where the frequency offset is estimated, while the frequency offset is removed but the DC offset is not removed for the preamble part and the payload part after the part used for frequency offset estimation . In this case, there may arise a problem that the DC offset affects the demodulation circuit in the subsequent stage even though the estimation accuracy of the frequency offset is high.

Figure 4 shows an example of a receiver circuit for solving this problem. In the structure of the receiver shown in FIG. 1, after removing the DC offset from the received baseband signal input to the differential filter 5, using the frequency offset estimated by the frequency offset estimator 6, by the frequency offset corrector 7 Subjecting the DC offset comprising part of the received baseband signal following the part for frequency offset estimation to frequency offset compensation. On the other hand, in the configuration of the receiver shown in FIG. 4, after removing the DC offset by the differential filter 5, the frequency of oscillation by the local oscillator 11 is inverted based on the frequency offset estimated by the frequency offset estimator 6. Phase of the local frequency. Therefore, it is possible to simultaneously remove the influence of the DC offset and the frequency offset from the DC offset including section of the received baseband signal following the section for frequency offset estimation.

Also in this case, differential filter 5 inputs the detection signal to frequency offset estimator 6 upon detection of a rapid change in DC offset. As described above, the frequency offset estimator 6 does not perform frequency offset estimation on the sampled output from the differential filter 5 when a detection signal is input, thereby avoiding the influence of the DC offset transmitted through the differential filter 5 .

FIG. 5 is a structural example of another receiver capable of removing the influence of DC offset from the received baseband signal portion after the portion used for frequency offset estimation (after LTS).

In the receiver shown in FIG. 5, a predetermined period of a preamble signal in a received baseband signal is divided and input to a differential filter 5 to remove a DC offset, and a frequency offset estimator 6 estimates a frequency offset. Once a rapid change in DC offset is detected, the differential filter 5 inputs the detection signal to the frequency offset estimator 6 . As described above, when a detection signal is input, the frequency offset estimator 6 does not perform frequency offset estimation on the sampled output from the differential filter 5 in order to avoid the influence of the DC offset transmitted through the differential filter 5 .

In parallel to the frequency offset estimation process, a DC offset estimator 9 estimates the DC offset of the received baseband signal, and a DC offset corrector 10 removes the DC offset from the received baseband signal. Then, based on the high-precision frequency offset value estimated after the DC offset has been removed, the frequency offset corrector 7 performs frequency offset on the portion of the received baseband signal after the portion for frequency offset estimation from which the DC offset has been removed. shift compensation.

Typically, the DC offset is estimated by averaging the converted digital baseband signal. Therefore, if the DC offset changes rapidly due to a change in the gain of the low-noise amplifier 2 or the like, the average value is no longer useful, and the continuation of the DC offset estimation will cause accuracy to deteriorate. Therefore, upon detection of a rapid change in DC offset, differential filter 5 inputs the detection signal to frequency offset estimator 6 and DC offset estimator 9 . The DC offset estimator 9 excludes the estimation data estimated before the detection signal is input, and then re-estimates the DC offset to prevent a decrease in estimation accuracy.

FIG. 6 shows another structural example of a receiver capable of removing the influence of a DC offset from a portion of a received baseband signal following a portion for frequency offset estimation (after LTS).

In the receiver shown in FIG. 6 , the mixer 3 downconverts the received signal using a direct conversion architecture by multiplying the received signal by a local frequency f _L0 generated by a local oscillator 11 . The final received baseband signal includes a DC offset caused by self-mixing of the local signal, etc. (see part (a) of FIG. 32 ). The DC offset is removed by passing it through a high pass filter (HPF) 12 . Since the high-pass filter 12 allows all signals to pass through, the cut-off frequency of the high-pass filter 12 is set relatively low so as not to cut off near-DC signals in OFDM symbols. The baseband signal transmitted through the high-pass filter 12 is converted into a digital signal by an AD converter (ADC) 4 .

A high-pass filter 12 with a low cut-off frequency _fc ensures good demodulation characteristics in subsequent stages, but has a low response to changes in the DC offset. Therefore, if a DC offset change occurs due to gain switching of the low-noise amplifier 2, etc., the influence of the change remains for a long time, and the DC offset is continuously transmitted through the high-pass filter 12 (see part (b) of FIG. 32 ). To deal with this situation, the predetermined period of the preamble in the received baseband signal is divided into two branches, and one branch is input to a differential filter 5 with a higher cutoff frequency to remove residual DC offset. As shown in part (c) of Fig. 32, the differential filter blocks the residual DC offset and only outputs a strong pulse waveform when changing the gain at the beginning of the preamble _t5 .

Then, frequency offset estimator 6 estimates a frequency offset based on the autocorrelation value output by difference filter 5 . Frequency offset corrector 7 removes frequency offset from the received baseband signal.

Once a rapid change in DC offset is detected, the differential filter 5 inputs the detection signal to the frequency offset estimator 6 . If the pulse waveform is input to the frequency offset estimator 6, MSE may increase. Therefore, as described above, when a detection signal is input, the frequency offset estimator 6 does not perform frequency offset estimation on the sampled output from the differential filter 5 to avoid the influence of the DC offset transmitted through the differential filter 5 .

As described above, the threshold value for the differential filter 5 to detect a rapid change of the DC offset can be calculated based on the number of gain changes and the received signal level. For example, a received signal strength indicator (RSSI) circuit may be provided in a subsequent stage of the high-pass filter 12 to detect a received signal level.

The DC offset estimator 9 estimates the DC offset of the portion of the received baseband signal following the portion used for frequency offset estimation, and the DC offset corrector 10 removes the DC offset from the received baseband signal.

The differential filter 5 inputs the detection signal to a DC offset estimator 9 upon receiving a rapid change in DC offset. As described above, the DC offset estimator 9 excludes the estimation data estimated before the input of the detection signal, and re-estimates the DC offset to prevent a decrease in estimation accuracy.

Then, based on the high-precision frequency offset value estimated after removing the DC offset, the frequency offset corrector 7 performs frequency offset on the part of the received baseband signal after the part for frequency offset estimation (DC offset removed). shift compensation.

FIG. 7 shows a structural example of still another receiver capable of removing the influence of DC offset from the portion of the received baseband signal after the portion for frequency offset estimation (after LTS).

In the receiver shown in FIG. 7 , the mixer 3 downconverts the received signal using a direct conversion architecture by multiplying it by a local frequency f _L0 generated by a local oscillator 11 . The resulting reception baseband signal is transmitted through the high-pass filter 12 in the subsequent stage to remove a DC offset caused by self-mixing of the local signal or the like from the reception baseband signal. Since the high-pass filter 12 allows all signals included in the packet to pass, the cut-off frequency f _c of the high-pass filter 12 is set relatively low so as not to cut off near-DC signals in OFDM symbols. Then, the received baseband signal is converted into a digital signal by an AD converter (ADC) 4 .

A high-pass filter 12 with a low cut-off frequency _fc ensures good demodulation characteristics in subsequent stages, but has a low response to changes in the DC offset. Therefore, if a DC offset change occurs due to gain switching of the low noise amplifier 2, etc., the influence of the change remains for a long time, and the DC offset is continuously transmitted through the high-pass filter 12 (see part (b) of FIG. 32 ) . To solve this situation, the predetermined period of the preamble signal in the received baseband signal is divided into two branches, and one branch is input to the differential filter 5 with a higher cutoff frequency to remove the residual DC offset. As shown in part (c) of Fig. 32, the differential filter 5 blocks the residual DC offset, and outputs only a strong pulse waveform when changing the gain at the beginning of the preamble _t5 .

The phase of the local frequency oscillated by the local oscillator 11 is inverted based on the estimated frequency offset. Therefore, it is possible to simultaneously remove the influence of the DC offset and the frequency offset from the DC offset including section of the received baseband signal following the section for frequency offset estimation.

Once a rapid change in DC offset is detected, the differential filter 5 inputs the detection signal to the frequency offset estimator 6 . If the pulse waveform is input to the frequency offset estimator 6, the MSE increases. Therefore, as described above, when a detection signal is input, the frequency offset estimator 6 does not perform frequency offset estimation on the sampled output from the differential filter 5 to avoid the influence of the DC offset transmitted through the differential filter 5 .

The threshold value for the differential filter 5 to detect a rapid change of the DC offset can be calculated based on the number of gain changes and the received signal level. As described above, for example, an RSSI circuit may be provided in the subsequent stage of the high-pass filter 12 to detect the reception signal level.

In a first embodiment of the present invention, the frequency offset is estimated using a preamble that repeatedly transmits the same OFDM signal symbol twice. In the preamble structure of FIG. 15 , signal processing for packet detection, coarse timing detection, and frequency offset estimation and correction is performed using the STS. In this signal processing, characteristic degradation is very sensitive to DC offset.

In the receiver shown in FIG. 6, the influence of the DC offset is removed by using the output signal of the differential filter 5, which is a kind of high-pass filter, to ensure high-precision frequency offset estimation. Packet detection and rough timing detection can also be performed using the output signal of the high-pass filter to prevent characteristic degradation due to DC offset.

FIG. 8 shows a configuration example of a peripheral synchronization circuit that performs frequency offset estimation and correction, packet detection, and rough timing detection using the output of a high-pass filter.

The synchronization circuit shown in FIG. 8 includes a path leading to a high-pass filter 21 and a DC offset estimator 25 for each of the I-axis and Q-axis input signals, and specifically opens or closes two switches 26 and 27 to switch between the two paths to switch between.

STS has a relatively large subcarrier spacing of 1.25MHz between subcarriers closest to DC. In view of this, the cutoff frequency of the high-pass filter 21 is set high to ensure desired response characteristics to DC offset changes. Therefore, the effect of DC offset can be suppressed with minimum SNR loss.

During the synchronization process, a DC offset estimator 25 estimates a DC offset from the received baseband signal. The STS has a period of 0.8 μs, and the DC offset can be estimated using a moving average or the like. If the first four symbols _t1 to _t4 of the STS are used for automatic gain control, DC offset processing, etc., the synchronization circuit can obtain a maximum estimated time of six symbols _t5 to _t10 , i.e., 4.8 microseconds, And ensure high precision DC offset estimation.

During the period before the end of the STS, the IQ input is connected to the path leading to the high-pass filter 21, and the frequency offset estimator 22 and the packet detection and coarse timing detector 23 respectively extract the DC offset from the received baseband signal with Frequency offset estimation, as well as packet detection and coarse timing detection are performed with high precision.

At the end of the STS, switches 26 and 27 are switched on/off and switch the IQ input from the path leading to the high pass filter 21 to the path leading to the DC offset corrector. The output signal of the coarse timing detector 23 is used at the end of the STS.

The DC offset estimator 25 performs DC offset estimation using a sufficiently long time before the end of the STS, thereby realizing high-precision DC offset correction. After the end of the STS, there is no path leading to the high-pass filter 21 . That is, DC offset removal is not performed on the portion of the received baseband signal after the LTS with a short subcarrier spacing using the high-pass filter 21 with a high cutoff frequency, and degradation in SNR is not considered.

Also, during the period before the end of the STS, the frequency offset corrector 24 uses the frequency offset estimated by the frequency offset estimator 22 to perform frequency offset correction on the portion of the received baseband signal after the end of the STS.

In the structure of the synchronous circuit shown in FIG. 8, the switching time between the path leading to the high-pass filter 21 and the path leading to the frequency offset corrector is determined using a rough timing detection signal. In view of the processing delay of the digital circuit, it is possible to switch paths before the end of the STS.

In the configuration of the synchronous circuit shown in FIG. 9, the switches 26 and 27 are not directly switched in response to the detection signal of the coarse timing detector 23, but a switch controller 28 for controlling switching of the switches 26 and 27 is additionally provided. FIG. 10 shows an example of a method for adjusting switching timing using switch controller 28 .

The switch controller 28 uses a moving average of the correlation values of the input signal to determine the switching timing. The switching controller 28 sets a predetermined threshold, and outputs timing as a control signal to the switches 26 and 27 when the moving average exceeds the threshold. As shown in Figure 10, by setting the threshold, the switching timing can be flexibly adjusted within the period from packet detection to coarse timing.

A structural example of the high-pass filter 21 is shown in FIGS. 11A and 11B . FIG. 11A shows a structure using a difference filter, and FIG. 11B shows a structure using a moving average based DC offset estimator and corrector.

FIG. 12 shows an example of the structure of the DC offset estimator 25 . In the structure shown in FIG. 12 , the estimated DC offset value obtained at the switching timing of the switches 26 and 27 can be held using the output signal of the switch controller 28 or the coarse timing detector 23 . Therefore, the DC offset corrector can perform DC offset correction on the portion of the received baseband signal after the end of the STS using the high-precision DC offset value estimated at the end of the STS.

The synchronization circuits shown in Fig. 8 and Fig. 9 are configured to use only STS in the preamble structure specified in IEEE 802.11a, to perform frequency offset correction in consideration of DC offset, and to arbitrarily use LTS after the end of STS .

Typically, after performing coarse frequency offset correction using STS, fine frequency offset correction and channel estimation are performed using LTS. Therefore, an accurate frequency offset is estimated in STS to achieve more accurate channel estimation.

FIG. 13 shows a configuration example of a peripheral synchronization circuit including a circuit module that performs frequency offset correction and channel estimation within a long preamble period. In the example shown in FIG. 13 , the frequency offset estimator 31 and the frequency offset corrector 32 for LTS are provided independently from the frequency offset estimator 22 and the frequency offset corrector 24 for STS, and also A channel estimator 33 is provided in a subsequent stage. Frequency offset estimator 22 performs coarse frequency offset estimation using short preambles (wherein, since the first and second short preambles _t1 and _t2 may be affected by multipath, the third and following short preambles Synchronization codes t ₃ to t ₁₀ are used for estimation). Frequency offset estimator 31 performs fine frequency offset estimation using long preambles _T1 and _T2 .

At the end of the short preamble period, the IQ input switches from the path leading to the high pass filter 21 to the path leading to the DC offset corrector. After the LTS, each DC offset corrector subtracts from each IQ input signal the precise DC offset estimated using a time long enough before the end of the STS to correct the DC offset. Then, the frequency offset corrector 24 corrects the frequency offset estimated by the frequency offset estimator 22 during the period before the end of the STS.

The frequency offset estimator 31 performs fine frequency offset estimation upon receiving the portion of the received baseband signal after the LTS from which the DC offset estimated using the STS and the offset have been removed. The frequency offset corrector 32 removes the frequency offset estimated by the frequency offset estimator 31 from the portion of the received baseband signal after the LTS.

The channel estimator 33 performs channel estimation with higher accuracy using the received baseband signal from which the minor (residual) frequency offset has been removed using the LTS.

In the circuit structure shown in FIG. 13 , a circuit module for performing frequency offset estimation and removal on the part of the received baseband signal after the LTS is additionally provided. Optionally, the frequency offset estimator 22 and the frequency offset corrector 24 may be configured to estimate and remove the frequency offset in the portion of the received baseband signal after the LTS, respectively. Fig. 14 shows a configuration example of a peripheral synchronization circuit in the latter case.

At the end of the STS, the IQ input switches from the path leading to the high-pass filter 21 to the path leading to the DC offset corrector. After the LTS, each DC offset corrector subtracts from each IQ input signal the precise DC offset estimated using a sufficient time before the end of the STS to correct the DC offset. Then, the frequency offset corrector 24 corrects the frequency offset estimated by the frequency offset estimator 22 during the period before the end of the STS.

Additionally, at the end of the STS, a switch 27 is opened to create a path for returning the output of the frequency offset corrector 24 to the frequency offset estimator 22 .

The frequency offset estimator 22 receives the portion of the received baseband signal after the LTS from which the DC offset estimated using the STS and the frequency offset have been removed, and further estimates the frequency offset. The frequency offset corrector 24 removes the frequency offset estimated by the frequency offset estimator 22 from the portion of the received baseband signal after the LTS.

The channel estimator 33 performs channel estimation with higher accuracy using the received baseband signal from which the minor (residual) frequency offset has been removed using the LTS.

A method has been described for reducing the influence of changing the DC offset in the frequency offset estimation process when the DC offset changes according to the gain change of the low noise amplifier in an OFDM direct conversion receiver.

OFDM direct conversion receivers also have problems with IQ imbalance and DC offset caused by self-mixing of local signals. The IQ imbalance is caused by the phase difference between the local signals input to the I-axis and Q-axis mixers and the amplitude difference between the mixers. Similar to DC offset, IQ imbalance causes degradation of frequency offset estimation accuracy, and also affects decoding characteristics. The method of estimating frequency offset in the presence of IQ imbalance and the presence of time-varying DC offset will be described in detail below.

In the receiver structure shown in FIG. 1, the DC offset level varies according to the gain change of the low noise amplifier 2. When the varying DC offset is equal to or greater than a predetermined value, frequency offset estimation is not performed on symbols output from the differential filter 5 (see FIG. 22 ), thereby reducing the influence of varying DC offset in the frequency offset estimation process. However, the receiver shown in Figure 1 does not fully take into account the effects of IQ imbalance.

OFDM direct conversion receivers have problems with IQ imbalance and DC offset caused by self-mixing of local signals. The direct conversion architecture does not use the IF signal in the digital domain, and the IQ quadrature demodulation is not performed in the digital domain but in the analog domain. Thus, IQ imbalance is caused by an imbalance between the in-phase (I) component and the quadrature-phase (Q) component. Specifically, the IQ phase imbalance is caused by a non-90-degree phase difference between the local signals input to the I-channel and Q-channel mixers, and the IQ gain imbalance is caused by the gain difference between the signals in the I-channel and Q-channel. balance.

Figure 23 shows the causes of IQ imbalance. In Figure 23, the local signal from a single local oscillator is split into two branches, and the phase of one branch is shifted by 90° to generate cosine and sine signals. If the two signals have a phase difference of more than 90° or have different amplitudes, the frequency-converted baseband signal will be distorted. This phenomenon is known as IQ imbalance. The influence of distortion is transmitted through the differential filter, and therefore, the IQ imbalance causes degradation of frequency offset estimation accuracy.

If the phase difference between the cosine signal and the sine signal is represented by θ and the amplitude difference is represented by λ (expressed in decibels (dB)), then the I and Q components of the local signal at the local frequency f _c are represented by the following formula, and is input to the corresponding mixer:

I component: (1+α)cos(2πf _c t-θ/2)

Q component: -(1-α)sin(2πf _c t+θ/2)

where α is expressed by the following equation using the amplitude difference λ:

$\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; 10</span>}}^{\mathrm{<span\; class="notranslate">\; \&lambda;</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 20</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; 10</span>}}^{\mathrm{<span\; class="notranslate">\; \&lambda;</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 20</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}}$

The components of the local signal are frequency multiplied with the received signal r(i) using corresponding mixers. The complex transmission symbols are set as X _n =a _n +jb _n .

$\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span>\frac{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 2</span>}}\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 2</span>}}{\overset{\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}<span\; class="notranslate">\; [</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; b</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \{</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;n</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; -</span>$

$<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; b</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \{</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;n</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; ]</span>$

$\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \{</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span>$

$<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\frac{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 2</span>}}\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 2</span>}}{\overset{\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}<span\; class="notranslate">\; [</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; b</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \{</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;n</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span>$

$<span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; b</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \{</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;n</span>}{\mathrm{<span\; class="notranslate">\; f</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; ]</span>$

The received baseband signal without considering the IQ imbalance is determined by the above equation (7), while the received complex baseband signal under the influence of the IQ imbalance is determined by the following equation (11):

$\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 11</span>}<span\; class="notranslate">\; )</span>$

$\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

in

$\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

$\mathrm{<span\; class="notranslate">\; \&psi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

In Equation (11), i represents the number of samples in the short preamble, and a superscript asterisk (*) represents the complex conjugate.

Therefore, the differential signal output from the differential filter 5 under the influence of IQ imbalance is determined by the following equation (12):

$\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 12</span>}<span\; class="notranslate">\; )</span>$

The frequency offset estimator 6 estimates the frequency offset based on the differential signal determined by the above equation (12). The multiplier 205 multiplies the differential signal by the differential signal delayed by N/4 samples to obtain a frequency offset estimation vector for each sample. The autocorrelation value (i.e., the frequency offset estimation vector) including the IQ imbalance output from the differential filter 5 is represented by the following equation (13):

$\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span>{<span\; class="notranslate">\; |</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; |</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{<span\; class="notranslate">\; |</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; |</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;fN</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; r</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;fN</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&phi;\&psi;</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;fN</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{<span\; class="notranslate">\; |</span>\mathrm{<span\; class="notranslate">\; \&psi;</span>}<span\; class="notranslate">\; |</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{<span\; class="notranslate">\; |</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; |</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;fN</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 13</span>}<span\; class="notranslate">\; )</span>$

The frequency offset information represented by the above equation (13) has four items. As shown in Figure 24, these terms are represented as vectors in complex space.

The first term in equation (13) is a vector that depends only on the frequency offset. That is, when the received baseband signal does not include IQ imbalance, only the first term is output from the multiplier 205, and the frequency offset can be estimated from the angle of the vector.

The second to fourth terms in Equation (13) are terms resulting from IQ imbalance, and these terms lead to degradation of frequency offset estimation accuracy. In the fourth term, the value |ψ| ² is negligibly small (it should be understood that ψ is small since α is about 0.1 and θ is about 0.05°). The second and third terms are complex conjugate pairs, which are added to produce only the real number component, which is considered to be the main cause of the degradation of frequency offset estimation accuracy.

A multiplier 205 determines a cross-correlation result for each short preamble, and an adder 206 is used to add the cross-correlation results for all short preambles to estimate the frequency offset Δf. The distortion included in the sum of the cross-correlation results depends on the total number of samples of the synchronization signal pattern.

FIG. 21 shows a configuration example of differential filter 5 and frequency offset estimator 6 in which time-varying DC offset is considered although the influence of IQ imbalance in frequency offset estimation is not sufficiently considered. FIG. 25 shows a structural example of differential filter 5 and frequency offset estimator 6 that suppress the influence of IQ imbalance. In FIG. 25 , differential filter 5 includes delay unit 301 and adder 302 . Frequency offset estimator 6 includes delay unit 303 , complex conjugate calculation circuit 304 , multipliers 305 and 306 , coefficient calculation circuit 307 , adder 309 , storage element 310 , and phase detection circuit 311 .

An operation of performing more accurate frequency offset estimation by suppressing the influence of IQ imbalance and time-varying DC offset in the frequency offset estimation process will be described below with reference to FIG. 25 .

The signal received by the antenna is transmitted through the bandpass filter 1 and only the desired OFDM signal is amplified through the low noise amplifier 2 . The amplified signal is multiplied by a local signal from a local oscillator 11 using a multiplier 3, and converted into a baseband signal. The received baseband signal is converted into a digital signal by an AD converter 4 . If the amplitude difference and the phase difference caused by the IQ imbalance are represented by α and θ, respectively, the received baseband signal is represented by Equation (11) above.

The differential filter 5 includes a delay unit 301 and an adder 302 . The AD converted signal obtained by the AD converter 4 is input to the input terminal of the delay unit 301 . The AD conversion signal is also input to the first input terminal of the adder 302, and the output of the delay unit 301 is inverted and input to the second input terminal of the adder 302 for subtraction therebetween. Therefore, differential filter 5 processes the received baseband signal according to equation (12) above.

Frequency offset estimator 6 in the subsequent stage includes delay unit 303 , complex conjugate calculation circuit 304 , multipliers 305 and 306 , coefficient calculation circuit 307 , adder 309 , storage element 310 , and phase detection circuit 311 . The output of the adder 302 is input to the delay unit 303 and the first terminal of the multiplier 305 . The delay unit 303 delays the input signal by N/4 (=16) samples corresponding to the short preamble length, and inputs the delayed signal to the complex conjugate calculation circuit 304 in the subsequent stage. The output of the complex conjugate calculation circuit 304 is input to the second input terminal of the multiplier 305 . Therefore, in the frequency offset estimator 6, the operation given by the above equation (13) is performed on each of the short preambles t ₁ , t ₂ , and so on.

The first term in equation (13) is a vector including the frequency offset Δf. The vectors are added using the adder 309 and the storage element 310 , and the rotation angle of the added vector is detected by the phase detection circuit 311 . Therefore, the value of the frequency offset Δf is estimated.

As described earlier, the second to fourth terms in Equation (13) are terms resulting from IQ imbalance, and these terms cause degradation in frequency offset estimation accuracy. As described above, in the fourth term, the absolute value of the value ψ is negligibly small. The second and third terms are a complex conjugate pair, and these terms are added to generate only real number components that will be the main cause of frequency offset estimation accuracy degradation.

The second and third terms are vectors that depend on the short-preamble r(i) pattern and frequency offset, and the directions of the vectors are not fixed. Over a significant number of samples, the estimated frequency offset value for the preamble symbol output from multiplier 305 is summed over a sufficient number of samples using storage element 310 and adder 309 to increase equation (13) The first term in , while reducing the distortion components in the second to fourth terms relative to the first term. The phase of the vector is detected by the phase detection circuit 311 in the subsequent stage. Therefore, a more accurate frequency offset can be estimated.

If the gain of the low noise amplifier 2 is changed during frequency offset estimation performed by the frequency offset estimator 6, the IQ imbalance component included in the estimated frequency offset when a large gain is set increases. Therefore, if the estimated frequency offsets over multiple preamble symbols are simply added, it may be difficult to sufficiently reduce the proportion of the IQ imbalance component.

Generally, in a receiver, a large gain is determined for the low noise amplifier 2 at the start of signal detection, and thereafter, the gain is changed to a lower gain according to the power rate of the received signal. Specifically, the automatic gain control circuit sets a large gain for the low noise amplifier 2 at the start of signal detection, and determines a desired gain using the first to fourth short preambles _t1 to _t4 . The gain is switched to a lower gain at the beginning of the fifth short preamble _t5 . The gain switching level is about 20dB.

It is assumed that the first and second short preambles t ₁ and t ₂ are not used for frequency offset estimation in consideration of the influence of multipath. In this case, for example, as shown in FIG. 26 , amplitudes of frequency offset estimation vectors representing frequency offsets (output from the multiplier 305 ) before and after changing the gain of the low noise amplifier 2 in the complex space are significantly different from each other. . That is, the absolute value of the output of the multiplier 305 with respect to the short-preamble symbols _t3 and _t4 is larger, while the absolute value of the output of the multiplier 305 with respect to the short-preamble symbol codes _t5 and _t10 is smaller. The output of multiplier 305 for preambles _t3 and _t4 is only 15 samples. Therefore, the second and third terms included in the output of the multiplier 305 with respect to the preambles _t3 and _t4 in the equation (13) remain in the storage element 310 having a value larger than that of the first term Composite vector of items.

In the structure shown in FIG. 21, the sum of cross-correlation results of all short-preambles _t3 to _t10 is used for frequency offset estimation. The larger the number of samples used to calculate the autocorrelation, the smaller the distortion dependent on the second and third terms in equation (13). However, if _the gain _of the LNA 2 is changed, there remains in the estimated vector a dependence on the second and the distortion vector of the third term. As a result, the estimated composite vector obtained by adding the 79-sample sequence of the output of multiplier 205 corresponding to short-preamble symbols _t5 and _t10 to the output of storage element 207 (see FIG. 27 ) causes a frequency shift Estimated degradation of accuracy.

On the contrary, in the structure of the frequency offset estimator 6 shown in FIG. 25, when the corresponding preamble symbol is received, the frequency offset estimated for each preamble symbol is adjusted according to the gain set in the low noise amplifier 2. The weighted frequency offsets are weighted and the weighted frequency offsets are added to obtain the final frequency offset value. Therefore, after changing the gain of the low noise amplifier 2 by a larger factor (or not using the samples until the gain of the frequency offset estimate is changed), the IQ included in the estimated frequency offset value can be relatively reduced by weighting the samples unbalanced components, and can ultimately result in a more accurate frequency offset.

Specifically, the absolute value of the output signal of the multiplier 305 is input to the coefficient calculation circuit 307 , and the corresponding coefficient is input to the multiplier 306 . The coefficient calculation circuit 307 calculates a weighting factor corresponding to the absolute value of the output signal of the multiplier 305 using either of the first or second method described below. The calculation of the weighting factor corresponding to the absolute value of the output signal of the multiplier 305 is basically equivalent to the calculation of the weighting factor corresponding to the gain amplitude set in the low noise amplifier 2 .

The first method will be described below.

Determine the threshold. If the absolute value of the output signal of the multiplier 305 exceeds the threshold, the coefficient is set to zero. If the absolute value does not exceed the threshold, set the coefficient to 1. For example, the threshold is determined from Received Signal Strength Indication (RSSI). In this case, the frequency offset is estimated according to the following equation (14):

$\mathrm{<span\; class="notranslate">\; \&Delta;f</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; arg</span>}<span\; class="notranslate">\; \{</span>\underset{\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 5</span>}\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{\overset{\mathrm{<span\; class="notranslate">\; ION</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; /</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 14</span>}<span\; class="notranslate">\; )</span>$

In other words, in the first method, the frequency offset is estimated from the short preambles _t5 to _t10 after the gain of the low noise amplifier 2 is changed. In this case, the number of samples used to calculate autocorrelation increases, and the number of distortion terms depending on IQ imbalance can be reduced.

The second method will be described below.

The reciprocal of the absolute value of the output signal of the multiplier 305 is used as a coefficient. In this case, the frequency offset is estimated according to the following equation:

$\mathrm{<span\; class="notranslate">\; \&Delta;f</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; arg</span>}<span\; class="notranslate">\; \{</span>\underset{\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{\overset{\mathrm{<span\; class="notranslate">\; 3</span>}\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\frac{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>}{<span\; class="notranslate">\; |</span>\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; |</span>}<span\; class="notranslate">\; +</span>\underset{\mathrm{<span\; class="notranslate">\; j</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 5</span>}\mathrm{<span\; class="notranslate">\; N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}}{\overset{\mathrm{<span\; class="notranslate">\; 10</span>}\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; /</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;N</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 15</span>}<span\; class="notranslate">\; )</span>$

As shown in FIG. 26, it is assumed here that the gain of the low noise amplifier 2 is changed instantaneously between short preambles _t4 and _t5 . Through this processing, the influence of distortion components appearing in the output of the adder 302 due to IQ imbalance can be reduced.

Fig. 28 shows the estimated resultant vector obtained by the frequency offset estimator 6 using the first method. As shown in FIG. 28, the estimated vectors obtained using the short-preamble symbols _t3 and _t4 before the gain reduction of the low noise amplifier 2 are deleted. Thus, unlike the example shown in FIG. 27 , the estimated vectors of a sufficient number of samples are summed such that the first term in equation (13) is increased while decreasing the second through fourth terms relative to the first term Distortion component.

FIG. 29 shows the frequency offset estimation accuracy obtained using the first method compared to that obtained using the structure shown in FIG. 21 (mean square error versus normalized frequency offset value).

In the receiver shown in FIG. 1 , differential filter 5 and frequency offset estimator 6 are configured in the same manner as shown in FIG. 25 . Therefore, accurate frequency offset estimation can be achieved with simple signal processing in the presence of IQ imbalance and time-varying DC offset.

second embodiment

The direct conversion OFDM receiving apparatus according to the first embodiment uses a technique that can accurately estimate a frequency offset by simple signal processing even in the presence of IQ imbalance and time-varying DC offset. The apparatus according to the first embodiment aims to provide a proper frequency offset estimation, but not to perform IQ imbalance correction together with the frequency offset correction.

However, IQ imbalance correction as well as simple frequency offset correction basically fall within the scope of the present invention. An OFDM receiving apparatus for correcting IQ imbalance will be described below without departing from the scope of the present invention.

Fig. 33 and Fig. 34 show the structure of an OFDM receiving apparatus for realizing the above functions according to the second embodiment of the present invention. 33 and 34 correspond to FIGS. 1 and 21, respectively, and components similar to those shown in FIGS. 1 and 21 are denoted by the same reference numerals.

As shown in FIGS. 33 and 34 , the OFDM receiving apparatus according to the second embodiment is configured such that a signal output from differential filter 5 is input to frequency offset estimator 6 and IQ imbalance estimator 1000 .

The output signal from the differential filter 5 is expressed by the above equation (12), and the signal delayed by N/4 (=16) samples with respect to the output signal is determined by the following equation:

$\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;\&eta;\&gamma;</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; \&eta;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 16</span>}<span\; class="notranslate">\; )</span>$

where η=(r(i)−r(i−1)), and γ=exp(j2πΔf(N/4).

A signal N/4 samples ahead of the signal determined by equation (12) is represented by the following equation:

$\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; r</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="j"\; filenames="A20071010305900691.png,A20071010305900721.png"\; state="\{\{state\}\}">j</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;\&eta;</span>}{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; \&eta;</span>}}^{<span\; class="notranslate">\; *</span>}\mathrm{<span\; class="notranslate">\; \&gamma;</span>}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 17</span>}<span\; class="notranslate">\; )</span>$

In equations (16) and (17), the value γ is obtained from equation (13) above, and is determined by the operation of the phase detection circuit 208 in the frequency offset estimator 6 . The value d, determined by the phase detection circuit 208, is fed back to the IQ imbalance estimator 1000 to reduce one unknown from equations (16) and (17), and equations (16) and (17) can be used as the value d , a function of φ and η. As a result, it should be understood that three equations (ie, equations (12), (16), and (17)) are obtained with respect to three unknown variables (φ, η, and d), and solutions for all variables can be obtained.

The three samples corresponding to each of equations (12), (16), and (17) undergo the operations given by equations (18) and (19) below:

$\frac{\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; \&eta;</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 18</span>}<span\; class="notranslate">\; )</span>$

$\frac{\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&phi;\&eta;</span>}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 19</span>}<span\; class="notranslate">\; )</span>$

Therefore, the relationship expressed by the following equation (20) can be obtained from equations (18) and (19):

$\frac{{\mathrm{<span\; class="notranslate">\; \&psi;</span>}}^{<span\; class="notranslate">\; *</span>}}{{\mathrm{<span\; class="notranslate">\; \&phi;</span>}}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; \&gamma;</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}{{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; \&gamma;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; N</span>}}{\mathrm{<span\; class="notranslate">\; 4</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 20</span>}<span\; class="notranslate">\; )</span>$

Since the values φ and ψ are represented by the following equations from equation (11) above:

$\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

$\mathrm{<span\; class="notranslate">\; \&psi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

Thus, by approximating the above equations, the values φ and η are represented by the following equations:

$\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; \&ap;</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

$\mathrm{<span\; class="notranslate">\; \&psi;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; \&ap;</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}$

As mentioned above, in the above equation, α and θ represent:

(i) α=amplitude values of I component and Q component; and

(ii) θ = Phase difference between cosine signal and sine signal.

IQ imbalance occurs when the I and Q components of a signal have the following two relationships:

(a) Generate a local signal input to the mixer 3 for frequency conversion by splitting the output signal from the phase-locked loop (PLL) into two signals and passing one signal through a 90° phase shifter (i.e., the local oscillator 11's output signal). If the output signal from the PLL is a high-frequency signal, the phase shifter does not have a perfect 90° phase shift (ie, the I and Q components of the signal are not orthogonal to each other), resulting in a phase difference θ.

(b) A difference in amplitude occurs between the I component and the Q component at the input of the A/D converter due to loss caused by the phase shifter, gain error between amplifiers with respect to the I and Q components, etc., and the value α is not 0.

From these relationships, the values α and θ are determined based on the three samples d(i-N/4), d(i), and d(i+N/4), and the received complex baseband signal is corrected based on the determined values α and θ, The IQ imbalance is thereby corrected. In the OFDM receiving apparatus according to the second embodiment, IQ imbalance estimator 1000 determines values α and θ (that is, IQ imbalance estimator 1000 estimates IQ imbalance), and IQ imbalance corrector 1100 converts the received complex The baseband signal is multiplied by a correction coefficient corresponding to the determined value to perform IQ imbalance correction.

A specific technique for determining the values α and θ using the IQ imbalance estimator 1000 will be described below. Substituting the appropriate expressions for φ and ψ into equation (20), yields the following equation:

$\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}}{{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; =</span>\frac{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; j</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; twenty\; one</span>}<span\; class="notranslate">\; )</span>$

Solving equation (21) for the real and imaginary parts yields the following equations:

$\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; twenty\; two</span>}<span\; class="notranslate">\; )</span>$

$\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; twenty\; three</span>}<span\; class="notranslate">\; )</span>$

According to equations (22) and (23), the determined values α and θ satisfy the following relationship:

${\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}{\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}$

$\mathrm{<span\; class="notranslate">\; \&alpha;</span>}<span\; class="notranslate">\; =</span>\frac{<span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\sqrt{{<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 4</span>}{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}}{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; twenty\; four</span>}<span\; class="notranslate">\; )</span>$

$\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; (</span>\frac{<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&epsiv;</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}\mathrm{<span\; class="notranslate">\; \&alpha;</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 25</span>}<span\; class="notranslate">\; )</span>$

Here, ε is determined by Equation (20), and the variable d in Equation (20) is obtained as an output signal from the difference filter 5 . Since the value γ is determined by the phase detection circuit 208, it should be understood that the value ε is determined from the output signal from the difference filter 5 (i.e., the value d), thus determining the signal fed back from the phase detection circuit 208 (i.e., value γ) and values α and θ.

In the second embodiment, IQ imbalance estimator 1000 performs the above-described operations to determine values α and θ, and outputs correlation coefficients corresponding to the determined values to IQ imbalance corrector 1100 . As a result, the IQ imbalance caused in the received complex baseband signal is corrected by multiplying the received complex baseband signal by the correction coefficient using the IQ imbalance corrector.

The IQ imbalance corrector 1100 can be placed anywhere between the ADC 4 and the DFT 8. It has been found empirically that an IQ imbalance corrector 1100 arranged upstream of the frequency offset corrector 7 provides improved reception characteristics. In addition, experience has shown that when the IQ imbalance correction is performed toward the upstream of the differential filter 5 with respect to the branch point x and the correction signal is distributed to the differential filter 5, the IQ imbalance correction is performed differently from the downstream with respect to the branch point x Compared to improved reception characteristics. In view of this, although the OFDM receiving apparatus according to the second embodiment is configured such that the IQ imbalance corrector 1100 is not disposed between the ADC 4 and the branch point x (see FIG. 33 ), if the position of the IQ imbalance corrector 1100 is changed , the desired advantages can still be achieved. Therefore, the present invention can be implemented regardless of the location of the IQ imbalance corrector 1100 .

The correction coefficient may be determined using any method based on the values α and θ. For example, IQ imbalance estimator 1000 may experimentally determine correction coefficients corresponding to determined values α and θ, and may store a table having experimental values and correspondences between values α and θ in IQ imbalance estimator 1000 . Alternatively, the correction coefficients can be determined directly from the values α and θ. In this case, the following method can be used. That is, according to equation (11) above, the received complex baseband signal is determined by the following equation:

$\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j\&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; j</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; =</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

$<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; j</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>$

Therefore, a correction coefficient satisfying the following expression is determined, and the received complex baseband signal is multiplied by the determined correction coefficient to correct the IQ imbalance:

$\left(\begin{array}{c}{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\\ {\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\end{array}\right)<span\; class="notranslate">\; =</span>\left(\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\\ <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& \mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\end{array}\right)\left(\begin{array}{c}{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\\ {\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\end{array}\right)$

$\left(\begin{array}{c}{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\\ {\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\end{array}\right)<span\; class="notranslate">\; =</span>{\left(\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\\ <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; sin</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}& \mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&alpha;</span>}\mathrm{<span\; class="notranslate">\; cos</span>}\frac{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\end{array}\right)}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}\left(\begin{array}{c}{\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; I</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\\ {\stackrel{<span\; class="notranslate">\; ^</span>}{\mathrm{<span\; class="notranslate">\; r</span>}}}_{\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\end{array}\right)<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 26</span>}<span\; class="notranslate">\; )</span>$

The structure and operation of other components are similar to those of the first embodiment.

Therefore, in the OFDM receiving apparatus according to the second embodiment, IQ imbalance estimator 1000 determines correction coefficients based on the received complex baseband signal from differential filter 5, and IQ imbalance corrector 1100 multiplies the received complex baseband signal by With the determined correction value, the IQ imbalance included in the received complex baseband signal is corrected.

By using this method, the OFDM receiving apparatus realizes the desired MSE characteristics shown in FIGS. 35 and 36 . The experimental values shown in Fig. 35 and Fig. 36 represent the MSE obtained when α = 0.05 and θ = 5° without changing the LAN 2 gain. Figure 35 shows the MSE in the estimate for a value of α, and Figure 36 shows the MSE in an estimate for a value of θ.

Although the IQ imbalance correction method for the configuration shown in FIGS. 1 and 21 of the first embodiment has been described, IQ imbalance correction can also be realized in other device configurations. Other examples of the structures of the difference filter 5 and the frequency offset estimator 6 will be described below.

FIG. 37 shows another example of the structure of the differential filter 5 and the frequency offset estimator 6. In the configuration shown in FIG. 37 , an IQ imbalance estimator 1000 is added to the configuration of the differential filter 5 and the frequency offset estimator 6 shown in FIG. 25 . In FIG. 37, components similar to those shown in FIG. 25 are denoted by the same reference numerals.

As shown in FIG. 37, according to the second embodiment, with the structure shown in FIG. 34, the output signal from the differential filter 5 (that is, the delay unit 301 and the adder 302) is input to the IQ imbalance estimator 1000, and The signal from the phase detection circuit 311 corresponding to the value γ is fed back to the IQ imbalance estimator 1000 . The IQ imbalance estimator 1000 performs operations given by equations (16) to (25) based on an input signal, and determines an imbalance correction coefficient. The IQ imbalance corrector 1100 multiplies the received complex baseband signal by a determined correction coefficient to correct the IQ imbalance induced in the received complex baseband signal. Operations of other components are similar to those shown in FIG. 25, and thus detailed descriptions thereof are omitted.

Modification of the second embodiment

first modification

As shown in FIGS. 38 to 41 , various modifications can be made to the OFDM receiving apparatus shown in FIG. 33 . FIG. 38 shows an OFDM reception device configured to add a structure for IQ imbalance correction to the OFDM reception device shown in FIG. 4 . FIG. 39 , FIG. 40 , and FIG. 41 show OFDM receiving apparatuses configured to add a structure for IQ imbalance correction to the OFDM receiving apparatuses shown in FIG. 5 , FIG. 6 , and FIG. 7 , respectively.

In any of the configurations shown in FIGS. 38 to 41 , (i) the output signal from the difference filter 5 (see FIG. 34 ) is input to the IQ imbalance estimator 1000, and (ii) the value corresponding to the value γ is The signal is fed back to the IQ imbalance estimator 1000 from the phase detection circuit 208 . The IQ imbalance estimator 1000 performs operations given by equations (16) to (25) based on the input signal to determine correction coefficients. The IQ imbalance corrector 1100 corrects the IQ imbalance induced in the received complex baseband signal based on the correction coefficient. Since the structure for IQ imbalance correction is added to the OFDM receiving apparatus shown in FIG. 4, other structures and operations of the OFDM receiving apparatus shown in FIG. 38, FIG. 39, FIG. 40, and FIG. 4. The OFDM receiving devices shown in Fig. 5, Fig. 6, and Fig. 7 are similar.

As described above, in any of the configurations shown in FIGS. 38 to 41 , the IQ imbalance corrector 1100 can be placed anywhere between the ADC 4 and the DFT 8 . However, in the structures shown in FIGS. 38 to 41, the following method is used to improve reception characteristics.

In the OFDM receiving apparatus shown in FIG. 38 and FIG. 41, the IQ imbalance corrector 1100 is provided between the ADC 4 and the branch point toward the differential filter 5, so that the IQ imbalance corrected signal can be input to the differential filter. device 5.

In the OFDM receiving apparatus shown in FIGS. 39 and 40 , the signal output from the DC offset corrector 10 is subjected to IQ imbalance correction after frequency offset correction for the following two reasons.

The first reason is that reception characteristics are improved when IQ imbalance correction is performed after DC offset correction is performed, compared to the case when IQ imbalance correction is performed before DC offset correction is performed.

The second reason is that reception characteristics are improved when frequency offset correction is performed after IQ imbalance correction is performed, compared to a case where frequency offset correction is performed before IQ imbalance correction is performed.

The synchronization circuits shown in FIGS. 8 and 9 can also be modified to add structures for IQ imbalance correction. FIGS. 42 and 43 show circuit configurations of synchronous circuits configured to add a structure for IQ imbalance correction to the synchronous circuits shown in FIGS. 8 and 9 , respectively. In FIGS. 42 and 43 , components performing the same functions and operations as those shown in FIGS. 8 and 9 are denoted by the same reference numerals as those shown in FIGS. 8 and 9 . The synchronization circuits shown in FIGS. 42 and 43 will be described below.

In the synchronization circuit shown in FIG. 42 , an IQ imbalance corrector 1100 is provided between the DC offset corrector and the frequency offset corrector 24 . The output signal from HPF 21 is input to frequency offset estimator 22 and packet detector and coarse timing detector 23, and is also input to IQ imbalance estimator 1000. With the OFDM receiving apparatus shown in FIG. 33, the signal corresponding to the value γ from the frequency offset estimator 22 is input to the IQ imbalance estimator 1000, and then the IQ imbalance estimator 1000 performs the equation given by the equation ( 16) to (25) given operations to determine the correction coefficient. The determined correction coefficients are supplied from the IQ-imbalance estimator 1000 to the IQ-imbalance corrector 1100 to perform IQ-imbalance correction on the received complex baseband signal based on the correction coefficients.

As mentioned above, the synchronization circuit shown in FIG. 8 corresponding to FIG. 42 includes a path leading to the high-pass filter 21 and the DC offset estimator 25 for each I-axis and Q-axis input signal, and turns both on and off exclusively. switches 26 and 27 to switch between the two paths. In the case of activating the path leading to the HPF 21 among these paths, the IQ imbalance estimator 1000 determines a correction coefficient. Other structures (for example, switching control of the switches 26 and 27) are similar to the synchronous circuit shown in FIG. 8, and thus a detailed description thereof is omitted.

The synchronization circuit shown in Fig. 43 will be described below. In the synchronization circuit shown in FIG. 43, the switches 26 and 27 are not directly switched in response to the detection signals of the packet detector and the coarse timing detector 23, but a switch controller 28 is additionally provided. In this configuration as well, the IQ imbalance corrector 1100 is provided between the DC offset corrector and the frequency offset corrector 24, and performs IQ imbalance correction on the received complex baseband signal.

As shown in FIGS. 42 and 43 , instead of performing DC offset correction, IQ imbalance correction, and frequency offset correction using only STS, a function of more accurate frequency offset correction using LTS can be realized. 44 and 45 show the circuit configuration of the peripheral synchronization circuit for performing this function. 44 and 45 are diagrams showing a structure of a synchronous circuit configured to add a structure for IQ imbalance correction to the synchronous circuit shown in FIGS. Components performing the same functions and operations are denoted by the same reference numerals as shown in FIGS. 13 and 14 .

In the synchronization circuit shown in FIG. 44 , IQ imbalance estimator 1400 and IQ imbalance corrector 1500 for LTS are provided independently from IQ imbalance estimator 1200 and IQ imbalance corrector 1300 for STS. The IQ-imbalance estimator 1200 for STS determines correction coefficients for performing coarse IQ-imbalance correction using a short preamble. The IQ-imbalance estimator 1400 for LTS determines correction coefficients for performing fine IQ-imbalance correction using long-preambles T ₁ and T ₂ .

As mentioned above, at the end of the short preamble period, the IQ input is switched from the path leading to the high pass filter 21 to the path leading to the DC offset corrector. After the LTS, the IQ imbalance corrector 1300 performs IQ imbalance correction by multiplying the received complex baseband signal by a correction coefficient determined before the end of the STS. Thereafter, the IQ imbalance corrector 1500 corrects the minor (residual) IQ imbalance using the LTS.

In the circuit structure shown in FIG. 44 , a circuit module for performing IQ imbalance correction on the portion receiving the complex baseband signal after the LTS is provided separately. Alternatively, as shown in FIG. 45 , the IQ imbalance estimator 1600 and the IQ imbalance corrector 1700 may perform IQ imbalance correction using the STS, and may also perform IQ imbalance correction using the LTS.

The invention has been described in detail with reference to specific embodiments of the invention. However, it should be understood that various modifications and changes can be made to the embodiments by those skilled in the art without departing from the scope of the present invention.

Although the embodiments described herein are in the context of a wireless communication system conforming to the IEEE 802.11 standard, the scope of the present invention is not limited thereto. The receiver according to the embodiment of the present invention can also be used in a wireless communication system in which the same OFDM symbol is repeatedly transmitted in the preamble part, wherein the DC subcarrier is set as a null symbol to achieve accurate frequency offset estimation. Not only the application of wireless LAN but also various digital communication technologies based on OFDM transmission mode such as terrestrial digital broadcasting system, fourth generation mobile communication system, and power line carrier communication system also fall within the scope of the present invention.

Although the DC offset problem induced in direct conversion receivers can be overcome by embodiments of the present invention, the scope of the present invention is not limited thereto. Receivers that downconvert the RF received signal using other frequency conversion methods can be used to deal with DC offset and frequency offset issues.

It should be understood that the present invention has been disclosed in the form of exemplary embodiments, and the disclosure in this specification is not intended to limit the scope of the present invention. Rather, the true scope of the invention should be determined from the appended claims.

Those skilled in the art should understand that, according to design requirements and other factors, there can be various modifications, combinations, recombinations and improvements, all of which should be included within the scope of the claims of the present invention or equivalents.

Claims (31)

1. A wireless communication device for receiving a packet formed by a signal modulated by Orthogonal Frequency Division Multiplexing, comprising: A band-pass filter extracts an OFDM signal of a desired frequency band; a low noise amplifier having a gain controlled according to the strength of the received signal to amplify the OFDM signal of the desired frequency band; A frequency converter for down-converting the amplified OFDM signal into a baseband signal; an analog-to-digital converter converting the baseband signal into a digital signal; a first high-pass filter to remove a DC offset from said baseband signal corresponding to a predetermined preamble portion of said packet; a frequency offset estimator estimating a frequency offset from sampled signals constituting said baseband signal from which said DC offset has been removed by said first high-pass filter; a frequency offset corrector that removes the estimated frequency offset from the baseband signal; and A demodulator demodulates subcarrier signals arranged in a frequency domain from the baseband signal compensated for the frequency offset. 2. The wireless communication device of claim 1, wherein the first high pass filter comprises a differential amplifier. 3. The wireless communication apparatus of claim 1 , wherein the first high-pass filter inputs a detection signal to the frequency offset estimator upon detection of a rapid change in DC offset; and The frequency offset estimator does not perform frequency offset estimation on a sample signal obtained from the first high-pass filter when the detection signal is input, of the sample signals. 4. The wireless communication device of claim 1, wherein the OFDM signal input to the wireless communication device does not include a DC subcarrier; and The frequency offset estimator estimates an offset using a preamble that transmits two OFDM symbols. 5. The wireless communication device according to claim 4, wherein an OFDM symbol is composed of n subcarriers, When the i-th sample of the time waveform of the two transmitted OFDM symbols is denoted by s(i), the sample of the first transmitted OFDM symbol is represented by {s(0), s(1),...,s(n-1)} means that the second transmitted OFDM symbols are sampled by {s(n), s(n+1),..., s(2n-1)}, the frequency offset is represented by Δf, and the DC offset is represented by D, the baseband signal is given by the following equation (1), the first high-pass filter The device performs the operation given by the following equation (2) on the baseband signal, and outputs the sampled signal d(i), and The frequency offset corrector uses the sampled signal d(i) to perform the operation given by the following equation (3) to estimate the frequency offset Δf: r(i)=s(i)exp(j2πΔfi)+D ...(1) d(i)=r(i+1)-r(i) ＝s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi) ...(2) $\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span>}$ $<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; exp</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; j</span>}\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;\&Delta;f</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 3</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; .</span>$ 6. The wireless communication device according to claim 5, wherein when the absolute value of the output sampled signal d(i) is due to the i-th and ( When the DC offset between i+1) samples changes greatly, according to the following equation: d(i)=r(i+1)-r(i) ＝{s(i+1)exp(j2πΔf(i+1))-s(i)exp(j2πΔfi)}+{D(i+1)-D(i)} ...(4) The first high-pass filter outputs a detection signal in which no frequency offset is estimated to the frequency offset estimator by performing the operation given by the equation (3) on the i-th sample. 7. The wireless communication device according to claim 1, wherein the frequency converter converts the amplified OFDM signal of the desired frequency band using a local frequency generated by a local oscillator according to a direct conversion mode converted to baseband signal, and The phase of the local frequency oscillated by the local oscillator is inverted according to the frequency offset estimated by the frequency offset estimator. 8. The wireless communication device of claim 1, further comprising: a DC offset estimator that estimates a DC offset in the digital baseband signal converted by the analog-to-digital converter; and A DC offset corrector removes the estimated DC offset from the converted digital baseband signal. 9. The wireless communication apparatus of claim 8, wherein the first high pass filter inputs a detection signal to the DC offset estimator upon detection of a rapid change in DC offset; and The DC offset estimator excludes an estimated DC offset value determined before inputting the detection signal, and re-estimates a DC offset. 10. The wireless communication device of claim 1 , further comprising a second high-pass filter for filtering the baseband signal output from the frequency converter, Wherein, the analog-to-digital converter converts the baseband signal transmitted through the second high-pass filter into a digital signal. 11. The wireless communication device according to claim 10, wherein a cutoff frequency of the second high-pass filter is set lower than a cutoff frequency of the first high-pass filter. 12. The wireless communication apparatus of claim 8, further comprising a detector that performs packet detection and coarse timing detection using an output signal of the first high-pass filter. 13. The wireless communication device of claim 12, further comprising a dedicated connection of the output of the analog-to-digital converter to a path leading to the first high-pass filter or a path leading to the DC offset corrector switch. 14. The wireless communication apparatus of claim 13 , wherein when the detector detects the end of the predetermined preamble portion for frequency offset estimation, the switch switches the analog-to-digital converter to The output of is switched from a path leading to the first high pass filter to a path leading to the DC offset corrector. 15. The wireless communication apparatus according to claim 14 , wherein the frequency offset estimator uses the first high-pass filter from which the predetermined preamble portion has been removed within a period of time before the end of the predetermined preamble portion. Estimated frequency offset of said baseband signal at said DC offset. 16. The wireless communications apparatus of claim 14 , wherein the DC offset estimator estimates a DC offset for a period of time before the end of the predetermined preamble portion, and The DC offset corrector corrects the estimated DC offset from the portion of the baseband signal after the end of the predetermined preamble portion. 17. The wireless communications apparatus of claim 14 , wherein the frequency offset estimator estimates a frequency offset for a period of time before the end of the predetermined preamble portion, and The frequency offset corrector corrects the estimated frequency offset from the portion of the baseband signal after the end of the predetermined preamble portion. 18. The wireless communication device according to claim 13, further comprising determining a moving average of the correlation value of the output signal of the analog-to-digital converter, and controlling switching of the switch when the moving average exceeds a predetermined threshold switch controller. 19. The wireless communication apparatus of claim 13 , wherein the received packet includes a short preamble portion consisting of a short training sequence with a relatively large subcarrier spacing and a long training sequence with a relatively small subcarrier spacing. the long preamble portion of the training sequence, and At the beginning of the long-preamble portion after the end of the short-preamble portion, the switch switches the output of the analog-to-digital converter from a path leading to the first high-pass filter to the path leading to the DC offset corrector. 20. The wireless communication apparatus of claim 19 , wherein the frequency offset estimator estimates a frequency offset in the short preamble portion, and the frequency offset corrector estimates a frequency offset from the long preamble portion remove the estimated frequency offset in the code part, and the DC offset estimator estimates a DC offset in the short preamble portion, and the DC offset corrector removes the estimated DC offset from the long preamble portion, The wireless communication device also includes: a second frequency offset estimator, from the portion of the baseband signal following the long preamble portion from which the frequency offset and the DC offset estimated in the short preamble portion have been removed Estimated frequency offset in , and A second frequency offset corrector that removes the frequency offset estimated by the second frequency offset estimator from the portion of the baseband signal following the long preamble portion. 21. The wireless communication apparatus of claim 19 , wherein the frequency offset estimator estimates a frequency offset in the short preamble portion, and the frequency offset corrector estimates a frequency offset from the long preamble portion The estimated frequency offset is removed from the code part, the DC offset estimator estimates a DC offset in the short preamble portion, and the DC offset corrector removes the estimated DC offset from the long preamble portion, feeding back to the frequency offset estimation part of the baseband signal after the long preamble part from which the frequency offset and the DC offset estimated in the short preamble part have been removed to estimate a frequency offset from said portion of said baseband signal following said long-preamble portion, and The frequency offset corrector removes the estimated frequency offset from the portion of the baseband signal following the long-preamble portion. 22. The wireless communication apparatus according to claim 20 or 21, further comprising a channel estimator from which the frequency offset estimated in the short preamble part and the DC The channel is estimated in the baseband signal offset and corrected for said frequency offset estimated in said portion of said baseband signal following said long-preamble portion. 23. The wireless communication device of claim 1, further comprising: an in-phase and quadrature phase imbalance estimator estimating in-phase and quadrature phase imbalance from said sampled signal from which said DC offset has been removed by said first high-pass filter; and An in-phase and quadrature phase imbalance corrector corrects said in-phase and quadrature phase imbalance from said baseband signal. 24. The wireless communication apparatus of claim 23, wherein the in-phase and quadrature phase imbalance corrector performs in-phase and quadrature phase imbalance correction on the baseband signal that has been corrected for the DC offset. 25. The wireless communication apparatus according to claim 23, wherein said frequency offset corrector is derived from said in-phase and quadrature phase imbalance corrected by said in-phase and quadrature phase imbalance corrector The estimated frequency offset is removed from the baseband signal. 26. The wireless communications apparatus of claim 1 , wherein a plurality of preamble symbols used for frequency offset estimation are transmitted, and The frequency offset estimator adds the estimated frequency offsets for each of the preamble symbols to obtain a final estimated frequency offset value. 27. The wireless communication device of claim 26, further comprising a gain controller that adjusts the gain of the low noise amplifier, Wherein, when the corresponding preamble symbol is received, the frequency offset an estimator weights each of the frequency offsets estimated for the preamble symbols according to the gain set in the low noise amplifier, and adds the weighted frequency offsets to obtain a final frequency offset transfer value. 28. The wireless communication apparatus of claim 26, wherein the frequency offset estimator calculates weighting factors based on an absolute value of the frequency offset estimated for each of the preamble symbols, by the A weighting factor weights the frequency offsets, and adds the weighted frequency offsets to obtain a final frequency offset value. 29. The wireless communication apparatus according to claim 28, wherein the frequency offset estimator operates by applying a weighting factor of 0 to frequency offsets whose absolute values exceed a predetermined threshold and applying a weighting factor of 1 to frequency offsets whose absolute values do not exceed the predetermined threshold. The frequency offset estimated for the preamble symbol is weighted by the predetermined threshold frequency offset, and the weighted frequency offsets are added to obtain a final frequency offset value. 30. The wireless communications apparatus of claim 29, wherein the frequency offset estimator determines the predetermined threshold based on received signal strength. 31. The wireless communication apparatus of claim 28, wherein the frequency offset estimator calculates the frequency offset pair by applying a weighting factor formed by the inverse of the absolute value of the frequency offset to the frequency offset pair. The frequency offsets estimated by the preamble symbols are weighted, and the weighted frequency offsets are added to obtain a final frequency offset value.

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