TWI880789B - Method of controlling full-bridge resonant converter - Google Patents

Abstract

A method of controlling a full-bridge resonant converter includes steps of: operating a first upper switch in a first control action during a first time interval; operating a first lower switch in a second control action during the first time interval; operating a second upper switch in a third control action during the first time interval; operating a second lower switch in a fourth control action during the first time interval; operating the first upper switch in the fourth control action during a second time interval after the first time interval; operating the first lower switch in the third control action during the second time interval; operating the second upper switch in the second control action during the second time interval; operating the second lower switch in the first control action during the second time interval.

Description

Control method of full-bridge resonant converter

The present invention relates to a control method of a full-bridge resonant converter, and more particularly to an interleaved control method of a full-bridge resonant converter having asymmetric and non-complementary characteristics.

Due to the global energy shortage, countries around the world are gradually paying attention to the investment and development of green energy-related industries. With the vigorous development of electrification and intelligent technology in recent years, coupled with the formulation of relevant environmental protection regulations, various car manufacturers are also actively developing hybrid electric vehicles (PHEV) and pure electric vehicles (BEV) to reduce carbon emissions and promote sustainable development of the industry. For electric vehicles, the on-board charger is one of the important components, mainly responsible for converting the AC power of the power grid system into DC power to charge the battery pack. According to the current specifications of the automotive market, the energy conversion demand of the on-board charger is approximately 7KW-22KW. In order to meet the requirements of this specification, the design needs to use a full-bridge converter topology as the main circuit architecture. For applications where the battery pack has a wide range of voltage charging curves, in order to reduce the weight and volume of the on-board charger and increase the power density, the full-bridge LLC resonant converter topology with high conversion efficiency and adjustable output voltage is the first choice in design and is widely used.

Therefore, how to design a control method for a full-bridge resonant converter to solve the problems and technical bottlenecks existing in the prior art is an important topic studied by the inventors of this case.

The purpose of the present invention is to provide a control method for a full-bridge resonant converter. The input side circuit of the full-bridge resonant converter includes a first bridge arm and a second bridge arm, wherein the first bridge arm includes a first upper switch and a first lower switch, and the second bridge arm includes a second upper switch and a second lower switch. The control method includes: operating the first upper switch in a first time period as a first control action, wherein the first control action is an alternating action between an off state and an on state in the first time period; operating the first lower switch in a first time period as a second control action, wherein the second control action is an alternating action between an on state in the first time period and an off state; operating the second upper switch in the first time period as a third control action, wherein the third control action is an alternating action between an on state in the second time period and an off state, wherein the on state in the second time period is an alternating action between an on state in the first time period and an off state. The time of the on-state is greater than the time of the on-state in the first time period; operating the second lower switch in the first time period is a fourth control action, wherein the fourth control action is an alternating action between the off state and the on-state in the second time period; operating the first upper switch in the second time period following the first time period is a fourth control action; operating the first lower switch in the second time period is a third control action; operating the second upper switch in the second time period is a second control action; and operating the second lower switch in the second time period is the first control action.

In one embodiment, in the first time period, the first period of conduction state of the second control action of the first lower switch partially overlaps with the second period of conduction state of the third control action of the second upper switch.

In one embodiment, in the first time period, the first time period of the first control action of the first upper switch is partially overlapped with the second time period of the fourth control action of the second lower switch.

In one embodiment, in the second time period, the first period of conduction state of the second control action of the second upper switch partially overlaps with the second period of conduction state of the third control action of the first lower switch.

In one embodiment, in the second time period, the first period of the conduction state of the first control action of the second lower switch partially overlaps with the second period of the conduction state of the fourth control action of the first upper switch.

In one embodiment, in the first time period, the first upper switch and the first lower switch are hard-cut operations.

In one embodiment, during the first time period, the second upper switch and the second lower switch are operated in a soft-cut manner.

In one embodiment, during the second time period, the first upper switch and the first lower switch are operated in a soft-cut manner.

In one embodiment, during the second time period, the second upper switch and the second lower switch are hard-cut.

Thus, the control method of the full-bridge resonant converter proposed by the present invention has the following characteristics and advantages: 1. There is no need to modify the existing hardware mechanical components and increase the development cost. 2. It can effectively and significantly reduce the loss caused by MOSFET switching, improve efficiency, and improve the problem of component temperature rise. 3. Actively staggered control of MOSFET to disperse heat loss, prevent temperature rise from concentrating on specific components, and extend service life. 4. Replace the burst mode operation mode and apply it to light load or low voltage conditions to effectively reduce output transient ripple. 5. Compared with the traditional PWM operation method, better EMC effect can be obtained by improving the hard switching of MOSFET. 6. Compared with the traditional PSM operation method, there is no need to use phase shift control, and software development is simple.

In order to further understand the technology, means and effects adopted by the present invention to achieve the intended purpose, please refer to the following detailed description and drawings of the present invention. It is believed that the purpose, features and characteristics of the present invention can be understood in depth and in detail. However, the attached drawings are only provided for reference and explanation, and are not used to limit the present invention.

The technical content and detailed description of the present invention are described as follows with reference to the accompanying drawings.

The primary side of the full-bridge LLC resonant converter is composed of two arms with four power switches (MOSFET), resonant inductor, transformer leakage inductor, transformer magnetizing inductor and resonant capacitor. Its main operation mode is pulse frequency modulation (PFM). Under the condition of fixed switching duty cycle, the switching frequency of MOSFET is modulated to achieve conversion power and output voltage control. The resonant tank design and component specifications are selected according to the output specification requirements. The gain curve of the resonant converter is obtained through design and the switching frequency operating range is determined. The gain curve shows that under light load or low voltage conditions, the MOSFET switching frequency needs to be adjusted toward high frequency to obtain a stable output voltage.

Due to the limitation of related component characteristics, the upper limit of switching frequency operation will be limited. Therefore, when the switching frequency needs to be adjusted to a level higher than the limit, the controller will usually directly turn off the MOSFET until the voltage drops and the MOSFET switching frequency also drops to the operating range. The MOSFET will continue to switch and continue to transfer energy to the output. This operation mode is called burst mode, but this operation mode will cause larger output transient ripples and intermittent energy supply, resulting in output current oscillation. Therefore, the use of duty modulation operation mode to replace or improve the impact of burst mode has been widely discussed and applied in recent years.

When the full-bridge LLC resonant converter uses the traditional duty modulation method under light load or low voltage conditions, the flexible switching characteristics will be lost, resulting in increased MOSFET switching loss, decreased efficiency, and increased MOSFET component temperature, which in turn causes component damage and reduced product life. The technical feature of the present invention is that it does not require modification of the existing hardware circuit topology, is more efficient than the traditional method, and can effectively reduce the MOSFET component temperature, increase the service life of vehicle components, and extend the mechanical life of water-cooled pumps, water-cooled fans, etc.

The present invention uses innovative control methods and strategies to significantly reduce the switching loss of the full-bridge LLC resonant converter. At the same time, this control strategy can also disperse the temperature accumulated by the MOSFET components, avoiding the burden on the water cooling system due to high temperature, and the need to limit the output capacity (de-rating) due to excessive temperature, thereby improving the practicality of the product and the consumer experience. In addition, when the load changes, the output voltage and current will not be unstable and oscillating due to mode switching. The present invention uses simulation analysis to determine the feasibility of its concept, and conducts actual product testing and waveform measurement comparison observation, and further uses temperature rise testing to prove the perfect improvement effect of this technology.

Please refer to FIG. 1, which is a circuit diagram of the full-bridge resonant converter of the present invention. The full-bridge resonant converter shown in FIG. 1 is more specifically a full-bridge bidirectional CLLC resonant converter, which includes a DC bus side and a battery side. Since the full-bridge bidirectional CLLC resonant converter has a bidirectional power flow structure, the DC bus side or the battery side can be used as the input side, and the other side is the output side.

Since the full-bridge bidirectional CLLC resonant converter is a symmetrical power conversion architecture, for the convenience of explanation, the DC bus side is the input side, and the battery side is the output side. Therefore, the full-bridge resonant converter shown in FIG1 includes an input side circuit 10, an output side circuit 20, and a resonant circuit 30. The input side circuit 10 includes a first bridge arm Lg1 and a second bridge arm Lg2 connected in parallel to the first bridge arm Lg1. The first bridge arm Lg1 includes a first upper switch Q1 and a first lower switch Q2; the second bridge arm Lg2 includes a second upper switch Q3 and a second lower switch Q4. The first upper switch Q1 and the first lower switch Q2 are connected to the first node N1, and the second upper switch Q3 and the second lower switch Q4 are connected to the second node N2. The switches Q1-Q4 are controlled by control signals generated by a controller (or control circuit, not shown). Specifically, the controller generates a first upper switch control signal S1 to control the first upper switch Q1, generates a first lower switch control signal S2 to control the first lower switch Q2, generates a second upper switch control signal S3 to control the second upper switch Q3, and generates a second lower switch control signal S4 to control the second lower switch Q4. In addition, the input side circuit 10 further includes an input capacitor Ck.

As shown in FIG1 , the output side circuit 20 includes a first bridge arm Lga and a second bridge arm Lgb connected in parallel to the first bridge arm Lga. The first bridge arm Lga includes a first upper switch Qa and a first lower switch Qb; the second bridge arm Lgb includes a second upper switch Qc and a second lower switch Qd. The first upper switch Qa and the first lower switch Qb are connected to a first node Na, and the second upper switch Qc and the second lower switch Qd are connected to a second node Nb. The switches Qa-Qd are controlled by a control signal generated by the controller. In addition, the output side circuit 20 further includes an output filter capacitor Cf.

The MOSFET switching switch of the full-bridge CLLC resonant converter mainly uses PFM switching to perform energy modulation control. When the load is light or the voltage is low, the duty modulation control strategy will be used to replace or improve the impact of the burst mode. In the duty mode, the parasitic capacitance voltage on the MOSFET switch enters the soft switching area after the discharge is completed. However, due to the long dead time, the parasitic capacitance voltage is recharged due to the resonant tank current phase change, resulting in the loss of flexible switching. The parasitic capacitance ringing voltage before the MOSFET switch is turned on will cause hard switching when the MOSFET is turned on, which will cause serious energy loss, component temperature rise and electromagnetic radiation interference (EMC). Therefore, the present invention proposes an innovative control strategy that will effectively reduce and improve the MOSFET hard switching loss caused by the full-bridge CLLC resonant converter operating in duty mode when the load is light or the voltage is low, such as pulse width modulation (PWM) or phase shift modulation (PSM) mode, and disperse the component loss to reduce the temperature of the MOSFET switch component.

Please refer to Figure 2, S1 is the input side upper arm MOSFET drive signal waveform, S2 is the input side lower arm MOSFET drive signal waveform, Vds1 is the input side upper arm MOSFET parasitic capacitance voltage waveform, Vds2 is the input side lower arm MOSFET parasitic capacitance voltage waveform, Ipri is the input side resonant current waveform. The energy transfer status when the MOSFET is turned on can be observed. Through the drive signal waveforms S1, S2 and the parasitic capacitance voltage waveforms Vds1, Vds2, it can be observed that the MOSFET operates in a hard switching state in the PWM mode. Due to the ringing state of Vds when the MOSFET is turned on (S1, S2 low to Therefore, it can be known that in the PWM operation mode, the four MOSFETs in both arms of the input side are all operated in a hard switching state, which will cause serious energy loss and component temperature rise.

In the actual machine test, the full-bridge CLLC resonant converter uses PWM mode to implement the duty modulation method, further verifying the loss and component temperature rise caused by the hard switching of MOSFET. Under the test output voltage of 350V, ambient temperature of 25 degrees, and cooling water temperature of 25 degrees, when the load decreases, the MOSFET enters the duty mode range. Due to the loss caused by hard switching, the MOSFET component temperature rises sharply to the protection point. Although this method improves the impact caused by the burst mode, it brings about a decrease in efficiency and may require additional costs to solve the problem of component temperature rise.

Please refer to Figure 3, which shows the full-bridge CLLC resonant converter using the PSM mode to implement duty modulation, and observes the MOSFET switching state through waveforms. The switching state of the MOSFET in the PSM operation mode can be observed through the drive signal waveforms S1, S4 and the parasitic capacitance voltage waveforms Vds1, Vds4. This operation mode drives the MOSFET to turn on when the Vds1 voltage is zero, and operates in a soft switching state. However, when the MOSFET is turned on (S4 low to high), the lagging arm will be forced to short-circuit at Vds4 of 800V. Therefore, it can be known that in the PSM operation mode, the four MOSFETs in the two arms on the input side, although the two MOSFETs in the leading arm operate in soft switching, the two MOSFETs in the lagging arm still operate in hard switching, and the Vds4 voltage is higher than the PWM modulation method. Compared with the PWM modulation method, although it can reduce some losses and improve efficiency, the problem of temperature rise of the hard switching MOSFET components still exists.

Please refer to Figure 5, which is a flow chart of the control method of the full-bridge resonant converter of the present invention. The control method of the full-bridge resonant converter of the present invention is not limited to controlling the full-bridge bidirectional CLLC resonant converter shown in Figure 1 mentioned above. For the convenience of explanation, the description of the control method of Figure 5 is based on the full-bridge bidirectional CLLC resonant converter shown in Figure 1 as an example. Furthermore, in conjunction with Figure 4, it is a waveform diagram of the current, voltage and control signal of the control method of the full-bridge resonant converter of the present invention. As mentioned above, the input side circuit 10 includes a first bridge arm Lg1 and a second bridge arm Lg2. The first bridge arm Lg1 includes a first upper switch Q1 and a first lower switch Q2; the second bridge arm Lg2 includes a second upper switch Q3 and a second lower switch Q4. The controller generates a first upper switch control signal S1 to control the first upper switch Q1, generates a first lower switch control signal S2 to control the first lower switch Q2, generates a second upper switch control signal S3 to control the second upper switch Q3, and generates a second lower switch control signal S4 to control the second lower switch Q4.

The control method includes: operating the first upper switch Q1 in the first time interval T1 as a first control action, wherein the first control action is an alternating action between an off state and a first time period on state (step S10). As shown in FIG4 , the first time interval T1 is a time interval between time t1 and time t2. In addition, the second time interval T2 is a time interval between time t2 and time t3. And, during the operation process, the first time interval T1 and the second time interval T2 are performed alternately. The first control action is that the first upper switch control signal S1 turns off the first upper switch Q1 at a low level, and then turns on the first upper switch Q1 at a high level for a time To1 that continues the first time period on state, and the off state and the first time period on state are alternating actions.

At the same time, operating the first lower switch Q2 in the first time interval T1 is a second control action, wherein the second control action is an alternating action between the on state and the off state in the first time period (step S20). As shown in FIG. 4 , the second control action is that the first lower switch control signal S2 continues to be on for a time To1 of the first time period, and then turns off the first lower switch Q2 at a low level, and the on state and the off state in the first time period are alternating actions.

At the same time, operating the second upper switch Q3 in the first time interval T1 is a third control action, wherein the third control action is an alternating action between the on state in the second time period and the off state (step S30). It is worth mentioning that the time To2 of the on state in the second time period is greater than the time To1 of the on state in the first time period. As shown in FIG. 4 , the third control action is that the second upper switch control signal S3 continues to turn on the second upper switch Q3 at a high level for the time To2 of the on state in the second time period, and then turns off the second upper switch Q3 at a low level, and the on state in the second time period and the off state are alternating actions.

At the same time, operating the second lower switch Q4 in the first time interval T1 is a fourth control action, wherein the fourth control action is an alternating action between the off state and the on state in the second time period (step S40). As shown in FIG. 4 , the fourth control action is that the second lower switch control signal S4 turns off the second lower switch Q4 at a low level, and then turns on the second lower switch Q4 at a high level for a time To2 of the on state in the second time period, and the off state and the on state in the second time period are alternating actions.

Then, the first upper switch Q1 is operated in the second time interval T2 following the first time interval T1 as the fourth control action (step S50). As shown in FIG4 , the fourth control action is that the first upper switch control signal S1 turns off the first upper switch Q1 at a low level, and then turns on the first upper switch Q1 at a high level for a time To2 of the second time period conduction state, and the turn-off state and the second time period conduction state are alternately operated.

At the same time, the first lower switch Q2 is operated in the second time interval T2 as the third control action (step S60). As shown in FIG4, the third control action is that the first lower switch control signal S2 continues to be in the on state of the second period for a time To2 at a high level to turn on the first lower switch Q2, and then turns off the first lower switch Q2 at a low level, and the on state and the off state of the second period are alternately operated.

At the same time, the second upper switch Q3 is operated in the second time interval T2 as the second control action (step S70). As shown in FIG4, the second control action is that the second upper switch control signal S3 continues to be turned on at a high level for a time To1 of the first period of the on state, and then turns off the second upper switch Q3 at a low level, and the on state and the off state in the first period of time are alternately operated.

At the same time, the second lower switch Q4 is operated in the second time interval T2 as the first control action (step S80). As shown in FIG4, the first control action is that the second lower switch control signal S4 turns off the second lower switch Q4 at a low level, and then turns on the second lower switch Q4 at a high level for a time To1 of the first period of conduction state, and the turn-off state and the first period of conduction state are alternately operated.

According to the above description, it can be known that: in the first time interval T1, the first time segment conduction state of the second control action of the first lower switch Q2 partially overlaps with the second time segment conduction state of the third control action of the second upper switch Q3. In the first time interval T1, the first time segment conduction state of the first control action of the first upper switch Q1 partially overlaps with the second time segment conduction state of the fourth control action of the second lower switch Q4. In the second time interval T2, the first time segment conduction state of the second control action of the second upper switch Q3 partially overlaps with the second time segment conduction state of the third control action of the first lower switch Q2. In the second time interval T2, the first time segment conduction state of the first control action of the second lower switch Q4 partially overlaps with the second time segment conduction state of the fourth control action of the first upper switch Q1. Therefore, the interleaved control method of the full-bridge resonant converter with asymmetric and non-complementary characteristics proposed by the present invention can observe the soft (soft) or hard-cut operation of each switch from the parasitic capacitance voltage waveform of each switch. In the first time interval T1, the first upper switch Q1 and the first lower switch Q2 are in hard-cut operation, and in the second time interval T2, the first upper switch Q1 and the first lower switch Q2 are converted to soft-cut operation. Correspondingly, in the first time interval T1, the second upper switch Q3 and the second lower switch Q4 are in soft-cut operation, and in the second time interval T2, the second upper switch Q3 and the second lower switch Q4 are converted to hard-cut operation.

Therefore, the present invention proposes a novel intelligent switching method (smart-PWM) to modulate duty. Compared with the traditional PWM and PSM operation modes, it is proved through practice that this control strategy can effectively reduce switching loss and improve efficiency in the duty modulation range of a full-bridge CLLC resonant converter or a full-bridge LLC resonant converter with a lighter load or a low voltage, and it can also improve the problem caused by the temperature rise of MOSFET components. The technical features of the present invention are described below: 1. Different from the switching method of the traditional full-bridge phase-shift converter, the two MOSFETs in one arm can be operated in a flexible switching without using phase shift, which effectively reduces the MOSFET switching loss. In the application of the full-bridge LLC resonant converter, the efficiency of the other arm can also be improved; 2. Different from the switching method of the traditional full-bridge converter, the first upper switch Q1, the first lower switch Q2 and the second lower switch Q4, the second upper switch Q3 on the input side will be driven to conduct in an asymmetrical manner; 3. The first upper switch Q1, the first lower switch Q2 or the second lower switch Q4, The second upper switch Q3 uses a non-complementary PWM mode for duty modulation control; 4. The first upper switch Q1, the first lower switch Q2 and the second lower switch Q4, the second upper switch Q3 on the input side are driven to conduct using an interleaved control method, and the number of interleavings is further controlled to achieve a temperature rise dispersion balance effect; 5. The input side resonant current Ipri waveform phase timing is the same as the traditional PWM mode duty modulation method, and will not cause transformer saturation or excitation offset; 6. When the load or output changes, the PFM and PWM modulation modes switch, which will not cause abnormal transient phenomena.

The switching state of the MOSFET in the smart-PWM operation mode of the present invention can be observed through the drive signal waveforms S1-S4 and the parasitic capacitance voltage waveforms Vds1-Vds4. It can be found that the smart-PWM operation mode is similar to the PSM operation mode, which can drive the MOSFET to turn on when the Vds voltage is zero in one arm, so that it operates in a soft switching state. It is also observed that when the Vds voltage is in a ringing state when the other arm operating in hard switching drives the MOSFET to turn on, its Vds voltage is lower than that in the PSM operation mode, and the switching loss is smaller, which can obtain better efficiency improvement results. By observing the switching of the smart-PWM operation mode of the present invention, it can be found that the two arms on the primary side are operated in a flexible switching state in turn, so the temperature rise problem of the single-arm MOSFET element in the PSM operation mode can be effectively improved, and even the number of flexible switching operations of the two arms in turn can be planned and actively controlled through temperature sensing to achieve the best temperature rise improvement effect. Furthermore, the dynamic stability of the smart-PWM of the present invention during mode switching can be observed, and it will not cause abnormal instability of the controller and output current, and can stably control the output after the mode conversion.

According to the measured temperature rise of MOSFET components, the smart-PWM of the present invention has a significant improvement effect compared with the traditional PWM mode duty modulation method. Under the conditions of an ambient temperature of 25 degrees, a cooling water temperature of 25 degrees and the same output load, after using the smart-PWM mode duty modulation method of the present invention, the MOSFET component temperature has been greatly improved from the original situation of being greater than 125 degrees and unable to achieve thermal stability to 72 degrees thermal balance stability. Further tests have verified that under the worst conditions of an ambient temperature of 85 degrees, a cooling water temperature of 65 degrees and an output voltage of 246V, an excellent performance of 115.5 degrees thermal balance stability can be obtained.

In summary, the present invention has the following features and advantages:

1. No need to modify existing hardware and mechanical components, and no need to increase development costs.

2. It can effectively and significantly reduce the loss caused by MOSFET switching, improve efficiency, and improve component temperature rise problems.

3. Active interleaving controls MOSFET to disperse heat loss, prevent temperature rise from concentrating on specific components, and extend service life.

4. Replace the burst mode operation mode and apply it to light load or low voltage conditions to effectively reduce output transient ripple.

5. Compared with the traditional PWM operation mode, better EMC effect can be obtained by improving the hard switching of MOSFET.

6. Compared with the traditional PSM operation mode, there is no need to use phase shift control, and software development is simple.

The above description is only a detailed description and drawings of the preferred specific embodiments of the present invention, but the features of the present invention are not limited thereto and are not intended to limit the present invention. The entire scope of the present invention shall be subject to the following patent application scope. All embodiments that conform to the spirit of the patent application scope of the present invention and similar variations thereof shall be included in the scope of the present invention. Any changes or modifications that can be easily thought of by anyone familiar with the art within the field of the present invention shall be covered by the following patent scope of the present case.

10: Input side circuit 20: Output side circuit 30: Resonance circuit Lg1: First bridge arm Lg2: Second bridge arm Q1: First upper switch Q2: First lower switch Q3: Second upper switch Q4: Second lower switch S1: First upper switch control signal S2: First lower switch control signal S3: Second upper switch control signal S4: Second lower switch control signal N1: First node N2: Second node Lga: First bridge arm Lgb: Second bridge arm Qa: First upper switch Qb: First lower switch Qc: Second upper switch Qd: Second lower switch Na: First node Nb: Second node Ck: Input capacitor Cf: Filter capacitor Vds1-Vds4: Parasitic capacitor voltage Ipri: input side resonant current T1: first time interval T2: second time interval To1: first time interval conduction time To2: second time interval conduction time t1-t3: time S10-S80: steps

FIG1 is a circuit diagram of the full-bridge resonant converter of the present invention.

FIG. 2 is a waveform diagram of current, voltage and control signal of the full-bridge resonant converter of the present invention in PWM operation mode.

FIG3 is a waveform diagram of current, voltage and control signal of the full-bridge resonant converter of the present invention in the PSM operation mode.

FIG4 is a waveform diagram of current, voltage and control signal of the control method of the full-bridge resonant converter of the present invention.

FIG5 is a flow chart of the control method of the full-bridge resonant converter of the present invention.

S10-S80: Steps

Claims (9)

A control method for a full-bridge resonant converter, wherein an input side circuit of the full-bridge resonant converter includes a first bridge arm and a second bridge arm, wherein the first bridge arm includes a first upper switch and a first lower switch, and the second bridge arm includes a second upper switch and a second lower switch, and the control method includes: Operating the first upper switch in a first time interval as a first control action, wherein the first control action is an alternating action of an off state and a first time period on state; Operating the first lower switch in the first time interval as a second control action, wherein the second control action is an alternating action of the first time period on state and the off state; Operating the second upper switch in the first time interval is a third control action, wherein the third control action is an alternating action between the on state and the off state in the second time interval, wherein the time of the on state in the second time interval is greater than the time of the on state in the first time interval; Operating the second lower switch in the first time interval is a fourth control action, wherein the fourth control action is an alternating action between the off state and the on state in the second time interval; Operating the first upper switch in a second time interval following the first time interval is the fourth control action; Operating the first lower switch in the second time interval is the third control action; Operating the second upper switch in the second time interval is the second control action; and Operating the second lower switch in the second time interval is the first control action. A control method for a full-bridge resonant converter as described in claim 1, wherein in the first time period, the first period conduction state of the second control action of the first lower switch partially overlaps with the second period conduction state of the third control action of the second upper switch. A control method for a full-bridge resonant converter as described in claim 1, wherein in the first time period, the first period of conduction state of the first control action of the first upper switch partially overlaps with the second period of conduction state of the fourth control action of the second lower switch. A control method for a full-bridge resonant converter as described in claim 1, wherein in the second time period, the first period conduction state of the second control action of the second upper switch partially overlaps with the second period conduction state of the third control action of the first lower switch. A control method for a full-bridge resonant converter as described in claim 1, wherein in the second time period, the first period conduction state of the first control action of the second lower switch partially overlaps with the second period conduction state of the fourth control action of the first upper switch. A control method for a full-bridge resonant converter as described in claim 1, wherein in the first time period, the first upper switch and the first lower switch are hard-cut operations. A control method for a full-bridge resonant converter as described in claim 1, wherein in the first time period, the second upper switch and the second lower switch are in soft-cut operation. A control method for a full-bridge resonant converter as described in claim 1, wherein in the second time period, the first upper switch and the first lower switch are in soft-cut operation. A control method for a full-bridge resonant converter as described in claim 1, wherein in the second time period, the second upper switch and the second lower switch are hard-cut operations.