TWI905747B - Interleaved half-bridge phase-shift LLC resonant converter with wide output voltage range - Google Patents

Abstract

An interleaved half-bridge phase-shift LLC resonant converter with a wide output voltage range includes an inverter circuit module, a resonant circuit module connected to the inverter circuit module, a rectifier circuit module connected to the resonant circuit module, and a signal logic circuit module connected to the inverter circuit module. The resonant circuit module includes an upper resonant circuit, an upper transformer connected to the upper resonant circuit, a lower resonant circuit connected to the upper resonant circuit, and a lower transformer connected to the lower resonant circuit for outputting an output voltage. The signal logic circuit module provides two control signals—one for the leading arm and one for the lagging arm—for the control operation frequency, the duty cycle, and the phase shift. The rectifier circuit module can rectify the output voltage to output an operating voltage.

Description

Interleaved half-bridge phase-shift LLC resonant converter with wide output voltage range

This invention relates to a power converter, and more particularly to an interleaved half-bridge phase-shift LLC resonant converter with a wide output voltage range.

LLC resonant converters are a common type of isolated power converter. In addition to their simple structure and advantages of high efficiency and high power density over a wide load range, they can also achieve zero-voltage switching (ZVS) for MOSFETs and zero-current switching (ZCS) for diodes. As a result, they are widely used in rapidly developing electric vehicles in recent years to minimize losses and improve efficiency when electric vehicles perform necessary power switching during driving.

However, when the switching frequency of an LLC resonant converter is higher than the resonant frequency, the rectifier diodes cannot achieve zero-current switching. Furthermore, the relatively flat voltage gain curve in this frequency range makes output voltage regulation difficult. Therefore, LLC resonant converters are typically designed to operate at switching frequencies less than or equal to the resonant frequency to achieve zero-voltage switching, zero-current switching, and good output voltage regulation. Furthermore, when the operating frequency deviates significantly from the resonant frequency, the increased circulating current in the resonant circuit greatly reduces the conversion efficiency. This necessitates that the LLC resonant converter strike a balance between conversion efficiency and operating range to meet the requirements. Directly adjusting the voltage of the LLC resonant converter via frequency modulation will cause one of the flexible switching characteristics—zero-voltage switching or zero-current switching—to fail, thus increasing losses and reducing efficiency.

Therefore, the purpose of this invention is to provide a resonant converter that can maintain flexible switching characteristics while providing a wide output voltage range.

Therefore, the resonant converter of the present invention includes an inverter circuit module adapted to be connected to a power source for providing an input voltage, a resonant circuit module connected downstream of the inverter circuit module, a rectifier circuit module connected downstream of the resonant circuit module for rectifying the output voltage to output an operating voltage, and a signal logic circuit module connected to the inverter circuit module.

The inverter circuit module includes a front arm and a rear arm.

The resonant circuit module includes an upper resonant circuit, an upper transformer connected to the upper resonant circuit, a lower resonant circuit connected to the upper resonant circuit, and a lower transformer connected to the lower resonant circuit and used together with the upper transformer to transform the input voltage and output an output voltage.

The signal logic circuit module is used to provide two control signals to the forearm and the lag arm respectively. These control signals are used to control the operating frequency, working cycle, and phase displacement of the forearm and the lag arm respectively.

The advantage of this invention is that the control signals generated by the signal logic circuit module can properly control the phase shift between the leading arm and the trailing arm of the inverter circuit module, so that the power signal waveform can be completely superimposed before and after voltage switching. In this way, while providing a wide output voltage range, it still maintains the flexible switching characteristics, thereby reducing power loss and improving operating efficiency.

Referring to Figures 1 and 2, an embodiment of the resonant converter of the present invention is shown. The embodiment includes an inverter circuit module 2 adapted to be connected to a power supply 1 for providing an input voltage, a resonant circuit module 3 connected downstream of the inverter circuit module 2, a rectifier circuit module 4 connected downstream of the resonant circuit module 3 for rectifying the output voltage to output an operating voltage, a signal logic circuit module 5 connected to the inverter circuit module 2, and an isolation circuit module 6 connected between the signal logic circuit module 5 and the inverter circuit module 2.

Referring to Figure 3, this drive signal circuit uses the Texas Instruments IC UC3879 to generate the control drive signal. To prevent signal interference, as shown in Figure 3(a), the Broadcom IC HCPL-3120 is used for signal isolation. Its internal pinout and usage are shown in Figures 3(a) and 3(b). This prevents signal interference when the converter is powered on, which could lead to a short circuit and damage to the circuit. The isolation circuit module 6 consists of four optocoupler drivers 61, four resistors 62, and two diodes 63. That is, a drive isolator of a half-bridge converter is formed by two optically coupled drivers 61, a diode 63 and a capacitor to form a boost charge pump to constitute a bootstrap circuit.

Referring again to Figures 1 and 2, the inverter circuit module 2 includes a leading arm 21 and a trailing arm 22. The leading arm 21 has two first power switching switches 211, and the trailing arm 22 has two second power switching switches 221. The leading arm 21 is driven by the positive edge of an effective pulse width modulation (PWM) signal, and the trailing arm 22 is driven by the positive edge of an PWM signal whose phase shift lags behind the leading arm 21 switch. Furthermore, each of the first power switching switches 211 and each of the second power switching switches 221 has a built-in intrinsic diode and parasitic capacitor.

The resonant circuit module 3 includes an upper resonant circuit 31, an upper transformer 32 connected to the upper resonant circuit 31, a lower resonant circuit 33 connected to the upper resonant circuit 31, and a lower transformer 34 connected to the lower resonant circuit 33 and used together with the upper transformer 32 to transform the input voltage and output an output voltage. Specifically, the resonant circuit module 3 is composed of two half-bridge LLC circuits.

The upper resonant circuit 31 has an upper first inductor 311, an upper capacitor 312 connected in series with the upper first inductor 311, and an upper magnetizing inductor 313 inherent in the upper transformer 32 itself; the lower resonant circuit 33 has a lower first inductor 331, a lower capacitor 332 connected in series with the lower first inductor 331, and a lower magnetizing inductor 333 inherent in the lower transformer 34 itself. Furthermore, the upper transformer 32 has an upper primary side 321 and an upper secondary side 322, while the lower transformer 34 has a lower primary side 341 and a lower secondary side 342. The upper primary side 321 of the upper transformer 32 is connected in parallel with the lower primary side 341 of the lower transformer 34, while the upper secondary side 322 of the upper transformer 32 is connected in series with the lower secondary side 342 of the lower transformer 34.

The rectifier circuit module 4 includes three sub-circuits 41, each connected only to the upper transformer 32, each connected only to the lower transformer 34, and connected to both the upper transformer 32 and the lower transformer 34, and an output filter capacitor 42 connected in parallel with the sub-circuits 41. Each sub-circuit 41 has two rectifier transistors 411, and the sub-circuits 41 can provide the voltage level _VO required by the load.

The signal logic circuit module 5 has pre-programmed logic to provide two control signals to the lead arm 21 and the lag arm 22, respectively. These control signals are used to control the operating frequency, working cycle, and phase shift of the lead arm 21 and the lag arm 22, respectively. Specifically, regarding the phase shift of the lead arm 21 and the lag arm 22, the signal logic circuit module 5 includes an adjustment circuit for receiving a reference voltage set according to the voltage level _VO , and adjusting the phase shift between the lead arm 21 and the lag arm 22 accordingly. In addition, the signal logic circuit module 5 is also used to adjust the lag time of the signal between any of the first power switching switches 211 and any of the second power switching switches 221. In this embodiment, in order to facilitate intuitive setting of the reference voltage according to the load requirements, the reference voltage is a voltage that is proportional to the voltage level _VO shift.

Referring to Figure 3, this illustrates the operating mode of the isolation circuit module 6. When the low-end drive switch in the forearm switch is activated... On and high-side drive switch End of power supply capacitor , Charging, however power supply via diode (63) Conduction, for capacitor , Charging to provide high-end drive switches in advanced forearm switches. Required voltage. When the high-side drive switch in the forearm switch... If conduction occurs, the low-side driver switch... Cut-off. There is a lag time between the high-end and low-end drive switch signals in the advanced forearm 21 to prevent the switch from opening. With switch Simultaneous conduction caused a short circuit in the forearm switch, resulting in its burnout. Additionally, the low-end drive switch in the hind arm switch... On and high-side drive switch Deadline, at the same time Power supply to capacitor , Charging, however The power supply passes through the diode. (63) Conduction, for capacitor , Charging to provide high-end drive switches in the lag arm switch. Required voltage. High-side drive switch in the lag arm switch. When conduction occurs, the low-side driver switch... Cut-off. There is a lag time between the high-end and low-end drive switch signals in the lag arm 22 to prevent the switch from opening. With switch Simultaneous conduction caused the trailing arm switch to short-circuit and burn out.

Referring to Figure 4 and Figure 2, and in conjunction with the control signals provided by the signal logic circuit module 5 to the inverter circuit module 2, the resonant circuit module 3 in this embodiment, when working with the rectifier circuit module 4, will operate in multiple states. The first half-cycle operating mode will be explained below, while the second half-cycle will be generally symmetrical.

Operating state 1 [t0,t1]: One of the second power switching switches 221 of the lag arm 22 is turned on, while the other second power switching switch 221 and the first power switching switches 211 are all turned off. At this time, the first power switching switches 211 of the lead arm 21 will be in a lag period. In conjunction with the upper first inductor 311 of the upper resonant circuit 31 of the resonant circuit module 3, the parasitic capacitances of the first power switching switches 211 will release energy and store energy respectively. By making proper use of this lag period, the parasitic capacitances can complete the energy release before the first power switching switches 211 are turned on again, thus achieving zero voltage switching (ZVS).

Operating State Two [t1,t2]: Continuing from Operating State One, the first power switching switch 211 corresponding to the upper resonant circuit 31 is turned on. At this time, the corresponding parasitic capacitor has released energy, allowing the turned-on first power switching switch 211 to complete zero-voltage switching. At this time, the current of the upper first inductor 311 of the upper resonant circuit 31 and the upper magnetizing inductor 313 of the upper transformer 32 itself continues to rise, while the current of the lower resonant circuit 33 decreases. The resulting current difference allows energy to be transferred to the lower resonant circuit 33. At this time, the rectifier transistors 411 of some of the sub-circuits 41 are turned on to provide load energy, so that the current flowing through the other part of the rectifier transistors 411 drops to zero, thereby achieving zero current switching (ZCS).

Operating State 3 [t2,t3]: Continuing from Operating State 2, the second power switching switch 221 corresponding to the lower resonant circuit 33 is turned off, causing the second power switching switches 221 of the lag arm 22 to be in idle time. At this time, the second power switching switches 221 cooperate with the lower first inductor 331 of the lower resonant circuit 33, so that the parasitic capacitance of the second power switching switches 221 performs energy release and energy storage respectively. At this time, the lower resonant circuit 33 will still transmit current to the upper resonant circuit 33 due to the current difference with the upper resonant circuit 31. In conjunction with the conduction of some of the rectifier transistors 411, the upper magnetizing inductance 313 of the upper transformer 32 of the upper resonant circuit 31 is clamped.

Operating state four [t3,t4]: The second power switching switch 221 corresponding to the upper resonant circuit 31 is further turned on. At this time, the corresponding parasitic capacitor has completed energy release, so it can be turned on under zero-voltage switching. At this time, the energy is still transferred to the lower resonant circuit 33 through the current difference between the upper resonant circuit 31 and the lower resonant circuit 33. As long as an appropriate idle time is provided, the rectifier transistors 411 corresponding to the lower resonant circuit 33 will provide load energy after being turned on. With the energy release of the aforementioned parasitic capacitor, the effect of zero-voltage switching can be achieved. Next, since the upper magnetizing inductance 313 of the upper transformer 32 in the upper resonant circuit 31 and the lower magnetizing inductance 333 of the lower transformer 34 in the lower resonant circuit 33 are clamped by the output voltage and do not participate in the resonance, the current difference between the upper resonant circuit 31 and the lower resonant circuit 33 gradually disappears. At this time, the current of some of the rectifier transistors 411 also drops to zero, thus achieving the condition for zero current switching.

Operating state 5 [t4,t5]: Both the first power switching switch 211 and the second power switching switch 221 corresponding to the upper resonant circuit 31 are turned on, and when energy will no longer be supplied to the lower resonant circuit 33, the lower transformer 34 will transfer energy through the current difference between the lower first inductor 331 and the lower magnetizing inductor 333. At this time, the upper transformer 32 does not provide power, so the upper first inductor 311, the upper magnetizing inductor 313, and the upper capacitor 312 resonate. When the rectifier transistors 411 of the corresponding sub-circuit 41 of the lower transformer 34 provide load power, the current of some of the rectifier transistors 411 corresponding to the upper transformer 32 drops to zero, thus achieving zero-current switching. The next operating state is entered when the first power switching switch 211 corresponding to the upper resonant circuit 31 is turned off.

Referring to Figure 5 and Figure 2, in conjunction with the control signals provided by the signal logic circuit module 5 to the inverter circuit module 2, the resonant circuit module 3 in this embodiment, when working with the rectifier circuit module 4, will operate in multiple states. The first half-cycle operating mode will be explained below, while the second half-cycle will be generally symmetrical.

Operating state 1 [t0,t1]: One of the second power switching switches 221 of the lag arm 22 is turned on, while the other second power switching switch 221 and the first power switching switches 211 are all turned off. At this time, the first power switching switches 211 of the lead arm 21 will be in a lag period. In conjunction with the upper first inductor 311 of the upper resonant circuit 31 of the resonant circuit module 3, the parasitic capacitances of the first power switching switches 211 will release energy and store energy respectively. By making proper use of this lag period, the parasitic capacitances can complete the energy release before the first power switching switches 211 are turned on again, thus achieving zero voltage switching (ZVS).

Operating State Two [t1,t2]: Continuing from Operating State One, the first power switching switch 211 corresponding to the upper resonant circuit 31 is turned on. At this time, the corresponding parasitic capacitor has released energy, allowing the turned-on first power switching switch 211 to complete zero-voltage switching. At this time, the current of the upper first inductor 311 of the upper resonant circuit 31 and the upper magnetizing inductor 313 of the upper transformer 32 itself continues to rise, while the current of the lower resonant circuit 33 decreases. The resulting current difference allows energy to be transferred to the lower resonant circuit 33. At this time, the rectifier transistors 411 of some of the sub-circuits 41 are turned on to provide load energy, so that the current flowing through the other part of the rectifier transistors 411 drops to zero, thereby achieving zero current switching (ZCS).

Operating State 3 [t2,t3]: Continuing from Operating State 2, the second power switching switch 221 corresponding to the lower resonant circuit 33 is continuously turned on. At this time, the second power switching switch 221, together with the lower first inductor 331 and the lower capacitor 332 of the lower resonant circuit 33, continuously resonate. However, at this time, the current of the upper first inductor 311 of the upper resonant circuit 31 and the upper magnetizing inductor 313 of the upper transformer 32 itself continues to rise, while the current of the lower resonant circuit 33 continues to increase, and in conjunction with the conduction of some of the rectifier transistors 411, load energy is provided, so that the current flowing through the other part of the rectifier transistors 411 still drops to zero.

Operating State 4 [t3,t4]: Continuing from Operating State 3, the second power switching switch 221 corresponding to the lower resonant circuit 33 is turned off, causing the second power switching switches 221 of the lag arm 22 to be in idle time. At this time, the second power switching switches 221 cooperate with the lower first inductor 331 of the lower resonant circuit 33, so that the parasitic capacitance of the second power switching switches 221 performs energy release and energy storage respectively. At this time, the lower resonant circuit 33 will still have current transferred to the upper resonant circuit 31 due to the current difference between the two circuits. With an appropriate idle time, the rectifier transistors 411 can be turned on and provide load energy.

Operating State 5 [t4,t5]: Both the first power switching switch 211 and the second power switching switch 221 corresponding to the upper resonant circuit 31 are turned on, and energy continues to be supplied to the lower resonant circuit 33, making the current flowing through the upper first inductor 311 and the upper magnetizing inductor 313 of the upper transformer 32 equal. At the same time, the current flowing through the lower first inductor 331 and the lower magnetizing inductor 333 of the lower transformer 34 is made equal. At this time, the current of the rectifier transistor 411 of the upper transformer 32 and the lower transformer 34 drops to zero, thus achieving zero current switching and failing to provide load energy. The system will enter the next operating state once the first power switching switch 211 corresponding to the upper resonant circuit 31 is turned off.

Referring to Figures 4 to 6 in conjunction with Figure 1, the voltage gain characteristic curve presented in Figure 6 shows that the vertical axis represents the voltage gain of the resonant circuit module 3. First, a standardized frequency _fn is set for the switching frequency, then a first resonant frequency _fr1 and a second resonant frequency _fr2 are set. As shown in Figure 6, from the ratio between the standardized frequency _fn and the first resonant frequency _fr1 , it can be seen that when the switching frequency equals the first resonant frequency _fr1 , the voltage gain curve will operate at the resonant point, at which point the resonant circuit module 3 will reach a resonant state.

At this point, while exhibiting flexible switching characteristics, the input impedance becomes inductive, and the inductor current becomes a continuous waveform with relatively symmetrical amplitude. Therefore, the voltage gain of this embodiment is 1 under a wide range of loads, resulting in excellent conversion efficiency. However, if the switching frequency is greater than the first resonant frequency f _{_r1} and less than the second resonant frequency f_{_r2}, at best, the required flexible switching characteristics cannot be maintained; at worst, the normal operation of this embodiment will be affected.

It should be noted that if the switching frequency is between the second resonant frequency _fr2 and the first resonant frequency _fr1 , although the flexible switching characteristic can be maintained, the switching frequency will also increase with the increase of the voltage required by the load. As the amplitude of the inductor current decreases, the switching frequency will also deviate from the first resonant frequency _fr1 , which will lead to a decrease in voltage gain. Therefore, the signal logic circuit module 5 detects the feedback voltage signal of the load and generates a corresponding control signal. In addition to making the switching frequency equal to the first resonant frequency f _r1 , the control signal also includes adjusting the phase shift required between the leading arm 21 and the trailing arm 22 by the adjustment circuit of the signal logic circuit module 5 according to the voltage level _VO of the load.

As shown in Figures 7 and 8, the phase shifts are 0 degrees and 180 degrees, respectively, allowing the upper transformer 32 and the lower transformer 34 to provide energy in parallel and series connections, respectively. In the case shown in Figure 7, the voltage gain is 0.5; in the case shown in Figure 8, the voltage gain is 1. Therefore, the relationship between phase shift and voltage gain can be summarized.

Referring to Figures 9 to 11, for an actual output voltage of 250 volts, the voltage waveforms of output loads of 100 watts, 600 watts, and 1000 watts were directly measured. The first power switching switch 211 and its corresponding second power switching switch 221 are shown first. It can be observed that under the three load power conditions, the current in the upper first inductor 311 of the upper resonant circuit 31 and the current in the lower first inductor 331 of the lower resonant circuit 33 are... The current can completely discharge the parasitic capacitance of the first power switching switch 211 and the corresponding second power switching switch 221, thereby reducing conduction losses and the losses caused by the first power switching switch 211 and the corresponding second power switching switch 221. This can improve the efficiency of the converter and make the first power switching switch 211 and the corresponding second power switching switch 221 less prone to damage.

Referring to Figures 12 to 20, measurements were performed on the rectifier transistor 411 on one side of each of the sub-circuits 41, with corresponding output loads of 100 watts, 600 watts, and 1000 watts. The average voltage across the rectifier transistor 411 shows a reduction characteristic. Therefore, with reduced voltage stress on the rectifier transistor 411, its losses are relatively reduced, and zero-current switching characteristics can be achieved, thereby improving the conversion efficiency of this embodiment.

Referring to Figures 21 to 23, with an output voltage of 450 volts, the measured waveforms show output loads of 100 watts, 600 watts, and 1000 watts, demonstrating zero-voltage switching. Referring to Figures 24 to 29, measurements were performed on one side of the rectifier transistor 411 of two of these sub-circuits 41, with corresponding output loads of 100 watts, 600 watts, and 1000 watts. In all these measurements, zero-current switching is indeed achieved in the specific rectifier transistor 411.

In summary, in this embodiment of the resonant converter of the present invention, the resonant circuit module 3, in conjunction with the signal logic circuit module 5, inputs the control signals of the inverter circuit module 2, so that the on and off states of the first power switching switch 211 and the second power switching switch 221, in accordance with appropriate phase shifts and lag times, can adjust the output voltage under fixed-frequency operation, while simultaneously achieving zero-voltage switching and zero-current switching conditions. Therefore, the objective of the present invention is indeed achieved.

However, the above description is merely an example of the present invention and should not be used to limit the scope of the present invention. Any simple equivalent changes and modifications made in accordance with the scope of the patent application and the contents of the patent specification shall still fall within the scope of the present invention.

1. Vcc: Power Supply 2. Inverter Circuit Module 21. Lead Arm 211. First Power Switch 22. Lag Arm 221. Second Power Switch 3. Resonance Circuit Module 31. Upper Resonance Circuit 311. Upper First Inductor 312. Upper Capacitor 313. Upper Magnetizing Inductor 32. Upper Transformer 321. Upper Primary Circuit 322. : Upper secondary side 33: Lower resonant circuit 331: Lower first inductor 332: Lower capacitor 333: Lower magnetizing inductor 34: Lower transformer 341: Lower primary side 342: Lower secondary side 4: Rectifier circuit module 41: Sub-circuit 411: Rectifier transistor 42: Output filter capacitor 5: Signal logic circuit module 6: Isolation circuit module V _O : Voltage level; _fn : Standardized frequency; _fr1 : First resonant frequency; _fr2 : Second resonant frequency; _Q1 , _Q3 : High-side drive switches; _Q2 , _Q4 : Low-side drive switches; _C1 ~ _C8 : Capacitors; _D5 , _D6 : Diodes

Other features and effects of this invention will be clearly shown in the embodiments with reference to the figures, wherein: Figure 1 is a block diagram illustrating one embodiment of the resonant converter of this invention; Figure 2 is a circuit diagram illustrating the embodiment in conjunction with Figure 1; Figure 3 (including Figures 3a and 3b) is a schematic diagram illustrating an isolation circuit module for pulse width modulation signals provided by this embodiment; Figure 4 is a schematic diagram illustrating the operating mode of providing a 250-volt output voltage in this embodiment; Figure 5 is a schematic diagram illustrating the operating mode of providing a 450-volt output voltage in this embodiment; Figure 6 is a curve diagram illustrating the voltage... The gain characteristics are used to illustrate the operation of the resonant converter in embodiments with output voltages of 250V and 450V; Figures 7 and 8 are schematic diagrams of the circuit, illustrating the operation of one of the rectifier circuit modules in this embodiment; Figures 9 to 11 are waveform measurement diagrams, illustrating the measured waveforms of zero-voltage flexible switching at an output voltage of 250V and output loads of 100W, 600W, and 1000W respectively; Figures 12 to 14 are waveform measurement diagrams, illustrating the rectifier diode D at an output voltage of 250V and output loads of 100W, 600W, and 1000W respectively. _A. Measurement waveforms demonstrating zero-current flexible switching; Figures 15 to 17 are waveform measurement diagrams, illustrating the measurement waveforms of the rectifier diode _DC achieving zero-current flexible switching at an output voltage of 250 volts and output loads of 100 watts, 600 watts, and 1000 watts respectively; Figures 18 to 20 are waveform measurement diagrams, illustrating the measurement waveforms of the rectifier diode DC achieving zero-current flexible switching at an output voltage of 250 volts and output loads of 100 watts, 600 watts, and 1000 watts respectively. The measurement waveforms _for achieving zero-current flexible switching are shown in Figures 21 to 23, illustrating the measurement waveforms for achieving zero-current flexible switching at an output voltage of 450 volts and output loads of 100 watts, 600 watts, and 1000 watts, respectively. Figures 24 to 26 also show the measurement waveforms for achieving zero- _current flexible switching at an output voltage of 450 volts and output loads of 100 watts, 600 watts, and 1000 watts, respectively. Figures 27 to 29 further show the measurement waveforms for achieving zero-current flexible switching at an output voltage of 450 volts and output loads of 100 watts, 600 watts, and 1000 watts, respectively . _E achieves a measurement waveform with flexible switching at zero current.

2. Inverter Circuit Module 3. Resonance Circuit Module 4. Rectifier Circuit Module 5. Signal Logic Circuit Module 6. Isolation Circuit Module

Claims (6)

A resonant converter includes: an inverter circuit module adapted to be connected to a power supply for providing an input voltage, and including an upper arm and a lower arm; a resonant circuit module connected downstream of the inverter circuit module, and including an upper resonant circuit, an upper transformer connected to the upper resonant circuit, and a transformer connected to the upper resonant circuit. The system includes a lower resonant circuit of the side resonant circuit, and a lower transformer connected to the lower resonant circuit for transforming the input voltage together with the upper transformer to output an output voltage. The upper transformer has an upper primary side and an upper secondary side, while the lower transformer has a lower secondary side connected in parallel with the upper primary side. The system comprises: a secondary side and a secondary side connected in series with the upper secondary side; a rectifier circuit module connected downstream of the resonant circuit module for rectifying the output voltage to output an operating voltage, and including three sub-circuits connected only to the upper transformer, only to the lower transformer, and simultaneously connected to both the upper and lower transformers, each sub-circuit having two rectifier transistors; and a signal logic circuit module connected to the inverter circuit module for providing two control signals to the upper arm and the lower arm respectively, the control signals being used to control the operating frequency, working cycle, and phase shift of the upper arm and the lower arm respectively. The resonant converter as described in claim 1, wherein the leading arm of the inverter circuit module has two first power switching switches and the trailing arm has two second power switching switches, and the signal logic circuit module is further configured to adjust the lag time of the signal between any of the first power switching switches and any of the second power switching switches. The resonant converter as described in claim 2, wherein the upper resonant circuit of the resonant circuit module has an upper first inductor, an upper capacitor connected in series with the upper first inductor, and an upper magnetizing inductor contained in the upper transformer itself, and the lower resonant circuit has a lower first inductor, a lower capacitor connected in series with the lower first inductor, and a lower magnetizing inductor contained in the lower transformer itself. The resonant converter as described in claim 1 further includes an isolation circuit module connected between the signal logic circuit module and the inverter circuit module. The resonant converter as described in claim 1, wherein the signal logic circuit module includes an adjustment circuit for receiving a reference voltage to adjust the phase shift between the leading arm and the trailing arm. The resonant converter as described in claim 1, wherein the rectifier circuit module further includes an output filter capacitor connected in parallel with the sub-circuits.