KR0161051B1 - Digital TV Transmitter - Google Patents

Abstract

An object of the present invention is to provide a transmission and reception system that transmits more information in the same frequency band and solves the problem that the amount of transmission information cannot be increased when a frequency band is limited in a transmission device for transmitting a digital signal. In the transmitter 1, the n-value first data string, the p-value second data string, and the third data string are modulated by the modulator 4 performing m-value QAM modulation. The n-value data of the first data string is allocated and the QAM modulated signal of the modified m-value is transmitted. In the first receiver 23, the first data string having an n value by the demodulator 25, the first data string and the second data string by the second receiver 33, and the first data string and the second data by the third receiver 43. By demodulating the data string and the third data string, a transmission device for demodulating the data of the n-value first data string can be obtained even in a receiver having only n-value demodulation capability when the m-value modified multi-valued modulated wave is received.

Description

Digital TV Transmitter

1 is a block diagram showing the entire system of a transmission apparatus according to the first embodiment of the present invention.

2 is a block diagram of a transmitter (1) of Embodiment 1 of the present invention.

3 is a vector diagram of a transmission signal of Embodiment 1 of the present invention.

4 is a vector diagram of a transmission signal of Embodiment 1 of the present invention.

5 is a diagram of code allocation to signal points of Embodiment 1 of the present invention.

6 is a coded diagram for signal grouping according to the first embodiment of the present invention.

Fig. 7 is a diagram of coding to signal points of the signal point group in Example 1 of the present invention.

8 is a coded diagram of a signal point group and a signal point of Embodiment 1 of the present invention.

9 is a threshold state diagram of a group of signal points of a transmission signal according to the first embodiment of the present invention.

10 is a vector diagram of a modified 16-valued QAM according to the first embodiment of the present invention.

11 is a relationship diagram between the antenna radius r2 and the transmission power ratio n of the first embodiment of the present invention.

Fig. 12 is a diagram of signal points of modified 64-valued QAM in Embodiment 1 of the present invention.

13 is a relation diagram between the antenna radius r3 and the transmission power ratio n of the first embodiment of the present invention.

14 is a vector diagram of a signal group and a sub-signal point group of a modified 64-valued QAM according to the first embodiment of the present invention.

FIG. 15 is an explanatory diagram of the ratio A ₁ , A ₂ of the modified 64 value QAM according to the first embodiment of the present invention. FIG.

FIG. 16 is a relation diagram of antenna radii r ₂ and r ₃ and transmission power ratios n ₁₆ and n ₆₄ in Embodiment 1 of the present invention. FIG.

17 is a block diagram of a digital transmitter of Embodiment 1 of the present invention.

18 is a signal space diagram of 4PSK modulation of Embodiment 1 of the present invention.

19 is a block diagram of a first receiver in a first embodiment of the present invention.

20 is a signal space diagram of 4PSK modulation of Embodiment 1 of the present invention.

21 is a block diagram of a second receiver of Embodiment 1 of the present invention.

22 is a signal vector diagram of a modified 16-valued QAM according to the first embodiment of the present invention.

23 is a signal vector diagram of a modified 64-valued QAM according to the first embodiment of the present invention.

24 is a flowchart of Embodiment 1 of the present invention.

25 (a) is a signal vector diagram of an eight-valued QAM according to the first embodiment of the present invention.

(b) is a signal vector diagram of a 16-valued QAM according to the first embodiment of the present invention.

Fig. 26 is a block diagram of a third receiver of Embodiment 1 of the present invention.

27 is a view of a signal point of a modified 64 value QAM according to the first embodiment of the present invention.

28 is a flowchart of Embodiment 1 of the present invention.

29 is an overall configuration diagram of a transmission system according to a third embodiment of the present invention.

30 is a block diagram of a first image encoder of Embodiment 3 of the present invention.

Fig. 31 is a block diagram of a first image decoder of Embodiment 3 of the present invention.

32 is a block diagram of a second image decoder of Embodiment 3 of the present invention.

33 is a block diagram of a third image decoder of Embodiment 3 of the present invention.

34 is an explanatory diagram of time multiplexing of the D ₁ , D ₂ , and D ₃ signals according to the third embodiment of the present invention.

35 is an explanatory diagram of time multiplexing of D ₁ , D ₂ , and D ₃ signals according to Embodiment 3 of the present invention.

36 is an explanatory diagram of time multiplexing of D ₁ , D ₂ , and D ₃ signals according to Embodiment 3 of the present invention.

Fig. 37 is a configuration diagram of the whole system of the transmission device in accordance with the fourth embodiment of the present invention.

Fig. 38 is a vector diagram of signal points of modified 16 QAM in Embodiment 3 of the present invention.

Fig. 39 is a vector diagram of signal points of modified 16 QAM in Example 3 of the present invention.

Fig. 40 is a vector diagram of signal points of modified 64 QAM in Example 3 of the present invention.

Fig. 41 is a signal constellation diagram on the time side of Embodiment 3 of the present invention.

42 is a signal constellation diagram on a time axis of the TDMA system according to the third embodiment of the present invention.

Fig. 43 is a block diagram of a carrier reproducing circuit of Embodiment 3 of the present invention.

44 is a principle diagram of carrier reproduction according to the third embodiment of the present invention.

45 is a block diagram of a carrier modulation circuit of an inverse modulation method of Embodiment 3 of the present invention.

Fig. 46 is a signal point arrangement diagram of the 16 QAM signal of Embodiment 3 of the present invention.

Fig. 47 is a signal point arrangement diagram of a 64 QAM signal according to Embodiment 3 of the present invention.

FIG. 48 is a block diagram of a carrier reproduction circuit of the 16 multiplication method of Embodiment 3 of the present invention. FIG.

FIG. 49 is an explanatory diagram of time multiplexing of D _V1 , D _H1 , D _V2 , D _H2 , D _V3 , and D _H3 signals in Example 3 of the present invention. FIG.

FIG. 50 is an explanatory diagram of TDMA time multiplexing of D _V1 , D _H1 , D _V2 , D _H2 , D _V3 , and D _H3 signals according to Embodiment 3 of the present invention. FIG.

FIG. 51 is an explanatory diagram of TDMA time multiplexing of D _V1 , D _H1 , D _V2 , D _H2 , D _V3 , and D _H3 signals in Embodiment 3 of the present invention; FIG.

Fig. 52 is a diagram of a reception disturbance area according to a conventional embodiment in the fourth embodiment of the present invention.

Fig. 53 is a reception disturbance area diagram in the case of the hierarchical broadcasting system according to the fourth embodiment of the present invention.

Fig. 54 is a block diagram of a reception disturbance in a conventional manner in Embodiment 4 of the present invention.

Fig. 55 is a reception disturbance area diagram in the case of the hierarchical broadcasting system according to the fourth embodiment of the present invention.

56 is a reception disturbance area diagram of two digital broadcasting stations in accordance with the fourth exemplary embodiment of the present invention.

Fig. 57 is a signal point arrangement diagram of a modified 4ASK signal in accordance with the fifth embodiment of the present invention.

Fig. 58 is a signal point arrangement diagram of a modified 4ASK in Embodiment 5 of the present invention.

Fig. 59 (a) is a signal point arrangement diagram of a modified 4ASK in Embodiment 5 of the present invention.

(b) is a signal point arrangement diagram of modified 4ASK in Embodiment 5 of the present invention.

60 is a signal point arrangement diagram of a modified 4ASK signal in case of a low C / N value in Embodiment 5 of the present invention.

61 is a diagram of transmitters of 4VSB and 8VSB in Example 5. FIG.

62A is a spectroscopic diagram of an ASK signal, that is, a multi-value VSB signal before filtering, according to a fifth embodiment of the present invention.

(b) is a frequency distribution diagram of an ASB signal in Example 5 of the present invention.

63 is a block diagram of receivers of 4VSB, 8VSB, and 16VSB in Embodiment 5 of the present invention.

64 is a block diagram of a video signal transmitter in Embodiment 5 of the present invention.

65 is a block diagram of the entire TV receiver in the fifth embodiment of the present invention.

66 is a block diagram of another TV receiver in Embodiment 5 of the present invention.

67 is a block diagram of a satellite and terrestrial TV receiver according to a fifth embodiment of the present invention.

Fig. 68 (a) is a signal point arrangement diagram of 8 VSBs in Examples 5 and 6.

(b) is a signal point arrangement diagram of 8 VSBs in Examples 5 and 6. FIG.

(c) is a signal-time waveform diagram of 8VSB in Examples 5 and 6. FIG.

69 is another block diagram of an image encoder according to a fifth embodiment of the present invention.

70 is a block diagram of an image encoder of one separation circuit according to a fifth embodiment of the present invention.

71 is a block diagram of an image decoder according to a fifth embodiment of the present invention.

Fig. 72 is a block diagram of an image decoder of one synthesizer in the fifth embodiment of the present invention.

73 is a timing diagram of the transmission signal of the fifth embodiment according to the present invention;

74A is a block diagram of an image decoder according to a fifth embodiment of the present invention.

(b) is a time arrangement diagram of a transmission signal of a fifth embodiment according to the present invention;

75 is a time diagram of transmission signals of a fifth embodiment according to the present invention;

76 is a time arrangement diagram of a transmission signal of a fifth embodiment according to the present invention;

77 is a timing diagram of the transmission signal of the fifth embodiment according to the present invention;

78 is a block diagram of an image decoder according to a fifth embodiment of the present invention.

79 is a time diagram of transmission signals of the third layer according to the fifth embodiment of the present invention.

80 is a block diagram of an image decoder of Embodiment 5 according to the present invention;

81 is a timing diagram of the transmission signal of the fifth embodiment according to the present invention;

Fig. 82 is a block diagram of an image decoder of D ₁ according to the fifth embodiment of the present invention.

83 is a frequency-time diagram of a frequency modulated signal of Example 5 according to the present invention.

84 is a block diagram of a magnetic recording and reproducing apparatus of a fifth embodiment according to the present invention;

Fig. 85 is a diagram showing the relationship between C / N and layer number of Example 2 according to the present invention.

86 is a relation diagram of transmission distance and C / N of the second embodiment of the present invention.

87 is a block diagram of a transmitter of Embodiment 2 according to the present invention;

88 is a block diagram of a receiver of Embodiment 2 according to the present invention;

Fig. 89 is a relationship diagram of a C / N-error rate of Example 2 according to the present invention.

90 is a disturbance diagram of the third layer of Example 5 according to the present invention;

91 is a block diagram of the reception disturbance area in the fourth layer according to the seventh embodiment of the present invention.

92 is a layer transmission diagram of Embodiment 7 according to the present invention.

93 is a block diagram of a separation circuit of a seventh embodiment according to the present invention;

FIG. 94 is a block diagram of a synthesis part of a seventh embodiment in accordance with the present invention; FIG.

95 is a diagram illustrating a transmission hierarchy structure of Embodiment 7 according to the present invention.

Fig. 96 is a reception diagram of a digital TV broadcast of the conventional method.

97 is a reception state diagram of digital TV hierarchical broadcasting of the seventh embodiment according to the present invention;

FIG. 98 is a transmission hierarchy structure diagram of Embodiment 7 according to the present invention; FIG.

99 is a vector diagram of 16SRQAM of Example 3 according to the present invention;

100 is a vector diagram of 32SRQAM of Example 3 according to the present invention;

101 is a relationship diagram of a C / N-error rate of Example 3 according to the present invention.

102 is a relationship diagram of a C / N-error rate of Example 3 according to the present invention.

103 is a relationship diagram of shift amount n and C / N required for transmission according to the third embodiment of the present invention.

104 is a relationship diagram of shift amount n and C / N required for transmission according to the third embodiment of the present invention.

105 is a relationship diagram between a distance from a transmission antenna and a signal level in terrestrial broadcasting according to the third embodiment of the present invention.

106 is a service area diagram of 32SRQAM of Embodiment 3 according to the present invention;

Fig.107 is a service area diagram of 32SRQAM of Embodiment 3 according to the present invention.

108 is a frequency distribution diagram of a conventional TV signal.

(b) is a frequency distribution diagram of a conventional two-layer TV signal.

(c) is a figure which shows the threshold of Example 3 of this invention.

(d) is a frequency distribution diagram of a carrier group of OFDM in the second layer of Example 9. FIG.

(e) is a diagram showing three threshold values of OFDM in the third layer of Example 9;

FIG. 109 is a TV signal timing chart according to the third embodiment of the present invention. FIG.

110 is a principle diagram of C-CDM of Embodiment 3 according to the present invention;

111 is a code allocation diagram of Embodiment 3 according to the present invention.

112 is a code allocation diagram when the 36QAM of the third embodiment of the present invention is extended.

113 is a modulation signal frequency diagram of Embodiment 5 of the present invention;

Fig. 114 is a block diagram of a magnetic recording and reproducing apparatus according to the fifth embodiment of the present invention.

115 is a block diagram of a transceiver of a cellular phone according to an eighth embodiment of the present invention.

116 is a block diagram of a base station of Embodiment 8 according to the present invention;

FIG. 117 is a distribution diagram of a communication capacity and traffic in a conventional manner. FIG.

FIG. 118 is a distribution diagram of communication capacity and traffic of embodiment 8 according to the present invention;

FIG. 119 (a) is a time slot arrangement diagram of a conventional method.

(b) is a time slot diagram of Example 8 according to the present invention;

FIG. 120A shows a conventional TDMA time slot arrangement.

(b) is a TDMA time slot arrangement diagram of Embodiment 8 according to the present invention;

121 is a block diagram of a transceiver of Layer 1 of Embodiment 8 according to the present invention;

122 is a block diagram of a transceiver of a layer 2 of Embodiment 8 according to the present invention;

123 is a block diagram of an OFDM transceiver in accordance with a ninth embodiment of the present invention.

124 is an operation principle diagram of an OFDM scheme of Embodiment 9 according to the present invention.

125 is a frequency diagram of a modulation signal of the conventional method.

(b) is a frequency arrangement diagram of a modulated signal of Example 9 according to the present invention;

FIG. 126A is a diagram showing an unweighted state of OFDM in Example 9. FIG.

(b) is a diagram showing two subchannels of two-layer OFDM weighted by the transmission power in Example 9;

(c) is the frequency distribution diagram of OFDM which doubled the carrier spacing in Example 9;

(d) is a frequency distribution diagram of OFDM of non-weighted carrier interval in Example 9. FIG.

127 is a block diagram of a transceiver of Embodiment 9 according to the present invention;

Fig. 128 (a) is a block diagram of a trellis encoder (ratio 1/2) in Examples 2, 4, and 5.

(b) is a block diagram of a trellis encoder (ratio 2/3) in Examples 2, 4, and 5;

(c) is a block diagram of a trellis encoder (ratio 3/4) in Examples 2, 4 and 5. FIG.

(d) is a block diagram of a trellis encoder (ratio 1/2) in Examples 2, 4 and 5. FIG.

(e) is a block diagram of a trellis encoder (ratio 2/3) in Examples 2, 4, and 5;

(f) is a block diagram of a trellis encoder (ratio 3/4) in Examples 2, 4 and 5. FIG.

FIG. 129 is a time arrangement diagram of an invalid trial period and a guard period in Example 9. FIG.

FIG. 130 is a relationship between C / N versus error rate of the conventional example and Example 9. FIG.

131 is a block diagram of the magnetic recording and reproducing apparatus of the fifth embodiment;

FIG. 132 is a view showing the recording format of the track on the magnetic tape and the head of the fifth embodiment; FIG.

133 is a block diagram of a transceiver of Embodiment 3. FIG.

FIG. 134 is a frequency diagram showing a broadcasting system of a conventional example. FIG.

FIG. 135 is a relationship between service area and picture quality in the case of using the hierarchical transmission method of three layers in the third embodiment.

FIG. 136 is a frequency diagram showing a combination of the hierarchical transmission scheme and the FDM according to the third embodiment; FIG.

FIG. 137 is a block diagram of a transceiver in the case where the trellis encoding according to the third embodiment is used. FIG.

FIG. 138 is a block diagram of a transceiver in case of transmitting part of low frequency signals in OFDM in the ninth embodiment; FIG.

139 is a signal point arrangement diagram of 8-PS-APSK in Example 1. FIG.

140 is a signal point arrangement diagram of 16-PS-APSK in Example 1. FIG.

Fig.141 is a signal point arrangement diagram of 8-PS-PSK in the first embodiment.

142 is a signal point arrangement diagram of 16-PS-PSK (PS type) in Example 1. FIG.

Fig. 143 is a relation between the radius of the satellite antenna and the transmission capacity in the first embodiment.

144 is a block diagram of a weighted OFDM transceiver according to a ninth embodiment;

Fig.145 (a) is a waveform diagram of guard time and symbol time hierarchy type OFDM when the multipath is short in the ninth embodiment.

(b) is a waveform diagram of a guard time and a symbol time hierarchy type OFDM when the multipath is long in Example 9. FIG.

FIG. 146 is a principle diagram of guard time and symbol time hierarchy type OFDM in Example 9. FIG.

FIG. 147 is a subchannel arrangement diagram of a two-layer transmission scheme by power weighting in the ninth embodiment;

FIG. 148 is a relationship between D / U conversion, multipath delay time and guard time in Example 9. FIG.

149 (a) is a timeslot diagram of each layer in the ninth embodiment.

(b) is a time distribution diagram of guard time of each layer in Example 9. FIG.

(c) is a time distribution diagram of guard time of each layer in Example 9. FIG.

150 is an explanatory diagram of a three-layer hierarchical transfer method for multipath in the relationship diagram between the multipath delay time and the transmission rate of the ninth embodiment;

FIG. 151 is an explanatory diagram of a hierarchical broadcasting method having a two-dimensional matrix structure in a relationship diagram between a delay time and a CN value when the GTW-OFDM and C-CDM (or CSW-OFDM) of the ninth embodiment are combined;

FIG. 152 is a time alignment diagram of TV signals of three layers in each time slot when the GTW-OFDM and C-CDM (or CSW-OFDM) of the ninth embodiment are combined;

FIG. 153 shows a hierarchical broadcasting of a three-dimensional matrix structure in a relationship diagram between a multipath signal delay time, a CN value, and a transmission rate when the GTW-OFDM and C-CDM (or CSW-OFDM) of the ninth embodiment are combined; Diagram of the scheme.

154 is a frequency distribution diagram of Power-Weighted-OFDM of Example 9;

FIG. 155 is a layout view of the three-layer TV signal on a time axis in each time slot when the Guard-Time-OFDM and C-CDM are combined.

156 is a block diagram of a transmitter and a receiver in Embodiments 4 and 5. FIG.

157 is a block diagram of a transmitter and a receiver in Embodiments 4 and 5. FIG.

158 is a block diagram of a transmitter and a receiver in Embodiments 4 and 5. FIG.

Fig.159 (a) is a signal point arrangement diagram of 16 VSB in the fifth embodiment.

(b) is a signal point arrangement diagram (8VSB) of 16VSB in Example 5. FIG.

(c) is 16VSB signal point arrangement diagram (4VSB) in Example 5. FIG.

(d) is a signal point arrangement diagram (16VSB) of 16VSB in Example 5. FIG.

160 (a) is a block diagram of an ECC encoder in Examples 5 and 6. FIG.

(b) is a block diagram of the ECC encoder in Examples 5 and 6. FIG.

161 is an overall block diagram of a VSB receiver in a fifth embodiment;

162 is a view showing a transmitter in a fifth embodiment;

Figure 163 shows the error rate / CN value curves of 4VSB and TC-8VSB of the example.

FIG. 164 shows the error rate curves of subchannel 1 and subchannel 2 of Example 4 VSB and TC-8VSB.

FIG. 165 (a) is a block diagram of a Reed Solomon encoder in Examples 2, 4, and 5. FIG.

(b) is a block diagram of the Reed Solomon encoder in Examples 2, 5 and 6. FIG.

FIG. 166 is a flowchart of Reed Solomon error correction and calculation in Examples 2, 4, and 5. FIG.

FIG. 167 is a block diagram of a deinterleave portion in Embodiments 2, 3, 4, 5, and 6. FIG.

FIG. 168 (a) is an illustration of the interleaved and deinterleaved tables in Example 2, 3, 4, and 5. FIG.

(b) is a figure which shows the interleave distance in Example 2, 3, 4, 5. FIG.

FIG. 169 is a comparison of redundancies of 4-VSB, 8-VSB, and 16-VSB in Example 5. FIG.

170 is a block diagram of a TV receiver for receiving high rank signals in Embodiments 2, 3, 4, and 5. FIG.

FIG. 171 is a block diagram of a transmitter and a receiver in Embodiments 2, 3, 4, and 5. FIG.

172 is a block diagram of a transmitter and a receiver in Embodiments 2, 3, 4, and 5.

173 is a block diagram of an ASK magnetic recording and reproducing apparatus according to the sixth embodiment.

* Explanation of symbols for main parts of the drawings

1: transmitter 4: modulator

6 antenna 6a: ground antenna

10: satellite 12: repeater

23: first receiver 25: demodulator

33: second receiver 35: demodulator

43: third receiver 51: digital transmitter

85: signal point 91: first split signal point group

401: First image decoder 703: Receivable area of SRQAM

The present invention relates to a digital TV transmission apparatus for transmitting a digital signal by modulating a carrier wave.

Recently, the digital transmission device has been used in various fields. In particular, the progress of digital video transmission technology is amazing.

Among them, the transmission method of the digital TV has recently attracted attention. Currently, digital TV transmitters are only partially used for relaying between broadcasting stations. However, in the near future, it is planned to be deployed to terrestrial and satellite broadcasting, and is being reviewed in each country.

In order to meet the demands of advanced consumers, it is necessary to improve the quality and quantity of contents of broadcasting services such as HDTV broadcasting, PCM music broadcasting, information providing broadcasting and fax broadcasting in the future. In this case, it is necessary to increase the amount of information in the limited frequency band of TV broadcasting. The amount of information that can be transmitted in this band increases according to the technical limitations of that era. For this reason, it is desirable to be able to extend the amount of information transmission by changing the receiving system according to the times.

However, when viewed from the point of broadcasting, publicity is important and securing the vested interests of all viewers for a long time becomes important. When starting a new broadcast service, it is a requirement to be able to enjoy the service by an existing receiver or receiver. The compatibility of the receiver or receiver and the compatibility of broadcast between the old and new, present and future new and old broadcasting service are the most important.

In the future, new transmission standards, such as digital TV broadcasting standards, require the scalability of the information volume to meet the demands of the future society and technological progress, and the compatibility and compatibility of existing receivers.

Here, the transmission method of the TV broadcast which has been proposed so far will be described in terms of scalability and compatibility.

First, as a satellite broadcasting method of digital TV, a signal compressed by NTSC-TV signal to about 6Mbps is multiplexed by TDM method using 4-value PSK modulation, and one transponder uses 4-20 channels of NTSC TV program or one channel. A method of broadcasting HDTV has been proposed. In addition, as a terrestrial broadcasting method of HDTV, a method of compressing one channel HDTV video signal into data of about 15Mbps and performing terrestrial broadcasting using 16 or 32QAM modulation method has been considered.

First of all, in the satellite broadcasting method, the currently proposed broadcasting method simply uses a conventional transmission method, and thus uses NTSC frequency bands for several channels for program broadcasting other than HDTV. For this reason, there was a problem in that it is not possible to receive and broadcast NTSC programs of multiple channels in the broadcast time zone of HDTV programs. It can be said that there was no compatibility and compatibility between the receiver and the receiver between NTSC and HDTV broadcasting. Further, it can be said that the scalability of the amount of information transmission required according to the conventional technical progress is not considered at all.

Next, the terrestrial broadcasting method of the conventional HDTV, which is currently under consideration, is merely broadcasting the HDTV signal in the conventional modulation method such as 16QAM or 32QAM. In the case of the existing analogue broadcasting service, there is always a bad reception condition that is obstructed by a building, a lowland area, or an adjacent TV station even in the broadcasting service area. In these areas, the quality of the existing analog broadcasting deteriorates, but the video can be reproduced and the TV program can be viewed. However, in the conventional digital TV broadcasting system, there is a serious problem that no video can be reproduced at all in such an area, and no TV program can be viewed at all. This included the inherent challenges of digital TV broadcasting and could be fatal to the spread of digital TV broadcasting. This is due to the fact that the positions of signal points of a modulation method such as a conventional QAM are arranged at equal intervals. There is no conventional method of changing or modulating the arrangement of signal points.

Disclosure of Invention The present invention solves the above-mentioned problems, and in particular, it is an object of the present invention to provide a transmission apparatus that significantly reduces the compatibility between NTSC broadcasting and HDTV broadcasting in satellite broadcasting, and the unreceivable area in the service area in terrestrial broadcasting. It is done.

In order to achieve the above object, the transmission device of the present invention comprises an input means for compressing an image signal, image compression means for compressing the image signal into a digital image compression signal, and applying a miscorrection correction code to the digital image compression signal, A transmission signal is sent by a transmitter having a error correction encoder for generating an encoded signal, modulation means for modulating the error correction coded signal into an n-value VSB modulated signal, and a transmission means for transmitting the modulation signal, Means for receiving a signal, demodulation means for demodulating the transmission signal into a received digital signal, error correction means for miscorrecting the received digital signal into a miscorrected digital signal, and converting the error corrected digital signal into an image output signal. 12. An apparatus for transmitting an image signal by a receiver having an image extension section that extends and an output means for outputting the image output signal. Is a VSB signal is characterized in that transmission or reception.

This configuration inputs a first data string and a second data string having n-value data as input signals, and generates a modulated m-value QAM system modulated wave having a m-value signal point on a vector diagram by a modulator of the transmitting apparatus. This signal point group is divided into n sets of signal point groups, and the signal point group is assigned to each of n pieces of data in the first data string, and the m / n signal points or sub-signal point groups in the signal point group are assigned to the second data string. Each data is allocated, trellis-coded, modulated, and transmitted by the transmitter. In some cases, the third data can also be transmitted.

Next, the receiving device having the p value demodulator of pm receives the transmission signal, first divides the signal point of p point into n sets of signal point groups for the signal point of point p on the signal space diagram. Demodulate and reproduce the signal. Next, the second data string of p / n value is demodulated in correspondence with the signal points of p / n points in the corresponding signal point group, and the first data and the second data are demodulated and reproduced. At this time, the first data string and / or the second data string are trellis coded. In the p = n receiver, the n group of signal point groups are reproduced, and only the first data string is demodulated and reproduced corresponding to n values.

When the same signal is received from the transmitting apparatus by the above operation, the receiver having the demodulation capability of the large antenna and the multivalue can demodulate the first data string and the second data string. At the same time, the receiver having the demodulation capability of the small antenna and the Sochi can receive the first data string. In this way, a compatible transmission system can be constructed. In this case, the first data string is assigned to a low-frequency TV signal such as NTSC or a low-frequency component of an HDTV, and the second data string is assigned to a high-frequency TV signal such as a high-frequency component of an HDTV. Receivers with multiple-value demodulation capabilities can receive HDTV signals. This makes digital broadcasting compatible with NTSC and HDTV possible.

EMBODIMENT OF THE INVENTION Hereinafter, the Example of this invention is described, referring drawings.

According to an embodiment of the present invention, a recording device comprising a transmitter comprising a transmitter and a receiver for transmitting a digital signal such as a digital HDTV signal, and a digital signal such as an HDTV signal on a recording medium such as magnetic tape, and recording and reproducing. Both of the apparatus will be described.

However, the operation principle of the digital modulation and demodulation unit and the error correction encoder, the encoder of the image encoding such as the decoder and the HDTV signal, and the decoder are common to the transmission apparatus and the recording / reproducing apparatus. It is basically the same technique. Therefore, in each embodiment, the present invention will be described using a block diagram of either the transmission apparatus or the recording / reproducing apparatus in order to explain efficiently. In the configuration of each embodiment of the present invention, any method may be used as long as it is a multi-value digital modulation method in which signal points are placed on the constellation such as QAM, ASK, and PSK. However, one modulation method will be described. .

Example 1

1 shows an overall system diagram of a transmission apparatus of Embodiment 1 according to the present invention. A transmitter 1 having an input unit 2, a separate circuit unit 3, a modulator 4, and a transmitter unit 5 transmits a plurality of multiplexed input signals by means of the separation circuit 3 to the first data string, D ₁ and D. It is divided into two data strings D ₂ and a third data string D ₃ , and is output by the modulator 4 as a modulated signal from the transmission section 5, by the antenna 6, and by the transmission path 7. Sent to the satellite (10). This signal is received by the antenna 11 in the satellite 10, amplified by the repeater 12, and transmitted back to the earth by the antenna 13.

Transmission radio waves are sent to the first receiver 23, the second receiver 33, and the third receiver 43 by the transmission paths 21, 31 and 41. First, the first receiver 23 inputs from the input unit 24 via the antenna 22, and only the first data string is demodulated by the demodulator 25 and output from the output unit 26. In this case, there is no demodulation capability of the second data string and the third data string.

In the second receiver 33, the first data string and the second data string are demodulated by the demodulator 35 to output the signal output from the input unit 34 via the antenna 32, and one synthesizer 37 The data is synthesized into a data string and output from the output unit 36.

In the third receiver 43, the input from the antenna 42 enters the input unit 44, and the demodulator 45 demodulates three data strings of the first data string, the second data string, and the third data string to form a synthesizer ( By 47), it becomes one data group and is output from the output unit 46.

Even when receiving the same frequency band from the same transmitter 1 as described above, the amount of information that can be received differs depending on the performance difference of the demodulators of the three receivers described above. This feature makes it possible to simultaneously transmit three pieces of information compatible with the performance of a receiver whose performance is different in one radio band. For example, in the case of transmitting three digital TV signals of the same program, NTSC, HDTV, and super-resolution HDTV, the super HDTV signal is separated into a low frequency component, a high frequency component, and a super high frequency component, and each of the first data is first data. Corresponding to the column, the second data sequence, and the third data sequence group, it is possible to simultaneously broadcast triple digital TV signals having compatible medium resolution, high resolution, and ultra high resolution in one channel frequency band.

In this case, it is possible to receive an NTSC-TV signal in a small-chamber demodulated receiver using a small antenna, an HDTV signal in a medium-demodulated receiver using a medium antenna, and an ultra-high definition HDTV in a multi-valued demodulator using a large antenna. Referring to FIG. 1 again, the digital transmitter 51 for NTSC digital TV broadcasting inputs only the same data as the first data group from the input unit 52, modulates it with the modulator 54, and transmits the transmitter 55. And by antenna 56 to satellite 10 by transmission path 57 and back to earth by transmission path 58.

In the first receiver 23, the demodulator 25 demodulates the data corresponding to the first data string by the demodulator 25 for the received signal from the digital transmitter 1. Similarly, the second receiver 33 and the third receiver 43 demodulate data groups having the same contents as the first data string. That is, the three receivers can also receive digital broadcasting such as digital general TV broadcasting.

Then, each part is demonstrated.

2 is a block diagram of the transmitter 1.

The input signal enters the input unit 2 and is separated into three digital signals of the first data string signal, the second data string signal, and the third data string signal in the separation circuit 3.

For example, when a video signal is input, the low frequency component of the video signal is allocated to the first data string signal, the high frequency component of the video signal to the second data string signal, and the ultrahigh frequency component of the video signal to the third data string signal. do. The three separate signals are input to the modulation input unit 61 inside the modulator 4. Here, there is a signal point position modulation / change circuit 67 which modulates or changes the position of the signal point based on an external signal, and modulates or changes the position of the signal point in accordance with the external signal. In the modulator 4, amplitude modulation is performed on each of two orthogonal carriers to obtain a multivalued QAM signal. The signal from the modulation input section 61 is sent to the first AM modulator 62 and the second AM modulator 63. One of the carriers from the carrier generator 64, which is COS (2πfct), is AM modulated by the first AM modulator 62, is sent to the synthesizer 65, and the other carrier is sent to the π / 2 phaser 66. After 90 ° or more, sent to the second AM modulator 63 in the state of sin (2πfct) and receives multi-value amplitude modulation, it is synthesized by the synthesizer 65 with the second AM modulated wave and transmitted by the transmitter 5. It is output as a transmission signal. Since this system itself is generally practiced in the related art, detailed description of the operation is omitted.

Figure 3 illustrates the operation using the first upper limit of the signal space diagram of a 16-value general QAM. All signals generated by the modulator 4 can be expressed as a composite vector of two vectors of the vector 81 of two orthogonal carriers A cos 2πfct and the vector 82 of B sin 2πfct. When the leading end of the synthesis vector from the zero point is defined as signal points, in the case of 16-value _{QAM a 1, a 2, a} 3, 4 of the amplitude value value of a ₄ 4 and _{b 1, b 2, b 3} , b 4 A total of 16 signal points can be set by combining the amplitude values of the values. In the first upper limit of FIG. 3, four signals of C ₁₁ signal point 84 of signal point 83, C ₁₂ , C ₂₂ of signal point 85, and C ₂₁ of signal point 86 exist.

C ₁₁ is a composite vector of vectors 0-a ₁ and vectors 0-b ₁ , and C ₁₁ = a ₁ cos 2πfct-b ₁ sin 2πfct = A cos (2πfct + dπ / 2).

Here, the distance between 0-a ₁ on the Cartesian coordinates of FIG. 3 is defined as A ₁ , a ₁ -a ₂ between A ₂ , 0-b ₁ between B ₁ , and b ₁ -b ₂ between B ₂ and B ₂ . And display on the drawing.

As shown in the entire vector diagram of FIG. 4, there are 16 signal points in total. Therefore, by making each point correspond to four bits of information, four bits of information transfer can be performed in one cycle, that is, one time slot.

Fig. 5 shows an example of general allocation when each point is expressed in binary.

Naturally, the farther the distance between each signal point is, the easier it is to distinguish on the receiver side. Therefore, in general, the distance between the signal points should be arranged as far as possible. If the distance between specific signal points is close, the receiver becomes difficult to distinguish between the two points, and the error rate is worse. Therefore, it is generally said that it is preferable to arrange at equal intervals as shown in FIG. Therefore, when the 16QAM is A ₁ = A _2/2 arrangement of the signal points is performed in general.

However, in the transmitter 1 of the present invention, data is first divided into a first data string, a second data string, and optionally a third data string. As shown in FIG. 6, 16 signal points or a group of signal points are divided into four signal point groups, and four data of the first data string are first assigned to each signal point group. That is, when the first data string is 11, any one of four signal points of the first signal point group 91 of the first data upper limit is transmitted, and in the case of 01, the second signal point group 92 of the second upper limit is transmitted. , One signal point among four signal points among the third signal point group 93 of the third upper limit and the fourth signal point group 94 of the fourth upper limit in the case of 10, according to the value of the second data string. Select and send. Next, two bits of the second data string, four values of data in the case of 16QAM, four bits of data in the case of 64 values of QAM, and data of 16 values are included in each of the divided signal point groups of (91), (92), (93), and (94). 4 signal points or sub-signal point groups are allocated as shown in FIG. Any upper limit is subject to placement. The assignment of signal points to (91) (92) (93) (94) is preferentially determined by 2-bit data of the first data group. In this way, two bits of the first data string and two bits of the second data string can be transmitted completely independently. The first data string may also be demodulated by the 4PSK receiver if the antenna sensitivity of the receiver is greater than or equal to a predetermined value. If the antenna has higher sensitivity, both the first data group and the second data group can be demodulated by the modified 16QAM receiver of the present invention.

8 shows an example of allocation of 2 bits of the first data string and 2 bits of the second data string.

In this case, by dividing the HDTV signal into a low frequency component and a high frequency component, and assigning a low frequency video signal to the first data string and a high frequency video signal to the second data string, a 4PSK receiving system generates an NTSC equivalent image of the first data string. In the 18QAM or 64QAM receiving system, both the first data stream and the second data stream can be reproduced, and these can be added to obtain an HDTV image.

However, when the distance between signal points is the same distance as shown in Fig. 9, there is a critical distance between the portion indicated by the diagonal line in the first upper limit as viewed from the 4PSK receiver. If the critical distance is A _TO , it may be the amplitude of A _TO as long as 4PSK is sent. However, to send 16QAM while maintaining A _TO requires 3A _{TO, which} is three times the amplitude. That is, compared with the case of transmitting 4PSK, nine times as much energy is required. Sending a 4PSK signal point in 16QAM mode without any consideration is inefficient in power usage. In addition, reproduction of the carrier becomes difficult. In the case of satellite transmission, the available power is limited. Such poor power utilization systems are not realistic until the satellite transmit power is increased. It is expected that a large number of 4PSK receivers will be released in the future when digital TV broadcasting starts. Once popularized, it is impossible to increase their reception sensitivity because problems of receiver compatibility arise. Therefore, the transmission power of the 4PSK mode does not decrease. For this reason, when a similar 4PSK signal point is sent in the 16QAM mode, it is expected that a method of lowering the transmission power than the conventional 16QAM is required. Otherwise, you will not be able to transmit with limited satellite power.

The characteristic of the present invention is that the transmission power of the similar 4PSK type 16QAM modulation can be lowered by lowering the distance of the four divided signal point groups shown by reference numerals 91 to 94 as shown in FIG.

Here, in order to clarify the relationship between the reception sensitivity and the transmission output, the reception method of the digital transmitter 51 and the first receiver 23 will be described.

First, the digital transmitter 51 and the first receiver 23 are general transmission apparatuses and perform video transmission including data transmission or broadcasting. As shown in FIG. 17, the digital transmitter 51 is a 4PSK transmitter and removes an AM modulation function from the transmitter 1 of the multi-valued QAM described in FIG. The input signal is input to the modulator 54 via the input unit 52. The modulator 54 modulates the input signal into two signals by the modulating input unit 121 to modulate the first-phase phase modulation circuit 122 that phase modulates the reference carrier and a carrier that is 90 degrees out of phase with the reference carrier. The two-phase phase modulation circuit 123 is sent, and these phase modulation waves are synthesized by the combiner 65 and transmitted by the transmitter 55.

The modulation signal space diagram at this time is shown in FIG.

In general, it is common sense that four signal points are set and the distance between signal points is equally spaced in order to increase the power utilization efficiency. As an example, the case where signal point 125 is defined as (11), signal point 126 as (01), signal point 127 as (00) and signal point 128 as (10) is shown. . In this case, in order for the 4PSK first receiver 23 to receive satisfactory data, a predetermined amplitude value is required at the output of the digital transmitter 51. Referring to FIG. 18, when the first receiver 23 defines the minimum amplitude of the transmission signal required to receive the signal of the digital transmitter 51 at 4 PSK, that is, the distance between 0-a ₁ as A _TO , the lowest transmission limit is shown. When transmitting at an amplitude A _TO or more, the first receiver 23 becomes receivable.

Next, the first receiver 23 will be described. The first receiver 23 receives the transmission signal from the transmitter 1 or the 4PSK transmission signal from the digital transmitter 51 through the repeater 12 of the satellite 10 and receives the small signal from the small antenna 22. The demodulator 24 regards the received signal as a 4PSK signal and demodulates it. The first receiver 23 is originally designed to receive signals of 4PSK or 2PSK of the digital transmitter 51, and to receive signals such as digital TV broadcasting and data transmission.

FIG. 19 is a block diagram of a first receiver, which receives radio waves from the satellite 10 at the antenna 22 and inputs the signals from the input unit 24, and then the carrier reproduction circuit 131 and? / 2 A carrier and an orthogonal carrier are reproduced by 132, and components orthogonal to each other are respectively independently detected by the first phase detection circuit 133 and the second phase detection circuit 134, and the timing wave extraction circuit 135 Are independently identified by the timeslots, and two independent demodulated signals by the first identification / regeneration circuit 136 and the second identification / regeneration circuit 137 are generated by the first data string regeneration unit 232. It is demodulated into a data string and output by the output unit 26.

Here, the reception signal will be described using the vector diagram of FIG. On the basis of the 4PSK transmission wave of the digital transmitter 51, the signals received by the first receiver 23 are four signals of (151) to (154) in FIG. 20 under ideal conditions without any transmission distortion or noise. It can be represented by a dot.

In practice, however, the received signal points are distributed in a certain range around the signal points under the influence of noise in the transmission path and amplitude distortion or phase distortion of the transmission system. If the signal point is separated from the signal point, the error rate cannot be distinguished from the adjacent signal point, and the error rate gradually increases, and if it exceeds a certain setting range, the data cannot be restored. Even in the worst case, the distance between adjacent signal points may be taken to demodulate within the set error rate. This distance is defined as 2A _R0 . When the limit reception input of 4PSK is applied, the signal point 151 is shown in Fig. 20 at | 0-a _R1 | A _R0 , | 0-b _R1 | If the transmission system is set to enter the first discriminating area 155 indicated by an oblique line of A _R0 , it can be demodulated later when the carrier is reproduced. If the lowest radius value set by the antenna 22 is r0, it can be received by all systems if the transmission output is set to a certain level or more. The amplitude of the transmission signal in FIG. 18 is set so as to be the 4PSK minimum reception amplitude, A _R0 of the first receiver 23. This transmission minimum amplitude value is defined as A _TO . As a result, if the radius of the antenna 22 is greater than or equal to r0, even if the reception condition is worst, the first receiver 23 can demodulate the signal of the digital transmitter 51. When receiving the modified 16QAM or 64QAM of the present invention, it is difficult for the first receiver 23 to reproduce the carrier. For this reason, as shown in FIG. 25 (a), when the transmitter 1 arranges and transmits 8 signal points at the position of the angle ((pi) / 4 + n (pi) / 2), a carrier wave can be reproduced by a four-multiplier system. Further, as shown in FIG. 25 (b), when 16 signal points are arranged on an extension line of an nπ / 8 angle, the signal points are degenerate by employing the carrier reproduction method of the 16 multiplication method in the carrier reproduction circuit 131. The carrier of the pseudo 4PSK type 16QAM modulated signal can be easily reproduced. In this case, a signal point of the transmitter 1 may be set and transmitted such that A ₁ / (A ₁ + A ₂ ) = tan (π / 8). Consider a case where a QPSK signal is received. Like the signal point position modulation / modification circuit 67 of the transmitter of FIG. 2, the signal point position may superimpose modulation such as AM on the signal point position of the QPSK signal of FIG. In this case, the first receiver 23 demodulates the position modulation signal or the position change signal of the signal point such as PM and AM. The first data string and the demodulation signal are output from the transmission signal.

Next, returning to the transmitter 1 and using the vector diagram of FIG. 9 to describe the 16PSK transmission signal of the transmitter ₁ , the amplitude A ₁ in the horizontal vector direction of the signal point 83 is shown in FIG. The 4PSK minimum transmission power A _TO of the digital transmitter 51 is made larger. Then, the signals of the first upper limit signal points 83, 84, 85, and 86 of FIG. 9 enter the fourteenth PSK receivable area 87 indicated by the diagonal lines. When these signals are received by the first receiver 23, these four signal points enter the first discriminating area of the reception vector diagram in FIG. Accordingly, the first receiver 23 determines that the signal point 151 of FIG. 20 is the signal point 151 of FIG. 20 regardless of which of the signal points 83, 84, 85 and 86 of FIG. Demodulate in a timeslot. As shown in FIG. 8, this data is (11) of the first divided signal point group 91 of the transmitter 1, that is, (11) of the first data string. Similarly, in the case of the second upper limit, the third upper limit, and the fourth upper limit, the first data string is demodulated. That is, the first receiver 23 demodulates only two bits of data of the first data string among the plurality of data strings of the modulated signal from the transmitter 1 of 16QAM or 32QAM or 64QAM. In this case, since the signals of the second data sequence and the third data sequence are all included in the first to fourth divided signal point groups 91, demodulation of the signals of the first data sequence is not affected. However, the regeneration of the carrier is affected, so a countermeasure as described later is required.

If the output of the satellite repeater is not limited, it can be realized by general 16 to 64 QAM of the conventional signal lighting distance method as shown in FIG. However, unlike the ground transmission as described above, in satellite broadcasting, the launch cost increases significantly when the weight of the satellite increases. Therefore, the transmission output is limited due to the output limit of the repeater of the main body and the power limit of the solar cell. This state will continue for some time unless the launch cost of the rockets has become inexpensive by technological innovation. The transmission power is 20W for communication satellites and 100W to 200W for broadcast satellites. Therefore, when 4PSK is transmitted to 16QAM of the signal lighting distance method as shown in FIG. 9, the amplitude of 16QAM is 2A ₁ = A _2, so 3A _{TO is} required and 9 times is required in terms of power. To be compatible, nine times as much power as 4PSK is required. In addition, if the first receiver of 4PSK is to be able to be received by a small antenna, it is difficult to obtain this much power in the currently planned satellite. For example, in a 40W system, 360W is required and cannot be economically realized.

In this case, if all receivers are antennas of the same size, the equidistant signal point method is most efficient if the same transmission power is used. However, considering a system combining receiver groups of antennas of different sizes, a new transmission scheme can be constructed.

Specifically, 4PSK increases the number of recipients by receiving them with a simple, low cost receiving system using a small antenna. Next, 16QAM is a high-performance using a medium-sized antenna but can be received by a high-cost multi-valued demodulation and reception system to provide high value-added services such as HDTV suitable for investment. This allows 4PSK and 16QAM, and in some cases 64DMA, to be transmitted hierarchically with a slight increase in transmit power.

For example, as shown in FIG. 10, the total transmission output can be lowered by taking signal point intervals such that A1 = A2. In this case, the amplitude A (4) for transmitting 4PSK can be represented by the vector 95, and becomes the square root of 2A ₁ ² . The overall amplitude A (16) can be represented by the vector 96 and becomes the square root of (A ₁ + A ₂ ) ² + (B ₁ + B ₂ ) ² .

That is, it is possible to transmit with twice the amplitude and four times the transmission energy when transmitting 4PSK. In the general receiver transmitting at the same distance signal point, the modified 16-value QAM cannot be demodulated, but can be received by the second receiver 33 by setting two thresholds A ₁ and A ₂ in advance. In the case of FIG. 10, the shortest distance of the signal points in the first divided signal point group 91 is A ₁ , and A ₂ / 2A ₁ compared with the distance between the signal points 2A ₁ of 4PSK. If the distance between the signal points of A ₁ = A ₂ is 1/2, and the same error rate is to be obtained, the reception sensitivity of twice the amplitude and the reception sensitivity of four times are required for energy. In order to obtain four times the reception sensitivity, the radius r2 of the antenna 32 of the second receiver 33 may be twice as large as that of the radius r1 of the antenna 22 of the first receiver 23, that is, r2 = 2r1. For example, if the antenna of the first receiver 23 is 30 cm in diameter, the antenna diameter of the second receiver 33 can be 60 cm. As a result, by demodulating the second data string, assigning it to the high frequency component of the HDTV enables new services such as HDTV to be performed on the same channel. The service content is multiplied so that the receiver can receive the service that is suitable for the investment of the antenna and receiver. Therefore, the high cost may be sufficient as the 2nd receiver 33. FIG. Since the lowest transmission power is determined for the 4PSK mode reception, the ratio of the transmission power of the modified 16APSK to the transmission power of 4PSK n16 and the antenna radius of the second receiver 33 are determined by the ratio of A ₁ and A ₂ in FIG. r2 is determined.

To calculate this optimization, the minimum required transmission energy of 4PSK is {(A ₁ + A ₂ ) / A ₁ } ² times. If we define this as n16, the distance between signal points when received by modified 16-value QAM is A ₂ , The signal-to-point distance when receiving at 4PSK is 2A ₁ , and the ratio of the distance between signal points is A ₂ / 2A _1, so the radius of the receiving antenna is r2. Curve 101 shows the relationship between the transmission energy ratio n16 and the radius r2 of the antenna 22 of the second receiver 23.

The point 102 is a case of transmitting 16QAM in the case of an equidistant signal point, and requires 9 times the transmission energy as described above, which is not practical. It can be seen from the graph that even if n16 is increased five times or more from FIG. 11, the antenna radius r2 of the second receiver 23 does not become very small.

In the case of satellites, the transmission power is limited, and it cannot take more than a certain value. From this, it becomes clear that n16 is preferably 5 times or less. This area is indicated by the oblique line of the area 103 in FIG. For example, within this area, for example, the point 104 is four times the transmit energy and the antenna radius r2 of the second receiver 23 is doubled. Also, point 105 has twice the transmission energy and r2 about five times. These are in a practical range.

If n ₁₆ is less than 5, then A ₁ and A ₂

Claim 10 when the distance between the dividing signal point group from the road as 2A (4), the maximum amplitude is 2A (16), and A (4) and A (16) -A (4) are proportional to A ₁ and A _2. So {A (16)} ² 5 {A (4)} ² may be used.

Next, an example using modified 64APSK modulation is shown. The third receiver 43 can perform 64-valued QAM demodulation.

The vector diagram of FIG. 12 is a case where the divided signal point group of the vector diagram of FIG. 10 is increased from 4 values to 16 values. In the first divided signal point group 91 in FIG. 12, signal points 170 and 4x4 = 16 values are arranged at equal intervals. In this case, A ₁ of transmission amplitude in order to make it compatible with 4PSK. Must be set to A _TO . If the radius of the antenna of the third receiver 43 is r3, the value of r3 in the case of defining the transmission and output signal n64 is obtained in the same manner.

The graph is the same as the radius r3-output multiple n of the 64 value QAM of FIG.

However, in the arrangement as shown in FIG. 12, when received by the second receiver 33, only two bits of 4PSK can be demodulated, so that the first, second, and third compatibility of the three receivers is established in the second receiver 33. It is desirable to have a function of demodulating the modified 16-value QAM from the modified 64-valued QAM modulated wave.

As shown in FIG. 14, compatibility of three receivers is achieved by grouping signal points of three layers. In the first upper limit only, the first divided signal point group 91 has been assigned two bits (11) of the first data string.

Next, the first partial signal point group 181 is allocated two bits (11) of the second data string. (01) is assigned to the second partial signal point group 182, (00) is assigned to the third partial signal point group 183, and (10) is assigned to the fourth partial signal point group 184. This is equivalent to FIG.

When the signal point arrangement of the third data string is described in detail using the first upper vector diagram of FIG. 15, for example, the signal points 201, 205, 209, and 213 are (11). Points 202, 206, 210, 214 to 01, signal points 203, 207, 211, 215 to 00, signal points 204, ( If 208, 212, and 216 are (10), two-bit data of the third data string can be transmitted independently of three layers of two-bit data independently of the first data and the second data.

In addition to sending 6 bits of data, as a feature of the present invention, receivers with different levels of performance can transmit 2 bits, 4 bits, and 6 bits of different amounts of data, and also the compatibility between transmissions of 3 layers. I can have it.

Here, a method of arranging signal points necessary to have compatibility in three-layer transmission will be described.

As shown in FIG. 15, in order to _first receive the data of the first data string in the first receiver 23, A ₁ Being A _TO has already explained.

Next, between the signal points of the second data string, for example, the signal points 91 of FIG. 10 and the signal points of 182, 183, and 184 of the partial-signal signal point group of FIG. It is necessary to secure the distance.

In FIG. 15, the case where 2 / 3A ₂ is dropped is indicated. In this case, the signal point distance between the first portion to the signal point group 181, the signal point 201 of the inner, 202 is the A _2/6. Received energy required when receiving by the third receiver 43 is calculated. In this case, if the radius of the antenna 32 is r ₃ and the required transmission energy is defined as n64 times the 4PSK transmission energy, r ₃ ² = (12r ₁ ) ² / (n-1).

This graph can be represented by the curve 221 of FIG. For example, in the case of points 222 and 223, when the transmission energy of 6 times the 4PSK transmission energy is obtained, the antenna of 8 times the radius, and the 9 times the transmission energy of the antenna 6 times, the first and second It can be seen that the third data stream can be demodulated. In this case, since the distance between signal points of the second data string is closer to 2 / 3A ₂ , r ₂ ² = (3r ₁ ) ² / (n-1), and the second receiver 33 is slightly lowered as shown by the curve 224. It is necessary to enlarge the antenna 32.

In this method, the first data stream and the second data stream are sent as long as the satellite transmit energy is small, and the first receiver 23 or the second receiver 33 is increased in the future when the satellite transmit energy is greatly increased. A large effect of both compatibility and development that the third data string can be sent without damaging or improving the received data can be obtained.

In order to explain the reception state, first, the second receiver 33 will be described. The first receiver 23 is configured to demodulate the 4PSK modulated signal of the digital transmitter 51 and the first data string of the transmitter 1 in a small antenna of the original radius r1. The second receiver 33 ), It is possible to completely demodulate the 16-value signal point shown in FIG. 10 of the transmitter 1, that is, the 2-bit signal of 16QAM of the second data string. A 4-bit signal can be demodulated in accordance with the first data string. In this case, the ratio of A ₁ and A ₂ differs depending on the transmitter. This data is set by the demodulation control unit 231 in FIG. 21, and a threshold is sent to the demodulation circuit. This enables AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 and the block diagram of the first receiver 23 in FIG. 19 are substantially the same. The difference is that first, the antenna 32 has a larger radius r2 than the antenna 22. For this reason, signals having a shorter distance between signal points can be discriminated. Next, the demodulator 35 has a demodulation control unit 231, a first data string regeneration unit 232, and a second data string regeneration unit 233. The first identification and reproduction circuit 136 has an AM demodulation function to demodulate the modified 16QAM. In this case, each carrier has a value of 4 values, and has a threshold of 0 level and ± 2 values. In the case of the present invention, since the modified 16QAM signal, as shown in the signal vector of FIG. 22, the threshold varies depending on the transmission output of the transmitter. Therefore, if the threshold value based on TH ₁₆ is referred to as evident from FIG.

Becomes

The A1, A2, or TH _16, and the demodulation information for the value m of the m-ary modulation is transmitted, including the from the transmitter 1, the first data string. The demodulation control unit 231 may also take a method of statistically processing the received signal to obtain demodulation information.

A method for determining the ratio of shift factors A ₁ / A ₂ will be described using FIG. 26. Changing A ₁ / A ₂ changes the threshold. The error increases as A ₁ / A ₂ set on the receiver side moves away from the value of A ₁ / A ₂ set on the transmitter side. The third receiver 43 controls the shift factor A ₁ / A ₂ in the direction in which the error rate decreases by feeding back the demodulation signal from the second data string regeneration unit 233 of FIG. 26 to the demodulation control unit 231. The circuit is simplified because the shift factor does not have to demodulate A ₁ / A ₂ . In addition, the transmitter does not need to send A ₁ / A ₂ and increases transmission capacity. This may be used for the second receiver 33. The demodulation control unit 231 has a memory 231a. When the channel of the TV broadcast stores different thresholds, that is, shift ratios, signal scores, and synchronization loops, and the channel is received again, the reception is stabilized quickly by calling this value.

If this demodulation information is unclear, demodulation of the second data string becomes difficult. It demonstrates using the flowchart of FIG. 24 (FIG. 24).

Even when demodulation information cannot be obtained, demodulation of the 4PSK in step 313 and the first data string in step 301 can be performed. In step 302, the demodulation information obtained by the first data string reproduction unit 232 is sent to the demodulation control unit 231. FIG. The demodulation control unit 231 demodulates 4PSK or 2PSK in step 313 when m is 4 or 2 in step 303. If NO, then at step 304, if m is 8 or 16, go to step 305; If NO, the flow advances to step 310. In Step 305 it is performed the operation of TH ₈ and TH _16. In step 306, the demodulation control unit 231 sends the threshold TH ₁₆ of the AM demodulation to the first identification playback circuit 136 and the second identification playback circuit 137. In step 307, 315, the demodulation of the modified 16QAM and the second data string are performed. Playback takes place. In step 308, the error rate is checked. If the error rate is bad, the flow returns to step 313 to perform 4PSK demodulation.

In this case, the signal points 85 and 83 in FIG. 22 are at an angle of cos (wt + nπ / 2), but the signal points 84 and 86 are not at this angle. Therefore, the carrier transmission information of the second data string is sent from the second data string reproduction unit 233 to the carrier reproduction circuit 131 in FIG. 21, and the carrier is set not to be extracted from the signal at the timing of the signal points 84 and 86. FIG. It is.

In the case where the second data string cannot be demodulated, the transmitter 1 intermittently sends a carrier timing signal by the first data string. Although the second data string cannot be demodulated by this signal, the signal points 83 and 85 can be known only by the first data string. For this reason, the carrier can be reproduced by sending the carrier transmission information to the carrier reproducing circuit 131.

Next, when the signal of the modified 64QAM as shown in FIG. 23 is sent from the transmitter 1, when it returns to the flowchart of FIG. 24, it is determined in step 304 whether m is not 16 and in step 310 whether m is 64 or less. If it is checked and if it is not the same distance signal point method in step 311, it will continue to step 312. In this case, if the distance between signal points TH ₆₄ for the modified 64QAM is obtained,

And is the same as TH ₁₆ . However, the distance between signal points becomes smaller.

When the distance between the signal points in the first subdivision signal point group 181 is A ₃ , the distance between the first subdivision signal point group 181 and the second subdivision signal point group 182 is (A ₂ -2A ₃ ), Standardizing is (A ₂ -2A ₃ ) / (A ₁ + A ₂ ). If this is defined as d ₆₄ , it cannot be discriminated when d ₆₄ is less than or equal to the discriminating capacity T _Z of the second receiver 33. In this case, if it is determined in step 314 d ₆₄ is outside the allowable range in Step 313 to enter the 4PSK mode. If it is within the discrimination range, the flow advances to step 305 to demodulate 16QAM in step 307. If the error rate is large in step 308, the 4PSK mode of step 313 is entered.

In this case, if the transmitter 1 transmits the modified 8QAM signal of the signal point as shown in Fig. 25 (a), since all signal points are on the angle of COS (2π ftn ° π / 4), four elements are used. By the double circuit, all carriers are degenerate in the same phase, so that the reproduction of the carrier becomes simple. In this case, two bits of the first data string can be demodulated even in a 4PSK receiver which is not considered, and one bit of the second data string can be reproduced by the second receiver 33, so that a total of three bits can be reproduced.

The signal point arrangement diagrams of Figs. 25A and 25B show signal points of C-CDM in the case where signal points shifted in the polar coordinate directions (r,?) Are added. The C-CDM above the Cartesian coordinates, i.e., the signal point shifted in the XY direction, is called the Cartesian coordinate system C-CDM. Call.

First, the signal point arrangement diagram of 8PS-APSK in Fig. 25 (a) adds one signal point shifted in the radial r direction in polar coordinates to each of the four signal points of QPSK. Thus, as shown in Fig. 25A, the APSK of the polar coordinate C-CDM having eight signal points from QPSK is realized. This is an abbreviation for shifted pole-APSK, which is called SP-APSK because it is an APSK that adds a signal shifted pole in polar coordinates. In this case, as shown in FIG. 139, the shift factor S ₁ can be used to define the coordinates of the signal point 85 added to the QPSK. The signal point of 8PS-APSK is the signal point ((S ₁ +1) r _{0 at} the position where the signal point 83 of the polar coordinates (r ₀ , θ ₀ ) of the standard QPSK is shifted by S ₁ , r ₀ in the radial r direction. , θ ₀ ) is added. In this way, subchannel 2 of 1 bit is added to subchannel 1 of 2 bits equal to QPSK.

As shown in the signal point arrangement diagram of FIG. 140, eight signal points of coordinates (r ₀ , θ ₀ ) and (r ₀ + S ₁ r ₀ , θ ₀ ) are shifted by S ₂ r ₀ in the radial r direction. By adding a signal point, a new signal point of (r ₀ + S ₂ r ₀ , θ ₀ ) and (r ₀ + S ₁ r ₀ + S ₂ r ₀ , θ ₀ ) is added. Since there are two kinds of arrangements, one-bit subchannel can be obtained. This is called 16PS-APSK, and has a 2-bit subchannel, a 1-bit subchannel 2, and a 1-bit subchannel 3. Since the 16-PS-APSK also has a signal point on θ = 1/4 (2n + 1) π, the second subchannel cannot be demodulated because the carrier can be reproduced by the conventional QPSK receiver described in FIG. The first subchannel of may be demodulated. As described above, the C-CDM method shifting in the polar coordinate direction can extend the amount of information transmitted while maintaining compatibility with the mainstream QPSK receiver in PSK, especially in current phase broadcasting. Therefore, it is possible to expand and maintain compatibility with satellite broadcasting having a large amount of information of multi-value modulation using APSK of the second generation without losing viewers of the first generation of satellite broadcasting using PSK.

The signal point in the case of Fig. 25B is above the angle = pi / 8 in polar coordinates. This is limited to four upper limit of the signal points of 16PSK, that is, three upper limit, or 12 signal points, of 16 signal points in total. By limitation, in the rough case, these three signal points can be regarded as one signal point and can be regarded as signal points of four QPSK as a whole. In this way, the first subchannel can be reproduced using the QPSK receiver in the same manner as in the above case.

These signal points are arranged on angles of θ = π / 4, θ = π / 4 + π / 8, θ = π / 4-π / 8. In other words, the signal point of shifting the signal point of QPSK on the angle π / 4 in the angular direction of polar coordinates is added. Since it is in the range of θ = π / 4 ± π / 8, it can be regarded as one signal point on θ = π / 4 of approximately QPSK. In this case, the error rate is slightly worse. However, since the QPSK receiver 23 shown in FIG. 19 can distinguish the signal points on four angles, demodulation can be performed, and 2-bit data is reproduced.

In the case of the angle shift C-CDM, when the angle is in the? / N phase, the carrier reproduction circuit can reproduce the carrier by the n-multiplier circuit as in the other embodiments. If there is no? / N phase, the carrier can be reproduced by sending a plurality of carrier information in a certain period as in the case of the other embodiment.

In addition, as shown in FIG. 141, if the angle in the polar coordinates between the signal points of QPSK or 8-PS-APSK is 2θ ₀ and the first difference angle shift factor is P ₁ , the signal points are divided into two and the angle θ By shifting by ± P ₁ θ _{0 in the} direction, it is divided into two signal points in the case of QPSK (r ₀ , θ ₀ + P ₁ θ ₀ ) and (r ₀ , θ ₀ -P ₁ θ ₀ ) The number doubles. In this way, one bit of subchannel 3 is added. This is called 8-PS-PSK of P = P ₁ . As shown in FIG. 142, the _{sum of} the signal points _obtained by shifting the signal points of the 8-PS-PSK by S _1r0 in the radial r direction is called 16-PS-APSK (P, S ₁ type). Subchannels 1 and 2 can be reproduced by 8PS-PSK having the same phase. But here comes back to FIG. 25 (b). Since the C-CDM using the angular shift of the polar coordinate system can be applied to the PSK as shown in FIG. 141, it can be used for the first generation satellite broadcasting. However, when used for satellite transfer of the second generation APSK, as shown in FIG. 142, the polar coordinate system C-CDM cannot uniformly space the signal points in the group. Therefore, the power utilization efficiency is bad. On the other hand, C-CDM with orthogonal display is not compatible with PSK.

The method of FIG. 25 (b) is compatible with both the rectangular coordinate system and the polar coordinate system. Since the signal is arranged on an angle of 16PSK, demodulation can be performed by 16PSK, and signal points are grouped, so that the QPSK receiver can also demodulate. In addition, since they are arranged on Cartesian coordinates, they can also be demodulated in 16-SRQAM. It is a method that can be extended while realizing compatibility between the polar coordinate system and the rectangular coordinate system C-CDM between three of QPSK, 16PSK, and 16-SRQAM.

Next, the third receiver 43 will be described. 26 is a block diagram of the third receiver 43, and is substantially the same configuration as the second receiver 33 of FIG. The difference lies in that the third data string reproducing unit 234 is added and that the identification reproducing circuit has eight values of discriminating ability. Since the radius r ₃ of the antenna 42 becomes larger than r ₂ , it is possible to demodulate a signal having a closer distance between signal points, for example, a 32-value QAM or a 64-value QAM. For this reason, in order to demodulate the 64 value QAM, the first identification reproduction circuit 136 needs to discriminate the level of 8 values from the detection signal wave. In this case, there are seven threshold levels. Since one of them is zero, there are three thresholds in one upper limit.

As shown in the signal space diagram of FIG. 27, three thresholds exist in the first upper limit.

As shown in FIG. 27, there are three normalized thresholds, TH 1 ₆₄ and TH 2 ₆₄ and TH 3 ₆₄ .

Can be displayed as

By this threshold value, AM demodulation of the received signal detected phase is demodulated in the same manner as the first data string and the second data string described with reference to FIG. As shown in FIG. 23, the third data string is taken four values, that is, two bits, by, for example, discriminating four signal points 201, 202, 203, and 204 of the first divided signal point group 181. FIG. In this way, demodulation of 6 bits, that is, modified 64-valued QAM, is possible.

Demodulation control unit 231 at this time is the first data string claim by the demodulating information included in the first data column of the reproducing unit _{(232), m, A 1} , A 2, so know the value of A ₃ the threshold value TH 1 ₆₄ TH 2 ₆₄ and TH 3 ₆₄ are calculated and sent to the first identification and playback circuit 136 and the second identification and playback circuit 137, whereby the modified 64 QAM demodulation can be surely performed. In this case, since the demodulation information is scrambled, only authorized recipients can demodulate 64QAM. 28 shows a flowchart of a demodulation control unit 231 of modified 64QAM. Only the differences in the flowchart of the 16-value QAM in FIG. 24 will be described. From step 304 in FIG. 28 to step 320, if m = 32, the 32 value QAM of step 322 is demodulated. If NO, it is determined in step 321 whether m = 64, and in step 323, since A ₃ is not a set value and cannot be reproduced, the flow proceeds to step 305, which is the same flowchart as in FIG. 24, and demodulates the modified 16QAM. Returning to step 323, if A ₃ is greater than or equal to the set value, the threshold value is calculated in step 324, and three threshold values are sent to the first and second identification and playback circuits in step 325, and the modified 64QAM is reproduced in step 326. In 327, the first, second, and third data are reproduced. If the error rate is large in step 328, 16QAM demodulation proceeds to step 305, and if small, 64QAM demodulation is continued.

Here, the carrier reproduction method important for demodulation will be described. The present invention is characterized in that the first data string of modified 16QAM or modified 64QAM is reproduced by the 4PSK receiver. In this case, when a normal 4PSK receiver is used, reproduction of the carrier becomes difficult and normal demodulation cannot be performed. To prevent this, some countermeasures are needed on the transmitter and receiver side.

There are two ways as a method according to the invention. The first method is a method of sending signal points on an angle of π / 4 intermittently (2n-1) based on a predetermined rule. The second method is a method in which almost all signal points are arranged and transmitted on an angle of nπ / 8.

In the first method, as shown in FIG. 38, signal points on four angles, π / 4, 3π / 4, 5π / 4, 7π / 4, for example, signal points 83, 85 ), The intermittent synchronous time slots 452, 452a, 452b, and 452c which are intermittently sent out of the time slot group 451 during the time chart of the transmission signal of FIG. Is set according to a certain rule. During this period, one signal point of the eight signal points on the angle is necessarily transmitted. Any other time slot transmits an arbitrary signal point. The transmitter 1 then arranges and transmits the above-mentioned rule for sending this time slot to the synchronization timing information unit 499 of the data shown in FIG.

In this case, the contents of the transmission signal will be described in more detail with reference to FIG. 41. The time slot group 451 including the synchronization time slots 452, 452a, 452b, and 452c includes one unit data. A column 491 constitutes Dn.

Since the synchronous time slots are intermittently arranged in accordance with the rules of the synchronous timing information in this signal, carrier reproduction can be facilitated by extracting the information in the synchronous time slots if this arrangement rule is known.

On the other hand, at the head of the frame of the data string 492, there is a synchronization area 493 indicated by S, which is composed of only synchronization time slots indicated by diagonal lines. This configuration increases the extraction information for reproduction of the carrier, so that the carrier reproduction of the 4PSK receiver can be surely and quickly performed.

This synchronization area 493 includes synchronization sections 496, 497, 498, and the like indicated by S1, S2, and S3, which contain a unique word for synchronization and the demodulation information described above. In addition, there is also a phase synchronization signal arrangement information unit 499 indicated by 1 _T , which contains information such as the information on the arrangement interval of the phase synchronization time slots and the information on the arrangement rule.

Since the signal point of an area of a phase-locked time slot is not to have only a specific phase carrier Content can be played in 4PSK receiver, the phase synchronization arrangement information 1 _T will provide excellent play, after the information obtained can surely reproduce the carrier wave have.

A demodulation information section 501 is provided next to the synchronization area 493 in FIG. 41, and contains demodulation information about a threshold voltage required for demodulating the modified multivalued QAM signal. Since this information is important for demodulation of multi-valued QAMs, the demodulation information 502 is included in the synchronization area as in the synchronization area 502 of FIG.

Fig. 42 is a signal arrangement diagram when a burst-shaped signal is sent by the TDMA method. The difference from FIG. 41 is that the guard time 521 is formed between the data string 491, Dn and another data string, and during this period, the transmission signal is not transmitted. At the head of the data string 492, a synchronizer 522 for synchronizing is formed. During this period, only the signal points of the above-described phases of (2n-1) π / 4 are transmitted. Therefore, the carrier can be reproduced even in a 4PSK demodulator. In this way, synchronization and carrier reproduction can be performed even in the TDMA method.

Next, the carrier reproduction method of the first receiver 23 of FIG. 19 will be described in detail with reference to FIG. 43 and FIG. The received signal input in FIG. 43 enters the input circuit 24, one of the demodulated signals synchronously detected by the synchronous detection circuit 541 is sent to the output circuit 542 for output, and the first data string is reproduced. . In the extraction timing control circuit 543, the phase synchronization unit arrangement information unit 499 of FIG. 41 is reproduced, and at which timing the signal of the phase synchronization unit of (2n-1)? / 4 comes in, the intermittent as shown in FIG. The in phase synchronous control signal 561 is sent. The demodulation signal is sent to the multiplication circuit 545 and multiplied by four to the carrier reproduction control circuit 544. Like the signal 562 of FIG. 44, the signal of true phase information 563 and other signals are included. As indicated by the oblique lines in the timing chart 564, a phase synchronization time slot 452 made up of a signal point having a phase of (2n-1)? / 4 is intermittently included. By sampling this by the carrier reproduction control circuit 544 using the phase synchronization control signal 564, the phase sample signal 565 is obtained. By sampling and holding this, a predetermined phase signal 586 is obtained. This signal is sent to the VCO 547 through the loop filter 546, the carrier is reproduced, and sent to the synchronous detection circuit 541. In this way, a signal point with a phase of (2n-1) [pi] / 4 as shown by the oblique line in FIG. 39 is extracted. Based on this signal, an accurate carrier can be reproduced by the triplex method. At this time, although a plurality of phases are reproduced, the absolute phase of the carrier can be specified by inserting a unique word after synchronization 496 of FIG.

In the case of transmitting the modified 64QAM signal as shown in FIG. 40, phase synchronization time slots 452 and 452b are applied only to signal points in the phase synchronization region 471 indicated by diagonal lines of approximately (2n-1) π / 4 phase. The sender sends a back. For this reason, although a carrier cannot be reproduced in a normal 4PSK receiver, the 4PSK 1st receiver 23 also has the effect that a carrier can be reproduced by equipping the carrier reproduction circuit of this invention.

The above is the case of using a coaster carrier circuit. Next, a case in which the present invention is used in an inverse modulation system carrier reproduction circuit will be described.

45 shows an inverse modulated carrier reproduction circuit of the present invention. The demodulated signal is reproduced by the synchronous detection circuit 541 by the received signal from the input circuit 24. On the other hand, the input signal delayed by the first delay circuit 591 is de-demodulated by the demodulation signal in the four-phase modulator 592 to become a carrier signal. The carrier signal passing through the carrier reproduction control circuit 544 is sent to the phase comparator 593. On the other hand, the reproduction carriers from the VCO 547 are delayed by the second delay circuit 594, and are compared in phase with the inverse modulation carrier signal described above by the phase comparator 593, and the phase difference signals are passed through the loop filter 546. 547, a carrier wave in phase with the reception carrier is reproduced. In this case, the extraction timing control circuit 543 samples the phase information of only the signal points in the region indicated by the oblique line in FIG. 39, similarly to the coaster-type carrier reproduction circuit in FIG. 43, so that the first receiver (16QAM, 64QAM, The carrier can be reproduced by the modulator of 4PSK.

Next, a description will be given of a method of reproducing a carrier by the 16 multiplication method. As shown in FIG. 46, the transmitter 1 of FIG. 2 arranges the signal point of modified 16QAM on the upper right side of n (pi) / 8, and performs modulation and transmission. On the first receiver 23 side of FIG. 19, the signal points of nπ / 8 phase as shown in FIG. 46 are degenerate to the first upper limit by the 16 multiplication circuit 661 as shown in FIG. The loop filter 546 and the VCO 547 can reproduce the carrier. By placing the unique words in the synchronization area, the absolute phase can be extracted from the 16 phases.

Next, the structure of the 16 multiplication circuit will be described. From the demodulated signal, the sum signal 662 and the difference circuit 663 produce the sum signal and the difference signal, and the multiplier 664 multiplies each other to produce cos2θ. The multiplier 665 produces sin2θ. These are multiplied by a multiplier 666 to form sin4θ.

Similarly from sin2θ and cos2θ, sin8θ is generated by the sum circuit 667, the difference circuit 668, and the multiplier 670. Cos8θ is formed by the sum circuit 671, the difference circuit 672, and the multiplier 673. By multiplying the multiplier 674 by making sin 16θ, 16 multiplication can be performed.

By the 16 multiplication method as described above, there is a great effect that the carriers of all signal points of the modified 16QAM signal having the signal point arrangement shown in FIG. 46 can be reproduced without extracting a specific signal point.

The carrier wave of the modified 64QAM signal having the arrangement as shown in Fig. 47 can also be reproduced. However, since some signal points are slightly shifted from the synchronization area 471, the error rate during demodulation increases.

There are two methods for this countermeasure. One is to not transmit the signal at the signal point out of the synchronization area, but the information amount is reduced, but the configuration is simplified. The other is to form a synchronous timeslot as described in FIG. During the period of the synchronization time slot in the time slot group 451, signal points such as the synchronous phase areas 471 and 471a of the nπ / 8 phase indicated by the diagonal lines can be precisely synchronized during this period. The phase error is reduced.

As described above, the 16 multiplication method has a great effect of reproducing the carrier of the modified 16QAM or modified 64QAM signal by the 4PSK receiver with a simple receiver configuration. In addition, when the synchronization time slot is further set, the effect of increasing the phase accuracy during carrier reproduction of the modified 64QAM can be obtained.

As described in detail above, by using the transmission apparatus of the present invention, a plurality of data can be simultaneously transmitted in a hierarchical structure in one radio band.

In this case, it is possible to demodulate the data amount suitable for the investment of the receiver by setting three layers of receivers having different reception sensitivity and demodulation capability for one transmitter. First, a receiver who purchases a small antenna and low resolution but low cost first receiver can demodulate and reproduce the first data stream. Next, a receiver who purchases a medium antenna and a high resolution second receiver can reproduce the first and second data streams. In addition, a person who purchases a large antenna and a high resolution third receiver, which is quite high cost, can demodulate and reproduce all of the first, second, and third data strings.

If the first receiver is a home digital satellite broadcast receiver, the receiver can be realized at a low price that can be accommodated by many general consumers. The second receiver initially requires a large antenna and is not acceptable to the general consumer because it is a high cost, but it is meaningful for those who want to watch HDTV. The third receiver requires a fairly large industrial antenna for the time until the satellite output increases, which is not practical for home use but initially suitable for industrial use. For example, if a high definition HDTV signal is sent and transmitted to movie theaters by satellite, the movie theater can be digitalized by video. In this case, the operating cost of a movie theater or video theater is also reduced.

As described above, when the present invention is applied to TV transmission, it is possible to provide three video quality image services in one frequency band of one radio wave, and also to be compatible with each other. In the embodiment, examples of 4PSK, modified 8QAM, modified 16QAM, and modified 64QAM are shown, but it can also be realized in 32QAM or 256QAM. It can also be performed at 8PSK, 16PSK, or 32PSK. In addition, although the embodiment has shown an example of satellite transmission, it goes without saying that the same can be realized in the ground transmission and the wire transmission.

The present invention can also be applied to four or eight ASK signals as shown in FIG. 58 or 68 (a) and (b).

Example 2

In Embodiment 2, the physical layer structure described in Embodiment 1 is further divided logically by differentiation of error correction capability, and a logical hierarchy structure is added. In the case of the first embodiment, each layer channel has a different electrical signal level, that is, a physical demodulation capability. On the other hand, in Example 2, logical reproducibility, such as error correction ability, differs. Specifically, for example, the data of the channel layer of the D _1, for example is divided into two of D _1-1 and D _1-2, it is one of the divided data, for example, error correction of data D _1-1 If the capacity is higher than the D _1-2 data and the error correction capability is differentiated, and the error demodulation ability of the data of D _1-1 and D _1-2 is different during demodulation playback, the C / N value of the transmission signal is lowered. Even at a signal level at which D _1-2 cannot be reproduced, D _1-1 can be stored within the set error rate to reproduce the original signal. This is a logical hierarchy.

In other words, by dividing the data of the modulation layer channel and differentiating the magnitude of the error-correction correction code such as the use of the error correction code and the bone code, a logical hierarchical structure according to the error correction capability is added and more detailed layer transmission is possible. Done.

Using this, the D ₁ channel extends to two subchannels of D _1-1 , D _1-2 , and the D ₂ channel to _two subchannels of D _2-1 and D _2-2 .

If this is explained using the C / N value of the input signal and FIG. 85 of the hierarchical channel number, the hierarchical channel D _1-1 can be reproduced with the lowest input signal. If this CN value is d, when CN = d, D _1-1 is reproduced, but D _1-2 , D _2-1 , and D _2-2 are not. Next, when CN = c or more, D _1-2 is regenerated thereon, when CN = b, D _2-1 is added, and when CN = a, D _2-2 is added. As the CN rises in this manner, the total number of hierarchical layers that can be reproduced increases. Conversely, as the CN goes down, the total number of layers that can be played back decreases. This is explained in the diagram of the transmission distance of FIG. 86 and the reproducible CN value. In general, as indicated by the solid line 86 in FIG. 86, as the transmission distance increases, the C / N value of the received signal decreases. Assuming that the distance from the transmission antenna at the point where CN = a described in FIG. 85 is La, the distance is Lb at CN = b, Lc at CN = c, Ld at CN = d, and Le at CN = e. Areas narrower than the distance from Ld can only reproduce D _1-1 channels as described in FIG. Receivable range of this D _1-1 is indicated by the diagonal area 862. As is apparent from the figure, the D _1-1 channel can reproduce in the widest area. Similarly, the D _1-2 channels can be reproduced in the region 863 within the distance Lc from the transmission antenna. Since the area 862 is also included in the range within the distance Lc, the D _1-1 channel can also be reproduced. Similarly, in the region 864, the D _2-1 channel can be reproduced, and in the region 865, the D _2-2 channel can be reproduced. In this manner, hierarchical transmission can be performed in which the transmission channel not accompanied by deterioration of the CN value is gradually reduced. By separating the data structure into a hierarchical structure and using the hierarchical transmission of the present invention, there is an effect that hierarchical transmission in which the amount of data gradually decreases as the C / N deteriorates, such as analog transmission, becomes possible.

Next, a specific configuration will be described. Here, embodiments of two physical layers and two logical layers will be described. 87 is a block diagram of a transmitter. Since it is basically the same as the block diagram of the transmitter of FIG. 2 described in Embodiment 1, the detailed description is omitted, except that an error correction coder is added. This is called an ECC encoder. The separation circuit 3 has four outputs of 1-1, 1-2, 2-1, 2-2, and input signals D _1-1 , D _1-2 , D _2-1 , D _2-2 The output is separated into 4 signals of. The D _1-1 and D _1-2 signals are inputted to the first ECC encoder 871a, and are transmitted to the primary ECC encoder 872a and the secondary ECC encoder 873a, respectively, and error definition is defined. The encoding is done.

Here, the main ECC encoder 872a has a stronger error correction capability than the secondary ECC encoder 873a. For this reason, as described in the graph of 85 degrees CN- layer channel, D _1-1 channel demodulating playback even in the low C / N value than D _1-2 D _1-1 channel is less than the reference error rate can be played on the have. D _1-1 has a logical hierarchy that is more resistant to C / N degradation than D _1-2 . The miscorrected D _1-1 and D _1-1 signals are synthesized by the synthesizer 874a into the D ₁ signal and input to the modulator 4. On the other hand, the D _2-1 and D _2-2 signals are combined by the main ECC encoder 872b in the _second ECC encoder 871b into the D ₂ signals, respectively, and are input by the modulator 4. The primary ECC encoder 872b has a higher error correction capability than the secondary ECC encoder 873b. In this case, the modulator 4 generates a hierarchical modulated signal from the D ₁ signal and the D ₂ signal and transmits it from the transmitter 5. As described above, the transmitter 1 of FIG. 87 has a physical layer structure of two layers, D ₁ and D ₂ by modulation described in the first embodiment. This explanation has already been made. Next, due to the differentiation of error correction capability, D _1-1 and D _1-2 or D _2-1 and D _2-2 have two logical layers.

Next, a state of receiving this signal will be described. 88 is a block diagram of a receiver. The basic configuration of the second receiver 33 that has received the transmission signal of the transmitter in FIG. 87 is substantially the same as that of the second receiver 33 described in FIG. The difference of adding ECC decoder 876a (876b) is shown. In this case, although the example of QAM modulation / demodulation is shown, ASK, PSK, and FSK modulation / demodulation may be sufficient.

By the way, in FIG. 88, the received signal is reproduced as a D ₁ , D ₂ signal by the demodulator 35 and D _1-1 and D _1-2 , D _2- by the separators 3a and 3b, respectively. Four signals, ₁ and D _2-2 , are generated and input to the first ECC decoder 876a and the second ECC decoder 876b. In the first ECC decoder 876a, the D _1-1 signal is miscorrected by the main ECC decoder 877a and sent to the synthesizer 37. On the other hand, the D _1-2 signal is miscorrected by the secondary ECC decoder 878a and sent to the synthesizer 37. Similarly, in the second ECC decoder 876b, the D _2-1 signal is miscorrected and corrected in the primary ECC decoder 877b, and the D _2-2 signal in the secondary ECC decoder 878b. It is input to the synthesizer 37. The error-corrected D _1-1 , D _1-2 , D _2-1 , D _2-2 signals become one signal in the synthesizer 37 and are output from the output unit 36.

In this case, because of the logic hierarchy structure, D _1-1 has a higher error correction capability than D _1-2 , and D _2-1 has a higher error correction capability than D _2-2 . Even when the value is lower, a predetermined error rate can be obtained and the original signal can be reproduced.

Specifically, a method of differentiating error correction capability between the main ECC decoder 877a and 877b of the high code gain and the secondary ECC decoder 878a and 878b of the low code gain will be described. If the secondary ECC decoder uses a standard inter-code encoding method such as the Reed Solomon code or the BCH code as shown in the drawing of the ECC decoder of FIG. 165 (b), the main ECC decoder Error correction using the trellis decoders 744p, 744q and 744r as shown in Figures 128 (d) (e) (f) or multiplication codes of both the Reed Solomon code and the Reed Solomon code. By using a coding method with a large sign-to-signal distance, the error correction capability, that is, the code gain, can be different. In this way, a logical hierarchy can be realized. Since various methods are known, the method of increasing the distance between codes is omitted for other methods. The present invention can basically be applied in any manner.

As shown in the block diagrams of FIGS. 160 and 167, the interleaver 744k is provided in the transmitter and the deinterleaver 759k and 936b in the receiver, and the interleaved table 954 in FIG. By interleaving and decoding in the deinterleave RAM 936x of the deinterleaver 936b, strong transmission is possible against burst errors of the transmission system, and the image is stabilized.

Here, the logical hierarchical structure will be described using the relationship diagram between C / N of FIG. 89 and error rate after error correction. In FIG. 89, the straight line 881 indicates the relationship between the C / N of the D _1-1 channel and the error rate, and the straight line 882 shows the relationship between the C / N of the D _1-2 channel and the error rate after correction. Is displayed.

The smaller the C / N value of the input signal is, the larger the error rate of the corrected data is. Below a certain C / N value, the error rate after error correction is not stored below the standard error rate Eth at the time of system design, and the original data is not reproduced normally. Incidentally, in FIG. 89, when C / N is gradually increased, as shown by the straight line 881 of the D _1-1 signal, when the C / N is e or less, demodulation of the D ₁ channel cannot be performed. e In the case of C / Nd, the D ₁ channel can be demodulated, but the error rate of the D _1-1 channel exceeds Eth and original data cannot be reproduced normally.

When C / N = d, since D _1-1 has a higher error correction capability than D _1-2 , the error rate after error correction is equal to or lower than Eth as indicated by the point 885d, and data can be reproduced. . Meanwhile, the D _1-2 is error correcting capability and error rate after correction and error rate after correction because high as D _1-1 is so low as D _1-1 can not be regenerated because more than Eth and E _2. Therefore, only D _1-1 can be played back in this case.

When C / N is improved and C / N = C, the error rate after error correction of D _1-2 reaches Eth as indicated by the point 885C, and thus becomes recyclable. At this point, the demodulation of the D _2-1 , D _2-2, or D ₂ channels is in an uncertain situation. With the improvement of C / N, it is possible to reliably demodulate the D ₂ channel at C / N = b '.

In C / N is enhanced at the time of the C / N = b, and reduced to Eth as shown by error rate point (885b) of the D _2-1, D _2-1 is able to play. At this time, since the error rate of D _2-2 is larger than Eth, it cannot be reproduced. As C / N = a and indicated by the point 885a, the error rate of D _2-2 is reduced to Eth, so that the D _2-2 channel can be reproduced.

In this way, by using the differentiation of error correction capability, the physical layer D ₁ and D ₂ channels are further divided into two logical layers of two layers, so that four layers of layer transmission can be achieved.

In this case, the data structure is a hierarchical structure capable of reproducing a part of the original signal even when high-layer data is missing, and in combination with the hierarchical transmission of the present invention, the data amount is gradually increased in accordance with C / N degradation, such as analog transmission. There is an effect that reduced hierarchical transmission is possible. In particular, the latest image compression technology is rapidly progressing. Therefore, when image compression data is hierarchical and combined with hierarchical transmission, images of much higher quality than analog transmission are transmitted between the same points and at the same time. Likewise, it is possible to receive in a wide area while lowering image quality according to the received signal level step by step. In this way, the effects of hierarchical transmission, which were not present in the conventional digital video transmission, can be obtained while maintaining high image quality by digital.

Also, the data most important for image extension of HDTV signals such as address data of image segment data, reference image data at the time of image compression, the scramble release data displayed by the descrambler of FIG. As the _1-1 , the ECC encoder 743a of the high-wood gain of FIGS. 88, 133, 170, and 172 is transmitted, and the ECC decoder 758 of the high-wood gain of the receiver 43 is obtained. Receive from In this system, even if the error rate of the signal is increased due to deterioration of the C / N, the error rate of the high rank data D _1-1 does not increase so much. The effect of full degradation is obtained. The modulation unit 750 and the demodulation unit 760 of FIG. 133 and FIG. 170 are graceful degradation even in the above-described 160QAM and 32QAM, as well as in the 4VSB of FIG. 57, 8VSB of FIG. 68 and 8PSK of FIG. The effect of is obtained.

Also, as shown in Figs. 133 and 156, the high-order data is misinterpreted by the ECC encoder 744a and trellis encoder 744b in the second data string input unit 744. By performing encoding and performing the low code gain error coding only with the ECC encoder 743a, the error rate of the high rank data and the low rank data at the time of reception can be greatly different. Because of this, high priority data can be received even in the case of a significant degradation of C / N of the transmission system. Therefore, even in applications where severe degradation of C / N such as a receiver with poor reception conditions such as a car TV receiver, Image quality deteriorates with deterioration. However, since the high-order data is reproduced, the arrangement information of the pixel blocks is reproduced, so that the image is not destroyed, and an image with deteriorated resolution and noise can be obtained, and a remarkable effect is obtained that the viewer can watch a TV program. Lose.

Example 3

A third embodiment of the present invention will be described below with reference to the drawings.

29 is an overall view of Embodiment 3. FIG. The third embodiment shows an example of using the transmission apparatus of the present invention in a digital TV broadcasting system, and inputs an ultra high resolution input video 402 to the input unit 403 of the first image encoder 401 and separates it. By the circuit 404, the first data string, the second data string, and the third data string are separated, and are compressed and output by the compression unit 405.

The other input images 406, 407, and 408 are accumulated and output by the second image encoders 409, 410, and 411 having the same configuration as that of the first image encoder 401, respectively. do.

Of these four sets of data, four sets of signals of the first data sequence are temporally multiplexed by the first multiplexer 413 of the multiplexer 412 such as a TDM scheme and sent to the transmitter 1 as the first data sequence. Lose.

All or part of the signal group of the second data string is multiplexed by the multiplexer 414 and sent to the transmitter 1 as the second data string. Further, all or part of the signal group of the third data string is multiplexed by the multiplexer 415 and sent to the transmitter 1 as the third data string.

In response, the transmitter 1 modulates three data strings by the modulator 4 as described in the first embodiment, and the transmitter 5 transmits the antenna 6 and the transmission path 7 to the satellite 10. The transmitter 12 is sent to the three types of receivers such as the first receiver 23 by the repeater 12.

The first receiver 23 receives the small-diameter antenna 22 having a radius r1 by the transmission path 21, reproduces only the first data string in the received signal in the first data reproducing unit 232, and reproduces the first image. The coder 421 reproduces and outputs low resolution image outputs 425 and 426 such as NTSC signals.

In the second receiver 33, the first data string and the second data string are received by the middle diameter antenna 32 having a radius r2, and are received by the first data string playback section 232 and the second data string playback section 233. The second image decoder 422 reproduces and outputs a high resolution video output 427 or video outputs 425 and 426 such as an HDTV signal.

The third receiver 43 receives from the large-diameter antenna 42 having a radius r3 and transmits the first data string regeneration unit 232, the second data string regeneration unit 233, and the third data string regeneration unit 234. By this, the first data stream, the second data stream and the second data stream are reproduced, and the video output 428 of ultra high resolution such as an HDTV for a video theater or a movie theater is output. Video outputs 425, 426, and 427 may also be output. The general digital TV broadcast is broadcast from the digital transmitter 51 and, when received by the first receiver 23, is output as a low-resolution video output 426 such as NTSC.

Next, the configuration will be described in detail based on the block diagram of the first image encoder 401 of FIG. The ultra high resolution video signal is input to the input unit 403 and sent to the separation circuit 404. In the separation circuit 404, four signals are separated by subband coding. The horizontal low pass filter 451 and the horizontal high pass filter 452, such as QMF, separate the horizontal low component and the horizontal high component, and the subsampling portions 453 and 454 each component has a sampling rate. After halving, the horizontal low pass components are vertical low pass filter 455 and vertical high pass filter 456, respectively, the horizontal low vertical low pass signal, omitted the H _L V _L signal and the horizontal low vertical vertical high pass signal, omitted. The sampling rate is separated into the H _L V _H signal, and the sampling rate is dropped by the sampling units 457 and 458 and sent to the compression unit 405.

Horizontal high-frequency component is a vertical low-pass filter 459 and the vertical by a high-pass filter 460, the horizontal high vertical low-band signal, not to H _H V _L signal and a horizontal high vertical high-band signal, not to H _H V _H signal The sampling rate is lowered by the subsampling units 461 and 462 and sent to the compression unit 405.

In the compression unit 405, the H _L V _L signal is optimally compressed by the first compression unit 471 such as DCT and output from the first output unit 472 as a first data string.

The H _L V _H signal is compressed by the second compression unit 473 and sent to the second output unit 464. The H _H V _L signal is compressed by the third compression unit 463 and sent to the second output unit 464. The H _H V _H signal is divided into a high resolution image symbol (H _H V _H 1) and an ultra high resolution image signal (H _H V _H 2) by a separation circuit 465, and H _H V _H 1 is a second output unit ( 464, H _H V _H 2 is sent to the third output unit 468.

Next, the first image decoder 421 will be described with reference to FIG. The first image decoder 421 inputs the output from the first receiver 23, the first data string, that is, D ₁ into the input unit 501, and decompresses the descrambling unit 502 by the decompression unit. 503, the aspect ratio is changed by the aspect ratio changing circuit 504 and the output unit 505 after extending to the H _L V _L signal described above. An image 507, an image 508 of a full screen of a wide TV, or an image 509 of a side panel screen of a wide TV. In this case, the types of two scanning lines of non-interlace or interlace can be selected. In the case of NTSC, 525 scan lines and 1050 by double drawing can be obtained. When the 4PSK general digital TV broadcast is received from the digital transmitter 51, the first receiver 23 and the first image decoder 421 can demodulate and reproduce the TV image.

Next, the second image decoder will be described using the block diagram of the second image decoder in FIG. First, the D ₁ signal from the second receiver 33 is input from the first input unit 521, is extended by the first extension unit 522, and doubles the sampling rate by the oversampling unit 523 and is vertical. The low pass filter 524 reproduces the H _L V _L signal. The D ₂ signal is input from the second input unit 530 and separated into three signals by the separation circuit 531, and the second extension part 532, the third extension part 533, and the fourth extension part ( 534 is stretched and descrambled respectively, and the sampling rate is doubled by the oversampling units 535, 536 and 537, and the vertical high pass filter 538, vertical low pass filter 539, Sent by the vertical high pass filter (54). The H _L V _L signal and the H _L V _H signal are added by the adder 525, are converted into a horizontal low pass video signal by the oversampling unit 541 and the vertical low pass filter 542, and sent to the adder 543. . The H _H V _L signal and the H _H V _L signal are added by the adder 526, and the horizontal high-pass video signal is added by the oversampling unit 544 and the horizontal high pass filter 545, and the HDTV is added by the adder 543. A high-definition video signal HD signal such as an image is output, and an image output 547 such as an HDTV is output from the output unit 546. In some cases, NTSC signals are also output.

33 is a block diagram of a third image decoder, in which the D ₁ signal is input from the first input unit 521 and the D ₂ signal is input from the second input unit 530, and the above-described sequence is performed by the high-definition image decoder 527. HD signal is played back. The D ₃ signal is input from the third input unit 551, stretched, descrambled, and synthesized by the ultra-high range image decoder 552 to reproduce the H _H V _H 2 signal. The signal is synthesized by the HD signal and the synthesizer 553, and becomes an ultra high definition TV signal and an S-HD signal, and an ultra high resolution video signal 555 is output from the output unit 554.

Next, a specific multiplexing method of the multiplexer 401 mentioned in the description of FIG. 29 will be described. 34 is a data array diagram, in which six NTSC channels L1, L2, L3, L4, L5, L6 and six HDTVs are arranged in the first data stream, D ₁ and the second data stream, D ₂ and the third data stream D ₃ . It shows how channels M1 to M6 and six S-HDTV channels H1 to H6 are arranged on the time axis during the period of T. FIG. 34 Turning first to place by the time-multiplexed with such a L1~L6 a D ₁ signal to the period T of the TDM scheme. The H _L V _L signal of the first channel is sent to the region 601 of D ₁ . Next, in the area 602 of the D ₂ signal, the difference information M1 between the HDTV and NTSC of the first channel in the time domain corresponding to the first channel, that is, the H _L V _H signal, the H _H V _L signal, and the H _H V. Sends the _H 1 signal. The S-HDTV difference information H1 of the first channel, that is, H _H V _H -2H1 described in FIG. 30, is sent to the area 603 of the D ₃ signal.

Here, a case where the TV pole of the first channel is selected will be described. First, a general receiver having a system of a small antenna, a first receiver 23, and a first image decoder 421 can obtain TV signals of NTSC or wide NTSC of FIG. Next, the specific receiver with the medium antenna, the second receiver 33, and the second image decoder 422 selects the first data string, the region 602 and the second data string, D of the first data string, D ₁ . Synthesize the signals in the _two regions 602 to obtain HDTV signals with the same program content as the NTSC programs on channel 1.

Partial receivers, such as a movie theater with a large antenna and multi-value demodulator 43 and a third image decoder 423, are the area 601 of D ₁ and the area 602 and D of D ₂ . The signals of the _three regions 603 are synthesized, and a super-resolution HDTV signal with a cinematic picture quality is obtained with the same program content as that of the NTSC of the channel 1. Other channels 2 to 3 are similarly played back.

35 is a configuration of another area. First, the first channel of NTSC is arranged in L1. This L1 is located at the position 601 of the one time region of the D ₁ signal, and contains information S11 including descrambling information between NTSCs and demodulation information described in the first embodiment at the head. Next, the first channel of the HDTV is divided into L1 and M1. M1 is the difference information between HDTV and NTSC, which contains on either side of the area of the D ₂ (602) and region 611. In this case, if the NTSC compressed signal of 6 Mbps is adopted and accommodated in L1, the bandwidth of M1 is 12 Mbps, which is doubled.

When L1 and M1 are added together, an 18 Mbps band can be demodulated and reproduced from the second receiver 33 and the second image decoder 422. On the other hand, it is possible to realize the HDTV compressed signal in the band of about 15Mbps using the compression method currently proposed. Thus, the arrangement of FIG. 35 enables simultaneous broadcasting of HDTV and NTSC on channel 1. In this case, HDTV cannot be played on channel 2. S21 is descrambling information of the HDTV. The super HDTV signal is broadcast by dividing it into L1, M1 and H1. Using the difference information in the region 603, 612 and 613 of D ₃ of a super HDTV, the case where the NTSC 6Mbps even set, are sent in total 36Mbps, the scanning lines of the cinema image quality, increasing the compression of about 2000 Super HDTV signals can also be broadcast.

FIG. 36 shows a case in which six time domains are occupied in D ₃ and a super HDTV signal is transmitted. When the NTSC compression signal is set to 6Mbps, it can transmit 54Mbps which is 9 times. For this reason, a higher quality Super HDTV can be transmitted.

The above is the case where one of the horizontal or vertical polarization planes of the radio waves of the transmission signal is used. Here, by using two polarized planes, horizontal and vertical, the frequency utilization efficiency is doubled. A description is given below.

FIG. 49 shows signal arrangement diagrams of horizontally polarized signal D _V1 and vertically polarized signal D _H1 of the first data string, and D _V2 and D _H2 of the second data string, and D _V3 and D _H3 of the third data string. In this case, a low-frequency TV signal such as NTSC is contained in the vertically polarized signal D _V1 of the first data string, and a high-frequency TV signal is contained in the horizontally polarized signal D _H1 of the first data string. Therefore, the first receiver 23 having only a vertically polarized antenna can reproduce a low pass signal such as NTSC. On the other hand, the first receiver 23 having vertically and horizontally bidirectionally polarized antennas can, for example, synthesize L1 and M1 signals to obtain an HDTV signal. In other words, when the first receiver 23 is used, since the NTSC can be played on one side and NTSC and HDTV on the other, the two systems are compatible.

50 shows a case of using the TDMA system, and the synchronization unit 731 and the guard unit 741 are formed at the head of each data burst 721. FIG. The synchronization information unit 720 is formed at the beginning of the frame. In this case, one channel is assigned to each timeslot group. For example, the first timeslot 750 may send NTSC, HDTV, and SuperHDTV of the same program of the first channel. Each timeslot 750-750e is completely independent. Therefore, when a specific broadcaster broadcasts in a TDMA manner using a specific timeslot, NTSC, HDTV, and Super HDTV can be broadcast independently from other stations. In addition, in the case where the receiving side also has a horizontally polarized antenna and has a first receiver 23, the HDSC can be reproduced if both sides of the NTSC TV signal are antennas. With the second receiver 33, a low resolution super HDTV can be reproduced. The third receiver 43 can completely reproduce the super HDTV signal. It is possible to construct a broadcast system compatible with the above. In this case, in the arrangement as shown in FIG. 50, the time multiplexing of the continuous signal as shown in FIG. 49 is possible instead of the burst-shaped TDMA scheme. In addition, when the signal arrangement as shown in FIG. 51 is used, HDTV signals with higher altitude can be reproduced.

As described above, according to the third embodiment, there is a remarkable effect that digital TV broadcasting compatible with three signals of ultra-high definition HDTV, HDTV, and NTSC-TV is possible. In particular, when transmitted to a movie theater or the like, there is a new effect that the image can be electronicized.

Here, the modified QAM according to the present invention will be referred to as SRQAM, and a specific error rate will be described.

First, the error rate of 16SRQAM is calculated. 99 is a vector diagram of signal points of 16SRQAM. In the first upper limit, in the case of 16QAM, the intervals of the 16 signal points such as the signal points 83a, 83b, 84a, 85, 86a, and the like are equally spaced, and both are 2δ.

The signal point 83a of 16QAM is at a distance δ from the I axis and the Q axis of the coordinate axis. In the case of 16SRQAM, when n is defined as a shift value, the signal point 83a shifts and moves the distance from the coordinate axis to the signal point 83 at the position of nδ. In this case n is

to be. The other signal points 84a and 86a are also shifted to move to the positions of the signal points 84 and 86. If the error rate of the first data string is Pe1

If the error rate of the second data string is Pe2

Becomes

Next, calculate the error rate of 36SRQAM or 32SRQAM. 100 is a signal vector diagram of 36SRQAM. In the first upper limit, the distance between signal points of 38QAM is defined as 2 ?.

The signal point 83a of 36QAM is at a distance δ from the coordinate axis. When the signal point 83a reaches 36SRQAM, the signal point 83a shifts to the position of the signal point 83, and the distance is nδ from the coordinate axis. Each signal point is shifted to become signal points 83, 84, 85, 86, 97, 98, 99, 100, and 101. When the signal point group 90 consisting of nine signal points is regarded as one signal point and received by the modified 4PSK receiver, and only the first data string D ₁ is reproduced, the error rate is set to Pe1. If the error rate at the time of reproducing the second data string D ₂ by discriminating nine signal points from each other,

Becomes

In this case, the C / N-error rate diagram in FIG. 101 shows an example in which the relationship between the error rate Pe and C / N of the transmission system is calculated. Curve 900 shows the error rate of 32QAM in the conventional manner for comparison. The straight line 905 indicates a straight line having an error rate of 10 ^-1.5 . When the shift amount n of SRQAM of the present invention is set to 1.5, the error rate of the first layer D ₁ becomes the curve 901a, and when the error rate is 10 ^-1.5 , the C / N value for the 32QAM of the curve 900 is 10. Even if it goes down 5dB, D ₁ can reproduce at the same error rate.

Next, the error rate of the second layer D ₂ when n = 1.5 is represented by a curve 902a. If the error rate is 10 ^-1.5 , the C / N cannot be reproduced at the same error rate unless the C / N is increased by 2.5 dB compared to the 32QAM indicated by the curve 900. Curves 901b and 902b indicate D ₁ and D ₂ when n = 2.0. Curve 902c represents D ₂ . When this clearance, compared to the 32QAM when the error rate according to the value of 10 ^-1.5 22n = 1.5,2.0,2.5 D ₁ respectively is improved 5,8,10dB, D ₂ is deteriorated 2.5dB.

In the case of 32SRQAM, when the shift amount n is changed, the C / N value of the first data string D ₁ and the second data string D ₂ required to obtain a predetermined error rate is shown in the relation diagram of shift amount n and C / N in FIG. Display. As apparent from FIG. 103, when n is 0.8 or more, it is understood that a difference in C / N values necessary for hierarchical transfer, i.e., transfer of the first data string D ₁ and the second data string D ₂ , is produced, resulting in the effect of the present invention. Can be. Therefore, 32SRQAM is effective under the condition of n0.85. In the case of 16SRQAM, the error rate is the same as the relationship between C / N and error rate in FIG.

In FIG. 102, the curve 900 indicates the error rate of 16QAM. Curves 901a, 901b, and 901c indicate error rates when n = 1.2, 1.5, 1.8 of the first data string D ₁ , respectively. Curves 902a, 902b, and 902c indicate error rates when n = 1.2, 1.5 and 1.8 of the second data string D ₂ , respectively.

The relationship diagram between the shift amount n and C / N in FIG. 104 shows the C / N values of the first data string D ₁ and the second data string D ₂ required to obtain a specific error rate when the shift amount n is changed in the case of 16SRQAM. It is displayed. As is apparent from FIG. 104, in case of 16SRQAM, n0.9 indicates that hierarchical transmission of the present invention is enabled. From the above, if n0.9, hierarchical transmission is established.

Specifically, an example in which the SRQAM of the present invention is applied to terrestrial broadcasting of digital TV is shown. 105 shows a relationship between the distance between the transmitting antenna and the receiving antenna and the signal level during terrestrial broadcasting. Curve 911 indicates the reception antenna signal level when the height of the transmission antenna is 1250ft. First, it is assumed that the required error rate of the transmission system in the digital TV broadcasting system under consideration is 10 ^-1.5 . The area 912 indicates the noise level, and the point 910 indicates the received threshold of the conventional 32QAM method at the point where C / N = 15 dB. At this point of L = 60 miles, digital HDTV broadcasting can be received. However, due to deterioration of the reception conditions such as the weather, C / N fluctuates by 5 dB in time. In a reception situation where the C / N is close to the threshold, when the C / N falls, there is a problem that the reception of the HDTV is suddenly disabled. In addition, due to the influence of the terrain or the building, a change of at least 10 dB is expected, and not all points within a radius of 60 miles can be received. In this case, unlike the analogue, the digital can not transmit the video completely. Therefore, the service area of the conventional digital TV broadcasting system is uncertain.

On the other hand, in the case of 8-VSB shown in 32SRQAM or FIG. 68 of the present invention, as described above, the layers of three layers are formed by the configuration of FIG. 133 and FIG. MPEG-1 low-level NTSC signals can be sent from Layer 1-1, D _1-1 , medium-resolution TV components can be sent from Layers _1-2 , and only high-band components of HDTV can be sent from Layer ₂ D2. have. For example, in FIG. 105, the service area of the first to second layers extends to 70 miles as shown by the point 910a, and the second layer retreats to 55 miles as shown at 910b. The service area diagram of 32SRQAM in FIG. 106 indicates the difference of the area of the service area in this case. FIG. 106 is a computer simulation that calculates FIG. 53 in more detail. In FIG. 106, areas 708, 703c, 703a, 703b, and 712 are the conventional service area of 32QAM, the service area of layer 1-1 layer D _1-1 , and the layer 1-2. The service area of D _1-2, the service area of the second layer D _{2, and} the service area of the adjacent analytical station are displayed. The data of the service area of the 32QAM of the conventional method uses the data disclosed in the past.

In the conventional 32QAM, a service area of nominally 60 miles can be set. In reality, however, the reception was very unstable near the reception limit due to changes in weather or terrain conditions.

However, by using the 36SRQAM of the present invention, the low-frequency TV component of the MPEG-1 grade is transmitted from the 1-1st layer D _1-1 , and the principal TV component of the NTSC grade is transmitted from the 1-2nd layer D _1-2 , By transmitting the high frequency TV component of the HDTV in the layer D ₂ , the radius of the service area of the HDTV of the high resolution grade is reduced by 5 miles as shown in FIG. 106, but the radius of the service area of the EDTV of the medium resolution grade is increased by 10 miles or more, and the low resolution LDTV is transmitted. The service area in Fig. 1 has the effect of expanding 18 miles. Fig. 107 shows the service area when the shift factor n or s = 1.8, and Fig. 135 shows the service area of Fig. 107 by the area.

Therefore, in the conventional method, by applying the SRQAM method of the present invention even in an unreceivable area existing in an area with poor reception conditions, at least in the set service area, most receivers are used in medium or low resolution grades. A transmission capable of receiving TV broadcasts becomes possible. Therefore, the use of the present invention in the area behind the building or in the low-level inaccessible area and adjacent analytical station that occurs in a normal QAM greatly reduces the unacceptable area, thereby increasing the number of recipients. have.

Secondly, in the conventional digital TV broadcasting system, only a receiver having an expensive HDTV receiver and a receiver can be received, so that only a part of receivers can be viewed even in the service area. However, in the present invention, the receiver having a conventional TV receiver of the conventional NTSC, PAL, or SECAM system can increase the number of digital TV broadcast programs, even NTSC grade or LDTV grade, by receiving only the digital receiver. This allows the recipient to watch the program with less economic burden. At the same time, as the total number of recipients increases, the TV sender can get more viewers, which creates a social effect of more stable management as a TV business.

Third, when the area of the receiving area of the low and medium resolution grade is n = 2.5, it is enlarged by 36% compared with the conventional method. Recipients increase with magnification. As the service area expands and the number of recipients increases, the business income of TV operators increases. As a result, the business risk of digital broadcasting is expected to decrease, and the spread of digital TV broadcasting is expected to accelerate.

As can be seen from the service area diagram of 32SRQAM in FIG. 107, the same effect is obtained even when N or s = 1.8. By changing the shift value n, each station is HDTV receiver and the NTSC TV according to the specific conditions or circumstances area, such as distribution of the receiver, and change the n, SRQAM D1 and D ₂ of the service area (703a) and (703b of ) Can be set to the optimum condition so that the receiver can obtain the maximum satisfaction and the broadcaster can obtain the maximum number of recipients. in this case

When, the above effects are obtained. So for 32SRQAM n is

Becomes

Likewise for the logger 16SRQAM n is

Becomes

In this case, when n is equal to or larger than 1.0 in 16 SRQAM, 32 SRQAM, and 64 SRQAM in the SRQAM system in which the first and second layers are shifted as in Figs. 99 and 100, the effect of the present invention is obtained in terrestrial broadcasting.

In the embodiment, the case where the video signal is transmitted is described. However, the audio signal is divided into a high frequency region or a high resolution region and a low frequency region or a low resolution region, respectively, using the transmission method of the present invention as the second data sequence and the first data sequence, respectively. When transmitted, the same effect is obtained. When used in PCM broadcasting, radio and mobile phones, the service area is expanded.

Further, in the third embodiment, as shown in FIG. 133, a subchannel by TDM is formed in combination with the time division multiplexing (TDM) method, and the ECC encoder 743a and ECC encoder 743b are provided. As indicated, by differentiating the code gain of error correction of each subchannel, it is possible to increase the subchannel of hierarchical transmission by giving a difference to the threshold of each subchannel. In this case, as shown in FIG. 137, the code gains of ECC encoders such as trellis encoders of two sub-channels of VSB-ASK signals of 4VSB, 8VSB, and 16VSB may be changed. The detailed description is the same as that in the description of FIG. 131 of the sixth embodiment and will be omitted.

The block diagram of FIG. 131 is a magnetic recording / playback apparatus and the block diagram of FIG. 137 is a transmission apparatus. By replacing the Daon converter of the U _P converter receiver of the transmitter of the transmitter with the magnetic head recording signal amplifying circuit and the magnetic head reproducing signal amplifying circuit of the magnetic recording / reproducing apparatus, respectively, it can be seen that both have the same configuration. Therefore, the configuration and operation of the modulation / demodulation part are exactly the same. Similarly, it can be seen that the magnetic recording and reproducing apparatus of FIG. 84 has the same configuration as that of the transmitting apparatus of FIG. If the configuration is to be simplified, FIG. 157 may be used. If it is desired to be simplified, the configuration may be similar to that of FIG.

In the simulation of FIG. 106, there is shown a case where there is a difference of 5 dB of coding gain between 1-1 subchannel D _1-1 and 1-2 subchannel D _1-2 . SRQAM is the application of the Constellation-Code Division Multiplex of the present invention called C-CDM to rectangle-QAM. C-CDM is a multiplexing method independent of TDM or FDM. The sub-channel is obtained by dividing the signal point code corresponding to the code. By increasing the number of signal points, the scalability of transmission capacity which is not available in TDM or FDM can be obtained. This is achieved while maintaining almost complete compatibility with conventional equipment. As such, C-CDM has an excellent effect.

However, although the embodiment using a combination of C-CDM and TDM is used, the combination of the frequency division multiplexing (FDM) also produces the same threshold mitigation effect. For example, when used for TV broadcasting, it is as shown in the frequency distribution diagram of the TV signal of FIG. Conventional analog broadcasting, for example, NTSC signal has the same frequency distribution as the spectrum (725).

The largest signal is the carrier 722 of the image. Color carrier 723 or voice carrier 724 is not very large. In order to avoid mutual interference, a method of dividing a digital broadcast signal into two frequencies by FDM is considered. In this case, as shown in the figure, the first carrier 726 and the second carrier 727 are divided to avoid the carrier 722 of the image, and the first signal 720 and the second signal 721 are sent, respectively. As a result, interference can be reduced. Hierarchical broadcasting by FDM is realized by transmitting a low resolution TV signal at a large output by the first signal 720 and transmitting a high resolution signal at a small output by the second signal 721 while avoiding interference.

Here, FIG. 134 shows a diagram in the case where conventional 32QAM is used. Since the subchannel A has a larger output, the threshold threshold 1 may be 4 to 5 dB smaller than the threshold threshold 2 of the subchannel B. Therefore, two-layer hierarchical broadcasting with a difference of 4 to 5 dB threshold is realized. In this case, however, when the level of the received signal is less than or equal to Threshold 2, all of the signals indicated by the diagonal lines of the second signal 721a, which occupy a large amount of information, cannot be received at all, and the first signal 720a having a small amount of information. Since only the image can be received, the second layer can receive only an image with significantly poor image quality.

However, in the case of using the present invention, as shown in FIG. 108, first of subchannel A is added to the first signal 720 using 32SRQAM obtained by C-CDM. The low resolution component is added to 1 of the subchannel A having a low threshold value. The second signal 721 is set to 32SRQAM, and the threshold of 1 of the subchannel B is adjusted to the threshold Threshold 2 of the second signal. If the signal level drops to Threshold 2, it will not be received. The area consists only of the second signal portion 721a, which is indicated by an oblique line, and since the 1 of the subchannel B can be received, the transmission amount does not decrease very much. Therefore, there is an effect that the image having good image quality in the second layer can be received even at the signal level of Threshold 2.

By transmitting components of normal resolution on one subchannel, the number of hierarchies is further increased, and the service area of low resolution is expanded. By inserting important information such as voice information, synchronization information, header of each data, etc. into the sub-channel having a low threshold value, this important information can be received reliably, thereby enabling stable reception. If the same technique is used for the second signal 731, the hierarchy of the service area is increased. When the number of scanning lines of HDTV is 1050, 775 service areas are added by C-CDM in addition to 525.

In this way, the combination of the FDM and the C-CDM has the effect of expanding the service area. In this case, although two subchannels are arranged by FDM, three subchannels may be arranged by dividing into three frequencies.

Next, the combination of TDM and C-CDM explains how to avoid interference. As shown in FIG. 109, the analog TV signal includes a horizontal retrace unit 732 and a video signal unit 731. FIG. The signal level of the horizontal retrace unit 732 is low, and it is not output to the screen even if disturbed during this period. Synchronize the digital TV signal with the analog TV signal and send important data to the horizontal retrace slots 733 and 733a of the period of the horizontal retracer 732, for example, a synchronization signal or the like, or a large amount of data at high output. You can send This has the effect of increasing the amount of data or increasing the output without increasing disturbance. Similar effects can be obtained by forming the vertical retrace synchronous slots 737 and 737a in synchronization with the periods of the vertical retrace units 735 and 735a.

110 is a principle diagram of C-CDM. FIG. 111 shows the code allocation of C-CDM of the 16QAM extension, and FIG. 112 shows the code allocation of the 32QAM extension. As shown in Figs. 110 and 111, 256QAM is divided into four layers of first, second, third, and fourth layers 740a, 740b, 740c, and 740d, respectively, 4,16,64,256. Has segments The signal point codeword 742d of 256QAM of the fourth layer 740d is 8111 11111111. It is divided into four codewords 741a, 741b, 741c and 771d by 2 bits, and the signal point area (1) of each of the first, second, third, and fourth layers 740a, 740b, 740c, and 740d. 11, 11, 11, and 11 are assigned to 742a, 742b, 742c, and 742d, respectively. In this way, two bits of subchannels, namely, subchannel 1, subchannel 2, subchannel 3, and subchannel 4 are generated. This may be done by signal point code division multiplexing. FIG. 111 shows a specific code layout of the 16QAM extended edition, and 112 shows an extended edition of 36QAM. C-CDM multiplexing is independent. Therefore, by combining with conventional frequency division multiplexing (FDM) or time division multiplexing (TDM), there is an effect that the subchannel can be further increased. In this way, a new multiplexing method can be realized by the C-CDM method. Although C-CDM has been described using Rectangle-QAM, other modulation schemes with signal points, such as other types of QAM, PSK, ASK, and frequency domain are considered as signal points, and FSK can be multiplexed as well.

For example, the error rate of subchannel 1 of 8PS-APSK described above is

Pe _2-8 of subchannel 2

The error rate of subchannel 1 of 16-PSAPSK (PS type) is

The error rate of subchannel 2 is

The error rate of subchannel 3 is

It can be represented as.

Example 4

A fourth embodiment of the present invention will be described below with reference to the drawings.

37 is an overall system diagram of the fourth embodiment. The fourth embodiment uses the transmission apparatus described in the third embodiment for terrestrial broadcasting, and has almost the same configuration and operation. The difference from FIG. 29 described in Embodiment 3 is that the transmitting antenna 6a is an antenna for terrestrial transmission, and that each antenna 22a, 32a, 42a of each receiver is an antenna for terrestrial transmission. It is only. Other operations are exactly the same, so duplicate descriptions are omitted. Unlike satellite broadcasting, in the case of terrestrial broadcasting, the distance between the transmitting antenna 6a and the receiver becomes important. Far-field receivers have weakened propagation, so they can't demodulate at all with just multi-valued QAM-modulated signals, so they can't watch the program.

However, in the case of using the transmission apparatus of the present invention, as shown in FIG. 37, the first receiver 23 having the antenna 22a at a long distance receives the modified 64QAM modulated signal or the modified 16QAM modulated signal and demodulates the signal in the 4PSK mode to perform the first data sequence. By reproducing the D ₁ signal, an NTSC TV signal is obtained. Therefore, even if the radio wave is weak, you can watch TV programs in medium resolution.

Next, in the second receiver 33 having the antenna 32a at the intermediate distance, the reaching radio wave is sufficiently strong, so that the second data string and the first data string can be demodulated from the modified 16 or 64QAM signal, and an HDTV signal is obtained. Therefore, the same TV program can be watched on HDTV.

On the other hand, the third receiver 43, which is near or has an ultra-sensitive antenna 42a, demodulates the _first , _second , and third data strings D ₁ , D ₂ , and D ₃ because the radio wave is a sufficient strength of the modified 64QAM signal. Ultra high resolution HDTV signals are obtained. You can watch the same TV program on a super HDTV with the same quality as a big movie.

In this case, the method of arranging frequencies can be described by reading the time multiplexing by using frequency distribution in FIG. 34, 35, and 36. FIG. 34 to help NTSC L1 of the D ₁ signal if a frequency is assigned to the channel 1 to 6 to a first channel, such as, the difference information of the HDTV to M1 of the first channel of the D ₂ signal, the first signal of the D ₃ By distributing the difference information of the ultra high definition HDTV on the H1 of the channel, NTSC, HDTV and the super resolution HDTV can be transmitted on the same channel. 35 and 36, if a D ₂ signal or D ₃ signal of another channel is permitted to be used, a higher quality HDTV or an ultra-high definition HDTV can be broadcast.

As described above, three digital TV terrestrial broadcasts compatible with each other can be broadcast using one channel or another channel D ₂ and D ₃ signal region. In the case of the present invention, the TV program of the same content in the same channel can be received in a wider area if the medium resolution.

As digital terrestrial broadcasting, HDTV broadcasting of 6MHZ band using 16QAM has been proposed. However, since these methods are not compatible with NTSC, the premise of adopting a thermal broadcast method in which the same program is transmitted on different channels of NTSC is assumed. In the case of 16QAM, it is expected that the service area that can be transmitted is narrowed. The use of the present invention in terrestrial broadcasting not only eliminates the need for a separate channel, but also has the effect that the broadcast service area is wide because a remote receiver can watch a program in medium resolution.

52 is a view showing a reception disturbance area during digital terrestrial broadcasting of a conventionally proposed HDTV, which is capable of receiving HDTV from a digital broadcasting station 701 of an HDTV using the conventionally proposed method. The reception allowable area 712 of the analog broadcast station 711 adjacent to 702 is displayed. In the overlapping section 713 in which both of them overlap, at least the HDTV cannot be stably received due to the radio wave disturbance of the analog broadcasting station 711.

53 shows a reception disturbance area diagram when the hierarchical broadcasting method according to the present invention is used. In the present invention, since the power utilization efficiency is low when the transmission power is the same as that of the conventional method, the high resolution receivable area 703 of the HDTV is slightly narrower than the receivable area 702 of the conventional method. However, there is a wider range of low resolution receivable regions 704 such as digital NTSC than the conventional receivable region 702. It consists of two areas mentioned above. In this case, the radio wave interference from the digital broadcasting station 701 to the analog broadcasting station 711 is the same level as the conventional system shown in FIG.

In this case, in the present invention, there are three areas of interference from the analog broadcasting station 711 to the digital broadcasting station 701. One is the first disturbance area 705 which neither HDTV nor NTSC can receive. The second is the second disturbance area 706, which is interrupted but can receive NTSC as before the disturbance, and is indicated by single-line radiation. In this case, since NTSC uses the first data string that can be received even if C / N is low, the influence range of interference is narrow even if C / N decreases due to the interference of the radio station 711.

The third is a third obstruction area 707, which can receive HDTV before interference but only NTSC after interference, is indicated by double diagonal lines.

As described above, the reception area of the HDTV before jamming is slightly narrower than the conventional method, but the reception range including NTSC is widened. In addition, the disturbance from the analogue broadcasting station 711 enables NTSC to receive the same program as that of the HDTV even in the area where the HDTV could not be received by the interference in the conventional system. In this way, there is an effect that the unreceivable area of the program is greatly reduced. In this case, by slightly increasing the transmission power of the broadcasting station, the receivable area of the HDTV becomes equivalent to the conventional method. In the conventional system, it is possible to receive a program in the quality of NTSC TV in a remote area where a program cannot be viewed at all or in an overlapping area with an analogue station.

In addition, although the example using the two-layer transmission method is shown, the three-layer transmission method can also be used, as shown in FIG. By separating and transmitting HDTV into three levels of images of HDTV, NTSC, and low resolution NTSC, the reception range of FIG. 53 is expanded from the second floor to the third floor, and the outermost layer becomes a large area, and cannot be received at all in two-layer transmission. In the first disturbance area 705, the program can be received by the quality of the low resolution NTSC TV. The above shows an example in which a digital broadcasting station interferes with analog broadcasting.

Next, an embodiment is shown under the rule condition that digital broadcasting does not interfere with analog broadcasting. The empty channel currently under consideration in the United States uses the same channel adjacently. For this reason, digital broadcasts that broadcast later should not interfere with existing analog broadcasts. Therefore, it is necessary to lower the transmission level of digital broadcasting than when transmitting on the condition of FIG. In this case, in the case of the conventional 16QAM or 4ASK modulation, as shown in the disturbance diagram of FIG. 54, the reception area 708 of the HDTV is significantly reduced because the reception area 713 represented by double diagonal lines is large. Throw it away. Sponsor decreases because service area becomes narrow and receiver decreases by that amount. Therefore, it is expected that the broadcasting business will be difficult to establish economically in the conventional method.

Next, Fig. 55 shows a case where the broadcasting method of the present invention is used. The high resolution receivable area 703 of the HDTV is slightly narrower than the receivable area 708 of the conventional method. However, a wider range of low resolution receivable areas 704, such as NTSC, is obtained than in the conventional system. The portion indicated by singlet lines indicates an area where the same program cannot be received at the HDTV level but can be received at the NTSC level. In the first disturbance area 705, neither HDTV nor NTSC can be received due to the interference from the analog broadcasting station 711.

As described above, in the hierarchical broadcasting of the present invention, in the hierarchical broadcasting of the present invention, the reception area of HDTV quality is slightly narrowed while the area in which the same program can be received in quality of NTSC TV increases. For this reason, the service area of the broadcasting station increases. The effect is to provide the program to more recipients. The broadcasting business of HDTV / NTSC TV can be established more economically and stably. In the future, when the ratio of digital broadcasting receivers increases, the interference rule to analog broadcasting is mitigated, thereby increasing the radio wave strength. At this point, the service area of HDTV can be enlarged. In this case, by adjusting the interval between the signal points of the first data string and the second data string, the reception region of the digital HDTV / NTSC and the reception region of the digital NTSC shown in FIG. 55 can be adjusted. In this case, by transmitting the information of this interval to the first data string as described above, it can be more stably received.

56 shows the degree of disturbance when switching to future digital broadcasting. In this case, unlike FIG. 52, the neighboring station is a digital broadcasting station 701a which performs digital broadcasting. Since the transmission power can be increased, the high resolution receivable area 703 such as HDTV can be extended to the receivable area 702 equivalent to analog TV broadcasting.

In the contention area 714 of both receiveable areas, the programs are not reproduced in the quality of the HDTV by a general directional antenna, but a program of a digital broadcasting station in the direction of the reception antenna is NTSC TV. I can receive it in the elegance of. In addition, when a very high directional antenna is used, a program of a broadcasting station in the directional direction of the antenna can be received in the quality of the HDTV. The low resolution receivable area 704 is wider than the standard receivable area 702 of the analog TV broadcasting, and the contention areas 715 and 716 of the low resolution receivable areas 704a of adjacent broadcasting stations are oriented in the direction of the antenna's directivity. Program of existing broadcasting station can be reproduced with quality of NTSC TV.

However, in the full-scale dissemination of digital broadcasting in the distant future, the rule conditions are further relaxed, and the hierarchical broadcasting of the present invention enables HDTV broadcasting in a wide service area. At this point, the hierarchical broadcasting method of the present invention ensures the wide range of HDTV reception range as the same as the conventional method, and it is possible to program NTSC TV even in remote areas or contention areas that cannot be received by the conventional method. Since it is possible to receive a signal, there is an effect that the defective part of the service area is greatly reduced.

Example 5

Example 5 is an example where the present invention is used for amplitude modulation, that is, ASK.

Fig. 57 shows a signal constellation diagram of ASK signals such as the four-value VSB signal of the fifth embodiment, and has four signal points 721, 722, 723, and 724. Figs. Fig. 68 (a) shows the signal point arrangement of eight values of VSB signals. In the case of 4 values, 2-bit data In case of 8 values, 4-bit data can be sent in one cycle. In the case of 4VSB, the signal points 721, 722, 723, and 724 may correspond to, for example, 00,01,10,11.

In order to perform hierarchical transmission according to the present invention, as shown in the signal point arrangement diagram of the four-level ASK such as the four-level VSB of FIG. 58, the signal points 721 and 722 are grouped into one group, that is, the first signal point group. The signal points 723 and 724 are defined as another group, the second signal point group 726, as 725. As shown in FIG. The interval between the two signal point groups is made wider than the interval of the signal points at equal intervals. That is, if the interval between the signal points 721 and 722 is L, the interval between the signal points 723 and 724 may be similarly L, but the interval L ₀ between the signal point 722 and the signal point 723 is set larger than L. do.

L ₀ L

Is set. This is a feature of the hierarchical transmission system of the present invention. However, depending on the design of the system, L = L ₀ may be temporarily or permanently depending on conditions or settings. In the case of an eight-value VSB, the signal point arrangements as shown in Fig. 58 (a) ((b) are obtained.

As shown in (a) of FIG. 59, one bit of data of the first data string D ₁ can be associated with two signal point groups. For example, if the first signal point group 725 is defined as 0 and the second signal point group 726 is defined as 1, a signal of 1 bit of the first data string may be defined. Next, the 1-bit signal of the second data string D ₂ is associated with two signal point groups in each signal point group. For example, if signal points 721 and 723 are D ₂ = 0 and signal points 722 and 724 are D ₂ = 1 as shown in FIG. 59 (b), the data of the second data string D ₂ is used. Can be defined. This case is also 1 bit / symbol.

By arranging the signal points in this way, hierarchical transmission of the present invention is enabled by the ASK method. The hierarchical transmission system is similar to the conventional equally spaced signal point method when the signal-to-noise ratio, that is, the C / N value is sufficiently high. However, when the C / N value is low, the second data string D ₂ cannot be reproduced by using the present invention even under conditions in which the data cannot be reproduced at all in the conventional system, but the first data string D ₁ can be reproduced. To explain this, the bad C / N state can be represented by the signal point arrangement of the ASK of 4VBS of FIG. That is, the signal points reproduced by the receiver are dispersed in a Gaussian distribution in a wide range of the distributed signal point areas 721a, 722a, 723a, and 724a due to noise, transmission distortion, and the like. In such a case, it is difficult to distinguish the signal point 721 and the signal point 722 by the slice level 2, and the signal point 723 and the signal point 724 by the slice level 4 to be difficult. In other words, the error rate of the second data string D ₂ becomes very high. However, as is apparent from the drawing, it is easy to distinguish the group of signal points 721 and 722 from the group of signal points 723 and 724. That is, the first signal point group 725 and the second signal point group 726 can be distinguished from each other. For this reason, the first data string D ₁ can be reproduced at a low error rate.

In this way, data streams D ₁ and D ₂ of two layers can be transmitted and received. Therefore, hierarchical transmission in which both the first data string D ₁ and the second data string D ₂ are in a bad C / N state and in the region where only the first data string D ₁ is reproduced in a state and region where the C / N is good. The effect is that you can.

61 is a block diagram of the transmitter 741. The input unit 742 includes a first data input unit 743 and a second data input unit 744. The carrier wave from the carrier generator 64 is amplitude-modulated in the multiplier 746 by an input signal obtained by combining the signal from the input unit 742 with the processing unit 745, and is equal to or larger than 4 in Fig. 62 (a). Value is the ASK signal. The 4ASK or 8ASK signal is also band limited by the filter 747 and becomes an ASK signal such as a Vestigial Side Band having a sideband in which carrier remains slightly as shown in FIG. Is output from

Here, the output waveform after passing through the filter will be described. 62 (a) is a frequency distribution diagram of an ASK modulated signal. As shown in the figure, there are shaft bands on both sides of the carrier. This signal is removed from the filter 747, i.e., the bandpass filter, with one carrier component leaving some carrier components as shown in the transmission signal 749 of FIG. 62 (b). This is called a VSB signal, and if fo is a modulation frequency band, it is known that the frequency utilization efficiency is good because it can be transmitted in a frequency band of about fo / 2. The ASK signal of FIG. 60 is originally 2 bits / symbols, but using the VSB method, 4VSB and 8VSB can transmit 16QAM, 32QAM 4bits / symbols and 5bits / symbols in the same frequency band.

Next, in the VSB receiver 751 shown in the block diagram of FIG. 63, the signal received from the terrestrial antenna 32a passes through the input unit 752, and the signal from the variable oscillator 754, which is varied by channel selection, and the mixer. In 753, they are mixed and converted to a low intermediate frequency. Next, the detector 755 detects the baseband signal by the LPF 756, and the identification player 757 has a slicer of 4 levels for 4VSB and an slicer of 8 levels for 8VSB. The data string D ₁ and the second data string D ₂ are reproduced and output from the first data string output section 758 and the second data string output section 759.

Next, the case where the TV signal is sent using the transmitter and the receiver will be described. 64 is a block diagram of the video signal transmitter 774. A high-definition TV signal such as an HDTV signal is input to the input unit 403 of the first image encoder 401, and is separated by the video separation circuit 404 such as a subband filter._LV_L, H_LV_H, H_HV_L, H_HV_HIt is separated into a high frequency TV signal and a low frequency TV signal. Since this content was explained using FIG. 30 in Example 3, detailed description is abbreviate | omitted. The separated TV signal is encoded by the compression unit 405 using a technique such as DPCMDCT variable length encoding used in MPEG or the like. Motion compensation is processed in the input unit 403. Compressed four image data are combined by the combining unit 771 to the first data string D. FIG._OneAnd second data string D₂Two data strings. In this case H_LV_L That is, the low frequency image signal is included in the first data string. The signal is inputted to the first data input unit 743 and the second data string input unit 744 of the transmitter 741 to receive amplitude modulation, and becomes an ASK signal such as VSB, and is broadcast from the ground antenna.

65 is a block diagram of the entire TV receiver of the digital TV broadcast. The broadcast signal of 4VSB or 8VSB received from the terrestrial antenna 32a is input to the input unit 752 of the receiver 751 in the TV receiver 781, and the arbitrary channel desired by the receiver by the VSB detection demodulation unit 760. Signal is tuned and demodulated, and the first data string D ₁ and the second data string D ₂ are reproduced and output from the first data string output section 758 and the second data string output section 759. Detailed descriptions will be omitted since they are duplicated. The D ₁ and D ₂ signals are input to the separating unit 776. The D ₁ signal is separated by the separator 777 and the H _L V _L compressed component is input to the first input unit 521. The other side is synthesized with the D ₂ signal by the synthesizer 778 and input to the second input unit 530. In the second image decoder, the H _L V _L compressed signal entering the first input unit 521 is extended by the first extension unit 523 into an H _L V _L signal, and the aspect ratio of the image synthesis unit 548 and the aspect ratio is increased. It is sent to the change circuit 779. When the original TV signal is an HDTV signal, the H _L V _L signal becomes a wide NTSC signal, and when the original signal is an NTSC signal, a low resolution TV signal having a lower quality than an NTSC such as MPEG1 is obtained.

In this explanation, since the original video signal is set as an HDTV signal, the H _L V _L signal is a wide NTSC TV signal. If the aspect ratio of the TV is 16: 9, the TV is output as the video output 426 via the output unit 780 with the aspect ratio of 16: 9. If the aspect ratio of the TV is 4: 3, the aspect ratio changing circuit 779 changes the aspect ratio of the aspect ratio from 16: 9 to a letter box or side panel format of the 16: 3 aspect ratio and interposes the output unit 780. And output as the video output 425.

On the other hand, the second data string D ₂ from the second data string output section 759 is synthesized with the signal of the separator 777 in the synthesizer 778 of the separator 776, so that the second image decoder It is input to the second input unit 530, separated by the compression signal of H _L V _H , H _H V _L , H _H V _H by the separating circuit 531, respectively, the second extension part 535, the third extension part 536, it is sent to the fourth extension portion 537, and is extended to become the original H _L V _H , H _H V _L , and H _H V _H signals. H _L V _L signals are added to these signals, input to the image synthesis unit 548, synthesized into one HDTV signal, and output from the output unit 546, and through the output unit 780, a video signal of the HDTV. It is output as 427.

The output unit 780 detects an error rate of the second data string of the second data string output unit 759 in the error rate detection unit 782, and automatically generates a predetermined time when the error rate is high for a predetermined time. A low-resolution video signal of the _L V _L signal is output.

In this manner, the hierarchical broadcast can be transmitted and received. If the transmission conditions are good, for example, for a broadcast having a close TV transmission antenna, both the first data string and the second data string can be reproduced, so that the program can be received in the quality of the HDTV. Also, for broadcasts far from the transmission antenna, the first data string is reproduced and a low resolution TV signal is output from the H _L V _L signal. As a result, the same program can be received in a wider area by the quality of HDTV or NTSC TV.

Also, as shown in the block diagram of the TV receiver of FIG. 66, if the function of the receiver 751 is reduced by only the first data string output unit 758, the receiver does not have to handle the second data string and the HDTV signal. do. As the image decoder, the first image decoder 421 described in FIG. 31 may be used. In this case, an image of the quality of an NTSC TV is obtained. In HDTV quality, the program cannot be received, but the cost of the receiver is significantly lower. Therefore, there is a possibility of wide spread. This system has the effect of being able to receive digital TV broadcasts by adding them as adapters without changing many receiving systems with conventional TV displays.

In addition, as shown in FIG. 66, when receiving scrambled 4VSB and 8VSB signals, the scramble release signal transmitted as 4VSB and 8VSB signals and the number of the descrambling number memory 502c in the descrambler 502 are given. The descrambling number combiner 502b inquires and releases the descramble only when it matches, so that the descrambling of the specific scramble program can be appropriately released.

With the configuration shown in FIG. 67, a receiver having the functions of the satellite broadcast receiver for demodulating the PSK signal and the terrestrial broadcast receiver for demodulating the VSB signal can be simply configured. In this case, the PSK signal received from the satellite antenna 32 is mixed in the mixer 786 with the signal from the oscillator 787, is converted to a low frequency and input to the input unit 34 of the TV receiver 781, It is input to the mixer 753 described in FIG. The PSK or QAM signal converted to the low frequency of the specific channel of the satellite TV broadcast is demodulated by the demodulator 35, and the data sequence D ₁ and D ₂ are decoded by the second image encoder 422 via the separation unit 789. Is reproduced as an image signal and output from the output unit 780. On the other hand, the digital terrestrial broadcast and the analog broadcast received by the ground antenna 32a are input to the input unit 752, and a specific channel is selected and detected by the mixer 753 in the same process as described with reference to FIG. This results in a baseband signal with only low frequencies. The demodulator enters the mixer 953 in the analog satellite TV broadcast. In the case of digital broadcasting, the data streams D ₁ and D ₂ are reproduced by the identification player 757, and a video signal is reproduced by the second image decoder 422 and output. In addition, in the case of receiving the analog TV broadcast of the ground and the satellite, the analog TV signal AM-demodulated by the image demodulation unit 788 is outputted from the output unit 780. With the configuration of FIG. 67, the mixer 753 can be shared by satellite broadcasting and terrestrial broadcasting. The second image decoder 422 can also be shared. When the ASK signal is used in digital terrestrial broadcasting, a receiver 755 and an LPF 756 similar to the conventional analog broadcasting can be used for AM demodulation. As described above, the configuration shown in Fig. 67 can greatly reduce the number of circuits by sharing the receiving circuit.

In the embodiment, the four-valued ASK signal was divided into two groups, and hierarchical transmission of each one bit of two layers of D ₁ and D ₂ was performed. However, when eight values of the ASK signal, i.e., 8-level VSB, shown in the signal point arrangement diagram of the 8VSB signal in Fig. 68 (a) and (b) are used, the total of 1 bit of each of the three layers of D ₁ , D ₂ , and D ₃ is Hierarchical transmission of bits / symbols can be performed. As shown in FIG. 68 (a), when the first bit encoding is first described, the signal points of the D ₃ signal are the signal points 721a and 721b, 722a and 722b, 723a and (723a). 723b), 724a and 724b are binary, i.e., 1 bit. Next, when encoding of the next 1 bit is described, the signal points of D ₂ are binary 1 bits of the signal point groups 721 and 722 and the signal point groups 723 and 724. The data of D ₃ is a binary one bit of the large signal point group 725 and 726. In this case, the four signal points 721, 722, 723, and 724 of FIG. 57 are divided into two signal points 721a, 721b, 722a, 722b, and 723a, respectively. By separating 723b, 724a and 724b, and reducing the distance between groups, hierarchical transmission of up to three layers is possible.

The video transmission of digital HDTVs such as the third floor using the three-layer hierarchical transmission system has been described in the third and fourth embodiments, and detailed descriptions of the operations are omitted.

Here, the effect of performing TV broadcasting by 8VSB in FIG. 68 will be described. While 8VSB has a large amount of transmission information, the error rate for the same C / N value is higher than 4VSB. However, when high-definition HDTV broadcasting is carried out, there is a large amount of error correction code because there is room in the transmission capacity, so that the error rate can be lowered, or hierarchical TV broadcasting can be performed in the future.

Here, the effects of 4VSB, 8VSB, and 16VSB will be described with comparison.

In the case of terrestrial broadcasting using the frequency band of NTSC or PAL, as shown in FIG. 136, in case of NTSC, there is a 6MHZ band limitation and a practical transmission band of about 5MHZ is allowed. In the case of 4VSB, since the frequency utilization efficiency is 4 bits / HZ, the data transmission capacity is substantially 5MHZ x 4 = 20Mbps. On the other hand, at least 15 Mbps to 18 Mbps are required for transmission of the digital HDTV signal. For this reason, in 4-VSB, since there is no room for data capacity, as shown in the comparison diagram of FIG. 169, redundancy for error correction and correction can only take 10 to 20% of the actual transmission amount of the HDTV.

Next, in the case of 8-VSB, since the frequency utilization efficiency is 6 bits / HZ, a data transmission capacity of 5 MHZ x 6 = 30 Mbps is obtained. As described above, the transmission of the HDTV signal requires 15 to 18 MHz, but in the case of the 8VSB modulation system, as shown in FIG. 169, an information amount of 50% or more of the actual transmission amount of the HDTV signal can be used for the error correction code. Therefore, under the condition of terrestrial broadcasting HDTV digital signals having the same data rate in the 6MHZ band, since 8VSB can add a larger error correction code, error curves 805 and 806 of FIG. 164 are provided. As shown in the figure, for the same C / N value of the transmission system, the TCM-8VSB having a higher error correction correction gain lowers the error rate after error correction than 4 VSB which has a lower correction error correction correction gain. Therefore, the 8VSB which is error coded with high-wood gain has the effect that the service area in TV terrestrial broadcasting is expanded than 4VSB. Clearly, 8VSB has a drawback that the circuit of the receiver becomes more complicated due to the increase in the error correction circuit. However, since the VSB and ASK methods are amplitude modulation methods, the circuit size of the original equalizer of the receiver is significantly smaller than that of the QAM modulation method including phase components. Therefore, even if an error correction circuit is added, the overall circuit size is not larger than that of the 32QAM system in the 8VSB system. Therefore, the 8VSB method realizes a digital HDTV receiver having a wide service area and appropriate circuit size.

In addition, examples of the specific error correction method are described in the fifth embodiment and the like, but the ECC 744a of the transceiver diagrams of the 131, 137, 156, and 157 of FIG. The trellis encoder 744b is used for transmission using the VSB modulator 750 of 4VSB, 8VSB, and 16VSB described in FIG. On the receiver side, digital data is reproduced by the level slicer 757 of 4, 8, 16 values from the 4VSB or 16VSB signal using the VSB demodulator 760 described with reference to FIG. After the error correction is performed by the trellis decoder 759b and the ECC decoder 759a of FIG. 84, Embodiment 131, 137, 156, and 157, which are described in FIGS. A digital HDTV signal is reproduced and output by the image expander of the image decoder 402.

The ECC encoder 744a uses the Reed Solomon encoder 744j and the interleaver 744k as shown in FIG. 160 (a) (b) described in Example 6, and the ECC decoder. A deinterleaver 759k and a Reed Solomon coder 759j are used for 759a. As described in the foregoing embodiment, interleaving makes the burst error resistant.

By adopting the trellis encoder shown in FIG. 128 (a) (b) (c) (d) (e) (f), the gain of the code can be further increased, and the error rate is lowered. In the case of 8 VSB, trellis encoder 744b and decoder 759b having a ratio of 2/3 can be used as shown in FIG.

In the embodiment, description has been made mainly using an example of transmitting a hierarchical digital TV signal. In the case of the hierarchical type, an ideal broadcast can be performed. However, since the image compression circuit and the circuit of the demodulator are complicated, it is not preferable in terms of cost at the start of the broadcast. As described earlier in Embodiment 5, non-hierarchical TV transmission is performed at signal point intervals L = L _0, i.e., at _equal intervals of 4VSB or 8VSB, with a simple configuration as shown in FIG. Thus, a simple TV broadcasting system of a circuit is realized. Then, it can be switched to hierarchical transmission of 8VSB at the stage of dissemination.

By the way, although 4VSB and 8VSB were demonstrated above, 16VSB and 32VSB are demonstrated in FIG.159 (a)-(d). FIG. 159 (a) shows a signal point arrangement diagram of 16 VSB. 159 (b) is grouped into groups 722a to 722h of two signal points and regarded as eight signal points, so that it can be treated as an 8VSB, so hierarchical transmission of two layers is realized. In this case, even if an 8VSB signal is intermittently sent by the time division multiplexing scheme, hierarchical transmission is realized. In this method, however, the maximum data rate is 2/3. In FIG. 159 (c), there are four groups 723a to 723d, and since they are treated as 4VSBs, the hierarchical layer is increased. In this case, even if the 4VSB signal is intermittently sent by the time division multiplexing method, the maximum data rate is lowered, but hierarchical transmission is realized. As described above, three-layer hierarchical VSB is realized.

In this manner, hierarchical transmission is realized in which data of 8VSB or 4VSB can be reproduced when C / N of 16VSB is deteriorated. Also, as shown in FIG. 159 (d), the 32VSB can be transmitted by doubling the signal point of 16VSB. If the capacity of 16VSB is to be expanded in the future, this method has an effect that a data capacity of 5 bits / symbol can be obtained while maintaining compatibility.

In summary, the configuration shown in the block diagram of the VSB receiver of FIG. 161 and the transmitter shown in the block diagram of the VSB transmitter of FIG.

Although mainly described using 4-VSB and 8-VSB, it is also possible to transmit using 16VSB as shown in FIG. 159 (a) (b) (c). In the case of 16VSB, in the case of terrestrial broadcasting, a transmission capacity of 40 Mbps is taken in the 6 MHZ band. However, since the data rate of the HDTV digital compressed signal is 15 to 18 Mbps when using the MPEG standard, the margin of transmission capacity becomes excessively large. As shown in FIG. 169, the redundancy is R ₁₆ = 100% or more, and the redundancy is too large for transmitting one channel of a digital HDTV, and the circuit is complicated, and the effect is low for 8VSB. And 16VSB for terrestrial broadcasting of 2 channel HDTV, the redundancy is the same as 4VSB and can only take about 10b. As described above, in 4-VSB, redundancy: R ₄ = 10 to 20%, and sufficient error correction cannot be made, so that the service area cannot be widened. As is apparent from FIG. 169, a redundancy degree of ₈ VSB: R ₈ = 50% is not sufficient for error correction. The circuit size of error correction is not so large that it can take the service area. Therefore, under the condition that digital HDTV terrestrial broadcasting is performed under the band limit of 6 to 8 MHz, as can be seen from FIG. 169, it can be seen that 8-level VSB is the most effective and most suitable VSB modulation method.

In the third embodiment, the image encoder 401 shown in FIG. 30 is described. However, the block diagram of FIG. 30 can be rewritten as in FIG. The content is exactly the same, so description is omitted. In this manner, the image encoder 401 has two separation circuits 404 and 404a of an image such as a subband filler. If they are referred to as the separating section 794, the circuit can be reduced by passing a signal twice through time division through one division circuit as shown in the block diagram of the separation section in FIG. To explain this, in the first cycle, the video signal of the HDTV or the super HDTV from the input unit 403 is compressed by the time base compression circuit 795, and the time axis is compressed so that the separation circuit 404 causes the H _H V _H -H, H _It is divided into four components, _H V _L -H, H _L V _H -H, and H _L V _L -H. In this case, the switches 765, 765a, 765b, and 765c are in one position, and the compression unit 405 has H _H V _H -H, H _H V _L -H, H _L V _H -H Outputs three signals. However, the signal of H _L V _L -H is input from the output 1 of the switch 765c to the input 2 of the time base compression circuit 795, and is sent to the separation circuit 404 at the second cycle, that is, at the time of time division processing, to separate the processing. The output is divided into four components, H _H V _H , H _H V _L , H _L V _H and H _L V _L. In the second cycle, the switches 765, 765a, 765b, and 765c change to the position of the output 2, so that four components are sent to the compression unit 405. In this manner, the separation circuit can be reduced by taking the configuration of FIG. 70 and time-division processing.

Next, when such three-layer hierarchical image transmission is performed, the receiver requires an image decoder as described in the block diagram of FIG. In other words, this is the same block diagram as in FIG. Although the processing capacity is different, two synthesizers 556 of the same configuration exist.

This configuration can be realized with one synthesizer in the same manner as the separation circuit of FIG. Referring to FIG. 72, five switches 765a, 765b, 765c, and 765d first switch the inputs of the switches 765, 765a, 765b, and 765c to 1 at timing 1. do. Then, from the first extension portion 522, the second extension portion 522a, the third extension portion 522b and the fourth extension portion 522c, respectively, H _L V _L , H _L V _H , H _H V _L , H The signal of _H V _H is input to a corresponding input portion of the synthesizer 556 via a switch, and is synthesized to form one video signal. This video signal is sent to switch 765d, outputted from output 1, and sent back to input 2 of switch 765c. This video signal is originally a signal of the H _L V _L -H component obtained by dividing a high resolution video signal. At the next timing 2, the switches 765, 765a, 765b and 765c are switched to input 2. In this way, the H _H V _H -H, H _H V _L -H, H _L V _H -H and H _L V _L -H signals are sent to the synthesizer 556 and synthesized to obtain one video signal. Lose. This video signal is output from the output unit 554 from the output 2 of the switch 765d.

In this manner, when the hierarchical broadcasting of the third layer is received, there is an effect of reducing the two synthesized recordings by time division processing.

However, this method first inputs the H _H V _H , H _H V _L , H _L V _H , and H _L V _L signals at timing 1, and synthesizes the H _L V _L -H signals. Then, at timing 2 different from timing 1, H _H V _H -H, H _H V _L -H, H _L V _H -H and the above H _L V _L -H signals are inputted, and the final video signal You are taking the same order as you get. Therefore, it is necessary to shift the timing of two groups of signals.

If the timing of the components of the original input signal is different or overlaps, a memory is provided in the switches 765, 765a, 765b, and 765c so as to be separated in time. It will be necessary to make adjustments. However, by transmitting the transmission signal of the transmitter separately in timing 1 and timing 2 as shown in FIG. 73, the time axis adjustment circuit is unnecessary on the receiver side. Therefore, there is an effect that the configuration of the receiver is simplified.

In FIG. 73, the first data string D ₁ of the transmission signal of D _{1 in} the time alignment diagram of FIG. 73 is displayed, and H _L V _L , H _L V _H , H _H V _L , H _H V _H in the D ₁ channel during the timing 1 period. The signal is timed, and the time alignment of the signal in the case of sending H _L V _H -H, H _H V _L -H, and H _H V _H -H in the D ₂ channel in the period ₂ is shown. In this way, by sending the transmission signals separated in time, there is an effect of eliminating the circuit configuration of the encoder of the receiver.

Next, the number of extension portions of the receiver is large. The method of reducing these numbers is demonstrated.

FIG. 74 (b) shows a time plot of data 810, 810a, 810b and 810c of the transmission signal. In this figure, different data 811 (811a) (811b) (811c) is transmitted between data. The target transmission data is then sent intermittently. Then, the second image decoder 422 shown in the block diagram of FIG. 74 (a) sequentially rotates the data string D ₁ to the decompression unit 503 via the first input unit 521 and the switch 812. Enter it. For example, after completion of input of the data 810, decompression processing is performed during the time of other data 811, and after completion of the processing of the data 810, the next data 810a is inputted. In this way, one expansion unit 503 can be shared by time and division by the same method as in the case of the synthesizer. In this way, the total number of kidney parts can be reduced.

75 is a time arrangement diagram when transmitting an HDTV. For example, if the H _L V _L signal corresponding to the NTSC component of the first channel of the broadcast program is H _L V _L (1), it is time-arranged at the position of data 821 represented by the thick line of the D ₁ signal. do. The H _L V _H , H _H V _L , and H _H V _H signals corresponding to the HDTV additional components of the first channel are arranged at the positions of the data 821a, 821b and 821c of the D ₂ signal. As a result, other data 822, 822a, 822b, and 822c, which are information of other TV programs, exist between all data of the first channel, so that the decompression processing of the decompression unit can be performed during this period. In this way, all the components can be processed in one extension part. This method can be applied when the processing of the stretcher is fast.

Similar effects can be obtained by arranging data 821, 821a, 821b and 821c in the D ₁ signal as shown in FIG. This is valid when transmitting / receiving using a layerless transmission such as 4PSK or 4ASK.

FIG. 77 shows a temporal arrangement diagram in the case where hierarchical broadcasting is performed using two-layer hierarchical transmission system, for example, three-layer video such as NTSC, HDTV, high-resolution HDTV, or low-resolution NTSC, NTSC, and HDTV. For example, in the case of broadcasting low-resolution NTSC, three-layer video of NTSC and HDTV, an H _L V _L signal corresponding to a low resolution NTSC signal is arranged in the data 821 in the D ₁ signal. In addition, signals of the respective components of the separated signals H _L V _H , H _H V _L , and H _H V _H are arranged at the positions of the data 821a, 821b, and 821c. The H _L V _H -H, H _H V _L -H, and H _H V _H -H signals that are separate signals of the HDTV are arranged in the data 823 and 823a and 823b.

Here, as shown in the block diagrams of FIG. 156 and 170, logical hierarchical transmission by differentiating the error correction capability described in Embodiment 2 is added to 4VSB and 8VSB. Specifically, H _L V _L uses the D _1-1 channel of the signal layer. As described in the second embodiment, the D _1-1 channel employs a miscorrection correction method having significantly higher correction capability than the D _1-2 channel. Since the D _1-1 channel has a higher redundancy compared to the D _1-2 channel but a low error rate after reproduction, the D _1-1 channel can be reproduced even under a condition where the C / N value is lower than that of the other data 821a, 821b and 821c. For this reason, the program can be reproduced in a low-resolution NTSC TV even in a case where reception conditions such as an area far from the antenna or in a vehicle of the car are bad. As described in Embodiment 2, in view of the error rate, the data 821 in the D _1-1 channel of the D ₁ signal is higher than the other data 821a, 821b, 821c in the D _1-2 channel. It is resistant to reception and differentiated, so the logical layer is different. As described in the second embodiment, the hierarchies of D ₁ and D ₂ may be referred to as physical hierarchies, and the hierarchical structure by differentiating the distance between error correction and correction codes may be said to be a logical hierarchical structure.

However, demodulation of the D ₂ signal requires a higher C / N value than the D ₁ signal. Therefore, in the reception condition where the C / N value of the remote location is the lowest, the H _L V _L signal, that is, the low resolution NTSC signal, is reproduced. In addition, in the reception condition where the C / N value is the next lowest, H _L V _H , H _H V _L , and H _H V _H are additionally reproduced and NTSC signals can be reproduced. In the HDTV signal it is reproduced since the C / N value is the high reception condition added to the H _L V _L by reproducing with _{H L V H -H, H H} V L -H, H H V H -H. In this way, three layers of broadcasting can be known. By using this method, the receiveable area described in FIG. 53 is enlarged from the second floor to the third floor as shown in the receive disturbance area diagram in FIG. 90, and the program receiveable area is further expanded.

FIG. 78 shows a block diagram of the third image decoder in the case of the time alignment of FIG. In terms of time, the configuration of the block diagram of FIG. 74 (a) is added to the configuration in which the third input unit 551 of the D ₃ signal is omitted from the block diagram of FIG.

Referring to the operation, the D ₁ signal is input from the input unit 521 and the D ₂ signal is input from the input unit 530 at the timing 1. Since each component such as H _L V _H is separated in time, these components are sequentially sent independently to the expansion unit 503 by the switch 812. This procedure will be explained using the timing chart of FIG. First, the compressed signal of H _L V _H of the first channel enters the decompression unit 503 and is decompressed. Next, the H _L V _H, H _H V _L, H _H V _H of the first channel is being processed height, via a switch (812a) is input to a predetermined input of the synthesizer 556 and the synthesis process, first, H _L The V _L -H signal is synthesized. This signal is input from the output 1 of the switch 765a to the input 2 of the switch 765 and to the H _L V _L input of the synthesizer 556.

Next, at timing 2, the H _L V _H -H, H _H V _L -H, and H _H V _H -H signals of the D ₂ signal are inputted as shown in the timing chart of FIG. Is expanded, the signals are input to a predetermined input of the synthesizer 556 via a switch 812a, and are synthesized to output an HDTV signal. The HDTV signal is output from the output 2 of the switch 765a via the output unit 554. As described above, transmission by the time arrangement of FIG. 77 has the effect of greatly reducing the number of the extension section and the synthesizer of the receiver. In addition, although the two time stages of the D ₁ and D ₂ signals are used in the timing diagram of FIG. 77, the TV broadcast of four layers can be performed by adding a high-definition HDTV using the above-described D ₃ signal.

FIG. 79 is a timing diagram of hierarchical broadcasting in which three layers of images are broadcast using three physical layers of D ₁ , D ₂ , and D ₃ . As is apparent from the figure, each component of the same TV channel is arranged so as not to overlap in time. 80 shows a receiver in which a third input unit 521a is added to the receiver described in the block diagram of FIG. By broadcasting by the time arrangement of FIG. 79, there is an effect that the receiver can be configured with a simple configuration as shown in the block diagram of FIG.

The operation is substantially the same as the timing diagram of FIG. 77 and the block diagram of FIG. For this reason, description is abbreviate | omitted. In addition, as shown in the temporal arrangement diagram of FIG. 81, all signals may be time-multiplexed to the D ₁ signal. In this case, the two data of the data 821 and the other data 822 increase the error correction capability as compared with the data 821a, 821b and 821c. For this reason, the hierarchy is higher than other data. As described above, it is physically one layer but logically two layers are layered. In addition, other data of another program channel 2 is inserted between the data of program channel 1. This enables serial processing on the receiver side, and the same effect as that of FIG. 79 is obtained.

In the case of the temporal arrangement diagram of FIG. 81, although it is a logical hierarchy, the error rate at the time of transmission of this data is reduced by reducing the transmission bit rate of data 821 and other data 822 to 1/2 or 1/3. As you go down, you can do physical layer transfers. In this case, the physical layer is three layers.

FIG. 82 is a block diagram of the image decoder 423 in the case of transmitting only the data string D ₁ signal, such as the time arrangement diagram in FIG. 81, and is simpler than the image decoder shown in the block diagram in FIG. It becomes a composition. Since the operation is the same as that of the image decoder described with reference to Fig. 80, the description is omitted.

As described above, when the transmission signal such as the time arrangement diagram in FIG. 81 is transmitted, the number of the expansion unit 503 and the synthesizer 556 can be greatly reduced as in the block diagram in FIG. Since the four components are input separately in time, several blocks are time-divided by a connection change depending on the image component which inputs the circuit block inside the synthesizer 556, i.e., the image combining unit 548 of FIG. The circuit can also be omitted by common use.

As described above, the receiver can be configured with a simple configuration.

In the fifth embodiment, the operation is described using ASK modulation, but most of the techniques described in the fifth embodiment can be used for the PSK or QAM modulation described in the first, second, and third embodiments.

The above embodiments can also be used for FSK modulation.

For example, when performing multi-valued FSK modulation of f1, f2, f3, and f4 as shown in FIG. 83, grouping is performed like the signal point arrangement in FIG. 58 of Example 5, and the signal point positions of each group are dropped. Hierarchical transmission is possible.

In FIG. 83, the frequency group 841 at frequencies f1 and f2 is defined as D ₁ = 0, and the frequency group 842 at frequencies f3 and f4 is defined as D ₁ = 1. If f1 and f3 are defined as D ₂ = 0, f2 and f4 as D ₂ = 1, as shown in the drawing, hierarchical transmission of 1 bit of D ₁ and D ₂ and a total of 2 bits is possible. do. For example, when C / N is high, D ₁ = 0 and D ₂ = 1 can be reproduced at t = t 3, and D ₁ = 1 and D ₂ = 0 can be reproduced at t = t ₄ . Next, when C / N is low, only D ₁ = 0 at t = t3, and only D ₁ = 1 at t = t4. In this way, hierarchical transmission of FSK can be performed. The hierarchical transmission method of this FSK may be used for hierarchical broadcasting of video signals described in the third, fourth and fifth embodiments.

The fifth embodiment of the present invention can also be used for the magnetic recording / playback apparatus shown in the block diagram as shown in FIG. Since the fifth embodiment is ASK, magnetic recording and reproduction can be performed.

84 shows a block diagram of a recording device / transmitter and a playback device / receiver.

In the block diagram of FIG. 84, the VSB-ASK modulation scheme of Embodiment 5 of the transmitter 1 and the receiver 43 replaces the transmitting circuit 5 of the transmitter 1 with the recording device magnetic recording signal amplifier 857a. Subsequently, the reception circuit 24 of the receiver 43 is replaced with the magnetic reproduction signal amplifier 857b to obtain a completely identical configuration. In the main text, since all ASK signals are VSB-ASK in the transmission apparatus, the VSB-ASK signal is abbreviated as an ASK signal.

Referring to FIG. 84, the HDTV signal is compressed in the picture encoder 401 and then divided into two pieces of data. The first data string is error coded in the ECC encoder 743a and the second data string. Is error coded in the ECC 744a, then trellis encoded by the trellis encoder 744b, and enters the modulator 750 of the VSB-ASK. In the case of the recording apparatus, the offset generator 856 applies an offset signal and then writes on the magnetic tape 855 by the recording circuit 853. In the case of the transmitter 1 of the transmission device, the offset generator 856 transmits the DC offset voltage to the ASK signal by the up-converter 5a. By DC offset, carrier reproduction in the receiver 43 becomes easy. The transmitted VSB-ASK signals of 4VSB, 8VSB, and 16VSB are received by the antenna 32b and input to the demodulator 852b via the receiving circuit 24.

On the other hand, the signal recorded by the recording apparatus is reproduced by the reproduction head 854a and sent to the demodulator 852b via the reproduction circuit 858 in the same manner.

The signal input to the demodulator 852b is demodulated by the ASK demodulator 852b such as VSB described above via the filter 858a of the demodulator 852b. The first data sequence of the demodulation signal is miscorrected by the ECC decoder 785a, and the second data sequence is miscorrected by the trellis decoder 759b. The image decoder 402 extends the video signal and outputs an HDTV, TV signal or SDTV signal.

With the addition of trellis encoder, the circuit becomes complicated, but the error rate is lowered, the transmission distance of the transmission device is extended, and the image quality of the recording / reproducing device is improved. In this case, the filter 858a of the receiver 43 uses a comb-type filter 760a having a filter characteristic excluding the main carrier, the video carrier or the audio carrier of the analog TV signal as shown in FIG. By doing so, the disturbance of the analog TV signal can be eliminated, and the error rate is lowered. In this case, the reception signal deteriorates if the filter is always placed even when there is no interference. To avoid this, as shown in FIG. 65, the analog TV filter 760a is turned ON only when the signal is degraded due to the disturbance of the analog TV by the error rate detection unit 782, and the OFF signal is turned off when there is no interference. The signal degradation caused by the filter can be prevented.

In the case of Fig. 84, the second data string among the first data string and the second data string has a lower error rate. Therefore, by transmitting / recording important high priority (HP) information such as the descramble information of FIG. 66 or the header information of the image data of each image block in the second data string, it is possible to stabilize the descramble or image signal reproduction of each image block. Can be.

Also, as shown in Figs. 137 and 172, the E of each time-divided subchannel in the 8VSB or 16VSB transmission apparatus is used, and the code gain of the error correction of the trellis decoder or ECC decoder. Change in the subchannel, and send high rank (HP) information in the subchannel with the higher gain. Since the error rate of the HP information is lowered, noise is generated to some extent in the transmission path, the signal is degraded, and even if the low priority (LP) information is destroyed, the data of the HP information is not destroyed. By transmitting the above-described descrambling information or header information such as the address of the data packet in image block units as the HP information, the scramble release is stable for a long time, and the viewer can watch the scrambled program stably. In addition, since destructive destruction of each image block is prevented, the entire image quality deteriorates even when the reception signal deteriorates, and the viewer can watch TV programs with a certain quality.

Example 6

In the sixth embodiment, an example in which the transfer and recording method of the present invention is applied to a magnetic recording / playback apparatus will be described. In Example 5, the present invention is applied to an ASK transmission method for multi-value transmission. However, the present invention can also be applied to a multi-value ASK recording type magnetic recording and reproducing apparatus as shown in the block diagram of FIG. . In addition to ASK, by applying the C-CDM method of the present invention to PSK, FCK, and QAM, hierarchical or non-hierarchical multi-value magnetic recording is possible.

First, a layering method will be described using an example in which the C-CDM method of the present invention is applied to a 16QAM or 32QAM magnetic recording / playback apparatus. 84 shows a block diagram when C-CDM is applied to 16QAM, 32QAM, 4ASK, 8ASK, 16ASK, and 8PSK. Hereinafter, QAM C-CDM multiplexing is referred to as SRQAM. 137 and 154 show block diagrams in the case of application to a transmission system such as a broadcast.

Referring to FIG. 84, the magnetic recording / reproducing apparatus 851 is configured to transfer input video signals such as HDTVs to the first image encoder 401a and the second image encoder 401b of the image encoder 401. The high frequency signal and the low frequency signal are separated and compressed, and a low frequency video signal such as an H _L V _L component is supplied to the first data string input unit 743 of the input unit 742, and the H data is transmitted to the second data string input unit 744. A high pass video signal including an _H V _H component and the like is input to the modulator 750 in the modulator 852. In the first data string input unit 743, an error correction code is added to the low-pass signal in the ECC unit 743a. Meanwhile, in the case of 16SRQAM, 32SRQAM, and 64SRQAM, the second data string input to the second data string input unit 744 is trellis encoder 744b shown in FIG. 128 (a) (b) (c). It is trellis coded at a ratio of 1/2, 2/3, and 3/4, respectively. For example, in the case of 64SRQAM, the first data string is 2 bits and the second data string is 4 bits. For this reason, 3-bit data was made into 4 bits using the trellis encoder 744b as shown in FIG. 128 (c). A trellis encoder with a ratio of 3/4 is performed. In the case of 4ASK, 8ASK and 16ASK, trellis codes of 1/2, 2/3 and 3/4 are used. In this way, redundancy is increased, data rate is lowered, and error correction ability is increased, so the error rate of the same data rate can be lowered. As a result, the actual amount of information transmission of the recording / reproducing system or transmission system increases. In the case of the 8-VSB transmission apparatus described in Embodiment 5, since it is 3 bits / symbols, the trellis encoder 744g and trellis decoder 744q having the ratio 2/3 indicated in 128 (b) (e). ), And the overall block diagram is shown in FIG. However, the trellis encoder is not used for the first data string originally having a low error rate in the block diagram of FIG.

Although the distance between codes is smaller in the second data string than the first data string and the error rate is poor, the error rate is improved by trellis encoding the second data string. The configuration in which the trellis encoding circuit of the first data string is omitted has the effect of simplifying the entire circuit. Since the operation of the modulation is almost the same as that of the transmitter of FIG. 64 in Embodiment 5, the detailed description is omitted. The signal modulated by the modulator 750 is AC biased by the bias generator 856 to be amplified by the amplifier 875a in the recording / reproducing circuit 853 and on the magnetic tape 855 by the magnetic head 854. Is written on.

The format of the recording signal is a frequency of 2f _{C, which} is twice as large as f _C while information is recorded in the main signal 859 of, for example, 16SRQAM having a carrier of frequency f _C as shown in the recording signal frequency arrangement diagram of FIG. The pilot f _P signal 859a with is recorded at the same time. Since an AC bias is added by the bias signal 859b of frequency f _BIAS , recording distortion is reduced. Since hierarchical recording of two layers of the three layers shown in FIG. 113 is performed, two threshold values that can be recorded and reproduced are Th-1-2 and Th-2. According to the C / N value of the recording and reproducing, if the signal 859 is all the second floor, only the D ₁ is recorded and reproduced.

When 16 SRQAM is used as the main signal, the signal point arrangement is as shown in FIG. In the case of using 36SRQAM, the result is as shown in FIG. When 4ASK and 8ASK are used, the arrangements are the same as those in Figs. 58 and 68 (a) and (b). When reproducing this signal, the main signal 859 and the pilot signal 859a are reproduced from the magnetic head 854 and amplified by the amplifier 857b. From this signal, the pilot signal f _P of 2f _C is frequency-separated by the filter 858A of the carrier reproduction circuit 858, and the carrier of f _C is reproduced by the 1/2 divider 858b to demodulate 760. Is sent to. The demodulator 760 demodulates the main signal using the reproduced carrier wave. The case of using the high C / N value is a high magnetic recording tape 855, such as for HDTV, it is both the D ₁ and D ₂ are demodulated in the demodulator 760 because the discrimination of each of the signal points of the 16 points are easily . The entire signal is reproduced by the image decoder 402. In the case of the HDTV VTR, for example, a high bit rate TV signal of a 15Mbps HDTV is reproduced. The lower the C / N value, the lower the videotape. At present, the difference between commercial VHS tape and broadcast high C / N tape is more than 10dB C / N. When the video tape 855 having a low C / N value is used, since the C / N value is low, it is difficult to discriminate all 16 or 36 signal points. For this reason, the first data string D ₁ can be reproduced, but the 2-bit, 3-bit or 4-bit data string of the second data string D ₂ cannot be reproduced. Only the 2-bit data string of the first data string is reproduced. When the hierarchical HDTV image signal of two layers is recorded and reproduced, the low pass image signal of the low rate of the first data string is reproduced because the high pass image signal is not reproduced on the low C / N tape. Specifically, for example, a 7 Mbps wide NTSC TV signal is output.

As shown in the block diagram of FIG. 114, the second data string output unit 759, the second data string input unit 744, and the second image decoder 402b are omitted, and only the first data string D ₁ is omitted. A low bit rate dedicated recording and reproducing apparatus 851 having a modulator such as modified QPSK to be modulated and demodulated can also be set as one product type. This apparatus can perform recording and reproduction of only the first data string. That is, it is possible to record and reproduce the image signal of the wide NTSC grade. When the video tape 855 for outputting a high C / N value on which a high bit rate signal such as the above HDTV signal is recorded is reproduced by this low bit rate magnetic recording and reproducing apparatus, only the D ₁ signal of the first data string is reproduced. , A wide NTSC signal is output, and the second data string is not reproduced. That is, when the video tape 855 on which the same hierarchical HDTV signal is recorded is reproduced, the HDTV signal can be reproduced by the recording / reproducing apparatus of one complicated structure, and the wide NTSC TV signal by the recording / reproducing apparatus of one simple structure. That is, in the case of the two-layer hierarchy, there is a great effect that the full compatibility of four combinations of tapes having different C / N values and models having different recording and reproducing data rates is realized. In this case, the NTSC dedicated device is remarkably simple compared to the HDTV dedicated device. Specifically, for example, the circuit size of the decoder of the EDTV is 1/6 as compared with that of the HDTV decoder. Therefore, a low functional group can be realized at a significantly low cost. As described above, since two types of recording and reproducing apparatuses having different recording and reproducing capabilities of HDTV and EDTV can be realized, there is an effect that models of a wide range of prices can be set. In addition, the user can freely select from high-priced C / N high tapes to low-cost low C / N tapes according to the required quality. In this way, scalability can be achieved while maintaining full compatibility, and future compatibility can be secured. Therefore, it is also possible to realize the specification of the recording / reproducing apparatus which does not obsolete in the future. As another recording method, hierarchical recording by phase modulation described in Embodiments 1 and 3 can also be performed.

Recording by the ASK described in Example 5 can also be performed. With the current two-value record as multiple values, as shown in Fig. 59 (c), (d) or 68 (a) (b), the signal points of four-valued ASK or eight-valued ASK are divided into two groups. The second and third floors can be layered.

The block diagram in the case of ASK is the same as in FIG. It becomes like FIG. 173. The combination of Trellis and ASK lowers the error rate. In addition to those described in the embodiments, hierarchical recording by many tracks on magnetic tape can also be performed. In addition, logical hierarchical recording by differentiating data by changing error correction capability can also be performed.

This section describes the compatibility with future standards. Usually, when setting the standard of a recording / reproducing apparatus such as a VTR, the standard is determined by using the highest C / N tape that is practically available. The recording characteristics of the tape are improved by Iljin Moonbo. For example, C / N is more than 10dB higher than the tape 10 years ago. In this case, when a new standard is set at a point in time where tape performance is improved in the future 10 to 20 years from now, it is very difficult to achieve compatibility with the old standard in the conventional method. For this reason, old and new standards were often incomplete or incompatible. However, in the case of the present invention, first, a standard for recording and reproducing the first data string or the second data string on the current tape is made. Next, when the present invention is adopted in advance when the C / N of the tape is improved in units of 10 dB, the data of the upper data layer, for example, the data of the third data stream, is added, and for example, three layers of 64SRQAM or 8ASK are added. The new standard is realized while the super HDTV VTR for recording and reproducing is fully compatible with the conventional standard. At the time when this future standard is realized, the magnetic recording and reproducing of the old standard can record and reproduce only the first data string and the second data string of the magnetic tape recorded to the third data string according to the present invention and the new standard. In the case of playback on the device, the third data stream cannot be reproduced, but the first and second data streams can be completely reproduced. For this reason, the HDTV signal is reproduced. This has the effect of expanding the amount of recorded data in the future while maintaining compatibility between the old and new standards.

Returning to the description of FIG. 84, during reproduction, the magnetic tape 855 is reproduced by the magnetic head 854 and the magnetic reproduction circuit 853, and sent to the side reporter 852. The demodulator performs almost the same operations as those in the first, third, and fourth embodiments, and thus description thereof is omitted. The demodulation unit 760 reproduces the first data string D ₁ and the second data string D ₂ , and the second data string is subjected to trellis decoders such as Viterbi decoders to obtain the code gain. High error correction, and the error rate is low. The D ₁ and D ₂ signals are demodulated by the picture decoder 402 to output a video signal of the HDTV.

The above uses the embodiment of the magnetic recording and reproducing apparatus having two hierarchies, or the embodiment of the three-layer magnetic recording and reproducing apparatus in which one logical layer is added to the two physical layers. Explain. Basically, the configuration is the same as in FIG. 84, but the first data sequence is divided into two subchannels by TDM to have a three-layer structure. As shown in FIG. 131, first, the HDTV signal is mid-range by the 1-1st image encoder 401c and the 1-2th image encoder 401d in the first image encoder 401a. It is divided into two data D _1-1 and D _1-2 which are video signals of the low and low bands, and are input to the first data string input unit of the input unit 742. The data sequence D _1-1 of the MPEG grade image quality is coded with a high error correction code in the ECC encoder 743a, and D _1-2 is a normal code gain in the ECC encoder 743b. It is a misconception-correction with D _1-1 and D _1-2 are time multiplexed by the TDM unit 743c to form one data string D ₁ . D ₁ and D ₂ are modulated by the C-CDM modulator 750 and hierarchically recorded by the magnetic head 854 into two layers on the magnetic tape 855.

At the time of reproduction, the recording signal reproduced by the magnetic head 854 is demodulated by the C-CDM demodulation unit 760 into D ₁ and D ₂ by the same operation as described with reference to FIG. The first data string D ₁ is demodulated in two sub-channels D _1-1 and D _1-2 in the TDM unit 758c of the first data output unit 758. Since D _1-1 is miscorrected in the ECC decoder 758a with high gain, the D _1-1 is demodulated even at a low C / N value as compared to D _1-2 . The LDTV is decoded and output by the coder 402c. On the other hand, D _1-2 has a higher C / N threshold than D _1-1 because the code gain is miscorrected in the ordinary ECC decoder 758b, so that reproduction cannot be performed unless the signal level is large. The demodulation is performed in the 1-2 picture decoder 402d, synthesized with D1-1, and an EDTV of a wide NTSC grade is output.

The second data string D ₂ is Viterbi-decoded by the trellis decoder 759b, miscorrected and corrected by the ECC 759a, and becomes a high-pass image signal by the second image decoder 402b. HD-1 is output by combining with _1-1 and D _1-2 . The threshold of C / N of D ₂ in this case is set larger than D _1-2 . Therefore, when the C / N value of the tape 855 is small, D _1-1, that is, LDTV, is reproduced. In the case of the normal C / N value tape 855, D _1-1 , D _1-2, or EDTV, is reproduced. When the tape 855 having a high C / N value is used, D _1-1 , D _1-2 , D _2, that is, HDTV signals are reproduced.

In this way, a three-layer magnetic recording / reproducing apparatus is realized. As described above, the C / N value and the cost of the tape 855 are correlated. In the present invention, since the user can record and reproduce image signals of three grades of image quality corresponding to three types of tape coasts, the user can select the grade of the tape according to the contents of the TV program to be recorded. It is effective.

Next, the effect of hierarchical recording in fast forward playback will be described. As shown in the recording track diagram of FIG. 132, on the magnetic tape 855, recording tracks 855a of azimuth A and recording tracks 855b of opposite azimuth angle B are recorded. As shown in the drawing, the recording area 855c is formed in the center of the recording track 855a as described above, and the other area is called the D _1-2 recording area 855d. At least one of these is formed according to the number of recording tracks. One frame of LDTV is recorded. The D ₂ signal of the high frequency signal is recorded in the D ₂ recording area 855e which is the entire area of the recording track 855a. When recording and reproducing at normal speed, this recording format has no label. However, during fast forward and reverse tape play, the magnetic head trace 855f at the azimuth A does not coincide with the magnetic track as shown in the figure. In the present invention shown in FIG. 132, a D _1-1 recording area 855c is formed in a narrow area of the tape center portion. Because of this, there is some certain probability, but this area is certainly reproduced. From the reproduced D _1-1 signal, it is possible to demodulate an image of the entire screen at the same time, although the LDTV quality is about MPEGI. In this way, in the case of fast forward playback, if a complete image of dozens of LDTVs is reproduced from the purchase in one second, the user can check the screen during fast forwarding.

On the contrary, during sending and reproducing, only a part of the magnetic track is traced as indicated by the head trace 855g. However, even in this case, when the recording / playback format shown in FIG. 132 is used, since the D _1-1 recording area can be reproduced, a moving image of the image quality of the LDTV grade is intermittently output.

In this way, in the present invention, since the LDTV grade image is recorded in a small area of a part of the recording track, the user can reproduce an intermittently almost complete still image of the quick transfer with the LDTV grade image quality at the time of forward and reverse forward. There is an effect that the screen can be easily checked during the high-speed search.

Next, a description will be given of a method corresponding to high-speed fast-forward reproduction. As shown in the lower right of FIG. 132, a D _1-1 recording area 855c is formed, and one frame of the LDTV is recorded, while at the same time a smaller area of D _1- is recorded in a part of the D _1-1 recording area 855c. _1, to form a D ₂ recording region (855h). Information on a part of one frame of LDTV is recorded in subchannel D _1-1 in this area. Is recorded in duplicate in the remaining information of LDTV the D ₂ recording region (855j) of the D _1-1 · D ₂ recording region (855h). Subchannel D ₂ has a data recording amount of three to five times that of subchannel D _1-1 . Therefore, information of one frame of LDTV on a tape having an area of 1/3 to 1/5 can be recorded in D _1-1 and D ₂ . Since the head trace can be recorded in the areas 885h and 885j, which are narrower areas, the time and area are 1/3 to 1/5 of the head trace time T _S1 . Therefore, even if the head is more inclined at a faster sending speed, the probability of tracing the entire area is increased. For this reason, a complete LDTV image is intermittently reproduced even at a speed of 3 to 5 times faster than D _1-1 .

This system has no function of reproducing the D ₂ recording area 855j in the case of two-layer VTRs, and therefore cannot be reproduced at high speed and fast. On the other hand, in the high-performance VTR of three layers, the image can be confirmed even when sending 3 to 5 times faster than the two layers. That is, not only the image quality according to the number of hierarchies, that is, the cost, but also the VTR having different maximum reproduction speeds that can be reproduced depending on the cost is realized.

In addition, although the hierarchical modulation method is used in the embodiment, even if it is a normal modulation method such as 16QAM, it is needless to say that the hierarchical picture encoding according to the present invention can be realized by performing hierarchical image encoding.

In the conventional non-hierarchical digital VTR recording method of compressing an image at high speed, image data is uniformly distributed, and therefore, the entire image of the same time of each frame cannot be reproduced at the time of fast forward reproduction. For this reason, only images whose time axis of each block on the screen are shifted can be reproduced. However, in the hierarchical HDTV VTR of the present invention, although it is an LDTV grade, there is an effect that it is possible to reproduce an image in which the time axis of each block on the screen is not shifted at the time of quick transmission and reproduction.

When hierarchical recording of the third layer of the HDTV of the present invention is performed, when the C / N of the recording / reproducing system is high, a high resolution TV signal such as an HDTV can be reproduced. When the C / N of the recording / reproducing system is low or when playing back with a low function magnetic reproducing apparatus, a TV signal of an EDTV grade such as wide NTSC or an LDTV grade such as low resolution NTSC is output.

As described above, in the magnetic reproducing apparatus using the present invention, even when the C / N is lowered or the error rate is increased, the same image can be reproduced with a lower resolution or lower image quality.

Example 7

Example 7 uses the present invention for video layer transmission of four layers. By combining the four-layer transmission scheme described in the second embodiment and the four-layer video data structure, as shown in the reception disturbance region diagram of FIG. As shown in the figure, a first receiving area 390a is formed at the innermost side, a second receiving area 890b, a third receiving area 890c, and a fourth receiving area 890d at the outer side thereof. The method of realizing these four layers will be described.

In order to realize the four layers, there are four physical layers by modulation and four logical layers by differentiation of error correction capability, but the former requires a large C / N because the C / N difference between layers is large. do. Since the latter is predicated on demodulation, the C / N difference between layers cannot be large. In reality, the four-layer hierarchical transmission is performed using two physical layers and two logical layers. First, a method of dividing the video signal into four layers will be described.

93 is a block diagram of the separation circuit 3. The separation circuit 3 is composed of an image separation circuit 895 and four compression circuits. Since the basic configuration of the separation circuits 404a, 404b and 404c is the same as the block diagram of the separation circuit 404 in the first image encoder 401 of FIG. 30, description thereof is omitted. The separation circuit 404a and the like separate the video signal into four signals, a low frequency component H _L V _L , a high frequency component H _H H _H, and an intermediate component H _H V _L , H _L V _L. In this case, H _L V _L is half the resolution of the original video signal.

However, the input video signal is divided into a high frequency component and a low frequency component by the image separation circuit 404a. Four components are output because they are divided in the horizontal and vertical directions. The dividing point of the high and low ranges is at the midpoint in this embodiment. Therefore, when the input signal is 1000 HDTV signals, the H _L V _L signal is a TV signal with 500 vertical signals and a half horizontal resolution.

The low frequency component H _L V _L signal is divided into two frequency components in the horizontal and vertical directions by the separation circuit 404c. Thus, the H _L V _L outputs are 250 vertical, for example, the horizontal resolution is 1/4. If this is defined as an LL signal, the LL component is compressed by the compression unit 405a and output as a D _1-1 signal.

On the other hand, the three components of the high frequency component of H _L V _L are synthesized into one LH signal by the synthesizer 772c, compressed by the compression unit 405b, and output as a D _1-2 signal. In this case, three compression units may be disposed between the separation circuit 404c and the combiner 772c.

The three components, H _H V _H , H _L V _H and H _H V _L of the high frequency component, become one H _H V _H -H signal by the synthesizer 772a. When there are 1000 vertical and horizontal compression signals, the signal has 500 to 1000 components in the horizontal and vertical directions. The separation circuit 404b separates the four components.

Therefore, 500 to 750 components in the horizontal and vertical directions are separated as the output of the H _L V _L. This is called the HH signal. The three components, H _H V _H , H _L V _H and H _H V _L , have 750 to 1,000 components, are synthesized by the synthesizer 772b, become HH signals, and are compressed by the compression unit 405d. _It is output as a _2-2 signal. On the other hand, the HL signal is output as a D _2-1 signal. Thus, for example, the LL or D _1-1 signal has, for example, zero or more 250 components, the LH or D _1-2 signal has 250 or more and 500 frequency components, and the HL or D _2-1 signal has 500 or more components. 750 or less components, HH, ie, the D _2-2 signal, have 750 or more and 1000 frequency components. This separation circuit 3 has the effect that a hierarchical data structure is possible. By using the separation circuit 3 of FIG. 93, the part of the separation circuit 3 in the transmitter 1 of FIG. 87 explained in the second embodiment can be replaced to perform four-layer hierarchical transmission.

In this way, by combining the hierarchical data structure and the hierarchical transmission, it is possible to realize image transmission in which image quality deteriorates step by step as the C / N deteriorates. This has a big effect of expanding the service area in broadcasting. The receiver which demodulates and reproduces this signal has the same configuration and operation as the second receiver of FIG. 88 described in the second embodiment. Therefore, the whole operation is omitted. However, since the video signal is handled, the configuration of the synthesizer 37 is different from that of data transmission. Here, the synthesizer 37 will be described in detail.

As described in the second embodiment using the block diagram of the receiver of FIG. 88, the received signal is demodulated and miscorrected so that D _1-1 , D _1-2 , D _2-1 , D _2-2 It becomes four signals and is input into the synthesizer 37.

FIG. 94 is a block diagram of the synthesizer 37. The input D _1-1 , D _1-2 , D _2-1 , D _2-2 signals are extended in the extension sections 523a, 523b, 523c and 523d, and described in the separation circuit of FIG. LL, LH, HL and HH signals. If the horizontal and vertical band of the original video signal is 1, LL is 1/4, LL + LH is 1/2, LL + LH + HL is 3/4, and LL + LH + HL + HH is It is a band of 1. The LH signal is separated by the separator 531a, synthesized with the LL signal in the image synthesizing unit 548a, and input to the H _L V _L terminal of the image synthesizing unit 548c. The description of the example of the image synthesizing unit 531a has been described with reference to the image decoder 527 in FIG. On the other hand, the HH signal is separated by the separator 531b and input to the image combining unit 548b. The HL signal is synthesized with the HH signal in the image synthesizing unit 548b, becomes a H _H V _H -H signal, and is separated by the separator 531c. The video signal is synthesized and output from the synthesizer 37 as a video signal. Then, the output unit 36 of the second receiver in FIG. 88 becomes a TV signal and is output. In this case, if the original signal is 1050 vertical and approximately 1000 HDTV signals, four TV signal images are received under the four reception conditions shown in the reception diagram of FIG.

The picture quality of the TV signal will be described in detail. The transmission layer structure diagram of FIG. 92 is shown by combining FIG. 91 and FIG. 86 into one. As described above, with C / N improvement, D _1-1 , D _1-2 , D _2-1 , and D _2-2 in the reception areas 862d, 862c, 862b, and 862a. A hierarchical channel that can be reproduced is added to increase the amount of data.

In the case of hierarchical transmission of a video signal, hierarchical channels of LL, LH, HL, and HH signals are reproduced with the improvement of C / N, as shown in FIG. Therefore, the image quality improves as the distance from the transmission antenna gets closer. LL signal when L = Ld, LL + LH signal when L = Lc, LL + LH + HL signal when L = Lb, and LL + LH + HL + HH signal when L = La. Therefore, if the band of the original signal is 1, the picture quality of the bands of 1/4, 1/2, 3/4, 1 is obtained in each reception area. If the original signal is 1000 HDTVs of vertical scan lines, 250, 500, 750 and 1000 TV signals are obtained. In this manner, hierarchical video transmission in which image quality deteriorates in stages is possible. Fig. 96 is a reception diagram in the case of conventional digital HDTV broadcasting. As is apparent from the figure, in the conventional method, reproduction of a TV signal becomes impossible at all when C / N is less than V ₀ . Therefore, even within the service area distance R, it cannot be received in a contention area with another station or behind a building as indicated by an X mark. FIG. 97 shows a reception state diagram of hierarchical broadcasting of HDTV using the present invention. As shown in FIG. 97, C / N = a at distance La, C / N = b at Lb, C / N = c at Lc, C / N = d at Ld and 250 in each receiving area. , 500, 750 and 1000 picture quality are obtained. Even within a distance La, C / N deteriorates, and there is an area where reproduction of HDTV quality itself is impossible. However, even in this case, the picture quality may be degraded, but playback may be possible. For example, 750 at point B behind the building, 250 at point D in the tank, 750 at point F, which receives ghosts, 250 at point G in the car, and 250 at point L, where it is competing with other countries. You can play back in quality. By using the hierarchical transmission of the present invention as described above, it is possible to receive in an area where reception and reproduction cannot be performed by the conventionally proposed method, and there is a remarkable effect that the service area of a TV station is greatly expanded. In addition, as shown in the hierarchical transmission diagram of FIG. 98, the program D, which is the same program as the local broadcasting program, is broadcasted on the channel D _{1-1, and the} channel D _1-2 , D _2-1 , D _2-2 By broadcasting other programs C, B, and A, it is possible to reliably broadcast the thermal broadcast of the program D in the entire region, and also obtain the effect of program diversification, serving as the thermal broadcast and serving the other three programs.

Example 8

An eighth embodiment will be described below with reference to the drawings. Embodiment 8 applies the hierarchical transmission method of the present invention to a transceiver of a cellular telephone system. In the block diagram of the transceiver of the cellular phone of FIG. 115, the voice of the caller input from the microphone 762 is compressed and encoded by the compression unit 405 into the data D ₁ , D ₂ , D ₃ of the above-described hierarchical structure. The time division unit 765 divides the time into predetermined time slots based on the timing. The modulator 4 receives the hierarchical modulation of the SRQAM and the like, rides on one carrier, and then passes the antenna through the antenna common unit 764. It can be transmitted from 22, received at a base station described later, and transmitted to another base station or telephone station to communicate with another telephone.

On the other hand, communication signals from other telephones are received by the antenna 22 as transmission radio waves from the base station. This received signal is demodulated as data of D ₁ , D ₂ , D ₃ in a hierarchical demodulator 45 such as SRQAM. The timing signal is detected by the timing circuit 767 from the demodulation signal, and the timing signal is sent to the time division unit 765. The demodulation time signals D ₁ , D ₂ , and D ₃ are extended by the decompression unit 503 to be audio signals, and are sent to the speaker 763 to be audio.

Next, as shown in the block diagram of the base station of FIG. 116, the base stations 771, 772 and 773 at the center of each of the three hexagonal or circular receiving cells 768, 769, and 770 are removed. A plurality of transceivers 761a to 761j similar to 115 degrees are provided to transmit and receive data of the same channel number as the number of transceivers. The base station controller 790 connected to each base station always monitors the traffic volume of the communication of each base station, and accordingly controls the entire system such as allocation of channel frequencies to each base station or control of the size of the receiving cell of each base station. Is done.

As shown in the conventional communication capacity traffic distribution diagram of FIG. 117, in the conventional digital communication methods such as QPSK, the transmission capacity of the Ach of the receiving cells 768 and 770 is as shown in the diagram of d = A. Frequency utilization efficiency The data 774d which combined the data 774a and 774b of 2 bits / Hz and the data 774c of the figure of d = B becomes 774d, and is the same frequency utilization efficiency of 2 bits / Hz at any point. . On the other hand, the actual urban portion has a high population density where the buildings are concentrated, such as dense places 775a, 775b, and 775c, and shows peaks as indicated by the data traffic 774e. In the rest of the world, the amount of communication is small. As to the data 774e of the actual traffic volume TF, the capacity of the conventional cellular telephone was the frequency efficiency of the same area of 2 bits / Hz, as indicated by the data 774d. That is, there is a disadvantage of efficiency in that the same frequency efficiency is applied to many places where there is little traffic. In the conventional method, an area with a lot of traffic is allocated a lot of frequency allocations and an increase in the number of channels or a smaller size of a receiving cell. However, in order to increase the number of channels, there was a limitation of the frequency spectrum. In addition, the multiplexing of conventional 16QAM, 64QAM and the like increased transmission power. Reducing the size of the receiving cell and increasing the number of cells causes an increase in the number of base stations and increases the installation cost.

Ideally, increasing the frequency efficiency in high-traffic areas and lowering the low-traffic areas can increase the overall system efficiency. By employing the hierarchical transmission method of the present invention, the above can be realized. This will be described using the communication capacity and the traffic distribution diagram in the eighth embodiment of the present invention of FIG. The distribution diagram of FIG. 118 shows the communication capacity on the A-A 'line of the receiving cells 770b, 768, 769, 770 and 770a in order from the top. The receiving cells 768 and 770 use the channel group A, and the receiving cells 770b and 769 and 770a use the frequencies of the channel group B that do not overlap with the channel group A. These channels are increased or decreased by the base station controller 790 of FIG. 116 according to the traffic volume of each receiving cell. By the way, in FIG. 118, d = A indicates the distribution of the communication capacity of the A channel. d = B is the communication capacity of B channel, d = A + B is the communication capacity of all channels, TF is the amount of communication traffic, and P is the distribution of buildings and populations. Since the receiving cells 768, 769, and 770 use a multilayered hierarchical transmission method such as SRQAM described in the previous embodiment, as shown in the data 776a, 776b, and 776c, In addition, 6 bits / Hz, which is three times the frequency utilization efficiency of QPSK 2 bits / Hz, can be obtained from the base station peripheral part. As it goes to the periphery, it decreases to 4 bits / HZ and 2 bits / Hz. If the transmission power is not increased, the area of 2 bits / Hz is narrower than the size of the QPSK receiving cell indicated by the dotted lines 777a, 777b, and 777c. However, by slightly increasing the transmission power of the base station, You can get the size. The 64SRQAM station transmits / receives the modified QPSK in which the shift amount of SRQAM is S = 1 at a distance from the base station, and transmits and receives at 16SRQAM and 64SRQAM in the vicinity. Therefore, the maximum transmission power does not increase compared to QPSK. The transceiver of 4SRQAM, as shown in the block diagram of FIG. 121, which has simplified the circuit, can also communicate with other phones while maintaining compatibility. The same applies to the case of 16SRQAM shown in the block diagram of FIG. Thus, there are three modulation modes of magnetism. For mobile phones, compact and lightweight is important. In the case of 4SRQAM, the call rate is higher because the frequency utilization efficiency is lowered, but the circuit is simplified, so it is suitable for users who require compact and lightweight. In this way, this method can be applied to a wide range of applications.

As described above, a transmission system having a different distribution of capacity such as d = A + B in FIG. 118 is possible. By installing the base station in accordance with the traffic volume of the TF, there is a great effect that the overall frequency utilization efficiency is improved. In particular, since the microcell method with a small cell can provide many sub-base stations, it is easy to install a sub-base station at a location with a lot of traffic.

Next, the data arrangement of each timeslot will be described using the time alignment diagram of the data in FIG. FIG. 119 (a) shows a conventional timeslot and FIG. 119 (b) shows a timeslot of the eighth embodiment. As shown in FIG. 119 (a), the conventional frequency transmission / reception scheme transmits a synchronization signal S in slot 780a of time at frequency A at the time of transmission, i.e., transmission from the base station to the own station, and slot 780b ( 780c and 780d send a transmission signal to the magnets of the A, B and C channels, respectively. Next, when transmitting from the up side to the base station, the synchronization signals a, b, and c channels are transmitted to the timeslots 781a, 781b, 781c, and 781d at frequency B, respectively.

In the present case because the 119 degree using the above 64SRQAM such as hierarchical transmission method, as illustrated in _{(b) D 1, D 2} , each of the 2-bit / Hz 3 of layer data of the D ₃ Have Since A ₁ and A ₂ data are sent to 16SRQAM, the data rate is approximately twice as indicated by slots 782b, 782c and slots 783b and 783c. If you send the same sound quality, you can spend half the time. Thus, timeslots 782b and 782c are half the time. In this way, twice as much transmission capacity is obtained in the region of the second layer in 776c in FIG. 118, i.e., in the vicinity of the base station. Similarly, in timeslots 782g and 783g, transmission and reception of E ₁ data is performed in 64SRQAM. Because it has a transmission capacity of about three times it is possible to secure the three channels of the three times of E _1, E _2, E ₃ in the same time slot. In this case, transmission and reception in the vicinity of the base station is required. In this way, there is an effect that up to about three times more calls are obtained in the same frequency band. In this case, however, such a call is made in the vicinity of the base station, which is actually lower than this number. In addition, the actual transmission efficiency drops to about 90%. In order to enhance the effect of the present invention, it is preferable that the regional distribution of the traffic volume and the transmission capacity distribution according to the present invention coincide. However, as shown by the drawing of TF of FIG. 118, in an actual city, the green zone is arrange | positioned around the building street. Even in the suburbs, fields and forests are located around residential areas. Therefore, the distribution is close to that of TF. Therefore, the effect of applying the present invention is great.

In the TDMA time slot diagram of FIG. 120, (a) is represented by the conventional method, and (b) is represented by the method of the present invention. As shown in FIG. 120 (a), the transmission of the A and B channels to the magnetism is performed in the timeslots 786a and 786b in the same frequency band, respectively, and A and B in the timeslots 787a and 787b, respectively. The channel transmits itself. As shown in Fig. 120B, in the present invention, in the case of 16SRQAM, the A ₁ channel is received in the slot 788a, and the A ₁ channel is transmitted in the slot 788c. The timeslot width is about 1/2. In the case of 64SRQAM, the D ₁ channel is received in the slot 788i, and the D ₁ channel is transmitted in the slot 788L. The timeslot width is about one third.

In particular, a line for reception of E ₁ of 16SRQAM in a time slot of one-half in the slot (788p) to lower the power consumption, transmission is carried out in a conventional time-slot of the slot 4SRQAM (788r). Since 4SRQAM consumes less power than 16SRQAM, there is an effect that power consumption during transmission is reduced. However, the longer the occupancy time, the higher the communication fee. It is effective when the battery is small and light type mobile phone or when the remaining battery is small.

As described above, since the transmission capacity distribution can be set in accordance with the actual traffic distribution, the actual transmission capacity can be increased. Since three or two transmissions can be selected by the base station or the local station, various effects can be obtained by lowering the frequency efficiency, lowering power consumption, or conversely, lowering the call rate or high degree of freedom. In addition, by a method such as 4SRQAM having a low transmission capacity, it is possible to easily set a circuit having a smaller circuit and a smaller cost. In this case, one of the features of the present invention is that transmission compatibility between all models can be achieved as described in the above embodiment. In this way, the transmission capacity can be increased, and a wide range of models can be developed from the microminiature machine to the high-performance machine.

Example 9

A ninth embodiment will be described below with reference to the drawings. Embodiment 9 applies the present invention to a D transmission scheme. A block diagram of the OFDM transceiver of FIG. 123 and an operation principle of OFDM of FIG. 124 are shown. OFDM, which is a kind of FDM, orthogonal to adjacent carriers, and thus has better efficiency in frequency bands than general FDM. It is also considered for digital music broadcasting and digital TV broadcasting because it is resistant to multipath interference such as ghost. As shown in the principle diagram of OFDM in FIG. 124, in the case of OFDM, the input signal is arranged by the serial-parallel conversion unit 791 on the frequency axis 793 at intervals of 1 / ts, and the subchannels 794a to 126 are arranged. Write (794e). The signal is inversely FFT transformed from the modulator 4 with the inverse FFT 40 to the time axis 799 to produce a transmission signal 795. During the period of the valid symbol period 796 of ts, this inverse FFT signal is transmitted, and a guide period 797 of tg is formed between each symbol.

The operation of the ninth embodiment in the case of transmitting / receiving HDTV signals using the hybrid block diagram of the OFDM-CCDM of FIG. 123 will be described. The input HDTV signal is separated by a picture encoder 401 into a hierarchical picture signal of three layers of low D _1-1 , (mid-low) D _1-2, and (high-mid-low) D ₂ . And input to the input unit 742. In the first data string input unit 743, the D _1-1 signal is ECC coded with high code gain, and the D _1-2 signal is ECC coded with normal code gain. D _1-1 and D _2-2 are time-division multiplexed by the TDM 743c, and are input to the D ₁ serial-parallel converter 791a of the modulator 852a as a D ₁ signal. The D ₁ signal becomes n parallel data and n C-CDM modulators 4a, 4b... It is input to the first input unit of.

On the other hand, the high frequency component signal D ₂ is ECC (Error Correction code) coded in the ECC unit 744a in the second data string input unit 744 of the input unit 742 and trellis encoder 744b in the trellis encoder 744b. Release-coded, input to the D ₂ serial-parallel converter 791b of the modulator 852a, and become n pieces of parallel data, and C-CDM modulators 4a, 4b,... It is input to the 2nd input part of. The C-CDM modulators 4a, 4b, 4c, ... by D ₁ data of the first input unit and D ₂ data of the second input unit. C-CDM modulation is performed with 16SRQAM and the like. These n C-CDM modulators have carriers of different frequencies, while adjacent carriers are denoted by (794a) (794b) (794c). It is on the frequency axis 793 while being orthogonal as indicated by. The n modulated signals C-CDM modulated in this way are mapped from the frequency axis dimension 793 to the time axis dimension 795 by the inverse FFT circuit 40, and the time signal 796a of the effective symbol length of ts is obtained. (796b) and so on. A guard time zone 797a of tg seconds is formed between the effective symbol time zones 796a and 796b to reduce multipath interference. This is expressed in time axis and signal level. The time axis-signal level diagram of FIG. 129, and tg of the guard time zone 797a is determined according to the use from the influence time of the multipath. By setting tg longer than the influence time of multipath such as TV ghost, the modulated signal from the inverse FFT circuit 40 at the time of reception becomes one signal by the parallel-serial converter 40b and is transmitted by the transmitter 5. It is transmitted as an RF signal.

Next, the operation of the receiver 43 will be described. A time axis symbol signal 796e shown in FIG. The received signal is inputted to the input unit 24 of FIG. 123, inputted to the demodulator 852b, digitized, developed into Fourier coefficients by the FFT unit 40a, and the time axis 799 as shown in FIG. From to the frequency axis 793a. The time axis symbol signal shown in FIG. 124 is converted into carriers 794a and 794b of the signal on the frequency axis. Since these carriers are orthogonal to each other, each of the modulated signals can be separated. 16 SRQAM and the like shown in FIG. 125 (b) are demodulated and sent to the respective C-CDM demodulators 45a and 45b. In each of the C-CDM demodulators 45a and 45b of the C-CDM demodulator 45 and the like, the hierarchical demodulation and the D ₁ and D ₂ sub-signals are demodulated, and the D ₁ parallel-serial converter 852e is used. D ₂ is a serial signal by a parallel-to-serial converter (852f) and is demodulated, the original D _1, D ₂ signals. In this case it uses the 125 (b) layer transmission scheme using a C-CDM as shown in, C / N values in the bad receiving conditions, D ₁ signal only is demodulated, in the good receiving condition D ₁ Both of the and D ₂ signals are demodulated. The demodulated D ₁ signal is demodulated in the output unit 780. Since compared with the D _1-2 signal is high the gain of the Code D _1-1 signal and error correction, the error signal of the D _1-1 signal is reproduced even in the worse condition than the receiving condition. The D _1-1 signal is a low-frequency signal of the LDTV by the 1-1st image decoder 402c, and the D _1-2 signal is a low-frequency signal of the EDTV by the 1-2-2 image decoder 402d. It is output as a signal.

The D ₂ signal is trellis decoded and is output as a high frequency component of the HDTV by the second image decoder 402b. LDTV is output only by the above-mentioned low-band signal, the EDTV signal of a wide NTSC grade is output by adding the mid-range component, and the HDTV signal is synthesized by adding the high-band component. As in the previous embodiment, it is possible to receive a TV signal of the image quality according to the received C / N. In the case of the ninth embodiment, by using a combination of OFDM and C-CDM, hierarchical transmission that cannot be realized by OFDM itself can be realized. As shown in the error rate C / N of FIG. 130, with respect to the curve 805 of the conventional OFDM-TCM modulated signal, in the C-CDM-OFDM method of the present invention, the subchannel 1 (807a) has a lower error rate. Subchannel 2 807b has a higher error rate. In this way, a hierarchical type is realized.

Since OFDM certainly receives interference signals of multipath during the guard period tg, it is strong in multipath such as TV ghost. Therefore, it can be used for digital TV broadcasting for TV receivers of automobiles. However, because it is not a hierarchical transmission, it cannot be received below a certain C / N threshold. By combining with the C-CDM of the present invention, two kinds of image reception (Craditional Degradation) that are resistant to multipath and deteriorate of C / N can be realized. When receiving TV in a car, not only multipath but also C / N values deteriorate. Therefore, the service area of a TV station does not become much wider by the multipath countermeasure alone. However, by combining with C-CDM of hierarchical transmission, even if the C / N deteriorates considerably, it can be received in the LDTV grade. On the other hand, in the case of a car TV, since the screen size is usually 100 or less, sufficient image quality can be obtained by LDTV grade. There is an effect that the service area of LDTV grade of automobile TV greatly expands. When OFDM is used for the entire band of HDTV, the circuit size of DSP increases in the present semiconductor technology. Thus, only D _1-1 of the low-frequency TV signal is indicated in OFDM. As shown in the block diagram of FIG. 138, two of the mid-range and high-range components D _1-2 and D ₂ of the HDTV are C-CDM multiplexed according to the present invention, and are transmitted in frequency band A by the FDM 40d. do. On the other hand, the signal received at the receiver side is frequency-separated by the FDM 40e, demodulated by the C-CDM demodulator 45 of the present invention, and the mid-range and high-band components of the HDTV are reproduced in the same manner as in FIG. In this case, the operation of the image decoder is omitted since it is the same as in Examples 1, 2 and 3.

A low-band signal, and then the D _1-1 signal is serial-to-parallel converter 791 receives the modulation of QPSK, 16QAM, inverse FFT unit 40 and a parallel signal in an OFDM modulator (852c) by the MPEG1 grade of HDTV in Is converted into a signal on the time axis and transmitted by the FDM 40d in frequency band B.

On the other hand, the signal received at the receiver 43 is frequency-separated in the FDM unit 40e, and becomes the signal of most of the frequency axes by the FFT 40a in the OFDM demodulator 852d, and each demodulator 45a ( 45b) and demodulated by, a D _1-1 signal by a parallel-to-serial converter (882a) is demodulated, a D _1-1 signal of LDTV grade, like to help 123 is outputted from the receiver 43.

In this way, hierarchical transmission in which only the LDTV signal is OFDM is realized. By using the method of FIG. 138, the complex circuit of OFDM may be only an LDTV signal. Compared to the HDTV signal, the LDTV signal has a bit rate of 1/20. Therefore, the circuit size of OFDM is 1/20, and the overall circuit size is small.

OFDM is a multipath strong transmission system, and its main purpose is to apply a large and varying multipath disturbance in mobile stations such as when receiving a mobile TV or a car TV or receiving a digital music broadcast in a car. For such applications, small screen sizes of less than 10 inches, 4 inches to 8 inches, are the mainstream. Therefore, OFDM modulation of the entire high-definition TV signal, such as HDTV and EDTV, is less effective for the cost cost, and it is sufficient to receive the LDTV grade TV signal for automobile TV. On the other hand, in fixed stations such as home TVs, the multipath is always constant, so it is easy to take a multipath countermeasure. For this reason, the effect of OFDM is not high except in the strong ghost area. The use of OFDM in the mid / high frequency components of HDTV is not a good measure in the situation where the OFDM circuit is large. Therefore, the OFDM scheme shown in FIG. 138 of the present invention uses only low-frequency TV signals, without losing the effect of OFDM, which significantly reduces the multipath interference of LDTV received in mobile stations such as automobiles. There is a significant effect that can be significantly reduced to less than 1/10.

In FIG. 138, only D _1-1 is OFDM modulated, but D _1-1 and D _1-2 may be OFDM modulated. In this case, since D _1-1 and D _1-2 can perform C-CDM two-layer transmission, hierarchical broadcasting resistant to multipath is realized even in a mobile body such as a car, and the LDTV and DSTV are received by the mobile body. The effect of the gradient graduation is that the image of the image quality according to the level or antenna sensitivity can be received.

In this way, hierarchical transmission of the present invention is enabled, and various effects described above are obtained. Since OFDM is resistant to multipath, the combination with hierarchical transmission of the present invention can achieve the effect of being strong in multipath and degrading data transmission grade due to deterioration of reception level.

As a method of realizing a hierarchical transmission method, as shown in FIG. 126 (a), each subchannel 794a to 794c of OFDM is the first layer 801a, and the subchannels 794d to 794f. The second layer 801b is formed, and the frequency band 802a with fg is formed in the middle, and the power difference 802b with Pg is formed as shown in FIG. The transmission power of 801a and the second layer 801b can be differentiated.

By using this, the power of the first layer 801a can be increased in a range that does not interfere with analog TV broadcasting as shown in FIG. 108 (d) described above. In this case, as shown in FIG. 108 (e), the threshold of the receivable C / N value of the first layer 801a is lower than that of the second layer 801b. Therefore, the first layer 801a can be received even in a region with a low signal level or a region with a lot of noise. As shown in FIG. 147, hierarchical transmission of two layers is realized. This is called Power-Weighted-OFDM (PW-OFDM). By combining the C-CDM method of the present invention described above with the PW-OFDM of this embodiment, as shown in Fig. 108 (e), the layer is increased to become three layers, and the effectable area is expanded. have.

In a specific circuit, as shown in FIG. 144, the first layer data is the inverse FFT 40 from the modulators 4a to 4c having a large amplitude via the first data string circuit 791a to the carriers f _{1 to} f ₃ . OFDM modulation is performed on the second layer data, and the second layer data is shifted by the inverse FFT 40 from the modulators 4d to 4f to the carriers f _{6 to} f ₈ via the second data string circuit 791b. It is transmitted by OFDM modulation.

The received signal is separated into a signal having a carrier of f _{1 to} f _n by the FFT 40a of the receiver 43, and the carriers f _{1 to} f ₃ are first data strings D by the demodulators 45a to 45c. ₁ and that is the first layer (801a) is demodulated, the carrier f ₆ ~f ₈ from the second data string D ₂ that is the second layer (801b) and demodulated.

Since the power of the first layer 801a is large, it can be received even in a region where the signal is weak. In this way, two-layer hierarchical transmission is realized by PW-OFDM. When PW-OFDM is combined with C-CDM, layers of three to four layers are realized. In addition, since the operation of FIG. 144 is the same as that of the block diagram of FIG. 123, description thereof is omitted.

Next, a layering method of the Time-Weighted-OFDM (TW-OFDM) method of the present invention will be described. Since the OFDM scheme has a guard time band tg, as described above, if the delay time t _M of the ghost, i.e., the multipath signal satisfies the conditional expression of t _M tg, the influence of ghost is eliminated. In fixed stations, such as TV receivers in homes, t _M is small and several constants, and is easy to erase. However, in the case of a mobile station such as a vehicle-mounted TV receiver, since there are many reflected waves, t _M is not only large and close to several tens of microseconds, but also difficult to erase because it changes with movement. Therefore, it is expected that hierarchization for multipath will be required.

In the hierarchical method of the present embodiment, as shown in FIG. 146, the guard time tga of the layer A is made larger than the guard time tgb of the layer B so that the symbols of the subchannels of the layer A are ghosted. Become stronger against In this way, hierarchical weighting realizes hierarchical transmission for multipath. This method is called Guard-Time-Wighted-OFDM (GTM-OFDM).

When the number of symbols of the symbol time Ts of the layer A and the layer B is set to the same number, the symbol time tsa of A is made larger than the symbol time tsb of B. As a result, if the carrier intervals of A and B are Δfa and Δfb on the frequency axis, respectively, ΔfaΔfb. For this reason, the error rate at the time of demodulating the symbol of A is lower than the symbol of B. In this way, the layering of two layers with respect to the multipath of the layer A and the layer B is realized by differentiating the weighting of the symbol time Ts. This method is called Carrier-Spacing-Weighted-OFDM (CSW-OFDM). By using GTW-OFDM, two-layer hierarchical transmission is achieved, and low-resolution TV signals are transmitted on the A-layer and high-band components are transmitted on the B-layer, so that low-resolution reception is possible even under conditions of high ghosting such as a vehicle-mounted TV receiver. Stable reception of the TV becomes possible. In addition, by combining GTW-OFDM with layering of layer A and layer C / N due to the difference in symbol time ts using CSW-OFDM, more stable reception can be achieved in a vehicle-mounted TV with a low reception signal level. A great effect can be realized. High resolution is not required for vehicle-mounted or portable TVs. Since the time ratio of the symbol time including the low resolution TV signal is small, lengthening only this guard time does not reduce the overall transmission efficiency much. Therefore, by using the GTW-OFDM of the present embodiment, a multipath countermeasure focusing on a low-resolution TV signal has almost no effect on transmission efficiency, so that a mobile station such as a portable TV or a vehicle-mounted TV is compatible with a fixed station such as a home TV. It has a great effect of realizing hierarchical TV broadcasting. In this case, by combining with CSW-OFDM or C-CDM as described above, hierarchization for C / N is added, thereby enabling reception of a more stable mobile station.

In detail, the influence of the multipath will be described. As shown in FIG. 145 (a), in the case of the multipaths 810a to 810d having a short delay time, signals of the first and second layers can be received. And demodulate the signal of the HDTV. However, as shown in FIG. 145 (b), in the case of long multipaths 811a to 811b, demodulation cannot be performed because the guard time and Tgb of the B signal of the second layer are short. In this case, since the guard time Tga of the A signal of the first layer is long, it is not affected by the multipath with a long delay time. As described above, since the B signal contains the high frequency component of the TV, and the A signal contains the low frequency component of the TV, for example, the vehicle-mounted TV can reproduce the LDTV. In addition, since the symbol time Tsa of the first layer is larger than Tsb, the first layer is also strong in deterioration of C / N.

In this way, the two-dimensional layering of OFDM is possible by a simple structure by differentiating the guard time from the symbol time. By combining guard time differentiation and C-CDM in the configuration as shown in FIG. 123, both hierarchies of multipath and C / N sequencing are achieved.

It demonstrates in detail using a specific example here.

The smaller the D / U ratio, the larger the reflected wave is and the larger the multipath delay time T _M is than the direct wave. For example, as shown in FIG. 148, at D / U30dB, the influence of the reflected wave becomes larger and becomes 30 µs or more. As shown in FIG. 148, by taking a Tg of 50 µs or more, reception can be performed even in the worst condition. Therefore, as specifically shown in FIG. 149 (a), during the 2ms period shown in FIG. 149 (b) for the TV signal 1see, each symbol is represented by the first layer 801a, the second layer 801b, It is divided into three hierarchical groups of the third layer 801c, and is shown in FIG. 149 (c). The guard time 797a, 797b, 797c of each group, i.e., Tga, Tgb, Tgc, is set by weighting, for example, 50 µs, 5 µs, 1 µs, for example, the hierarchy 801a (as shown in FIG. 150) ( Hierarchical broadcasting relating to the multipath of three hierarchies of 801b and 801c is realized. Of course, if GTW-OFDM is applied to all image quality, transmission efficiency is lowered. However, the multi-path countermeasure of the GTW-OFDM is applied only to the LDTV quality signal having a small amount of information, so that the overall transmission efficiency does not drop much. In the first layer 801a, since the guard time Tg is taken as 50 µs of 30 µs or more, the vehicle-mounted TV receiver can also receive. The circuit uses the one shown in the block diagram of FIG. In particular, the vehicle-mounted TV may have a LDTV grade image quality and may have a transmission capacity of about 1 Mbps of the MPEG1 class. Therefore, as shown in FIG. 149, the symbol time (796a) Tsa can be 2 Mbps by taking 200 μs for a period of 2 ms. According to the CSW-OFDM of the present invention, the transmission efficiency is slightly reduced, but the error rate is lowered. In particular, when the C-CDM of the present invention is combined with the GTW-OFDM, the effect is even greater because the transmission efficiency does not decrease. In Fig. 149, the symbol times 796a, 796b, and 796c are differentiated into 200 µs, 150 µs, and 100 µs for the same number of symbols. Therefore, hierarchical transmission is performed in which the error rate increases in the order of the first layer, the second layer, and the third layer.

At the same time, hierarchical transmission is realized for C / N. As shown in FIG. 151, the combination of CSW-OFDM and CSW-OFDM realizes two-dimensional hierarchical transmission of multipath and C / N. As mentioned above, the combination of the CSW-OFDM and the C-CDM of the present invention can be realized. In this case, there is an effect that the overall transmission efficiency is reduced. LDTV grade is also used in the first layer 801a, the first-second layer 851az, and the first-third layer 851a with a large multipath T _M and low C / N, for example, a vehicle-mounted TV receiver. Stable reception is possible. In the second layer 801b and the second to third layer 851b, like the fringe area of the service area, SDTV grades of standard resolution can be received in a fixed station in a reception area with a low C / N and a lot of ghosts. . In the third layer 801c, which occupies more than half of the service area, the C / N is high, the direct wave is large, and the ghost is low. In this way, two-dimensional hierarchical broadcasting of C / N and multipath is realized. This great effect is obtained by the combination of GTW-OFDM and C-CDM of the present invention or the combination of GTW-OFDM and CSW-C-CDM. Conventionally, a hierarchical broadcasting method for C / N has been proposed, but the present invention realizes two-dimensional matrix hierarchical broadcasting of C / N and multipath.

FIG. 152 shows the temporal arrangement diagram of TV signals of three layers of HDTV, SDTV, and LDTV in two-dimensional hierarchical broadcasting of three layers of C / N and three layers of multipath. As shown in the figure, LDTV is arranged in the slot 796a1 of the first layer of the A layer that is strongest in the multipath, and then in the slot 796a2 that is strong in the multipath or in the slot 796b1 that is resistant to C / N degradation. Place important HP signals such as SDTV sync signal and address signal. On the second and third floors of the B layer, a general signal of an SDTV, that is, an LP signal or an HP signal of an HDTV is disposed. On the C layer, high-frequency component TV signals such as SDTV, EDTV, and HDTV are arranged on the 1,2,3 layers.

In this case, the stronger the CN degradation or the multipath, the lower the transmission rate, so that the resolution of the TV signal is reduced, and as shown in FIG. 153, the present invention has no conventional effect of realizing three-dimensional graceful degradation. Obtained. FIG. 153 shows three-dimensional hierarchical broadcasting of the present invention by three parameters, CNR, multipath delay time and transmission rate.

The embodiment has been described using an example in which the two-dimensional hierarchical structure is obtained by the combination of the GTW-OFDM of the present invention and the C-CDM of the present invention or the combination of the GTW-OFDM and the CSW-C-CDM. Two-dimensional hierarchical broadcasting is also realized in combination of OFDM and Power-Weighted-OFDM, or in combination with GTW-OFDM and other CNR hierarchical transmission schemes.

FIG. 154 shows that the power of the carriers 794a, 794c, and 794e is smaller and transmitted than the carriers 794b, 794d, and 794f. Thus, two layers of Power-Weighted-OFDM are realized. The power of the carriers 795a and 795c orthogonal to the carrier 794a is similarly obtained by power-weighting the carriers 795b and 795d to obtain two layers. In total, four layers are obtained, but FIG. 154 shows an embodiment in the case of two layers. As shown in the figure, since the frequency distribution of the carriers is distributed, disturbances of other analog broadcasts in the same frequency band are distributed, thereby reducing the influence.

Also, as shown in FIG. 155, the hierarchical structure for the multipath of the third layer is obtained by taking a time arrangement in which the time widths of the guard times 797a, 797b, and 797c are changed for each symbol 796a, 796b, and 796c. The transmission is realized. In the time arrangement of FIG. 155, data of the A, B, and C layers are distributed on the time axis. For this reason, even when burst noise occurring at a specific time occurs, interleave the data of each layer to prevent data destruction and to stably demodulate the TV signal. In particular, by distributing data on the A-layer and interleaving, it is possible to greatly reduce the disturbance of burst noise generated from the ignition device of another vehicle generated when the vehicle-mounted TV is received.

The block diagrams of the specific ECC encoder 744j and the ECC decoder 749j in this case are shown in Figs. 160 (a) and (b), respectively. FIG. 167 shows a block diagram of the deinterleave unit 936b. The interleave table 954 processed in the deinterleave RAM 936a of the deinterleave unit 936b is shown in FIG. 168 (a), and the interleave distance L1 is shown in FIG. 168 (b).

By interleaving the data in this way, interference of burst noise can be reduced. As shown in the block diagram of the VSB receiver of FIG. 161 and the block diagram of the VSB transmitter of FIG. 162, the QAM or PSK transmission described in the 4VSB, 8VSB, or 16VSB transmission apparatus described in Embodiments 4, 5, 6, etc., or Embodiments 1, 2, etc. By using the apparatus, disturbance of burst noise can be reduced, so that TV reception with less noise can be performed in terrestrial broadcasting.

By performing the hierarchical broadcasting of the third layer by the method of FIG. 155, the A layer can reduce the disturbance of burst noise in addition to the above-described multipath and C / N degradation, so that the mobile station such as a vehicle-mounted TV receiver or a pocket TV can be used. This has the effect of stabilizing TV reception of the LDTV grade.

One feature of the hierarchical transmission scheme of the present invention is to improve frequency utilization efficiency, but in some receivers, the power utilization efficiency is considerably lowered. Therefore, it is not applicable to all transmission systems. For example, if the satellite communication system between specific receivers is the most economical way to change the equipment to the highest frequency utilization efficiency and the highest power utilization efficiency at that time. In this case, it is not necessary to use the present invention.

However, in the case of satellite broadcasting or terrestrial broadcasting, a hierarchical transmission method like the present invention is required. This is because the satellite broadcasting standard requires more than 50 years of durability. During this period, the broadcast standards remain unchanged, but with the technological innovation, the satellite's transmission power is dramatically improved. In the future decades later, broadcast stations must perform compatible broadcasts so that manufactured receivers can watch TV programs. By using the present invention, it is possible to obtain an effect of compatibility of existing NTSC broadcasting and HDTV broadcasting and scalability of future information transmission.

Although the present invention places more emphasis on frequency efficiency than power efficiency, it is not necessary to increase the power of the transmitter by setting various types of receivers, each of which has a design reception sensitivity set for each transmission stage. For this reason, even a satellite with a small current power can be transmitted sufficiently. In addition, even when the transmission power is increased in the future, transmission can be performed using the same standard, so that future expandability and compatibility with new and old receivers are obtained. As described above, when the satellite broadcasting standard is used, a remarkable effect can be obtained.

In addition, when the hierarchical transmission method of the present invention is used for terrestrial broadcasting, the present invention is easier to implement than satellite broadcasting because there is no need to consider power utilization efficiency at all. As described above, in the conventional digital HDTV broadcasting system, there is a remarkable effect of drastically reducing the unreceivable area in the service area, and the compatibility of the NTSC and HDTV receiver or receiver. In addition, there is an effect that the service area in the case of viewing from the sponsor of the TV program is substantially expanded. In addition, although the embodiment has been described using an example using a modulation scheme of QPSK, 16QAM, and 32QAM, it goes without saying that it can be applied to 64QAM, 128QAM, 256QAM, and the like. Moreover, it can be said that it can be applied to multi-value PSK, ASK, or FSK as demonstrated using drawing. Although an embodiment in which the present invention is combined with TDM has been described, it is also possible to transmit in combination with FDM, CDMA or spreading communication.

As described above, the present invention provides a signal input unit, a modulator for modulating a plurality of carriers of different phases by an input signal from the input unit, and generating a signal point having an m value that becomes a signal vector. In a transmission device comprising a transmitter for data transmission, an n-value first data string and a second data string are input, the signal is divided into n signal point groups, and each of the signal point groups is assigned to data of each of the first data strings. A demodulator for transmitting a signal by a transmitter that allocates and transmits each data of the second data group to each signal point in the signal point group, and demodulates the QAM modulated wave of the signal point of p value on the input portion of the transmission signal and on the signal space diagram. And a receiving device having an output unit and an output unit. The signal points are divided into n-value signal point groups, and the first data strings of each n-value signal point group are matched to each other. By demodulating and reproducing the data of the second data string having the p / n value at a signal point of approximately p / n value in the signal point group, and transmitting the data using a receiver, for example, the modulator 4 of the transmitter 1 N is assigned to the first data string, the second data string, and the third data string of the data value to the signal point group, and the QAM modulated signal of the modified m value is transmitted. In the first receiver 23, the demodulator 25 uses n Effect by demodulating the first data sequence of the value, the first data sequence and the second data sequence in the second receiver 33, and the first data sequence, the second data sequence, and the third data sequence in the third receiver 43. As a result, a transmitter having a compatible and highly developed signal capable of demodulating n-value data can be obtained even in a receiver having only n-value demodulation capability of a multi-modulation wave modulated with the maximum m-value data. When the distance between the signal point closest to the origin of the QAM system signal point and the I-axis or the Q-axis is f, hierarchical transmission is possible by shifting the signal point so that the distance is n1 nf. .

By transmitting the NTSC signal to the transmission system as the first data string and the difference signal between HDTV and NTSC as the second data string, digital broadcasting having both compatible with NSTC broadcasting and HDTV broadcasting and highly scalable information volume in satellite broadcasting In terrestrial broadcasting, there is a remarkable effect of expanding the service area and eliminating the unreceivable area.

Claims (9)

A terrestrial TV transmission apparatus for transmitting and receiving an n-value VSB image signal, comprising: input means for compressing an image signal, image compression means for compressing the image signal into a digital image compressed signal, and error correction code to the digital image compressed signal; An error correction encoder that applies a false correction code signal and trellis encodes the error correction code signal to obtain a trellis coded signal; A transmitting device including modulation means for modulating a VSB modulated signal to obtain an eight-value VSB signal, transmitting means for transmitting the eight-value VSB modulated signal, means for receiving the transmitted eight-value VSB signal, Demodulation means for demodulating an 8-value VSB signal into a VSB demodulation signal, a trellis decoder for trellis decoding the VSB demodulation signal to obtain a trellis decoded signal, and the trellis Error correction means for obtaining a mistaken-corrected digital image compression signal by error-correcting the call signal, image decompression means for decompressing the error-corrected digital image compression signal into a video output signal, and an output means for outputting the video output signal; Terrestrial TV transmitter, characterized in that consisting of a receiving device provided. 8. The received error correction means is divided into a high rank signal and a low rank signal, wherein the error correction means comprises a high code gain first error correction means and a low code gain second error correction means, And the first error correction means miscorrects the high priority signal. 3. The terrestrial TV transmitter as claimed in claim 2, wherein said high priority signal carries address data for a digital image compressed signal. The VSB demodulation signal according to claim 2, wherein the VSB demodulation signal of the high priority signal is trellis decoded by a trellis decoder, and is miscorrected and corrected by the first error correction means. Terrestrial TV transmission apparatus characterized in that the miscorrection by only error correction means. A transmitter for a terrestrial TV transmitter for transmitting an n-value VSB image signal, comprising: an input means for inputting an image signal, an image compression means for compressing the image signal into a digital image compressed signal, and an error correction code to the digital image compressed signal; A miscorrection correcting means for applying a miscorrection correcting signal, a trellis encoder for trellis encoding the miscorrection correcting signal to obtain a trellis encoded signal, and the trellis encoded signal having 8 VSBs And modulation means for modulating the modulated signal to obtain an eight-value VSB signal, and means for transmitting the eight-value VSB signal. A terrestrial TV transmitter for receiving a transmission signal of a terrestrial TV transmitter that compresses an image signal into a digital image compressed signal, trellis-codes an error correction signal applied with an error correction, and modulates it with an 8-value VSB signal. An apparatus comprising: means for receiving the transmitted eight-value VSB signal, demodulation means for demodulating the received eight-value VSB signal into a VSB demodulated signal, and a trellis decoded signal by trellis-decoding the VSB demodulated signal. A trellis decoder that obtains an error, error correction means for obtaining an error correction digital image compression signal by error-correcting the trellis decoded signal, and an image extension for extending the error correction digital image compression signal to a video output signal. Means and an output means for outputting the video output signal. 8. The received error value VSB signal is divided into a high rank signal and a low rank signal, and the error correction means includes a high code gain first error correction means and a low code gain second error correction means. And a first error correction means for error correction of the high priority signal. 8. The receiver according to claim 7, wherein said high priority signal repeats address data for a digital image compressed signal. The VSB demodulation signal according to claim 7, wherein the VSB demodulation signal of the high priority signal is trellis decoded by a trellis decoder, and is miscorrected and corrected by the first error correction means so that the VSB demodulation signal of the low priority signal is obtained. Receiver for terrestrial TV transmitter, characterized in that the error correction by only error correction means.