JP2019088039A - Motor controller - Google Patents

Abstract

To improve the control accuracy and stability of motor driving by creating a voltage feedback value close to a true value.SOLUTION: A phase corrector 10 of a motor controller 1-1 reads out a phase correction angle Δθe corresponding to an electric angular velocity ωe from a phase correction table 11 including the phase characteristic of an LPF 114. An accumulator 12 calculates an after-correction electric angle θe' by adding the phase correction angle Δθe to an electric angle θe. A coordinate converter 115 performs a coordinate conversion by applying a U-phase voltage feedback Vu, a V-phase voltage feedback Vv, and a W-phase voltage feedback Vw after LPF processing to a q-axis voltage feedback Vq and a d-axis voltage feedback Vd on the basis of the after-correction electric angle θe'. By doing this, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd become close to true values that compensate phase delays generated in the LPF 114.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a motor control device, and more particularly to a motor control device that converts three-phase voltage values of a stationary coordinate system into two-phase voltage values of a rotating coordinate system.

  Conventionally, a motor control device that performs vector control of a motor based on a q-axis voltage command and a d-axis voltage command is known (see, for example, Patent Document 1). This motor control device coordinates-converts the q-axis voltage command and the d-axis voltage command into a three-phase AC voltage command, and a gate signal (PWM) of PWM (Pulse Width Modulation: pulse width modulation control) based on the three-phase AC voltage command. Signal). The PWM inverter converts the DC voltage supplied from the power supply into a three-phase AC voltage based on the PWM signal, and supplies the three-phase AC voltage to the motor.

  In such a motor control device, voltage feedback is input from a voltage detector that detects a three-phase AC voltage supplied to the motor, and slip calculation control, constant output control, speed estimation, and a magnetic flux observer are performed using the voltage feedback. Perform various processes such as For this reason, it can be said that voltage feedback is important data that becomes a reference when the motor control device performs motor control.

  FIG. 7 is a block diagram showing a configuration example of a conventional motor control device. The motor control device 100 includes an electrical angular velocity calculator 110, an integrator 111, coordinate converters 112 and 115, a PWM controller 113, and an LPF (Low Pass Filter: low pass filter) 114.

  The electrical angular velocity calculator 110 calculates the electrical angular velocity ωe of the motor 101 and outputs the electrical angular velocity ωe to the integrator 111. For example, when the motor 101 is a synchronous motor, the electrical angular velocity calculator 110 receives an output signal of an encoder that detects the rotation of the motor 101, and calculates the electrical angular velocity ωe based on the output signal. When the motor 101 is an induction motor, the electrical angular velocity calculator 110 receives an output signal of an encoder for detecting the rotation of the motor 101, calculates a rotational frequency based on the output signal, and slips the rotational frequency. Are added, and the electric angular velocity ωe is obtained from the addition result.

  Furthermore, as described in the aforementioned Patent Document 1, the electrical angular velocity calculator 110 outputs the q-axis voltage command Vq * and the d-axis voltage command Vd * input by the coordinate converter 112, and the coordinate converter 115 outputs The electric angular velocity ωe may be calculated based on the q-axis voltage feedback Vq and the d-axis voltage feedback Vd.

  The integrator 111 receives the electrical angular velocity ωe from the electrical angular velocity calculator 110, integrates the electrical angular velocity ωe, and obtains the electrical angle θe. Then, the integrator 111 outputs the electrical angle θe to the coordinate converters 112 and 115.

  The coordinate converter 112 receives the q-axis voltage command Vq * and the d-axis voltage command Vd *, and receives the electrical angle θe from the integrator 111. Then, the coordinate converter 112 converts the two-phase q-axis voltage command Vq * and the d-axis voltage command Vd * of the rotating coordinate system into the three-phase U-phase voltage command Vu * of the stationary coordinate system based on the electrical angle θe. Coordinate conversion is performed to V-phase voltage command Vv * and W-phase voltage command Vw *. The coordinate converter 112 outputs the U-phase voltage command Vu *, the V-phase voltage command Vv * and the W-phase voltage command Vw * to the PWM controller 113.

  The PWM controller 113 receives the U-phase voltage command Vu *, the V-phase voltage command Vv * and the W-phase voltage command Vw * from the coordinate converter 112, and inputs a carrier from a carrier generator (not shown). Then, the PWM controller 113 compares the amplitudes of the U-phase voltage command Vu *, the V-phase voltage command Vv * and the W-phase voltage command Vw * with the carrier amplitude to obtain PWM of each phase according to the comparison result. Generate a signal. The PWM controller 113 outputs the PWM signal of each phase to a PWM inverter not shown.

  Thereby, the PWM inverter not shown turns on / off the gate of the semiconductor switching element of each phase according to the PWM signal of each phase from the PWM controller 113. Then, the PWM inverter converts a DC voltage supplied from a power supply (not shown) into a three-phase AC voltage (U-phase output voltage Vuo, V-phase output voltage Vvo and W-phase output voltage Vwo). The PWM inverter supplies the motor 101 with the U-phase output voltage Vuo, the V-phase output voltage Vvo and the W-phase output voltage Vwo.

  On the other hand, a not-shown voltage detector provided between the not-shown PWM inverter and the motor 101 detects a three-phase AC voltage supplied from the PWM inverter to the motor 101. Then, motor control apparatus 100 inputs three-phase AC voltage detected by a voltage detector (not shown) as three-phase AC voltage feedback (U-phase voltage feedback Vuo, V-phase voltage feedback Vvo and W-phase voltage feedback Vwo). .

  Hereinafter, for convenience of description, U-phase output voltage Vuo, V-phase output voltage Vvo and W-phase output voltage Vwo, U-phase voltage feedback Vuo, V-phase voltage feedback Vvo and W-phase voltage feedback Vwo have the same symbols. I will use it.

  The LPF 114 receives the U-phase voltage feedback Vuo, the V-phase voltage feedback Vvo, and the W-phase voltage feedback Vwo, and performs an LPF process on these voltage feedbacks. Then, the LPF 114 outputs the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the LPF processing to the coordinate converter 115.

  The coordinate converter 115 receives the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the LPF processing from the LPF 114, and also receives the electrical angle θe from the integrator 111.

  The coordinate converter 115 converts the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the three-phase LPF processing of the stationary coordinate system into two phases of the rotational coordinate system based on the electrical angle θe. Coordinate conversion is performed to q-axis voltage feedback Vq and d-axis voltage feedback Vd. Then, the coordinate converter 115 outputs the q-axis voltage feedback Vq and the d-axis voltage feedback Vd to components (not shown) (components that perform processing such as slip calculation control and constant output control).

  FIG. 8 (1) shows that in U-phase voltage command Vu *, V-phase voltage command Vv * and W-phase voltage command Vw * output from coordinate converter 112, voltage command Vuv * between U phase and V phase FIG. It can be seen from FIG. 8 (1) that the waveform of the voltage command Vuv * output from the coordinate converter 112 is a sine wave.

  FIG. 8 (2) is a diagram showing the waveform of the output voltage Vuvo between the U phase and the V phase at the U phase output voltage Vuo, the V phase output voltage Vvo and the W phase output voltage Vwo output from the PWM inverter. is there. It can be understood from FIG. 8 (2) that the waveform of the output voltage Vuvo output from the PWM inverter is a rectangular wave which is chopped along with the switching operation of the PWM inverter.

  As described above, the voltage detector (not shown) detects a rectangular wave as shown in FIG. 8B as the U-phase voltage feedback Vuo, the V-phase voltage feedback Vvo and the W-phase voltage feedback Vwo. Therefore, motor control apparatus 100 uses LPF 114 to cut the switching frequency components included in U-phase voltage feedback Vuo, V-phase voltage feedback Vvo and W-phase voltage feedback Vwo.

  FIG. 8 (3) shows the waveform of the voltage feedback Vuv between the U phase and the V phase in the U phase voltage feedback Vu, the V phase voltage feedback Vv and the W phase voltage feedback Vw after the LPF processing output from the LPF 114. FIG. Although the waveform of the voltage feedback Vuv output from the LPF 114 is a waveform in which the switching frequency component is cut from the output voltage Vuvo shown in FIG. 8 (2) from FIG. 8 (3), it is shown in FIG. As compared with the waveform of the voltage command Vuv *, it can be seen that the sine wave includes fine ticks remaining with the switching operation of the PWM inverter.

JP 2007-104777 A

  As described above, the motor control device 100 uses the LPF 114 to switch from the three-phase AC voltage feedback (U-phase voltage feedback Vuo, V-phase voltage feedback Vvo and W-phase voltage feedback Vwo) detected by the voltage detector. Cut frequency components.

  However, since the LPF 114 is generally configured by a hardware circuit, phase delay and amplitude attenuation (gain attenuation) due to frequency exist in the three-phase AC voltage feedback after the LPF processing.

  FIG. 9 is a diagram showing an example of phase and gain characteristics of the LPF 114, and shows characteristics when the LPF 114 is a second-order Butterworth LPF. The horizontal axis is frequency [Hz], the vertical axis is phase [°] and gain [dB], and the cutoff frequency is fc = 1 kHz.

  As shown in FIG. 9, it can be seen that, in the three-phase AC voltage feedback output from the LPF 114, phase delay and amplitude attenuation due to frequency exist. When phase lag and amplitude attenuation exist in the three-phase AC voltage feedback after the LPF processing, the accuracy of the q-axis voltage feedback Vq and the d-axis voltage feedback Vd coordinate-converted by the coordinate converter 115 is reduced.

  As described above, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd are used as reference data for slip calculation control, constant output control, and the like. Therefore, to stabilize control accuracy and control when driving the motor 101. It can be said that it is important data that affects.

  As described above, since there is phase delay and amplitude attenuation in the three-phase AC voltage feedback after the LPF processing, the accuracy of the voltage feedback value which is the q-axis voltage feedback Vq and the d-axis voltage feedback Vd is reduced. Therefore, in the conventional motor control device 100, the control accuracy in driving the motor 101 is lowered, and there is a problem that it is difficult to realize stable control.

  Therefore, the present invention has been made to solve the above-mentioned problems, and an object thereof is to generate a voltage feedback value close to a true value, and to provide a motor capable of improving control accuracy and stability of motor drive. It is in providing a control device.

  In order to solve the above problems, a motor control device according to claim 1 is an electric angular velocity for calculating an electric angular velocity of the motor in a motor control device that controls the motor based on a predetermined q-axis voltage command and d-axis voltage command. The q-axis voltage command and the integrator, which integrates the electrical angular velocity calculated by the electrical angular velocity calculator and calculates the electrical angle of the motor, and the electrical angle calculated by the integrator. A first coordinate converter for coordinate-converting the d-axis voltage command to a three-phase AC voltage command, and supplying the motor to the motor based on the three-phase AC voltage command coordinate-converted by the first coordinate converter The three-phase AC voltage is used as a three-phase AC voltage feedback from a PWM controller that generates a PWM signal for generating a three-phase AC voltage and a voltage detector that detects the three-phase AC voltage Using the phase correction table in which the LPF that performs the LPF processing on the three-phase AC voltage feedback and the phase characteristics of the LPF are reflected, and the electrical angular velocity and the phase correction angle are stored as a pair A phase corrector for specifying the phase correction angle corresponding to the electrical angular velocity calculated by the angular velocity calculator, and the phase correction angle specified by the phase corrector is added to the electrical angle obtained by the integrator And the adder for outputting the electrical angle after addition, and the 3-phase AC voltage feedback subjected to the LPF processing by the LPF based on the post-addition electrical angle outputted by the adder, the q-axis voltage feedback and and d) a second coordinate converter for coordinate conversion to d-axis voltage feedback.

  The motor control apparatus according to claim 2 is the motor control apparatus according to claim 1, further comprising gain characteristics reflecting the gain characteristics of the LPF, wherein the electrical angular velocity and the gain correction coefficient are stored as a pair. A gain corrector that specifies the gain correction coefficient corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator using a table, and the three-phase AC voltage feedback subjected to the LPF processing by the LPF A multiplier for multiplying the gain correction coefficient specified by the gain corrector and outputting a three-phase AC voltage feedback after multiplication, and the second coordinate converter includes the addition output from the adder. The post-multiplication three-phase AC voltage feedback output by the multiplier is converted to q-axis voltage feedback and d-axis voltage feedback based on the post electrical angle. To target conversion, characterized in that.

  In the motor control device according to claim 3, in the motor control device according to claim 1, the phase corrector reflects the phase characteristic of the LPF instead of the phase correction table, and the electric angular velocity and the phase correction are corrected. The phase correction angle corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator is specified using a mathematical expression in which a relationship between the angle and the angle is defined.

  In the motor control device according to claim 4, in the motor control device according to claim 2, the gain correction unit reflects the gain characteristics of the LPF, and the relationship between the electrical angular velocity and the gain correction coefficient is defined. The gain correction coefficient corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator is specified using the equation.

  As described above, according to the present invention, a voltage feedback value close to a true value can be generated, and control accuracy and stability of motor drive can be improved.

FIG. 2 is a block diagram showing a configuration example of a motor control device according to a first embodiment. FIG. 7 is a block diagram showing an example of configuration of a motor control device according to a second embodiment. It is a figure which shows the structural example of a phase correction table. It is a figure which shows the structural example of a gain correction table. It is a figure which shows the example of the gain characteristic and phase characteristic of LPF. FIG. 6 is a diagram showing a gain correction coefficient k and a phase correction angle Δθ e obtained from the examples of the gain characteristic and the phase characteristic of FIG. 5. It is a block diagram which shows the structural example of the conventional motor control apparatus. (1) is a figure which shows the waveform of voltage command Vuv * between the U-phase and V-phase output from a coordinate converter. (2) is a figure which shows the waveform of the output voltage Vuvo between U phase and V phase which are output from a PWM inverter. (3) is a figure which shows the waveform of the voltage feedback Vuv between the U-phase after the LPF process output from LPF, and V phase. It is a figure which shows the example of a phase of a LPF, and the characteristic of a gain. It is a figure which shows the structural example of an active type 1st-order LPF. It is a figure which shows the structural example of an active type 2nd-order LPF.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. The present invention corrects the electrical angle using a phase correction table reflecting the characteristics of the LPF 114, and based on the corrected electrical angle, the q-axis voltage feedback Vq and the d-axis voltage of the 3-phase AC voltage feedback after LPF processing Coordinate conversion to feedback Vd is characterized.

  As a result, an electrical angle for compensating for the phase delay occurring in the LPF 114 can be generated, so the q-axis voltage feedback Vq and the d-axis voltage feedback Vd become close to their true values. Therefore, the influence of the phase delay generated in the LPF 114 can be eliminated, and the control accuracy and stability of the motor drive can be improved.

  In the present invention, the electrical angle is corrected using the phase correction table reflecting the characteristics of the LPF 114, and the three-phase AC voltage feedback after LPF processing is corrected using the gain correction table reflecting the characteristics of the LPF 114. It is characterized in that coordinate conversion of the corrected three-phase AC voltage feedback into q-axis voltage feedback Vq and d-axis voltage feedback Vd is performed based on the later electrical angle.

  As a result, it is possible to generate an electrical angle for compensating the phase delay generated in the LPF 114 and three-phase AC voltage feedback for compensating the amplitude attenuation generated in the LPF 114. Therefore, the q-axis voltage feedback Vq and d axis are generated. The voltage feedback Vd becomes close to the true value. Therefore, the influence of the phase delay and the amplitude attenuation generated in the LPF 114 can be eliminated, and the control accuracy and stability of the motor drive can be further improved.

Example 1
First, the first embodiment will be described. The first embodiment uses the phase correction table to generate an electrical angle for compensating for the phase delay occurring in the LPF, and the 3-phase AC voltage feedback after the LPF processing is q-axis voltage feedback based on the electrical angle. It is an example which carries out coordinate conversion to Vq and d axis voltage feedback Vd.

  FIG. 1 is a block diagram showing an example of the configuration of a motor control device according to a first embodiment. The motor control device 1-1 includes an electrical angular velocity calculator 110, an integrator 111, coordinate converters 112 and 115, a PWM controller 113, an LPF 114, a phase corrector 10 and an adder 12. The phase corrector 10 includes a phase correction table 11.

  The electrical angular velocity calculator 110, the integrator 111, the coordinate converters 112 and 115, the PWM controller 113, and the LPF 114 are the same as the components shown in FIG. The electrical angular velocity calculator 110 outputs the electrical angular velocity ωe to the integrator 111 and the phase corrector 10.

The coordinate converter 112 converts the two-phase q-axis voltage command Vq * and the d-axis voltage command Vd * of the rotating coordinate system into the three-phase U-phase of the stationary coordinate system according to the following equation. Coordinates are converted into voltage command Vu *, V-phase voltage command Vv * and W-phase voltage command Vw *.

  FIG. 3 is a diagram showing a configuration example of the phase correction table 11. The phase correction table 11 is a table in which the phase characteristic of the LPF 114 with respect to the frequency is reflected, and the electric angular velocity ωe and the phase correction angle Δθe are stored in advance as a pair.

  The phase characteristic with respect to the frequency of the LPF 114 is known, and the amount of phase delay due to the frequency of the LPF 114 is also known. There is a relationship of ω = 2πf between the angular velocity ω and the frequency f. Therefore, the user can determine the relationship between the electrical angular velocity ωe and the phase correction angle Δθe according to the known phase characteristic with respect to the frequency of the LPF 114, and can configure the phase correction table 11.

  The phase correction angle Δθ e of the phase correction table 11 is an angle for correcting the phase delay of the U phase voltage feedback Vu, the V phase voltage feedback Vv and the W phase voltage feedback Vw generated in the LPF 114 to 0 °. For example, when the LPF 114 having the phase characteristic shown in FIG. 9 is used, since the phase at the cutoff frequency fc = 1 kHz is −90 °, the phase correction table 11 sets the electrical angular velocity ωe = 2πfc = 6283 rad / s. Correspondingly, a phase correction angle Δθe = 90 ° is stored.

  Referring back to FIG. 1, the phase corrector 10 receives the electrical angular velocity ωe from the electrical angular velocity calculator 110, and reads the phase correction angle Δθe corresponding to the electrical angular velocity ωe from the phase correction table 11. Then, the phase corrector 10 outputs the phase correction angle Δθ e to the adder 12.

  Here, instead of using the phase correction table 11, the phase correction unit 10 uses the mathematical expression in which the relationship between the electrical angular velocity ωe and the phase correction angle Δθe is defined, to obtain the phase correction angle Δθe corresponding to the electric angular velocity ωe. It may be calculated.

  The adder 12 receives the electrical angle θe from the integrator 111, receives the phase correction angle Δθe from the phase corrector 10, adds the phase correction angle Δθe to the electric angle θe, and corrects the corrected electric angle (after addition electric Find the angle θe ′. Then, the adder 12 outputs the corrected electrical angle θe ′ to the coordinate converter 115.

  The coordinate converter 115 receives the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the LPF processing from the LPF 114, and receives the corrected electrical angle θe ′ from the adder 12.

  The coordinate converter 115 converts the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the LPF processing of the stationary coordinate system into the rotational coordinate system based on the corrected electrical angle θe ′. Coordinate conversion is performed to 2-phase q-axis voltage feedback Vq and d-axis voltage feedback Vd.

  As a result, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd have values obtained by compensating for the phase delay generated in the LPF 114.

Here, the coordinate converter 115 performs coordinate conversion according to the following equation.

  As described above, according to the motor control device 1-1 of the first embodiment, the phase correction unit 10 uses the phase correction table 11 in which the phase characteristic of the LPF 114 is reflected, and the phase correction angle corresponding to the electrical angular velocity ωe Identify Δθe. The adder 12 adds the phase correction angle Δθe to the electric angle θe to obtain a corrected electric angle θe ′.

  The coordinate converter 115 converts the U-phase voltage feedback Vu, the V-phase voltage feedback Vv and the W-phase voltage feedback Vw after the LPF processing into the q-axis voltage feedback Vq and the d-axis voltage feedback Vd based on the corrected electrical angle θe ′. Coordinate conversion to

  As a result, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd become close to the true values obtained by compensating for the phase delay generated in the LPF 114. The processes such as slip calculation control and constant output control are performed using the q-axis voltage feedback Vq and the d-axis voltage feedback Vd close to the true value. Therefore, it is possible to improve control accuracy and stability of motor drive.

  The first embodiment can be particularly applied to the high-order LPF 114 in which phase delay is likely to occur in the used frequency band and amplitude attenuation is less likely to occur. For example, in FIG. 9, when the frequency range is 300 Hz or less as the use frequency band, phase delay occurs in the use frequency band, but the gain does not change and is constant, so amplitude attenuation hardly occurs.

  When such an LPF 114 is used, the motor control device 1-1 corrects the electrical angle θe using the phase correction table 11, and compensates only for the phase delay generated in the LPF 114. The q-axis voltage feedback Vq and d axis Generate voltage feedback Vd. As a result, it is possible to improve control accuracy and stability of motor drive only by compensating for the phase delay.

Example 2
Next, Example 2 will be described. The second embodiment generates a 3-phase AC voltage feedback after LPF processing for compensating for the amplitude attenuation generated in the LPF using the gain correction table, and based on the electrical angle generated using the phase correction table, This is an example of coordinate conversion of the three-phase AC voltage feedback into q-axis voltage feedback and d-axis voltage feedback.

  FIG. 2 is a block diagram showing a configuration example of a motor control device according to a second embodiment. The motor control device 1-2 includes an electrical angular velocity calculator 110, an integrator 111, coordinate converters 112 and 115, a PWM controller 113, an LPF 114, a phase corrector 10, an adder 12, a gain corrector 13 and a multiplier 15. -1, 15-2, and 15-3. The phase corrector 10 includes a phase correction table 11, and the gain corrector 13 includes a gain correction table 14.

  The electrical angular velocity calculator 110, the integrator 111, the coordinate converter 112, the PWM controller 113, the LPF 114, the phase corrector 10, and the adder 12 are the same as the components shown in FIGS.

  The electrical angular velocity calculator 110 outputs the electrical angular velocity ωe to the integrator 111, the phase corrector 10, and the gain corrector 13. The LPF 114 outputs the U-phase voltage feedback Vu after the LPF processing to the multiplier 15-1, outputs the V-phase voltage feedback Vv to the multiplier 15-2, and outputs the W-phase voltage feedback Vw to the multiplier 15-3. Do.

  FIG. 4 is a diagram showing an example of the configuration of the gain correction table 14. The gain correction table 14 is a table in which the gain characteristic with respect to the frequency of the LPF 114 is reflected, and the electric angular velocity ωe and the gain correction coefficient k are stored in advance as a pair.

  The gain characteristics for the frequency of the LPF 114 are known, and the amount of amplitude attenuation due to the frequency of the LPF 114 is also known. Therefore, the user can determine the relationship between the electrical angular velocity ωe and the gain correction coefficient k according to the known gain characteristics for the frequency of the LPF 114, and can configure the gain correction table 14. The gain correction coefficient k of the gain correction table 14 is a coefficient for correcting the amplitude attenuation of the U phase voltage feedback Vu, the V phase voltage feedback Vv and the W phase voltage feedback Vw generated in the LPF 114 to zero.

  Returning to FIG. 2, the gain corrector 13 receives the electrical angular velocity ωe from the electrical angular velocity calculator 110, and reads the gain correction coefficient k corresponding to the electrical angular velocity ωe from the gain correction table 14. Then, the gain correction unit 13 outputs the gain correction coefficient k to the multipliers 15-1, 15-2, and 15-3.

  Note that the gain correction unit 13 does not use the gain correction table 14 but uses a mathematical formula in which the relationship between the electrical angular velocity ωe and the gain correction coefficient k is defined to obtain the gain correction coefficient k corresponding to the electrical angular velocity ωe. It may be calculated.

  The multiplier 15-1 receives the U-phase voltage feedback Vu after the LPF processing from the LPF 114, and receives the gain correction coefficient k from the gain corrector 13. Then, the multiplier 15-1 multiplies the U phase voltage feedback Vu after the LPF processing by the gain correction coefficient k, and outputs the U phase voltage feedback Vu 'after correction (after multiplication) to the coordinate converter 115.

  Thus, the U-phase voltage feedback Vu 'after correction has a value obtained by compensating for the amplitude attenuation generated in the LPF 114.

  The multiplier 15-2 receives the V-phase voltage feedback Vv after the LPF processing from the LPF 114, and receives the gain correction coefficient k from the gain corrector 13. Then, the multiplier 15-2 multiplies the V phase voltage feedback Vv after the LPF processing by the gain correction coefficient k, and outputs the corrected V phase voltage feedback Vv ′ to the coordinate converter 115.

  As a result, the V-phase voltage feedback Vv 'after correction has a value obtained by compensating for the amplitude attenuation generated in the LPF 114.

  The multiplier 15-3 receives the W-phase voltage feedback Vw after the LPF processing from the LPF 114, and receives the gain correction coefficient k from the gain corrector 13. Then, the multiplier 15-3 multiplies the W phase voltage feedback Vw after the LPF processing by the gain correction coefficient k, and outputs the corrected W phase voltage feedback Vw ′ to the coordinate converter 115.

  Thus, the W-phase voltage feedback Vw 'after correction has a value obtained by compensating for the amplitude attenuation generated in the LPF 114.

  The coordinate converter 115 inputs the U-phase voltage feedback Vu ', the V-phase voltage feedback Vv' and the W-phase voltage feedback Vw 'after correction from the multipliers 15-1, 15-2, 15-3, and an adder. Input the corrected electrical angle θe ′ from 12.

  The coordinate converter 115 rotates the U-phase voltage feedback Vu ′, the V-phase voltage feedback Vv ′ and the W-phase voltage feedback Vw ′ after the three-phase correction of the stationary coordinate system on the basis of the corrected electrical angle θe ′. Coordinate conversion is performed to two-phase q-axis voltage feedback Vq and d-axis voltage feedback Vd of the system.

  As a result, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd have values obtained by compensating for the phase delay and the amplitude attenuation generated in the LPF 114.

  As described above, according to the motor control device 1-2 of the second embodiment, the gain correction unit 13 uses the gain correction table 14 in which the gain characteristic of the LPF 114 is reflected, and the gain correction coefficient corresponding to the electrical angular velocity ωe Identify k. The multipliers 15-1, 15-2, and 15-3 multiply the U phase voltage feedback Vu, the V phase voltage feedback Vv, and the W phase voltage feedback Vw after the LPF processing by the gain correction coefficient k, and the corrected U phase Voltage feedback Vu ', V-phase voltage feedback Vv' and W-phase voltage feedback Vw 'are obtained.

  The coordinate converter 115 corrects the U-phase voltage feedback Vu ′, the V-phase voltage feedback Vv ′ and the W-phase voltage feedback Vw ′ after the correction based on the corrected electrical angle θe ′ obtained by the adder 12 as the q-axis voltage. Coordinate conversion is performed to feedback Vq and d-axis voltage feedback Vd.

  As a result, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd become close to the true values obtained by compensating for the phase delay and the amplitude attenuation generated in the LPF 114. The processes such as slip calculation control and constant output control are performed using the q-axis voltage feedback Vq and the d-axis voltage feedback Vd close to the true value. Therefore, it is possible to further improve the control accuracy and stability of the motor drive.

  The second embodiment can be particularly applied to the low-order LPF 114 which is prone to phase delay and amplitude attenuation in the used frequency band. When such an LPF 114 is used, the motor control device 1-2 corrects the electrical angle θe using the phase correction table 11, and uses the gain correction table 14 to perform U-phase voltage feedback Vu, V after the LPF processing. The phase voltage feedback Vv and the W phase voltage feedback Vw are corrected. Then, the motor control device 1-2 generates the q-axis voltage feedback Vq and the d-axis voltage feedback Vd in which the phase delay and the amplitude attenuation generated in the LPF 114 are compensated. As a result, it is possible to further improve the control accuracy and stability of the motor drive by compensating the phase delay and the amplitude attenuation.

〔Concrete example〕
Next, the case where the LPF 114 is a first-order LPF and the case where it is a second-order LPF will be specifically described. Hereinafter, the amplitude of the sine wave in each frequency component is referred to as a gain.

FIG. 10 is a diagram showing a configuration example of an active type first-order LPF. The cutoff angular frequency ωc of this first-order LPF is represented by ωc = 1 / CR. C represents capacitance, and R represents resistance. Further, assuming that DC gain G ₀ = 1, the transfer function of the first-order LPF is expressed by the following equation.

Therefore, the gain | G (jω) | and the phase θ in the case of the first-order LPF are expressed by the following equations.

FIG. 11 is a diagram showing a configuration example of an active type second-order LPF. The cutoff angular frequency ωc of this second-order LPF is represented by ωc = 1 / √ (C ₁ R ₁ C ₂ R ₂ ), and the attenuation ratio ζ is ζ = (1⁄2ωcC ₁ ) (1 / R ₁ + 1 / It is represented by R ₂ ). C ₁ and C ₂ indicate capacitances, and R ₁ and R ₂ indicate resistance values.

Further, the transfer function of the second-order LPF is expressed by the following equation.

Therefore, the gain | G (jω) | and the phase θ in the case of the second-order LPF are expressed by the following equations.

Further, when the LPF 114 is a second-order LPF and the gain of the pass band is a Butterworth filter having a flat (hereinafter, referred to as second-order Butterworth LPF), the attenuation ratio 表 is expressed by the following equation.

Therefore, the gain | G (jω) | and the phase θ in the case of the second-order Butterworth LPF are expressed by the following equations.

Here, it is assumed that the switching frequency of the PWM inverter is set to fsw = 5 kHz, and the cutoff frequency of the LPF 114 is set to fc = 1 kHz. The cutoff angular frequency ωc is calculated by the following equation.

  FIG. 5 is a diagram showing an example of the gain characteristic and the phase characteristic of the LPF 114, and shows a case where the LPF 114 is a first-order LPF and a case where the second-order Butterworth LPF is. It is assumed that the rotation frequency fe (= ωe / 2π) of the PWM inverter is up to 300 Hz (the use rotation frequency band is up to 300 Hz).

  The gain | G (jω) | when the frequency f of the waveform input to the LPF 114 is a switching frequency fsw = 5000 Hz is 0.196 for the first-order LPF and 0.040 for the second-order Butterworth LPF. Therefore, it is understood that the gain | G (jω) | when the frequency f is the switching frequency fsw = 5000 Hz or more attenuates by about 80% or more in the first-order LPF and attenuates by 96% or more in the second-order Butterworth LPF.

  On the other hand, in the case of the rotational frequency fe = 300 Hz which is the maximum rotational frequency used of the PWM inverter, the gain | G (jω) | = 0.958 and the phase θ = −16.7 in the first-order LPF, and the second-order Butterworth LPF The gain | G (jω) | = 0.996, and the phase θ = −25.0. Therefore, when the frequency f of the waveform input to the LPF 114 is the rotational frequency fe = 300 Hz, which is the maximum frequency of use of the PWM inverter, the gain | G (jω) | It turns out that it is 16.7 degrees behind. Further, it is understood that in the second-order Butterworth LPF, the gain is attenuated by 0.4% and the phase θ is delayed by 25.0 °.

  Here, the user can calculate a gain correction coefficient k for correcting the gain attenuation occurring in the LPF 114 to 0 from the gain characteristics shown in FIG. Further, the user can calculate the phase correction angle Δθe for correcting the phase delay occurring in the LPF 114 to 0 from the phase characteristics shown in FIG.

  FIG. 6 is a diagram showing the gain correction coefficient k and the phase correction angle Δθ e obtained from the example of the gain characteristic and the phase characteristic of FIG. 5, and the gain correction for making the gain attenuation and the phase delay occurring at the LPF 114 be zero. The coefficient k and the phase correction angle Δθe are shown.

  When the LPF 114 is a first-order LPF, the electrical angular velocity ωe = 2πfe corresponding to the rotation frequency fe and the gain correction coefficient k shown in α1 of FIG. 6 are stored in the gain correction table 14 shown in FIG. Be done. Further, the electric angular velocity ωe = 2πfe corresponding to the rotational frequency fe and the phase correction angle Δθe shown in β1 of FIG. 6 are stored as a pair of data in the phase correction table 11 shown in FIG.

  On the other hand, when the LPF 114 is a second-order Butterworth LPF, the gain correction table shown in FIG. 4 shows electric angular velocity ωe = 2πfe corresponding to the rotational frequency fe and a gain correction coefficient k shown in α2 of FIG. Stored in Further, the electric angular velocity ωe = 2πfe corresponding to the rotational frequency fe and the phase correction angle Δθe shown in β2 of FIG. 6 are stored as a pair of data in the phase correction table 11 shown in FIG.

  Thus, by using the phase correction table 11 in which the data shown in FIG. 6 is stored, the phase correction angle Δθe read from the phase correction table 11 is added to the electric angle θe to correct the corrected electric angle θe ′ is obtained. Then, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd are obtained based on the corrected electrical angle θe ′. Thus, the q-axis voltage feedback Vq and the d-axis voltage feedback Vd become data in which the phase delay generated in the LPF 114 is compensated.

  Further, by using the gain correction table 14 in which the data shown in FIG. 6 is stored, the three-phase AC voltage feedback after LPF processing is multiplied by the gain correction coefficient k read from the gain correction table 14, A corrected three-phase AC voltage feedback is required. As a result, the corrected three-phase AC voltage feedback becomes data in which the gain attenuation generated in the LPF 114 is compensated.

  When the LPF 114 is a second-order Butterworth LPF, as shown in FIG. 5, the gain | G (jω) | is in the working rotation frequency band up to the rotation frequency fe = 300 Hz which is the use maximum rotation frequency of the PWM inverter. It is almost one. In this case, it is not necessary to consider the gain attenuation that occurs in the LPF 114. Therefore, when the LPF 114 is a second-order Butterworth LPF, the gain correction table 14 need not be used, and the first embodiment shown in FIG. 1 using only the phase correction table 11 is applied.

1, 100 Motor controller 10 Phase corrector 11 Phase correction table 12 Adder 13 Gain corrector 14 Gain correction table 15 Multiplier 101 Motor 110 Electric angular velocity calculator 111 Integrator 112, 115 Coordinate converter 113 PWM controller 114 LPF
Vq * q axis voltage command Vd * d axis voltage command Vq q axis voltage feedback Vd d axis voltage feedback Vu * U phase voltage command Vv * V phase voltage command Vw * W phase voltage command Vuv * Between U phase and V phase Voltage command Vuvo Output voltage between U phase and V phase U phase output voltage, U phase voltage feedback Vvo V phase output voltage, V phase voltage feedback Vwo W phase output voltage, W phase voltage feedback Vuv U phase and V Phase feedback Vu LPF after U phase feedback Vv LPF after V phase feedback Vw LPF after W phase feedback Vu 'after correction U phase feedback Vv' after correction V phase feedback Vw 'after correction W after correction Phase feedback ωe Electric angular velocity θe Electric angle Δθe Phase correction angle θe 'Corrected electric angle (added electric angle)
fe rotation frequency (= ωe / 2π)
k gain correction coefficient f frequency θ phase ω angular velocity G ₀ DC gain | G (jω) | gain fc cut-off frequency fsw switching frequency ωc cut-off angular frequency 減 衰 attenuation ratio

Claims (4)

In a motor control device that controls a motor based on a predetermined q-axis voltage command and a d-axis voltage command,
An electrical angular velocity calculator for calculating the electrical angular velocity of the motor;
An integrator that integrates the electrical angular velocity calculated by the electrical angular velocity calculator to obtain an electrical angle of the motor;
A first coordinate converter that coordinate-converts the q-axis voltage command and the d-axis voltage command into a three-phase AC voltage command based on the electrical angle obtained by the integrator;
A PWM controller that generates a PWM signal for generating a three-phase AC voltage to be supplied to the motor based on the three-phase AC voltage command subjected to coordinate conversion by the first coordinate converter;
An LPF that inputs the three-phase AC voltage as a three-phase AC voltage feedback from a voltage detector that detects the three-phase AC voltage, and performs an LPF process on the three-phase AC voltage feedback;
The phase correction angle corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator using a phase correction table in which the phase characteristic of the LPF is reflected and the electrical angular velocity and the phase correction angle are stored as a pair A phase corrector that identifies
An adder which adds the phase correction angle specified by the phase corrector to the electric angle obtained by the integrator, and outputs an electric angle after addition;
A second coordinate transformation for coordinate-converting the 3-phase AC voltage feedback subjected to the LPF processing by the LPF into q-axis voltage feedback and d-axis voltage feedback based on the post-addition electrical angle output by the adder And the
A motor control apparatus comprising: In the motor control device according to claim 1,
Furthermore, the gain corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator using a gain correction table in which the gain characteristic of the LPF is reflected and the electrical angular velocity and the gain correction coefficient are stored as a pair is stored. A gain corrector that specifies a correction factor;
The three-phase AC voltage feedback subjected to the LPF processing by the LPF is multiplied by the gain correction coefficient specified by the gain corrector, and a multiplier that outputs the multiplied three-phase AC voltage feedback is provided.
The second coordinate converter is
The three-phase AC voltage feedback output from the multiplier is coordinate-converted to q-axis voltage feedback and d-axis voltage feedback based on the post-addition electrical angle output from the adder. Motor control device. In the motor control device according to claim 1,
The phase corrector is
Instead of the phase correction table, the electrical angular velocity calculated by the electrical angular velocity calculator is calculated using a formula in which the phase characteristic of the LPF is reflected and the relationship between the electrical angular velocity and the phase correction angle is defined. A motor control apparatus characterized in that the corresponding phase correction angle is specified. In the motor control device according to claim 2,
The gain compensator is
The gain correction coefficient corresponding to the electrical angular velocity calculated by the electrical angular velocity calculator is specified using a mathematical expression in which the gain characteristic of the LPF is reflected and the relationship between the electrical angular velocity and the gain correction coefficient is defined. A motor control device characterized by.