JP2018182942A - Motor control device, sheet conveying device, and image forming apparatus - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor control apparatus that suppresses deterioration in current detection accuracy caused by reducing a step generated in a detected current waveform.SOLUTION: A current value generator 530a includes a detection control unit 901, a detection value generation unit 902, a correction value storage unit 904, and a correction control unit 903. The detection control unit outputs a control command to the detection value generation unit by a drive voltage Vα, and the detection value generation unit generates a current value Isns from a detection voltage value Vsns proportional to a motor current on the basis of the control command and carrier wave information. The correction control unit corrects the current value by adding a correction value read from a correction storage unit to the current value and outputs the corrected value as the corrected current value iα.SELECTED DRAWING: Figure 8

Description

  The present invention relates to a motor control device that controls a motor, a sheet conveying device, and an image forming apparatus.

  Conventionally, in drive control of a motor, there is known a control method of detecting a current value of a drive current flowing through a winding of the motor and controlling a drive current supplied to the winding based on the detected current value.

  FIG. 14 is a diagram showing an example of the configuration of a motor drive circuit 50 for driving a motor. As shown in FIG. 14, the motor drive circuit 50 has FETs Q1 to Q4 as switching elements, a winding L1 of the motor, and the like. Specifically, the FETs Q1 to Q4 form an H bridge circuit, and the winding L1 is connected to connect the connection point of the FETs Q1 and Q3 and the connection point of the FETs Q2 and Q4. Also, the drain terminals of the FETs Q1 and Q2 are connected to the 24V power supply terminal, and the source terminals of the FETs Q3 and Q4 are connected to one end of the resistor 200. Furthermore, the other end of the resistor 200 is connected to ground (GND). That is, the resistor is grounded. The motor has a winding of the first phase and a winding of the second phase, and a motor drive circuit is provided corresponding to each of the first phase and the second phase of the motor. Furthermore, it is assumed that the drive of the motor drive circuit is performed independently for each phase. The control of the motor is performed based on the current value detected in the first phase and the current value detected in the second phase.

  The FETs Q1 and Q4 are driven by the PWM signal PWM +, and the FETs Q2 and Q3 are driven by the PWM signal PWM-. Note that PWM + and PWM− are in opposite phase to each other. That is, when PWM + is 'H (high level)', PWM- is 'L (low level)'. When PWM- is 'H', PWM + is 'L'. When the PWM signal is 'H', the operation of the FET is on, and when the PWM signal is 'L', the operation of the FET is off.

  FIG. 15 is a time chart showing the relationship between the current value I of the drive current flowing through the PWM +, the PWM- and the winding L1.

  In FIG. 15, a period T1 is a period in which the drive current I flowing in the winding L1 is positive, that is, the drive current I flows in the direction of the arrow shown in FIG. Further, a period T2 is a period in which the drive current I flowing through the winding L1 is negative, that is, the drive current I flows in the reverse direction to the arrow shown in FIG.

  In the period T1, when the PWM + is 'H (high level)', the drive current flows in the order of the power supply, the FET Q1, the winding L1, the FET Q4, and the GND. Thereafter, when the PWM + becomes 'L (low level)', an induced electromotive force is generated in the winding L1 in the direction of blocking the change of the current. As a result, drive current flows in the order of GND, FET Q3, winding L1, FET Q2 and power supply. In addition, in the period T2, when PWM + is “L”, the drive current flows in the order of the power supply, the FET Q2, the winding L1, the FET Q3, and the GND. Thereafter, when PWM + becomes 'H', induced electromotive force is generated in the winding L1 in the direction of blocking the change of the current. As a result, drive current flows in the order of GND, FET Q4, winding L1, FET Q1 and power supply.

  The current value I is detected based on the voltage Vsns applied to the resistor 200. However, as described above, in the period T1, when PWM + is 'H', the drive current flows in the order of the power supply 1, the FET Q1, the winding L1, the FET Q4, and the GND. In addition, in the period T1, when PWM + is “L”, the drive current flows in the order of GND, the FET Q3, the winding L1, the FET Q2, and the power supply 1. That is, in the period T1, there are a case where the drive current flows in a direction from the power supply side to GND, and a case where the drive current flows in the direction from GND to the power supply side. The same applies to the period T2. Therefore, when a resistor is provided at the position shown in FIG. 14 and the drive current I is detected based on the voltage Vsns across the resistor, the direction of the actual current flowing through the resistor and the direction of the detected drive current are It does not match.

  In Patent Document 1, a resistor is provided between the motor drive circuit and the ground. Further, a switching means is provided for switching the polarity of the voltage Vsns across the resistor, and the switching means performs switching of the polarity of Vsns according to the PWM signal.

  In the configuration described in Patent Document 1, the switching element can respond to the switching between 'H' and 'L' of PWM + due to the short time interval for switching 'H' and 'L' of PWM +. It is conceivable that there is no case. In this case, although it is not necessary to switch the polarity of Vsns, the polarity of Vsns is switched, and the direction of the actual current flowing through the resistor may not match the direction of the detected drive current.

  Therefore, the applicant has proposed a configuration for detecting the current value in the longer one of the period (high period) in which PWM + is 'H' and the period (low period) in which 'L'. Specifically, as shown in FIG. 16, the applicant detects the current value during the high period when the duty ratio (the ratio of the high period to one cycle of PWM +) is 50% or more, and the duty ratio is 50 In the case of less than%, a configuration is proposed to detect the current value during the low period. By using such a configuration, it can be suppressed that the direction of the actual current flowing through the resistor and the direction of the detected drive current do not match.

  Furthermore, when detecting the current value by the above-described method, the applicant changes when the duty ratio changes from less than 50% to 50% or more, or when the DUTY ratio changes from 50% or more. A configuration is proposed to reduce the distortion of the current waveform that occurs when changing to a value less than%. Specifically, the applicant corrects the current value detected in the high period with the predetermined correction value C1, and corrects the current value detected in the low period with the predetermined correction value C2. Proposes a configuration for reducing distortion of the current waveform shown in FIG. The correction value C1 is a value for decreasing the detected current value, and the correction value C2 is a value for increasing the detected current value.

  FIG. 18 is a diagram for explaining a method of determining the correction values C1 and C2. FIG. 18 shows a current waveform in one cycle of the PWM signal having a duty ratio of 50%. The correction value C1 mentioned above is a value for reducing the detected current value by the difference value Δi1 shown in FIG. 18, and the correction value C2 mentioned above is shown in FIG. 18 for the detected current value. Is a value to be increased.

JP-A-8-99645

  When the current value is corrected by the method described above, for example, the current value detected when the duty ratio is 70% is determined based on the current value detected when the duty ratio is 50%. It is corrected by the correction values C1 and C2.

  However, the difference values Δi1 and Δi2 have different values depending on the duty ratio. Specifically, as the duty ratio increases, the difference value Δi1 decreases and the difference value Δi2 increases.

  Therefore, when the current value detected in the high period with the duty ratio of 70% is corrected by the correction value C1 that decreases the difference value Δi1, the value after correction becomes smaller than the value of the current that actually flows There is a possibility of In addition, when the current value detected in the low period with the duty ratio of 30% is corrected by the correction value C2 that increases the difference value Δi2, the value after correction becomes larger than the value of the current that actually flows There is a possibility of As a result, as shown in FIG. 19, the amplitude of the detected current waveform may be smaller than the amplitude of the current actually flowing through the motor winding.

  If control of the drive current flowing through the winding is performed based on a current waveform whose amplitude is smaller than the amplitude of the current actually flowing through the motor winding, the drive current flowing through the winding can not be accurately controlled. There is sex. As a result, there is a possibility that the control of the motor can not be performed with high accuracy.

  In view of the above problems, the present invention has an object to suppress the reduction in the amplitude of the waveform due to the reduction in the level difference generated in the waveform of the detected drive current.

In order to solve the above-mentioned subject, the motor control device concerning the present invention is:
A drive circuit including a plurality of switching elements forming an H-bridge circuit and to which a winding of a motor is connected;
A PWM signal that controls the on operation and the off operation of the plurality of switching elements, the first level signal being one of high level and low level, and the second level signal being the other of high level and low level Pulse generation means for generating a PWM signal composed of
Correction means for detecting the current value of the drive current flowing through the winding at a timing determined by the duty ratio of the PWM signal generated by the pulse generation means, and correcting the detected current value;
Have
The correction means detects the current value of the drive current during the first period in which the PWM signal is at the first level when the duty ratio is larger than a predetermined value, and the duty ratio is smaller than the predetermined value. Detects the current value of the drive current during a second period in which the PWM signal is at the second level,
Furthermore, the correction means corrects the current value detected in the high period in which the PWM signal is the high level based on a first correction value for decreasing the current value, and the PWM signal is at the low level. Correcting a current value detected in a low period based on a second correction value that increases the current value,
Furthermore, the correction means further corrects the current value corrected based on the first correction value or the second correction value, based on a third correction value that increases the absolute value of the current value.
The pulse generation means generates the PWM signal based on the current value corrected by the correction means based on the third correction value.

  According to the present invention, the current value corrected based on the first correction value or the second correction value is further corrected based on the third correction value that increases the absolute value of the current value. As a result, it is possible to suppress the reduction of the amplitude of the waveform due to the reduction of the step generated in the waveform of the detected drive current.

FIG. 1 is a cross-sectional view illustrating an image forming apparatus according to a first embodiment. FIG. 3 is a block diagram showing a control configuration of the image forming apparatus. It is a figure which shows the relationship between the two-phase motor which consists of A phase and B phase, and the rotating coordinate system represented by d axis | shaft and q axis | shaft. It is a block diagram showing composition of a motor control device concerning a 1st embodiment. It is a figure which shows the structure of a motor drive part. It is a figure explaining the composition where a PWM signal is generated. It is a figure explaining the method by which a PWM signal is generated. It is a figure which shows the structure of a current value generator. It is a flowchart explaining the method to determine a correction value. It is a figure which shows the waveform of the drive current in each duty ratio. It is a figure explaining the method to correct | amend the amplitude of the current waveform after correction | amendment. It is a flowchart which shows the control method of the motor which concerns on 1st Embodiment. It is a block diagram showing composition of a motor control device which performs speed feedback control. FIG. 6 is a diagram showing an example of a configuration of a motor drive circuit 50. 5 is a time chart showing the relationship between current value I of drive current flowing through PWM +, PWM− and winding L1. It is a figure which shows the relationship between a duty ratio and the electric current value detected. It is a figure which shows the waveform of the detected electric current. It is a figure explaining the method to determine a correction value. It is a figure which shows the amplitude of a current waveform.

  The preferred embodiments of the present invention will be described below with reference to the drawings. However, the shapes of the component parts described in this embodiment and their relative positions and the like should be appropriately changed according to the configuration of the apparatus to which the present invention is applied and various conditions, and the scope of the present invention is It is not a thing of the meaning limited to the following embodiment. In the following description, although the case where the motor control device is provided in the image forming apparatus will be described, the provision of the motor control device is not limited to the image forming apparatus. For example, it is also used in a sheet conveying apparatus for conveying a sheet such as a recording medium or an original.

First Embodiment
[Image forming apparatus]
FIG. 1 is a cross-sectional view showing the configuration of a monochrome electrophotographic copying machine (hereinafter referred to as an image forming apparatus) 100 having a sheet conveying apparatus used in the present embodiment. The image forming apparatus is not limited to a copying machine, and may be, for example, a facsimile machine, a printing machine, or a printer. Further, the recording method is not limited to the electrophotographic method, and may be, for example, an inkjet or the like. Furthermore, the format of the image forming apparatus may be either monochrome or color.

  The configuration and functions of the image forming apparatus 100 will be described below with reference to FIG. As shown in FIG. 1, the image forming apparatus 100 includes a document feeding device 201, a reading device 202, and an image printing device 301.

  The originals stacked on the original stacking unit 203 of the original feeding device 201 are fed one by one by the paper feed roller 204 and conveyed along the conveyance guide 206 onto the original glass table 214 of the reading device 202. Further, the document is conveyed by a conveying belt 208 at a constant speed, and is discharged by a discharge roller 205 to a discharge tray (not shown). Reflected light from the original image illuminated by the illumination 209 at the reading position of the reading device 202 is guided to the image reading unit 111 by the optical system including the reflecting mirrors 210, 211 and 212, and converted into an image signal by the image reading unit 111. Be done. The image reading unit 111 includes a lens, a CCD as a photoelectric conversion element, a drive circuit of the CCD, and the like. The image signal output from the image reading unit 111 is output to the image printing apparatus 301 after various correction processes are performed by the image processing unit 112 configured by a hardware device such as an ASIC. As described above, the document is read. That is, the document feeding device 201 and the reading device 202 function as a document reading device.

  Further, as a document reading mode, there are a first reading mode and a second reading mode. The first reading mode is a mode in which an image of a document conveyed at a constant speed is read by an illumination system 209 and an optical system fixed at a predetermined position. The second reading mode is a mode in which an image of a document placed on a document glass 214 of the reading device 202 is read by an illumination system 209 moving at a constant speed and an optical system. Usually, an image of a sheet-like document is read in a first reading mode, and an image of a bound document such as a book or a booklet is read in a second reading mode.

  Sheet storage trays 302 and 304 are provided inside the image printing apparatus 301. The sheet storage trays 302 and 304 can store different types of recording media. For example, A4 size plain paper is stored in the sheet storage tray 302, and thick sheet of A4 size is stored in the sheet storage tray 304. The recording medium is a medium on which an image is formed by the image forming apparatus. For example, a sheet, a resin sheet, a cloth, an OHP sheet, a label, and the like are included in the recording medium.

  The recording medium stored in the sheet storage tray 302 is fed by the paper feed roller 303, and is delivered to the registration roller 308 by the transport roller 306. Further, the recording medium stored in the sheet storage tray 304 is fed by the paper feed roller 305, and is fed to the registration roller 308 by the conveyance rollers 307 and 306.

  The image signal output from the reader 202 is input to an optical scanning device 311 including a semiconductor laser and a polygon mirror. In addition, the outer peripheral surface of the photosensitive drum 309 is charged by the charger 310. After the outer peripheral surface of the photosensitive drum 309 is charged, a laser beam corresponding to an image signal input from the reading device 202 to the light scanning device 311 is photosensitive from the light scanning device 311 via the polygon mirror and mirrors 312 and 313. The outer peripheral surface of the drum 309 is irradiated. As a result, an electrostatic latent image is formed on the outer peripheral surface of the photosensitive drum 309. For charging the photosensitive drum, for example, a charging method using a corona charger or a charging roller is used.

  Subsequently, the electrostatic latent image is developed by the toner in the developing device 314, and a toner image is formed on the outer peripheral surface of the photosensitive drum 309. The toner image formed on the photosensitive drum 309 is transferred onto the recording medium by a transfer charger 315 provided at a position (transfer position) facing the photosensitive drum 309. At the same time as the transfer timing, the registration roller 308 feeds the recording medium to the transfer position.

  As described above, the recording medium to which the toner image has been transferred is fed into the fixing device 318 by the conveyance belt 317, and is heated and pressurized by the fixing device 318 to fix the toner image on the recording medium. In this manner, the image forming apparatus 100 forms an image on the recording medium.

  When the image formation is performed in the single-sided printing mode, the recording medium having passed through the fixing unit 318 is discharged by the paper discharge rollers 319 and 324 to a paper discharge tray (not shown). When image formation is performed in the double-sided printing mode, the recording medium is subjected to the fixing process on the first surface of the recording medium by the fixing device 318, and then the recording medium is discharged from the discharge roller 319, the conveyance roller 320, and the reverse roller 321. Are transported to the reverse path 325. Thereafter, the recording medium is conveyed again by the conveyance rollers 322 and 323 to the registration roller 308, and an image is formed on the second surface of the recording medium by the method described above. Thereafter, the recording medium is discharged to a discharge tray (not shown) by discharge rollers 319 and 324.

  Further, when the recording medium on which the image is formed on the first surface is discharged to the outside of the image forming apparatus 100 with the face down, the recording medium having passed through the fixing unit 318 passes the discharge roller 319 and the conveyance roller 320 It is transported in the direction of Thereafter, the rotation of the conveyance roller 320 is reversed immediately before the trailing edge of the recording medium passes through the nip portion of the conveyance roller 320, and the recording medium is discharged from the paper discharge roller with the first surface of the recording medium facing downward. The image is discharged to the outside of the image forming apparatus 100 via 324.

  This completes the description of the configuration and functions of the image forming apparatus 100. The load in the present invention is an object driven by a motor. For example, various rollers (conveying rollers) such as the sheet feeding rollers 204, 303, and 305, the registration roller 308, and the discharging roller 319, the photosensitive drum 309, the conveying belts 208 and 317, the illumination system 209, the optical system, etc. Respond to the load. The motor control device of the present embodiment can be applied to a motor that drives these loads.

  FIG. 2 is a block diagram showing an example of a control configuration of the image forming apparatus 100. As shown in FIG. As shown in FIG. 2, the system controller 151 includes a CPU 151a, a ROM 151b, and a RAM 151c. In addition, the system controller 151 is connected to the image processing unit 112, the operation unit 152, an analog / digital (A / D) converter 153, a high voltage control unit 155, a motor control device 600, sensors 159, and an AC driver 160. . The system controller 151 can transmit and receive data and commands to and from each connected unit.

  The CPU 151a executes various sequences related to a predetermined image forming sequence by reading out and executing various programs stored in the ROM 151b.

  The RAM 151 c is a storage device. The RAM 151 c stores, for example, various data such as setting values for the high voltage control unit 155, command values for the motor control device 600, and information received from the operation unit 152.

  The system controller 151 transmits, to the image processing unit 112, setting value data of various devices provided inside the image forming apparatus 100, which are necessary for image processing in the image processing unit 112. Furthermore, the system controller 151 receives the signal from the sensors 159 and sets the setting value of the high voltage control unit 155 based on the received signal. The high voltage control unit 155 supplies necessary voltages to the high voltage unit 156 (the charger 310, the developing device 314, the transfer charger 315, and the like) according to the setting value set by the system controller 151. The sensors 159 include a sensor that detects a recording medium conveyed by the conveyance roller.

  The motor control device 600 controls the motor 509 that drives the load in accordance with the command output from the CPU 151a. Although only the motor 509 is illustrated as a motor of the image forming apparatus in FIG. 2, it is assumed that the image forming apparatus is actually provided with a plurality of motors. In addition, one motor control device may be configured to control a plurality of motors. Furthermore, although only one motor control device is provided in FIG. 2, it is assumed that a plurality of motor control devices are actually provided in the image forming apparatus.

  The A / D converter 153 receives a detection signal detected by the thermistor 154 for detecting the temperature of the fixing heater 161, converts the detection signal from an analog signal to a digital signal, and transmits the digital signal to the system controller 151. The system controller 151 controls the AC driver 160 based on the digital signal received from the A / D converter 153. The AC driver 160 controls the fixing heater 161 so that the temperature of the fixing heater 161 becomes a temperature required to perform the fixing process. The fixing heater 161 is a heater used for the fixing process, and is included in the fixing unit 318.

  The system controller 151 displays the operation screen for the user to set the type of recording medium to be used (hereinafter referred to as a paper type) and the like on the display unit provided in the operation unit 152. Control. The system controller 151 receives information set by the user from the operation unit 152, and controls an operation sequence of the image forming apparatus 100 based on the information set by the user. The system controller 151 also transmits information indicating the state of the image forming apparatus to the operation unit 152. The information indicating the state of the image forming apparatus is, for example, information on the number of formed images, the progress of the image forming operation, and jamming and double feeding of sheet materials in the document reading apparatus 201 and the image printing apparatus 301. The operation unit 152 displays the information received from the system controller 151 on the display unit.

  As described above, the system controller 151 controls the operation sequence of the image forming apparatus 100.

[Motor control unit]
Next, the motor control device in the present embodiment will be described. The motor control device in the present embodiment controls the motor using vector control.

<Vector control>
First, a method of the motor control device 600 according to the present embodiment performing vector control will be described using FIGS. 3 and 4. Although a motor such as a rotary encoder for detecting a rotational phase of a rotor of the motor is not provided in the motor in the following description, a sensor such as a rotary encoder may be provided.

  FIG. 3 shows a stepping motor (hereinafter referred to as a motor) 509 consisting of two phases of A-phase (first phase) and B-phase (second phase), and a rotating coordinate system represented by d-axis and q-axis. FIG. In FIG. 3, in the stationary coordinate system, an α axis which is an axis corresponding to the winding of the A phase and a β axis which is an axis corresponding to the winding of the B phase are defined. Also, in FIG. 3, the d-axis is defined along the direction of the magnetic flux produced by the magnetic poles of the permanent magnets used in the rotor 402, and is a direction 90 degrees counterclockwise from the d-axis (orthogonal to the d-axis The q axis is defined along the The angle between the α axis and the d axis is defined as θ, and the rotational phase of the rotor 402 is represented by the angle θ. In the vector control, a rotational coordinate system based on the rotational phase θ of the rotor 402 is used. Specifically, in vector control, a q-axis component (torque current component) that generates a torque in the rotor, which is a current component in a rotating coordinate system of a current vector corresponding to a drive current flowing through a winding, and a winding The d-axis component (excitation current component) that affects the strength of the magnetic flux passing through is used.

  The vector control is a motor by performing phase feedback control that controls the value of the torque current component and the value of the excitation current component so that the deviation between the command phase representing the target phase of the rotor and the actual rotational phase is reduced. Control method to control the Also, the motor is controlled by performing speed feedback control that controls the value of the torque current component and the value of the excitation current component so that the deviation between the commanded speed representing the target speed of the rotor and the actual rotational speed becomes small. There is also a way.

  FIG. 4 is a block diagram showing an example of the configuration of a motor control device 600 that controls the motor 509. As shown in FIG. The motor control device 600 according to this embodiment includes a motor control unit 157 that controls the motor using vector control, and a motor drive unit 158 that supplies a drive current to the winding of the motor to drive the motor. Motor control device 600 is configured of at least one ASIC, and executes each function described below.

  As shown in FIG. 4, the motor control unit 157 supplies drive current to the phase controller 502, current controller 503, coordinate inverse converter 505, coordinate converter 511, and motor winding as a circuit that performs vector control. PWM inverter 506 and the like. The coordinate converter 511 represents a current vector corresponding to the drive current flowing through the A-phase and B-phase windings of the motor 509 in the q-axis and d-axis from the stationary coordinate system represented by the α-axis and Coordinate conversion to a rotational coordinate system. As a result, the drive current flowing through the winding is represented by the current value (q-axis current) of the q-axis component which is the current value in the rotational coordinate system and the current value (d-axis current) of the d-axis component. The q-axis current corresponds to a torque current that causes the rotor 402 of the motor 509 to generate torque. The d-axis current corresponds to an excitation current that affects the strength of the magnetic flux passing through the winding of the motor 509, and does not contribute to the generation of torque of the rotor 402. The motor control unit 157 can independently control the q-axis current and the d-axis current. As a result, the motor control unit 157 can efficiently generate the torque necessary for the rotor 402 to rotate by controlling the q-axis current according to the load torque applied to the rotor 402.

  The motor control unit 157 determines the rotation phase θ of the rotor 402 of the motor 509 according to a method described later, and performs vector control based on the determination result. The CPU 151a generates a command phase θ_ref representing a target phase of the rotor 402 of the motor 509, and outputs the command phase θ_ref to the motor control unit 157 at a predetermined time cycle.

  The subtractor 101 calculates the deviation between the rotational phase θ of the rotor 402 of the motor 509 and the command phase θ_ref, and outputs the deviation to the phase controller 502.

  The phase controller 502 controls the q-axis current command value iq_ref and the d-axis so that the deviation output from the subtractor 101 becomes smaller based on proportional control (P), integral control (I), and differential control (D). The current command value id_ref is generated and output. Specifically, the phase controller 502 sets the q-axis current command value iq_ref and the d-axis current command value id_ref such that the deviation output from the subtractor 101 is 0 based on P control, I control, and D control. Generate and output The P control is a control method of controlling the value of an object to be controlled based on a value proportional to the deviation between the command value and the estimated value. Moreover, I control is a control method which controls the value of the control object based on the value proportional to the time integral of the deviation of a command value and an estimated value. Moreover, D control is a control method which controls the value of the object to control based on the value proportional to the time change of the deviation of a command value and an estimated value. The phase controller 502 in this embodiment generates the q-axis current command value iq_ref and the d-axis current command value id_ref based on PID control, but is not limited to this. For example, the phase controller 502 may generate the q-axis current command value iq_ref and the d-axis current command value id_ref based on PI control. When a permanent magnet is used for the rotor 402, the d-axis current command value id_ref that normally affects the strength of the magnetic flux passing through the winding is set to 0, but is not limited to this.

The drive current flowing through the A-phase and B-phase windings of the motor 509 is detected by the motor drive unit 158 according to a method described later. The current value of the drive current detected by the motor drive unit 158 is expressed by the following equation using the phase θe of the current vector shown in FIG. 3 as the current values iα and iβ in the stationary coordinate system. The phase θe of the current vector is defined as the angle between the α axis and the current vector. Also, I indicates the magnitude of the current vector.
iα = I * cos θe (1)
iβ = I * sin θe (2)

  The current values iα and iβ are input to the coordinate converter 511 and the induced voltage determiner 512.

The coordinate converter 511 converts the current values iα and iβ in the stationary coordinate system into the current value iq of the q-axis current and the current value id of the d-axis current in the rotating coordinate system by the following equations.
id = cosθ * iα + sinθ * iβ (3)
iq = −sin θ * iα + cos θ * iβ (4)

  The subtractor 102 receives the q-axis current command value iq_ref output from the phase controller 502 and the current value iq output from the coordinate converter 511. The subtractor 102 calculates a deviation between the q-axis current command value iq_ref and the current value iq, and outputs the deviation to the current controller 503.

  Further, the d-axis current command value id_ref output from the phase controller 502 and the current value id output from the coordinate converter 511 are input to the subtractor 103. The subtractor 103 calculates a deviation between the d-axis current command value id_ref and the current value id, and outputs the deviation to the current controller 503.

  The current controller 503 generates drive voltages Vq and Vd based on PID control so that the deviations become smaller. Specifically, the current controller 503 generates the drive voltages Vq and Vd so that the deviations become zero, respectively, and outputs the drive voltages Vq and Vd to the coordinate inverse transformer 505. That is, the current controller 503 functions as a voltage generation unit. Although the current controller 503 in the present embodiment generates the drive voltages Vq and Vd based on PID control, the present invention is not limited to this. For example, the current controller 503 may generate the drive voltages Vq and Vd based on PI control.

The coordinate inverse transformer 505 inversely converts the drive voltages Vq and Vd in the rotational coordinate system output from the current controller 503 into drive voltages Vα and Vβ in the stationary coordinate system according to the following equations.
Vα = cos θ * Vd-sin θ * Vq (5)
Vβ = sin θ * Vd + cos θ * Vq (6)

  The coordinate inverse transformer 505 outputs the inversely converted drive voltages Vα and Vβ to the induced voltage determiner 512, the PWM inverter 506, and the current value generator 530. The current value generator 530 will be described later.

  The PWM inverter 506 has a full bridge circuit. The full bridge circuit (motor drive circuit) has a configuration similar to that of the motor drive circuit 50 described with reference to FIG. That is, one end of the winding L1 is connected to the conducting wire connecting the FETs Q1 and Q3, and the other end is connected to the conducting wire connecting the FETs Q2 and Q4. In FIG. 18, winding L 1 is actually a winding provided in motor 509. That is, the winding L1 is provided outside the motor control device 157. The full bridge circuit is driven by a PWM (Pulse Width Modulation) signal based on the drive voltages Vα and Vβ input from the coordinate inverse converter 505. As a result, the PWM inverter 506 generates drive currents iα and iβ according to the drive voltages Vα and Vβ, and drives the motor 509 by supplying the drive currents iα and iβ to the windings of each phase of the motor 509. . In the present embodiment, the PWM inverter has a full bridge circuit, but the PWM inverter may be a half bridge circuit or the like.

Next, the configuration for determining the rotational phase θ will be described. For determining the rotational phase θ of the rotor 402, values of induced voltages Eα and Eβ induced in the A-phase and B-phase windings of the motor 509 by the rotation of the rotor 402 are used. The value of the induced voltage is determined (calculated) by the induced voltage determiner 512. Specifically, the induced voltages Eα and Eβ are input from the A / D converter 510 to the current value iα and iβ input to the induced voltage determiner 512, and from the coordinate inverse converter 505 to the induced voltage determiner 512. It is determined from the drive voltages Vα and Vβ by the following equation.
Eα = Vα-R * iα-L * diα / dt (7)
Eβ = Vβ-R * iβ-L * diβ / dt (8)

  Here, R is a winding resistance, and L is a winding inductance. The values of R and L are values specific to the motor 509 being used, and are stored in advance in a memory (not shown) or the like provided in the ROM 151 b or the motor control device 600.

  The induced voltages Eα and Eβ determined by the induced voltage determiner 512 are output to the phase determiner 513.

The phase determiner 513 determines the rotational phase θ of the rotor 402 of the motor 509 according to the following equation based on the ratio between the induced voltage Eα and the induced voltage Eβ output from the induced voltage determiner 512.
θ = tan ^ −1 (−Eβ / Eα) (9)

  In the present embodiment, the phase determiner 513 determines the rotational phase θ by performing the calculation based on the equation (9), but the present invention is not limited to this. For example, a table indicating the relationship between the induced voltage Eα and the induced voltage Eβ and the rotational phase θ corresponding to the induced voltage Eα and the induced voltage Eβ is stored in the ROM 151 b or the like, and the rotational phase θ is obtained by referring to the table. You may decide.

  The rotational phase θ of the rotor 402 obtained as described above is input to the subtractor 101, the coordinate inverse transformer 505, and the coordinate converter 511.

  The motor control device 157 repeatedly performs the control described above.

  As described above, the motor control device 157 in this embodiment performs vector control using phase feedback control for controlling the current value in the rotational coordinate system so that the deviation between the command phase θ_ref and the rotational phase θ becomes small. By performing the vector control, it is possible to suppress that the motor is out of step and that the motor noise increases and the power consumption increases due to the surplus torque. Further, by performing phase feedback control, it is possible to control the rotational phase of the rotor so that the rotational phase of the rotor becomes a desired phase. Therefore, in the image forming apparatus, image control on a recording medium is performed by applying vector control by phase feedback control to a motor that drives a load (for example, a registration roller) that needs to control the rotational phase with high accuracy. It can be done properly.

[Motor drive unit]
As described above, in drive control of the motor, the current value of the drive current flowing through the winding is detected, and the drive current flowing through the winding is controlled based on the detected current value. That is, in drive control of the motor, a configuration for detecting the current value of the drive current flowing through the winding and a configuration for supplying the drive current to the winding are required.

  FIG. 5 is a diagram showing an example of the configuration of the motor drive unit 158 in the present embodiment. As shown in FIG. 5, the motor drive unit 158 includes a PWM inverter 506 a, an A / D converter 510 a, and a current value generator 530 a in phase A. In addition, the motor drive unit 158 includes a PWM inverter 506b in phase B, an A / D converter 510b, and a current value generator 530b. The PWM inverter 506 shown in FIG. 4 includes a PWM inverter 506a and a PWM inverter 506b. Further, the A / D converter 510 shown in FIG. 4 includes an A / D converter 510a and an A / D converter 510b. Furthermore, the current value generator 530 shown in FIG. 4 includes a current value generator 530a and a current value generator 530b. Thus, the PWM inverter, the A / D converter, and the current value generator are provided corresponding to each of the A phase and the B phase of the motor 509, and are independently driven for each phase. Since the configuration of the PWM inverter 506a and the configuration of the PWM inverter 506b are the same, FIG. 5 shows a specific configuration of the PWM inverter 506a. The PWM inverter 506a amplifies a voltage signal of both ends of the motor drive circuit 50a, the PWM generator 203 which generates a PWM signal for controlling the on / off operation of a plurality of FETs provided in the motor drive circuit 50a, The amplifier 300 is provided.

  Further, as shown in FIG. 5, the motor control unit 157 determines the value of the induced voltage generated in the winding of the B phase and the induced voltage determiner 512a that determines the value of the induced voltage generated in the winding of the A phase. And an induced voltage determiner 512b. The induced voltage determiner 512 shown in FIG. 4 includes an induced voltage determiner 512a and an induced voltage determiner 512b.

  Hereinafter, a method of the motor drive unit 158 supplying a drive current to the A-phase winding and a method of detecting the current value of the drive current flowing to the A-phase winding by the motor drive unit 158 will be described. The B phase has the same configuration as that of the A phase, and therefore the description thereof is omitted. The description of the same parts as the contents already described in FIGS. 14 to 19 will be omitted.

<Method of supplying driving current>
First, a method in which the motor drive unit 158 supplies drive current to the winding will be described.

  FIG. 6 is a diagram for explaining a configuration in which the PWM generator 203 in the present embodiment generates a PWM signal. As shown in FIG. 6, the PWM generator 203 has a comparator 203a that compares the modulated wave with the carrier wave to generate a PWM signal. In the present embodiment, it is assumed that the PWM generator 203 generates a triangular wave carrier of a predetermined frequency. Specifically, for example, it is assumed that the PWM generator 203 generates a triangular wave carrier wave of a frequency corresponding to a cycle in which the CPU 151a outputs a command (command phase θ_ref) to the motor control device 600. Also, assuming that the period from the timing when the value of the triangular wave carrier becomes the minimum value to the timing when the value becomes the next minimum is one period, the waveform of one cycle of the triangular wave carrier (from the minimum value to the next minimum value) is The waveform is symmetrical with respect to the timing at which the value of the triangular wave carrier becomes the maximum value. Also, the triangular wave carrier in the A phase is synchronized with the triangular wave carrier in the B phase.

  FIG. 7 is a diagram for explaining how the PWM generator 203 generates a PWM signal. Hereinafter, a method of generating a PWM signal by the PWM generator 203 will be described with reference to FIGS. 5 to 7.

  As shown in FIG. 5, the drive voltage Vα output from the motor control unit 157 is input to the PWM generator 203. The PWM generator 203 uses the comparator 203a to compare the drive voltage Vα as a modulation wave with the triangular wave carrier, and 'H' for the period (high period) where the drive voltage Vα is larger than the triangular wave carrier, the drive voltage Vα is a triangular wave A period (low period) smaller than the carrier wave generates PWM + as 'L'. In addition, the PWM generator 203 generates PWM− in which the phase of PWM + is inverted. Thus, the PWM generator 203 functions as a pulse generator.

  As shown in FIG. 5, the PWM generator 203 outputs PWM + to the FETs Q1 and Q4, and outputs PWM- to the FETs Q2 and Q3. The on / off operation of the FETs Q1 to Q4 is controlled by PWM + and PWM-. As a result, it is possible to control the magnitude and direction of the drive current supplied to the A-phase winding L1.

  In this embodiment, when the drive voltage is 24 V, the duty ratio is 100%, when the drive voltage is 0 V, the duty ratio is 50%, and when the drive voltage is -24 V, the duty ratio is 0%. Do. That is, in the present embodiment, the drive voltage Vα is a value corresponding to the duty ratio of PWM +. In the present embodiment, the ratio of the high period to the period of PWM + is defined as the duty ratio, but the ratio of the low period to the period of PWM + may be defined as the duty ratio.

<Method of detecting drive current>
Next, a method of the motor drive unit 158 detecting the current value of the drive current flowing through the winding will be described.

  As described above, the drive current flowing in the winding is detected based on the voltage Vsns applied to the resistor 200. The amplifier 300 amplifies the signal of the voltage Vsns and outputs it to the A / D converter 510a. The A / D converter 510a converts the voltage Vsns from an analog value to a digital value, and outputs the converted value to the current value generator 530a. Further, the current value generator 530 a receives the drive voltage Vα output from the motor control unit 157 and information on the frequency and phase as information on the triangular wave carrier wave output from the PWM generator 203.

  FIG. 8 is a diagram showing the configuration of the current value generator 530a. The current value generator 530 a includes a detection control unit 901, a detection value generation unit 902, a correction control unit 903, a correction value storage unit 904, and a correction value generation unit 905.

  The detection control unit 901 controls the detection value generation unit 902 and the correction control unit 903 based on the drive voltage Vα output from the motor control unit 157.

  Specifically, when the drive voltage Vα is negative (the duty ratio is less than 50%), the detection control unit 901 controls the detection value generation unit 902 to sample the voltage value Vsns in the low period. The detected value generation unit 902 is configured such that the triangular wave carrier is initially poled after the PWM + generated by the PWM generator 203 falls (switched from 'H' to 'L') based on the command from the detection control unit 901. The voltage value Vsns is sampled at the timing when it becomes a value. Then, the detection value generation unit 902 inverts the polarity of the sampled voltage value Vsns to generate the current value Isns. The detection control unit 901 controls the detection value generation unit 902 to sample the voltage value Vsns in the high period when the drive voltage is 0 V or more (the duty ratio is 50% or more). The detected value generation unit 902 is configured such that the triangular carrier first reaches an extremum after the PWM + generated by the PWM generator 203 has risen (switched from 'L' to 'H') based on the command from the detection control unit 901. The voltage value Vsns is sampled at the timing when Then, the detected value generation unit 902 generates the current value Isns without inverting the polarity of the sampled voltage value Vsns. The detection value generation unit 902 may perform processing of inverting the polarity of the current value Isns after generating the current value Isns based on the sampled voltage value Vsns.

  As described above, by sampling the voltage value Vsns at the timing when the triangular wave carrier becomes the extreme value, it is possible to prevent the digital value from being sampled at the timing when the PWM signal rises or falls. As a result, noise generated due to switching of the switching element when the PWM signal rises or falls can be suppressed from being included in the sampled value.

<Method to correct current value Isns>
Next, a method of correcting the detected current value Isns will be described. In the present embodiment, by applying the following configuration to the motor drive unit 158, the level difference generated in the detected current waveform is reduced.

  FIG. 18 is a diagram for explaining a method of acquiring a correction value for correcting the current value. FIG. 18 shows the drive current I (t), sampling timings ts1 and ts2, and linear reference functions ref1 (t) and ref2 (t) used as a reference for correction of the current value Isns. In addition, t shows time. The reference function ref1 (t) indicates a case where the drive current I linearly increases during the H period of the PMW signal (PWM +), and the reference function ref2 (t) indicates the drive current I during the L period of the PMW signal (PWM +). Indicates a linear decrease. Note that FIG. 18 shows, as an example, a case where the PWM generation unit 203 generates a PWM signal with a duty ratio DR = 50%.

  As shown in FIG. 18, the value I (ts1) of the drive current at the sampling timing ts1 in the H period is higher by Δi1 than the reference value ref1 (ts1) indicated by the reference function. Therefore, the correction value C1 in the H period is determined as C1 = −Δi1 so as to decrease the detected value Isns of the drive current I by Δi1. Further, the value I (ts2) of the drive current at the detection timing ts2 in the L period is lower than the reference value ref2 (ts2) indicated by the reference function by Δi2. Therefore, the correction value C2 in the L period is set to C2 = Δi2 so as to increase the detection value Isns of the drive current I by Δi2. The correction values C1 and C2 are stored in advance in the correction value storage unit 904 shown in FIG.

  The detection control unit 901 instructs the correction control unit 903 which of the correction value C1 for the H period and the correction value C2 for the L period should be used to correct the detected value Isns. Specifically, when the duty ratio is 50% or more, the detection control unit 901 controls the correction control unit 903 to correct the current value Isns based on the correction value C1. Further, when the duty ratio is less than 50%, the detection control unit 901 controls the correction control unit 903 to correct the current value Isns based on the correction value C2.

  The correction control unit 903 corrects the detection value Isns generated by the detection value generation unit 902 in accordance with an instruction from the detection control unit 901. Specifically, the correction control unit 903 corrects the detection value Isns by adding the correction value C1 or C2 read from the correction value storage unit 904 to the detection value Isns, and outputs the corrected value as the current value Iα. Do.

<Method to determine correction value>
Next, a specific method of determining the correction values C1 and C2 will be described. As described above, in order for the correction values C1 and C2 to be determined, it is necessary to determine Δi1 and Δi2.

  In the present embodiment, Δi1 and Δi2 are determined based on the current value detected in the state where the duty ratio is 50%, but the duty ratio may not be 50%.

  FIG. 9 is a flowchart for explaining the process of acquiring the correction value. The process of acquiring the correction value in the present embodiment will be described below with reference to FIG. The processing of this flowchart is executed by the motor control device 600 that has received an instruction from the CPU 151a.

  First, in S101, the PWM generation unit 203 generates PWM signals (PWM + and PWM−) having a duty ratio of 50%, and outputs the PWM signals to the motor drive circuit 50. As a result, the drive current I shown in FIG. 14 is supplied to the coil L1 of the motor.

  Next, in S102, the current value generator 530a (the detected value generation unit 902) generates the voltage value Vsns at both the detection timing ts1 in the H period (high period) and the detection timing ts2 in the L period (low period). The sampling is performed to generate the detection value I (ts1) sns of the drive current at the detection timing ts1 and the detection value I (ts2) of the drive current at the detection timing ts2.

Next, in S103, the current value generator 530a (correction control unit 903) calculates the difference Idiff between the detection value I (ts1) and the detection value I (ts2) using the following equation (10).
Idiff = I (ts1) -I (ts2) (10)

  Thereafter, in S104, current value generator 530a (correction control unit 903) determines correction values C1 and C2 by calculating Δi1 and Δi2 shown in FIG. 18 based on difference Idiff calculated in S103. . Specifically, the current value generator 530a (correction control unit 903) determines the correction values C1 and C2 by dividing the difference Idiff calculated in S103 by 2 to calculate Δi1 and Δi2.

  Hereinafter, a method of determining the correction values C1 and C2 will be described. In the following description, the voltage E is a voltage applied to the coil L1 by the drive power source, R is a resistance component of the coil L1, and L is a reactance component (inductance) of the coil L1.

Δi1 and Δi2 shown in FIG. 18 are expressed by the following equations (11) and (12).
Δi1 = I (ts1) −ref1 (ts1) (11)
Δi2 = ref2 (ts2) −I (ts2) (12)

As described above, the reference functions ref1 (t) and ref2 (t) are linearly changing functions. The detection timing ts1 is the center timing of the H period, and the detection timing ts2 is the center timing of the L period. Therefore, as shown in FIG. 16, ref1 (ts1) and ref2 (ts2) become (E / R) / 2. As a result, the equations (11) and (12) are converted into the following equations (13) and (14).
Δi1 = − (E / R) / 2 + I (ts1) (13)
Δi2 = (E / R) / 2-I (ts2) (14)

As a result, the absolute value of the difference Idiff is expressed by the following equation (15) using the equations (13) and (14).
| Idiff | = | I (ts1) -I (ts2) | = Δi1 + Δi2 (15)

  As described above, the absolute value of the difference Idiff of the detection value Isns at the detection timings ts1 and ts2 is equivalent to the sum of Δi1 and Δi2.

Further, assuming that the start timings of the H period and the L period are each t = 0, the current value i1 (t) of the drive current at time t in the H period is expressed by the following equation (16). Further, the current value i2 (t) of the drive current at time t in the L period is expressed by the following equation (17).
i1 (t) = E / R * (1-e- ^{(R / L) t} ) (16)
i2 (t) = E / R * e ^{− (R / L) t} (17)

  Note that i1 (t) and i2 (t) correspond to transient solutions for the current flowing in the coil L1 when the coil L1 of the motor is approximated by a series circuit of its reactance component and resistance component. Expression (16) represents a transient current when the voltage E is applied to the start timing of the H period of the PWM signal. Further, equation (17) represents a transient current when the applied voltage is switched to 0 [V] at the start timing of the L period of the PWM signal in a state where the drive current has reached E / R by the application of voltage E. ing.

Further, assuming that the start timings of the H period and the L period are t = 0 respectively, the reference functions ref1 (t) and ref2 (t) are expressed by the following equations (18) and (19).
ref1 (t) = E / R / T1 * t (18)
ref2 (t) = E / R * (1- (t / T2)) (19)

  T1 is a value representing the length of the H period, and T2 is a value representing the length of the L period. When the duty ratio DR is 50%, T1 and T2 have the same value.

Furthermore, the difference between the current value i1 (t) of the drive current I and the reference function ref1 (t), that is, the difference i1diff (t) between the equation (16) and the equation (18) is represented by the following equation (20) Be done. Further, the difference between the current value i2 (t) of the drive current I and the reference function ref2 (t), that is, the difference i2diff (t) between the equation (17) and the equation (19) is given by the following equation (21) Be done.
i1diff (t) = E / R * (1-e- ^{(R / L) t-} t / T1) (20)
i2diff (t) =-E / R * (1-e- ^{(R / L) t-} t / T2) (21)

As described above, when the duty ratio DR is 50%, T1 and T2 have the same value. Therefore, the absolute value of i1diff (t) and the absolute value of i2diff (t) are equal (| i1diff (t) | = || i2diff (t) |). Further, since the detection timing ts1 is T1 / 2 and the detection timing ts2 is T2 / 2, when the duty ratio DR is 50%, the time from the start of the H period to the detection timing ts1 and L The time from the start of the period to the detection timing ts2 is equal. Therefore, the relationship shown in the following equation (22) can be obtained from the equations (13), (14), (20) and (21).
Δi1 = | i1diff (T / 2) | = | i2diff (T / 2) | = Δi2 (22)
That is, Δi1 = Δi2 is established.

From the above, it is possible to divide the absolute value (| Idiff |) of the difference between the detected value I (ts1) and the detected value I (ts2) by 2 as shown in the equations (15) and (22). , Δi1 and Δi2 can be calculated. Specifically, Δi1 and Δi2 can be calculated using the following equation (23).
| Idiff | / 2 = (Δi1 + Δi2) / 2 = Δi1 = Δi2 (23)

The current value generator 530a (correction control unit 903) determines the correction values C1 and C2 as in the following equations (24) and (25) based on Δi1 and Δi2 obtained as described above in S103. Do.
C1 = −Δi1 (24)
C2 = Δi2 (25)

  Next, in S105, the current value generator 530a (correction control unit 903) stores the determined correction values C1 and C2 in the correction value storage unit 904.

  As described above, the current value generator 530a acquires the correction value.

<Method for correcting the amplitude of the current waveform after correction>
As described above, in the present embodiment, the correction values C1 and C2 are determined based on the drive current in the state where the duty ratio is 50%. The current value detected in the H period (that is, detected when the duty ratio is 50% or more) is corrected by the correction value C1, and the current value detected in the L period (that is, the duty ratio is 50). The current value detected when it is less than% was corrected by the correction value C2.

  FIG. 10 is a diagram showing the waveform of the drive current at each duty ratio. FIG. 10A is a diagram showing the waveform of the drive current in the state where the duty ratio is 50%. FIG. 10B is a diagram showing the waveform of the drive current in the state where the duty ratio is 75%. FIG. 10C is a diagram showing the waveform of the drive current in the state where the duty ratio is 100%. As shown in FIG. 10, the difference value Δi between the current value I (ts1) of the drive current and the reference function ref1 (ts1) differs depending on the duty ratio. Specifically, as the duty ratio increases, the difference value Δi1 decreases. Therefore, when the current value detected in period H (that is, the current value detected at a duty ratio of 50% or more) is corrected using correction value C1 based on difference value Δi at a duty ratio of 50%, the corrected value becomes It may be smaller than the value of the current actually flowing. In addition, when the current value detected in the L period (that is, the current value detected at a duty ratio of less than 50%) is corrected using the correction value C2, the value after correction becomes larger than the value of the current actually flowing There is a possibility of As a result, as shown in FIG. 19, the amplitude of the detected current waveform may be smaller than the amplitude of the current actually flowing through the motor winding.

  So, in this embodiment, control of the drive current which flows into a winding performed based on the detected drive current is performed with high precision by applying the following composition.

FIG. 11 is a diagram for explaining a method of correcting the amplitude of the current waveform after correction. The current I actually flowing through the winding shown in FIG. 11 is expressed by the following equation (26).
I = (E * duty ratio−E / 2) * 2} / R (26)

  In addition, E is a power supply voltage and is a resistance value of the winding of R motor. Expression (26) is an expression representing a case where the current flowing through the winding is approximated to change linearly in accordance with the duty ratio.

  In the present embodiment, as an example, FIG. 11 shows the current I when the resistance value R of the motor winding is 12Ω and the power supply voltage E is 24V. Hereinafter, the case where the resistance value R of the motor winding is 12Ω and the power supply voltage E is 24 V will be described, but the values of the resistance R and the power supply voltage E are not limited to this.

  As shown in FIG. 11, the absolute value of the current value corrected based on the difference value Δi at a duty ratio of 50% is smaller than the absolute value of the current value of the actual current. This means that the amplitude of the current waveform after correction is smaller than the amplitude of the actual current waveform.

  In the present embodiment, the amplitude of the current waveform after correction is corrected by correcting the slope of the straight line representing the current after correction shown in FIG. Specifically, the amplitude of the current waveform after correction is corrected by correcting the inclination of the straight line representing the current after correction to the inclination of the straight line representing the actual current. The correction method will be described below.

In FIG. 11, when a point where the duty ratio is 0% and the current I is 0A is defined as the origin (0, 0), the straight line representing the actual current is point B (100, 2) and point C (50, 0). ) And a straight line passing through. Further, a straight line representing the current after correction is a straight line passing through the point A (100, 2-Δi) and the point C (50, 0). Therefore, if a correction coefficient for correcting the slope of the straight line representing the current after correction to the slope of the straight line representing the actual current is defined as G, the following equation (27) holds.
(2-Δi) / 50 * G = 2/50 (27)

Based on equation (27), the correction coefficient G is expressed by the following equation (28).
G = 2 / (2-Δi) (28)

  The correction control unit 903 further corrects by multiplying the current value corrected by the correction value C1 or C2 by the correction coefficient G thus obtained, and outputs the corrected current value as the current value iα. As shown in equation (28), the correction coefficient G is a value of 1 or more. Therefore, the absolute value of the current value can be increased by multiplying the current value corrected by the correction value C1 or C2 by the correction coefficient G. As a result, it can be suppressed that the amplitude of the waveform of the detected current actually becomes smaller than the amplitude of the current flowing through the motor winding. The process of determining the correction coefficient G is performed when the process of determining the correction values C1 and C2 is performed, and the determined correction coefficient G is stored in the correction value storage unit 904.

  FIG. 12 is a flowchart showing a motor control method by the motor control device 600. The control of the motor 509 in the present embodiment will be described below with reference to FIG. The processing of this flowchart is executed by the motor control device 600 that has received an instruction from the CPU 151a.

  First, in S201, when the enable signal 'H' is output from the CPU 151a to the motor control device 600, the motor control device 600 operates based on the command output from the CPU 151a. The enable signal is a signal that permits or prohibits the operation of the motor control device 600. When the enable signal is 'L (low level)', the CPU 151a prohibits the operation of the motor control device 600. That is, the motor control device 600 is stopped. When the enable signal is 'H (high level)', the CPU 151a permits the motor control device 600 to operate. That is, the motor control device 600 operates.

  Next, in S202, the CPU 151a controls the motor control device 600 so as to determine the correction values C1 and C2 and the correction coefficient G. As a result, the motor control device 600 determines the correction values C1 and C2 and the correction coefficient G by the method described above, and stores them in the correction value storage unit 904.

  Thereafter, in S203, the CPU 151a controls the motor control device 600 so as to drive the motor. As a result, the motor control device 600 starts driving of the motor.

  In S204, when the drive voltage output from the motor control unit 157 is 0 V or more (the duty ratio (DR) is 50% or more), the process proceeds to S205. In S205, the detection control unit 901 controls the detection value generation unit 902 so as to sample the voltage value Vsns in the high period and generate the detection value Isns. As a result, the detection value generation unit 902 samples the voltage value Vsns in the high period, and generates the detection value Isns.

  After that, in S206, the detection control unit 901 controls the correction control unit 903 to correct the generated detection value Isns with the correction value C1 stored in the correction value storage unit 904 in S202. As a result, the correction control unit 903 corrects the detection value Isns by adding the correction value C1 (C1 <0) to the acquired detection value Isns. Thereafter, the CPU 151a advances the process to step S207.

  If the drive voltage output from the motor control unit 157 is negative (duty ratio is less than 50%) in S204, the process proceeds to S209. In S209, the detection control unit 901 controls the detection value generation unit 902 so as to sample the voltage value Vsns in the low period and generate the detection value Isns. As a result, the detection value generation unit 902 samples the voltage value Vsns in the low period to generate the detection value Isns.

  Next, in S210, the detection control unit 901 controls the detection value generation unit 902 to perform an inversion process to invert the polarity of the generated detection value Isns (that is, the polarity of the voltage value Vsns). As a result, the detection value generation unit 902 inverts the polarity of the generated detection value Isns (that is, the polarity of the voltage value Vsns).

  After that, in S211, the detection control unit 901 controls the correction control unit 903 to correct the generated detection value Isns with the correction value C2 stored in the correction value storage unit 904 in S202. As a result, the correction control unit 903 corrects the detection value Isns by adding the correction value C2 (C1> 0) to the detection value Isns after the inversion processing. Thereafter, the CPU 151a advances the process to step S207.

  In S207, the correction control unit 903 corrects the current value corrected by the correction value C1 or C2 using the correction coefficient G determined in S202. Then, the correction control unit 903 outputs the corrected current value Iα.

  Thereafter, the motor control device 600 repeatedly performs the above-described control until the CPU 151a outputs the enable signal 'L' to the motor control device 600. The correction values C1 and C2 and the correction coefficient G may or may not be updated each time the driving of the motor is started as shown in FIG.

  As described above, in the present embodiment, the detected current value is corrected using the correction values C1 and C2 determined based on the current value in the state where the duty ratio is 50%. Specifically, when the duty ratio is 50% or more, the detected current value is corrected using the correction value C1, and when the duty ratio is less than 50%, the correction value C2 is used. Correct the detected current value. By using such a configuration, it is possible to reduce the level difference generated in the waveform of the detection current.

  Furthermore, in the present embodiment, the current value corrected by the correction values C1 and C2 is corrected by the correction coefficient G that increases the absolute value of the current value. As a result, it can be suppressed that the amplitude of the waveform of the current corrected by the correction values C1 and C2 is actually smaller than the amplitude of the current flowing through the motor winding.

  By applying these configurations, it is possible to control the drive current flowing through the winding with high accuracy, which is performed based on the detected drive current. As a result, highly accurate control of the motor can be performed.

  In the present embodiment, the correction coefficient G is determined based on the current value at a duty ratio of 50% and the current value at a duty ratio of 100%, but the present invention is not limited to this. For example, the correction coefficient G may be determined based on the current value at a duty ratio of 30% and the current value at a duty ratio of 70%.

  Further, in the present embodiment, the correction coefficient G is determined by approximating that the current flowing in the winding linearly changes in accordance with the duty ratio, but for example, the current flowing in the winding may The correction factor G may be determined by approximating to a function of

  Further, in the present embodiment, the polarity of the voltage in the direction from the power supply side to the GND side of the resistor 200 is positive, and the polarity of the current flowing from the connection point of FETQ1 and FETQ3 to the connection point of FETQ2 and FETQ4 is positive. But it is not this limitation. For example, the polarity of the voltage in the direction from the power supply side to the GND side of the resistor 200 may be positive, and the polarity of the current flowing from the connection point of the FET Q2 and the FET Q4 to the connection point of the FET Q1 and the FET Q3 may be positive. In this case, the polarity of the detected value detected in the high period is inverted, and the polarity of the detected value detected in the low period is not inverted.

  In the present embodiment, based on whether the duty ratio is 50% or more, it is determined whether the sampling of the voltage value Vsns is performed in the high period or in the low period, but it is not limited to this. For example, whether to sample the voltage value Vsns in the high period or in the low period may be determined based on whether the duty ratio is 70% or more. In this case, the correction values C1 and C2 are determined based on the current value in the state where the duty ratio is 70%.

  Moreover, in the present embodiment, detection and correction of the current value are performed based on the level of PWM +, but detection and correction of the current value may be performed based on the level of PWM−. In this case, the correction value C1 for the H period of PWM + is used as a correction value for the L period of PWM-, and the correction value C2 for L period of PWM + is used as a correction value for the H period of PWM- .

  Further, in the present embodiment, the current value generation unit 530 detects the current at the center timing of the H period or at the center timing of the L period, but the timing of detecting the current is limited to the center timing of the period. Absent. For example, the current value generation unit 530 may detect the current at a timing shifted from the center of the period.

  Further, in the present embodiment, a stepping motor is used as a motor for driving a load, but another motor such as a DC motor may be used. Further, the motor is not limited to a two-phase motor, and the present embodiment can be applied to other motors such as a three-phase motor.

Further, in the vector control in the present embodiment, the motor 509 is controlled by performing phase feedback control, but the present invention is not limited to this. For example, the motor 509 may be controlled by feeding back the rotational speed ω of the rotor 402. Specifically, as shown in FIG. 13, a speed determiner 514 is provided inside the motor controller, and the speed determiner 514 generates the rotational speed ω based on the time change of the rotational phase θ output from the phase determiner 513. decide. In addition, Formula (29) shall be used for determination of speed.
ω = dθ / dt (29)

  Then, the CPU 151a outputs a commanded speed ω_ref representing a target speed of the rotor. Furthermore, a speed controller 500 is provided inside the motor controller, and the q-axis current command value iq_ref and the d-axis current command value id_ref are generated so that the speed controller 500 reduces the deviation between the rotational speed ω and the command speed ω_ref. Output. The motor 509 may be controlled by performing such speed feedback control. In such a configuration, since the rotational speed is fed back, the rotational speed of the rotor can be controlled to be a predetermined speed. Therefore, in the image forming apparatus, the speed feedback control is applied to the motor for driving the load (for example, the photosensitive drum, the conveying belt, etc.) which needs to control the rotational speed to a constant speed in order to appropriately form the image on the recording medium. Apply the vector control used. As a result, image formation on the recording medium can be appropriately performed.

  Moreover, this embodiment is applied not only to the structure which has vector control but to what has a structure which detects electric current and feeds back the said detected value.

  Moreover, in this embodiment, although a permanent magnet is used as a rotor, it is not limited to this.

50 Motor drive circuit Q1 to Q4 FET
203 PWM generator 506 PWM inverter 509 stepping motor 530 current value generator

Claims (21)

A drive circuit including a plurality of switching elements forming an H-bridge circuit and to which a winding of a motor is connected;
A PWM signal that controls the on operation and the off operation of the plurality of switching elements, the first level signal being one of high level and low level, and the second level signal being the other of high level and low level Pulse generation means for generating a PWM signal composed of
Correction means for detecting the current value of the drive current flowing through the winding at a timing determined by the duty ratio of the PWM signal generated by the pulse generation means, and correcting the detected current value;
Have
The correction means detects the current value of the drive current during the first period in which the PWM signal is at the first level when the duty ratio is larger than a predetermined value, and the duty ratio is smaller than the predetermined value. Detects the current value of the drive current during a second period in which the PWM signal is at the second level,
Furthermore, the correction means corrects the current value detected in the high period in which the PWM signal is the high level based on a first correction value for decreasing the current value, and the PWM signal is at the low level. Correcting a current value detected in a low period based on a second correction value that increases the current value,
Furthermore, the correction means further corrects the current value corrected based on the first correction value or the second correction value, based on a third correction value that increases the absolute value of the current value.
A motor control apparatus characterized in that the pulse generation means generates the PWM signal based on the current value corrected by the correction means based on the third correction value.   The correction means may determine the third correction value based on a current value in a state in which the duty ratio is a first duty ratio and a current value in a state in which the duty ratio is a second duty ratio. The motor control device according to claim 1.   If the current value corrected based on the first correction value or the second correction value is 0, the current value after the current value is corrected based on the third correction value is 0 The motor control device according to claim 1 or 2, characterized in that:   The correction means multiplies the current value corrected based on the first correction value or the second correction value by the third correction value, based on the first correction value or the second correction value. The motor control device according to any one of claims 1 to 3, wherein the corrected current value is corrected.   The correction means determines the first correction value based on the current value of the drive current detected in the high period in the state where the duty ratio is the predetermined value, and the duty ratio is the predetermined value. The motor control device according to any one of claims 1 to 4, wherein the second correction value is determined based on a current value of the drive current detected in the low period in. The second correction value is a current value of the drive current detected in the high period when the duty ratio is the predetermined value, and the drive detected in the low period when the duty ratio is the predetermined value. It is a value calculated by dividing the absolute value of the difference between the current and the current value by 2.
The motor control device according to any one of claims 1 to 5, wherein the first correction value is a value obtained by inverting the polarity of the second correction value. The motor control device includes control means for controlling the motor based on the current value corrected by the correction means based on the third correction value.
The control means drives the drive circuit such that the deviation between the current value corrected by the correction means based on the third correction value and the current value of the drive current to be supplied to the winding is reduced. Having voltage generating means for generating a voltage;
7. The pulse generation means according to any one of claims 1 to 6, further comprising comparison means for generating the PWM signal by comparing the drive voltage generated by the voltage generation means with a triangular wave as a carrier wave. The motor control device according to any one of the preceding claims.   When the correction means detects the drive current in the high period, the drive current at a timing when the triangular wave first reaches an extreme value after the level of the PWM signal is switched from the low level to the high level. When the drive current is detected in the low period, the drive current is detected at a timing when the triangular wave first reaches an extreme value after the level of the PWM signal switches from the high level to the low level. The motor control device according to claim 7, wherein the motor control device detects the motor.   The invention is characterized in that the correction means does not invert the polarity of the current value of the drive current detected in the first period, but inverts the polarity of the current value of the drive current detected in the second period. The motor control device according to any one of 1 to 8.   The motor control device according to any one of claims 1 to 9, wherein the predetermined value is 50%. The drive circuit is
One end of the first switching element and one end of the second switching element are connected to the power supply,
One end of a third switching element is connected in series to the other end of the first switching element,
One end of a fourth switching element is connected in series to the other end of the second switching element,
A resistor is connected to the other end of the third switching element and the other end of the fourth switching element,
The resistor is grounded,
One end of the winding of the motor is connected to a lead connecting the first switching element and the third switching element, and the other end is connected to a lead connecting the second switching element and the fourth switching element The motor control device according to any one of claims 7 to 10, wherein the motor control circuit is a circuit as described above. The PWM signal generated by the comparison means is supplied to the first switching element and the fourth switching element,
The motor control device according to claim 11, wherein a PWM signal having a phase opposite to that of the PWM signal generated by the comparison means is supplied to the second switching element and the fourth switching element. . The control means
Current obtained by the correction means correcting the induced voltage induced in the first phase winding of the motor and the second phase winding of the motor by the rotation of the rotor of the motor based on the third correction value A means for determining an induced voltage based on the value;
Phase determining means for determining a rotational phase of a rotor of the motor based on the induced voltage of the first phase and the induced voltage of the second phase determined by the induced voltage determining means;
Have
The control means performs rotation based on the rotational phase determined by the phase determination means so that the deviation between the command phase representing the target phase of the rotor and the rotational phase determined by the phase determination means is reduced. Based on the value of the torque current component that causes the rotor to generate torque and the value of the excitation current component that affects the strength of the magnetic flux passing through the motor winding, which are the current components of the current value represented in the coordinate system. The motor control device according to any one of claims 1 to 12, wherein the drive current flowing through the winding is controlled. The control means
An induction voltage that is determined based on the current value corrected by the correction means, the induced voltage induced in the first phase winding of the motor and the second phase winding of the motor by the rotation of the rotor of the motor Voltage determination means,
Phase determining means for determining a rotational phase of a rotor of the motor based on the induced voltage of the first phase and the induced voltage of the second phase determined by the induced voltage determining means;
Speed determining means for determining the rotational speed of the rotor;
Have
The control means performs rotation based on the rotational phase determined by the phase determination means such that a deviation between a commanded speed representing a target speed of the rotor and the rotational speed determined by the speed determination means is reduced. Based on the value of the torque current component that causes the rotor to generate torque and the value of the excitation current component that affects the strength of the magnetic flux passing through the motor winding, which are the current components of the current value represented in the coordinate system. The motor control device according to any one of claims 1 to 12, wherein the drive current flowing through the winding is controlled. The motor control device
A first motor drive circuit to which a first phase winding of the motor is connected;
A second motor drive circuit to which a second phase winding of the motor is connected;
The motor control device according to any one of claims 1 to 14, comprising: A conveyance roller for conveying the sheet;
A motor for driving the transport roller;
The motor control device according to any one of claims 1 to 15,
Have
The sheet conveyance device, wherein the motor control device controls driving of a motor that drives the conveyance roller. A sheet conveying apparatus according to claim 16;
A document loading unit for loading documents;
Have
A document feeding apparatus characterized in that the sheet conveying device feeds the document stacked on the document loading unit. An original feeding device according to claim 17;
Reading means for reading the document fed by the document feeding device;
An original reading apparatus comprising: A sheet conveying apparatus according to claim 16;
Image forming means for forming an image on a recording medium;
Have
An image forming apparatus, wherein the image forming unit forms an image on the recording medium conveyed by the sheet conveying apparatus. An image forming apparatus for forming an image on a recording medium, comprising:
A motor that drives a load,
The motor control device according to any one of claims 1 to 15,
Have
The image forming apparatus, wherein the motor control device controls driving of a motor that drives the load. The image forming apparatus according to claim 20, wherein the load is a conveyance roller that conveys the recording medium.

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