JP2010284017A - AC motor control device - Google Patents

Abstract

In an AC motor control to which synchronous PWM is applied, current offset of the AC motor is suppressed with high responsiveness.
In synchronous PWM, a motor current Iv is sampled in synchronization with the phase of a carrier wave. When twelve current sampling points exist in one electric cycle of the motor current Iv, the offset amount Iof is calculated according to the average value of the latest twelve current sampling values (for one electric cycle). The When the current sampling value I13 is sampled, the offset amount Iof is calculated according to the average value of the current sampling values I2 to I13 instead of the average value of the current sampling values I1 to I12. Then, the control calculation based on the motor current is corrected so as to eliminate the offset amount Iof.
[Selection] Figure 8

Description

  The present invention relates to a control device for an AC motor, and more particularly to motor control by pulse width modulation control in which a carrier frequency is synchronized with a rotation frequency of an AC motor.

  Conventionally, a drive system using an inverter has been adopted for controlling an AC motor. For example, in an electric vehicle such as an electric vehicle, a hybrid vehicle, and a fuel cell vehicle, the output torque of a traveling AC motor (motor generator) is generally controlled by an inverter.

  In the AC motor control, when a DC component (offset) occurs in the motor current, a problem occurs due to an increase in the rotor eddy current. Therefore, it is required to control so that no current offset occurs.

  For example, in Japanese Patent Laid-Open No. 2001-29892 (Patent Document 1), an offset amount is calculated by applying a low-pass filter to a detected value of a drive current when an AC motor is driven, and based on this offset amount. A control configuration for correcting the duty ratio of the drive signal is described.

  Japanese Patent Application Laid-Open No. 8-149882 (Patent Document 2) describes a configuration in which an output current value of a current detector in a state where no drive current is supplied to a motor is stored as an offset value. This offset value is for detecting the offset of the current detector. Furthermore, Japanese Patent Application Laid-Open No. 2004-52403 (Patent Document 3) describes that a filter is provided in order to remove a ripple component of an AC fundamental frequency superimposed on a motor current that flows during rotation when detecting a motor current. ing. Japanese Patent Laid-Open No. 8-340691 (Patent Document 4) describes a control configuration for generating a control signal for an inverter in a state where an offset is applied to a voltage command value.

JP 2001-298990 A JP-A-8-149882 JP 2004-52403 A JP-A-8-340691

  In pulse width modulation (PWM) control, which is generally used for AC motor control using an inverter, on / off of switching elements of the inverter is controlled according to voltage comparison between a voltage command of each phase by current feedback control and a carrier wave of a predetermined frequency. Is done. Therefore, when the carrier frequency is increased, the number of times of switching per unit time increases, so that improvement in control accuracy can be expected, but power loss due to switching loss increases.

  However, if the frequency of the carrier wave (carrier frequency) is lowered in order to reduce the switching loss, securing the positive / negative symmetry of the voltage applied to the AC motor by PWM control becomes a problem. For this reason, so-called synchronous PWM is known in which the carrier frequency is an integral multiple (an integer of 1 or more) of the rotational frequency of the AC motor. According to the synchronous PWM, even if the number of carrier waves (number of carriers) included in the electrical angle of 360 ° (hereinafter also referred to as one electrical cycle) of the AC motor is reduced, it is easy to ensure the positive / negative symmetry of the pulse width voltage. Become. In particular, in synchronous PWM for a three-phase AC motor, it is generally set to an integer multiple of a multiple of 3.

  In the synchronous PWM, the electrical angle at which the switching element is turned on / off is fixed in one electrical cycle. On the other hand, in asynchronous PWM with a fixed carrier frequency, the carrier angle and the rotational frequency of the AC motor do not synchronize, so the electrical angle at which the switching element is turned on and off varies.

  Here, a delay (on / off delay) occurs between the on / off command timing of the switching element by the PWM control and the actual on / off timing due to the gate drive time by the drive circuit of the switching element. Alternatively, in order to stabilize on / off of the switching element such as suppression of a surge voltage, an on / off delay is actively provided by a technique such as changing a gate resistance.

  In asynchronous PWM with a high carrier frequency, the number of times of sampling within one electrical cycle is relatively large, and the on / off delay occurs at various electrical angles, so the influence of the on / off delay on the motor current has no DC component. Appears as current fluctuation. However, in synchronous PWM, the on / off delay occurs at a fixed electrical angle, and therefore there is a possibility that an offset (DC component) may occur in the motor current due to the influence of the on / off delay.

  As described above, according to Patent Document 1, the duty ratio of the drive signal can be corrected so as to eliminate the offset of the motor current. However, in Patent Document 1, since the offset amount is calculated by the low-pass filter, a certain period according to the filter time constant is required until the accurate offset amount is calculated. For this reason, when the amount of current offset changes during operation of the AC motor, it is difficult to suppress the current offset with high responsiveness.

  The present invention has been made to solve such problems, and an object of the present invention is to suppress current offset of an AC motor with high responsiveness in AC motor control to which synchronous PWM is applied. That is.

  An AC motor control device according to the present invention is an AC motor control device in which an applied voltage is controlled by an inverter, and includes a current detector and a motor control unit. The current detector is configured to detect the current of the AC motor. The electric motor control unit is configured to control the inverter according to a control calculation using the current detection value detected by the current detector. The electric motor control unit includes a PWM modulation unit, a carrier wave control unit, a current sampling unit, an extraction unit, and an offset amount calculation unit. The PWM modulation unit controls the pulse width modulation voltage applied from the inverter to the AC motor based on the voltage comparison between the phase voltage command and the carrier wave. The carrier wave control unit is provided to control to generate a carrier wave at a frequency k times (k: natural number) the frequency of the phase voltage command. The current sampling unit detects current by sampling the output of the current detector at a predetermined cycle according to the phase of the carrier wave when the carrier wave control unit controls the frequency of the carrier wave to k times the phase voltage command frequency. Configured to obtain a value. When the carrier frequency is controlled to be k times the frequency of the phase voltage command by the carrier control unit, the extraction unit calculates the current N (N: natural number) from a plurality of sequential current sampling values by the current sampling unit. The latest M current sampling values corresponding to the period are extracted. The offset amount calculation unit is configured to calculate the offset amount of the current detection value according to the average value of the M current sampling values extracted by the extraction unit.

  Preferably, the carrier wave control unit is further configured to control a phase difference of the carrier wave with respect to the phase voltage command. The offset amount calculation unit is configured to calculate an offset amount of the current detection value based on an average value of the M current sampling values and a correction amount corresponding to the phase difference.

  Preferably, the carrier wave control unit is further configured to variably control k in accordance with the operating state of the AC motor. The offset amount calculation unit is configured to calculate the offset amount of the current detection value based on the average value of the M current sampling values and the correction amount according to the phase difference and k.

More preferably, N is 1.
Preferably, the electric motor control unit further includes an offset correction unit configured to correct the control calculation based on the offset amount calculated by the offset amount calculation unit.

  According to the present invention, in an AC motor control to which synchronous PWM is applied, a current offset of the AC motor can be suppressed with high responsiveness.

1 is an overall configuration diagram of a motor drive control system to which an AC motor control device according to an embodiment of the present invention is applied. It is a wave form diagram explaining the basic operation of PWM control. It is a wave form diagram explaining the carrier wave in synchronous PWM. It is a functional block diagram explaining the structure of the PWM control by the control apparatus of the AC motor according to the embodiment of the present invention. It is a wave form diagram explaining on-off of the switching element in PWM control. It is a wave form diagram explaining the offset of the current sampling value in synchronous PWM. It is a conceptual diagram explaining the problem of offset amount calculation by a low-pass filter. FIG. 5 is a waveform diagram for explaining a function of a sampling value extraction unit shown in FIG. 4. It is a flowchart explaining the control processing procedure of detection and correction | amendment of the offset amount of the motor current in the alternating current motor control by this Embodiment. It is a wave form diagram for demonstrating the function by an offset correction part. It is a conceptual diagram which shows the actual current waveform containing a ripple component. It is a wave form diagram explaining the definition of a carrier phase. It is a conceptual diagram explaining the characteristic of the offset amount detection error with respect to a carrier phase. It is a flowchart explaining the control processing procedure of detection and correction | amendment of the offset amount of the motor current in the alternating current motor control by the modification 1 of this Embodiment. It is a conceptual diagram for demonstrating the example of selection of the control mode with respect to the driving | running state of an AC motor. It is a conceptual diagram explaining the characteristic of the offset amount detection error with respect to the number of synchronous pulses and a carrier phase. It is a flowchart explaining the control processing procedure of detection and correction | amendment of the offset amount of the motor current in the alternating current motor control by the modification 2 of this Embodiment.

  Embodiments of the present invention will be described below in detail with reference to the drawings. In the following description, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

(System configuration)
FIG. 1 is an overall configuration diagram of a motor drive control system to which an AC motor control device according to an embodiment of the present invention is applied.

  Referring to FIG. 1, motor drive control system 100 includes a DC voltage generation unit 10 #, a smoothing capacitor C0, an inverter 14, an AC motor M1, and a control device 30.

  For example, AC electric motor M1 generates torque for driving drive wheels of an electric vehicle (referred to as a vehicle that generates vehicle driving force by electric energy such as a hybrid vehicle, an electric vehicle, and a fuel cell vehicle). This is an electric motor for traveling. Alternatively, AC electric motor M1 may be configured to have a function of a generator driven by an engine, or may be configured to have both functions of an electric motor and a generator. Further, AC electric motor M1 may operate as an electric motor for the engine, and may be incorporated in a hybrid vehicle as one that can start the engine, for example. That is, in the present embodiment, the “AC motor” includes an AC drive motor, a generator, and a motor generator (motor generator).

  DC voltage generation unit 10 # includes a DC power supply B, system relays SR1 and SR2, a smoothing capacitor C1, and a converter 12.

  The DC power supply B is typically constituted by a rechargeable power storage device such as a secondary battery such as nickel metal hydride or lithium ion or an electric double layer capacitor. The DC voltage VL output from the DC power supply B and the input / output DC current Ib are detected by the voltage sensor 10 and the current sensor 11, respectively.

  System relay SR 1 is connected between the positive terminal of DC power supply B and power line 6, and system relay SR 1 is connected between the negative terminal of DC power supply B and ground line 5. System relays SR1 and SR2 are turned on / off by a signal SE from control device 30.

  Converter 12 includes a reactor L1, power semiconductor switching elements Q1, Q2, and diodes D1, D2. Power semiconductor switching elements Q 1 and Q 2 are connected in series between power line 7 and ground line 5. On / off of power semiconductor switching elements Q1 and Q2 is controlled by switching control signals SG1 and SG2 from control device 30.

In the embodiment of the present invention, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, or a power bipolar transistor is used as a power semiconductor switching element (hereinafter simply referred to as “switching element”). Etc. can be used. Anti-parallel diodes D1, D2 are arranged for switching elements Q1, Q2. Reactor L1 is connected between a connection node of switching elements Q1 and Q2 and power line 6. Further, the smoothing capacitor C 0 is connected between the power line 7 and the ground line 5.

  Inverter 14 includes a U-phase upper and lower arm 15, a V-phase upper and lower arm 16, and a W-phase upper and lower arm 17 provided in parallel between power line 7 and ground line 5. Each phase upper and lower arm is constituted by a switching element connected in series between the power line 7 and the ground line 5. For example, the U-phase upper and lower arms 15 are composed of switching elements Q3 and Q4, the V-phase upper and lower arms 16 are composed of switching elements Q5 and Q6, and the W-phase upper and lower arms 17 are composed of switching elements Q7 and Q8. Further, antiparallel diodes D3 to D8 are connected to switching elements Q3 to Q8, respectively. Switching elements Q3 to Q8 are turned on / off by switching control signals SG3 to SG8 from control device 30.

  Typically, AC motor M1 is a three-phase permanent magnet type synchronous motor, and is configured by commonly connecting one end of three coils of U, V, and W phases to a neutral point. Furthermore, the other end of each phase coil is connected to the midpoint of the switching elements of the upper and lower arms 15 to 17 of each phase.

  Converter 12 is basically controlled such that switching elements Q1 and Q2 are turned on and off in a complementary manner in each switching period. During the boosting operation, converter 12 boosts DC voltage VL supplied from DC power supply B to DC voltage VH (this DC voltage corresponding to the input voltage to inverter 14 is hereinafter also referred to as “system voltage”). This step-up operation is performed by supplying the electromagnetic energy stored in reactor L1 during the ON period of switching element Q2 to power line 7 via switching element Q1 and antiparallel diode D1.

Converter 12 steps down DC voltage VH to DC voltage VL during the step-down operation. This step-down operation is performed by supplying the electromagnetic energy accumulated in the reactor L1 during the ON period of the switching element Q1 to the power line 6 via the switching element Q2 and the antiparallel diode D2. The voltage conversion ratio (the ratio of VH and VL) in these step-up or step-down operations is controlled by the on-period ratio (duty ratio) of the switching elements Q1 and Q2 with respect to the switching period. If switching elements Q1 and Q2 are fixed to ON and OFF, respectively, VH = VL (voltage conversion ratio = 1.0) can be obtained.

  Smoothing capacitor C 0 smoothes the DC voltage from converter 12 and supplies the smoothed DC voltage to inverter 14. The voltage sensor 13 detects the voltage across the smoothing capacitor C 0, that is, the system voltage VH, and outputs the detected value to the control device 30.

  When the torque command value of AC motor M1 is positive (Trqcom> 0), inverter 14 responds to switching control signals SG3 to SG8 from control device 30 when a DC voltage is supplied from smoothing capacitor C0. AC motor M1 is driven so as to output a positive torque by converting a DC voltage into an AC voltage by switching operation of elements Q3 to Q8. Further, when the torque command value of AC electric motor M1 is zero (Trqcom = 0), inverter 14 converts the DC voltage into the AC voltage by the switching operation in response to switching control signals SG3 to SG8, and the torque is zero. The AC motor M1 is driven so that Thus, AC electric motor M1 is driven to generate zero or positive torque designated by torque command value Trqcom.

  Further, during regenerative braking of an electric vehicle equipped with motor drive control system 100, torque command value Trqcom of AC electric motor M1 is set to a negative value (Trqcom <0). In this case, the inverter 14 converts the AC voltage generated by the AC motor M1 into a DC voltage by a switching operation in response to the switching control signals SG3 to SG8, and converts the converted DC voltage (system voltage) to the smoothing capacitor C0. To the converter 12 via The regenerative braking here refers to braking with regenerative power generation when the driver operating the electric vehicle performs a footbrake operation, or regenerative braking by turning off the accelerator pedal while driving, although the footbrake is not operated. This includes decelerating (or stopping acceleration) the vehicle while generating electricity.

  Current sensor 24 detects a motor current (phase current) flowing through AC electric motor M <b> 1 and outputs the detected value to control device 30. Since the sum of instantaneous values of the three-phase motor currents iu, iv and iw is zero, two-phase motor currents (for example, V-phase current iv and W-phase current iw) are detected as shown in FIG. You may arrange so that.

  The rotation angle sensor (resolver) 25 detects the rotor rotation angle θ of the AC electric motor M 1 and sends the detected rotation angle θ to the control device 30. The control device 30 can calculate the rotation speed and the rotation frequency ωe of the AC motor M1 based on the rotation angle θ. Note that the rotation angle sensor 25 may be omitted by directly calculating the rotation angle θ from the motor voltage or current by the control device 30.

  The control device 30 is composed of an electronic control unit (ECU), and performs motor processing by executing software stored in advance by a CPU (Central Processing Unit) (not shown) and / or hardware processing by a dedicated electronic circuit. The operation of the drive control system 100 is controlled.

  As a typical function, the control device 30 includes the input torque command value Trqcom, the DC voltage VL detected by the voltage sensor 10, the DC current Ib detected by the current sensor 11, and the system voltage detected by the voltage sensor 13. Based on VH and motor currents iv and iw detected by current sensor 24, rotation angle θ from rotation angle sensor 25, etc., AC motor M1 outputs torque according to torque command value Trqcom by a control method described later. Next, the operation of the converter 12 and the inverter 14 is controlled. That is, switching control signals SG1 to SG8 for controlling converter 12 and inverter 14 as described above are generated and output to converter 12 and inverter 14.

  During the boosting operation of converter 12, control device 30 performs feedback control of system voltage VH and generates switching control signals SG1 and SG2 so that system voltage VH matches the voltage command value.

  In addition, when control device 30 receives signal RGE indicating that the electric vehicle has entered the regenerative braking mode from an external ECU, switching control signal SG <b> 3 to convert the AC voltage generated by AC motor M <b> 1 into a DC voltage. SG8 is generated and output to the inverter 14. Thereby, the inverter 14 converts the AC voltage generated by the AC motor M <b> 1 into a DC voltage and supplies it to the converter 12.

  Furthermore, when receiving a signal RGE indicating that the electric vehicle has entered the regenerative braking mode from the external ECU, control device 30 generates switching control signals SG1, SG2 so as to step down the DC voltage supplied from inverter 14. And output to the converter 12. As a result, the AC voltage generated by AC motor M1 is converted to a DC voltage, stepped down, and supplied to DC power supply B.

Next, the operation of PWM control will be described using FIG. 2 and FIG.
Referring to FIG. 2, in PWM control, each phase of AC electric motor M1 is pseudo-sine by controlling on / off of switching elements of each phase of inverter 14 based on voltage comparison between carrier wave 160 and voltage command 170. A pulse width modulation voltage 180 as a wave voltage is applied. The carrier wave 160 can be configured by a triangular wave or a sawtooth wave having a predetermined frequency. Below, a triangular wave is illustrated.

  In asynchronous PWM, the frequency of the carrier wave 160 (hereinafter referred to as the carrier wave frequency) does not change in synchronization with the rotational speed (rotational frequency) of the AC motor M1, and is fixed at a relatively high predetermined frequency at which electromagnetic noise is difficult to detect. Is done.

  On the other hand, as shown in FIG. 3, in synchronous PWM, it is synchronized with the rotational speed (rotational frequency) of AC electric motor M1, and becomes k times (k: an integer of 1 or more) of the rotational frequency of AC electric motor M1. In addition, the carrier frequency is controlled. As a result, in the synchronous PWM, the number of carriers 160 (hereinafter also referred to as the number of synchronous pulses) included in the electrical angle 360 degrees (one cycle) of the AC motor M1 is controlled to a constant value k.

Since phase voltage command 170 is also synchronized with the rotational frequency of AC electric motor M1, as a result, the frequency ratio between carrier wave 160 and phase voltage command 170 is also k: 1.

  In synchronous PWM, the positive / negative symmetry of the pulse width modulation voltage 180 (FIG. 2) can be ensured even if the number of carriers per electrical angle period (360 degrees) is reduced. For this reason, by applying synchronous PWM, the carrier frequency can be set lower than that of asynchronous PWM without impairing control stability, so by reducing the number of switching per unit time of each switching element of the inverter 14, Switching loss (power loss) can be reduced. Therefore, by applying synchronous PWM, it is possible to improve the fuel efficiency of the electric vehicle by improving the power conversion efficiency in the inverter 14 and to suppress the temperature rise of each switching element of the inverter 14.

  When k = 1, so-called rectangular wave voltage control is applied in which a rectangular wave voltage of one positive and negative pulse is applied in synchronization with the rotation frequency of the AC motor M1. Also in this case, the calculation of the offset amount of the motor current described below can be similarly applied.

  FIG. 4 is a functional block diagram illustrating a configuration of PWM control by the AC motor control device according to the embodiment of the present invention. Each functional block shown in FIG. 4 may be configured by a circuit (hardware) having a function corresponding to the block, or the control device 30 (ECU) performs software processing according to a preset program. You may implement | achieve by performing.

  Referring to FIG. 4, PWM control unit 200 includes current command generation unit 210, coordinate conversion units 220 and 250, voltage command generation unit 240, addition unit 255, PWM modulation unit 260, and carrier wave control unit 270. A PWM mode selection unit 280, a rotation frequency calculation unit 290, a current sampling unit 300, a current offset detection unit 310, and an offset correction unit 320. The current offset detection unit 310 includes a sampling value extraction unit 315 and an offset amount calculation unit 317.

  The current sampling unit 300 acquires the V-phase and W-phase current detection values Iv and Iw by sampling the output of the current sensor 24 in response to a sampling instruction generated at a constant period. As described above, the U-phase current detection value Iv is also obtained by Iv = − (Iu + Iv). Alternatively, the U-phase current sensor 24 may be further provided, and the U-phase current Iu may be obtained by sampling the output value of the current sensor 24.

  As described above, in the synchronous PWM, the current sampling instruction is generated in synchronization with the carrier wave 160. For example, it is assumed that a current sampling instruction is issued every half cycle of the carrier wave 160. On the other hand, when asynchronous PWM is selected, a sampling instruction is generated at a constant time interval corresponding to the control period in accordance with a clock signal (not shown).

  The PWM control unit 200 is configured to control the AC motor M1 according to a control calculation described later using the current detection values Iu, Iv, and Iw by the current sampling unit 300.

  Current command generation unit 210 generates a d-axis current command value Idcom and a q-axis current command value Iqcom according to torque command value Trqcom of AC electric motor M1 according to a table or the like created in advance.

  The coordinate conversion unit 220 is based on the current detection values Iv and Iw from the current sampling unit 300 by coordinate conversion (3 phase → 2 phase) using the rotation angle θ of the AC motor M1 detected by the rotation angle sensor 25. , D-axis current Id and q-axis current Iq are calculated.

  Deviation ΔId (ΔId = Idcom−Id) relative to the command value of the d-axis current and deviation ΔIq (ΔIq = Iqcom−Iq) relative to the command value of the q-axis current are input to the voltage command generation unit 240. The voltage command generation unit 240 obtains a control deviation by performing PI (proportional integration) calculation with a predetermined gain for each of the d-axis current deviation ΔId and the q-axis current deviation ΔIq, and a d-axis voltage command value corresponding to the control deviation. Vd # and q-axis voltage command value Vq # are generated.

  The coordinate conversion unit 250 converts the d-axis voltage command value Vd # and the q-axis voltage command value Vq # into the U-phase, V-phase, W-phase by coordinate conversion (2 phase → 3 phase) using the rotation angle θ of the AC motor M1. Each phase voltage command Vu #, Vv #, Vw # is converted.

  PWM mode selection unit 280 selects one of synchronous PWM and asynchronous PWM based on the operating state (rotational speed, output torque, temperature, etc.) of AC electric motor M1 and / or the operating state of inverter 14 (temperature of the switching element, etc.). A mode signal MD indicating the selection result is generated.

  The rotation frequency calculation unit 290 calculates the rotation frequency ωe of the AC motor M1 based on the output (rotation angle θ) of the rotation angle sensor 25.

  The carrier wave control unit 270 sets the carrier frequency fc based on the mode signal MD from the PWM mode selection unit 280, the rotation frequency ωe calculated by the rotation frequency calculation unit 290, and the synchronization pulse number k. The number of synchronization pulses k may be a fixed value, but may be variably set according to the operating state of AC motor M1 and / or inverter 14.

  The carrier wave control unit 270 sets the carrier wave frequency fc to a predetermined frequency when asynchronous PWM is selected. In general, the predetermined frequency is set to a frequency at which electromagnetic noise is relatively difficult to detect in consideration of an audible frequency band. Note that the carrier frequency fc may be changed according to the operating state of the AC motor M1. However, in such a change, the synchronization between the carrier wave and the rotation of the AC motor is not ensured.

  On the other hand, the carrier wave control unit 270 sets the carrier frequency fc based on the number k of synchronization pulses and the calculated rotation frequency ωe when the synchronous PWM is selected. As described with reference to FIG. 3, the carrier frequency fc in the synchronous PWM is set to fc = k · ωe. The carrier number k is set such that the carrier frequency fc in the synchronous PWM is lower than the carrier frequency fc in the asynchronous PWM. When a three-phase motor is used as the AC motor M1, k is a multiple of 3, so that k = 6 is set as an example.

  The adding unit 255 uses offset correction amounts ΔVu, ΔVv, and the like from an offset correction unit 320, which will be described later, from the phase voltage commands Vu #, Vv #, Vw # obtained according to the control calculation based on the current detection values Iu, Iv, Iw. Each final phase voltage command Vu, Vv, Vw is calculated by subtracting ΔVw. That is, Vu = Vu # −ΔVu, Vv = Vv # −ΔVv, and Vw = Vw # −ΔVw.

  The PWM modulation unit 260 generates the carrier wave 160 (FIGS. 2 and 3) according to the carrier frequency fc set by the carrier wave control unit 270, and also outputs the phase voltage commands Vu, Vv, and Vw from the addition unit 255 (FIGS. 2 and 3). The switching control signals SG3 to SG8 of the inverter 14 are generated in accordance with a voltage comparison between the phase voltage command 170 in FIG. By controlling on / off of the upper and lower arm elements of each phase of the inverter 14 according to the switching control signals SG3 to SG8, a pseudo sine wave voltage corresponding to the pulse width modulation voltage 180 of FIG. 2 is applied to each phase of the AC motor M1. The

  Note that the amplitude of the carrier wave 160 in PWM modulation corresponds to the input DC voltage (system voltage VH) of the inverter 14. However, if the amplitude of each phase voltage command Vu, Vv, Vw is converted to a value obtained by dividing the original amplitude value based on Vd #, Vq # by the system voltage VH, the amplitude of the carrier wave 160 used in the PWM modulator 260 is changed. Can be fixed.

  Here, the offset of the motor current in the synchronous PWM will be described with reference to FIGS.

  As shown in FIG. 5, in response to a switching control signal SG (generally indicating switching control signals SG1 to SG8 in FIG. 4) generated according to PWM control, a drive circuit (not shown) is connected to switching elements Q3 to Q3. By driving the gate voltage Vg of Q8, switching elements Q3 to Q8 are actually turned on / off. When the DC voltage VH is switched by the actual turning on / off of the switching element, a voltage pulse applied to each phase of the AC motor M1 is generated.

  At this time, the gate voltage Vg does not immediately change in accordance with the change in the switching control signal SG, depending on the gate drive time of the switching element drive circuit. Also, a design that actively reduces the switching speed by changing the gate resistance in order to suppress the surge voltage may be employed. Therefore, in practice, the voltage pulse controlled by the switching element that is turned on / off according to the gate voltage Vg is set to delay time (on / off delay) Td1, Td2 from the on / off command timing by the switching control signal SG, as shown in FIG. Will have.

  In asynchronous PWM with a high carrier frequency, a relatively large number of voltage pulses are generated at different electrical angles for each electrical cycle, so the effect of on / off delay appears as current fluctuation (pulsation) having no DC component.

  However, as described above, in synchronous PWM, a relatively small number of voltage pulses are generated at the same electrical angle for each electrical cycle. Therefore, as shown in FIG. 6, the direct current component ( There is a risk that an offset amount Iof) occurs. FIG. 6 shows a waveform of the V-phase current iv as an example.

  When such a current offset occurs, the inverter 14 is controlled according to the phase voltage commands Vu #, Vv #, Vw # calculated based on the current detection values Iu, Iv, Iw (FIG. 4). Feedback control acts so that the offset is maintained. As a result, the motor temperature rises due to an increase in the eddy current of the rotor (not shown) of AC electric motor M1, and there is a concern about the occurrence of demagnetization or a decrease in torque control accuracy due to a change in motor constant. For this reason, it is necessary to detect such an offset amount accurately and promptly and reflect it in Fordback control.

  When the offset amount is zero, the average value (integral value) of one electric cycle of the motor current (Iv in FIG. 6) is zero. Therefore, as described in Patent Document 1, the offset amount Iof can be obtained by applying a low-pass filter to the current sampling values at a plurality of current sampling points 400 provided in each electrical cycle. Then, by reflecting the obtained offset amount Iof in the feedback control, it is possible to generate a control command that cancels the current offset.

  However, a delay as illustrated in FIG. 7 occurs in the offset amount calculation by the low-pass filter. FIG. 7 shows a case in which the offset amount changes from Ia to Ib with the change of the motor state at time t0. However, even if the offset amount changes, the offset amount Iof calculated by the low-pass filter reaches Ib at time t1 when a time delay according to the time constant of the low-pass filter has elapsed. That is, during the time t0 to t1, the current offset cannot be properly eliminated, and a control error occurs. Thus, in the low-pass filter, it is difficult to suppress the current offset with high responsiveness when the offset amount changes during the operation of the AC motor M1.

  In the AC motor control according to the present embodiment, in the synchronous PWM control in which a current offset is likely to occur as described above, the offset amount of the motor current is calculated early and accurately by the current offset detection unit 310 shown in FIG. .

  Referring to FIG. 4 again, the current offset detection unit 310 includes a sampling value extraction unit 315 and an offset amount calculation unit 317.

  The sampling value extraction unit 315 obtains M current sampling values for the latest N cycles (N: natural number) from a plurality of current sampling values sequentially sampled each time a sampling instruction is generated for each phase of the motor current. To extract.

  Here, when the number of times of current sampling synchronized with the carrier wave 160 in the synchronous PWM is m (m: natural number), M = N × k × m. For example, current sampling is performed every half cycle of the carrier wave 160 (m = 2), and when the current sampling value for the latest one cycle is extracted (N = 1) in synchronous PWM with the number of synchronization pulses k = 6. , M = 12.

  Then, the offset amount calculation unit 317 obtains an average value of M current sampling values for N periods extracted by the sampling value extraction unit 315 for each of the U phase, the V phase, and the W phase, Phase offset amounts ΔIu, ΔIv, ΔIw are calculated.

  FIG. 8 is a conceptual diagram illustrating the operation of the sampling value extraction unit 315. FIG. 8 shows an operation when current sampling is performed every half cycle of the carrier wave 160 in the synchronous PWM with the synchronization number k = 6 as described above.

  Referring to FIG. 8, when the current sampling value of the latest electric cycle is extracted (N = 1), M = 12, as described above. Therefore, the sampling value extraction unit 315 extracts the latest 12 (M) current sampling values that are sequentially sampled every half cycle of the carrier wave 160. For example, when a new current sampling value I13 is sampled from the state where the current sampling values I1 to I12 of the V-phase current iv have been extracted, the sampling value extraction unit 315 responds to the current sampling and performs current sampling. Values I2 to I13 are newly extracted. In this way, each time current sampling is performed, the sampling value extraction unit 315 always extracts the current sampling values for the latest N cycles for each phase of the motor current.

  The offset amount calculation unit 317 obtains the offset amounts ΔIu, ΔIv, ΔIw of each phase motor current by obtaining an average of the M current sampling values extracted by the sampling value extraction unit 315. For example, in FIG. 8, ΔIv = (I1 + I2 +... + I11 + I12) / 12 immediately before the current sampling value I13 is acquired, and ΔIv = (I2 + I3 +... + I12 + I13) / 12.

  As described above, the offset amount calculation unit 317 calculates the offset amounts ΔIu, ΔIv, ΔIw, which are detected values of the offset amount Iof, according to the average value of the current sampling values for the latest N cycles for each phase of the motor current. be able to. In synchronous PWM, current sampling is performed at a timing synchronized with the carrier wave 160, so that current sampling values at two points whose electrical angles are separated by a half cycle (180 °) are included in one electrical cycle. . For this reason, the offset amount can be calculated easily and quickly by averaging the current sampling values for N cycles. If N = 1, a change in the offset amount can be detected most rapidly.

  Referring again to FIG. 4, offset correction section 320 has an offset correction amount for each phase voltage command for eliminating current offset based on offset amounts ΔIu, ΔIv, ΔIw obtained by offset amount calculation section 317. ΔVu, ΔVv, and ΔVw are calculated.

  The adder 255 uses the voltage corrections Vu #, Vv #, and Vw # of each phase obtained by the control calculation based on the current detection values Iu, Iv, and Iw including the current offset, as offset correction amounts ΔVu, By correcting with ΔVv and ΔVw, final voltage commands Vu, Vv and Vw for each phase are calculated.

  For example, the adding unit 255 calculates the offset correction amounts ΔVu, ΔVv, ΔVw by the following formulas (1) to (3) by PI calculation of the offset correction amounts ΔVu, ΔVv, ΔVw.

ΔVu = Kp × ΔIu + Ki × Σ (ΔIu) (1)
ΔVv = Kp × ΔIv + Ki × Σ (ΔIv) (2)
ΔVw = Kp × ΔIw + Ki × Σ (ΔIw) (3)
As shown in FIG. 10, the voltage command based on the current detection values Iu, Iv, and Iw (FIG. 9 illustrates the V-phase voltage command Vv #) is an ideal sine wave (average value = 0). is there. However, when the current detection values Iu, Iv, and Iw have offset amounts, the current offset remains even if the inverter 14 is controlled according to the voltage command Vv # as it is.

  Therefore, the offset amount is eliminated by correcting the original control calculation based on the detected current value using the offset correction amounts ΔVu, ΔVv, ΔVw calculated based on the offset amounts ΔIu, ΔIv, ΔIw of each current. The voltage commands Vu, Vv, and Vw for each phase can be calculated.

  FIG. 9 is a flowchart for explaining the calculation process procedure of the current offset amount in the AC motor control according to the embodiment of the present invention. The control process shown in FIG. 9 is executed for each control cycle with a current sampling instruction. Each step of the flowchart described below including FIG. 9 is realized by software processing (execution by the CPU of the stored program) or hardware processing (operation of the dedicated electronic circuit) by the control device 30.

  Referring to FIG. 9, control device 30 samples the current of each phase (U phase, V phase, W phase) in step S100. As described above, one of the U to W phases may be obtained not by directly sampling the output of the current sensor 24 but by calculation from other two-phase current sampling values. That is, the processing in step S100 corresponds to the function of the current sampling unit 300 in FIG.

  In step S110, the control device 30 extracts the current sampling values for the latest N cycles in each phase for the current sampling values sequentially sampled every time current sampling is instructed in step S100. That is, the processing in step S110 corresponds to the operation of the sampling value extraction unit 315 in FIG. As described above, N = 1 is preferable.

  Further, in step S120, the control device 30 calculates the current offset amounts ΔIu, ΔIv, ΔIw of each phase by averaging the current sampling values for N periods extracted in step S110. That is, the processing in step S120 corresponds to the function of the offset amount calculation unit 317 in FIG.

  In step S150, the control device 30 reflects the offset amount obtained in step S120 in the control calculation. That is, the processing in step S150 corresponds to the function of the offset correction unit 320.

  In the configuration of FIG. 4, the correction amounts Vu #, Vv #, Vw # of each phase voltage command are calculated by PI calculation of the offset amounts ΔIu, ΔIv, ΔIw of each phase current. The reflection is not limited to this configuration. That is, as described in Patent Document 1, the offset amount is calculated by the current offset detection unit 310 according to the present embodiment in addition to the correction of the offset amount by directly correcting the drive signal of the switching element based on the offset amount. ΔIu, ΔIv, and ΔIw can be reflected in the control calculation in an arbitrary manner.

  As described above, in the control apparatus for an AC motor according to the embodiment of the present invention, in the AC motor control to which the synchronous PWM in which the current offset is likely to occur is applied, the offset amount of the motor current is simply and easily reflected on the characteristics of the synchronous PWM. By calculating quickly, the current offset of the AC motor can be suppressed with high responsiveness.

[Modification 1]
In the first modification of the embodiment, a method for obtaining the current offset amount more accurately in consideration of the current detection error characteristic in the synchronous PWM will be described.

FIG. 11 is a conceptual diagram showing an actual current waveform including a ripple component.
Referring to FIG. 11, the actual motor current (in FIG. 11 exemplifies V-phase current iv) includes ripple component ΔIrp and therefore does not become a clean sinusoidal current (iv #). In addition, the ripple current is generated in response to turning on and off of the switching element in the inverter 14. Therefore, in synchronous PWM in which current sampling is performed at a timing synchronized with the carrier wave 160, the influence of the ripple component appears equally in each current sampling value. That is, since the ripple component is superimposed on each current sampling value, the offset amount may be calculated too large or too small only by the simple average value processing according to the first embodiment.

  The carrier control unit 270 controls the phase of the carrier 160 with respect to the voltage command 170 (hereinafter referred to as carrier phase) in addition to the frequency of the carrier 160. For example, the carrier phase is set in advance to an optimum value that minimizes the peak value of the motor current. Alternatively, the carrier phase may be changed according to the state (rotational speed or the like) of AC electric motor M1. In any case, the carrier phase θc is known in the synchronous PWM.

  In the following, it is defined that the state shown in FIG. 17 is the carrier phase θc = 0, and that the carrier phase θc increases when the carrier wave 160 is shifted to the left from the state of FIG.

  When the carrier phase θc changes, the pulse pattern of the line voltage of the AC motor M1 changes, so that the ripple current waveform also changes. For this reason, an error due to detection of a ripple component included in the average value of the M current sampling values extracted by the sampling value extraction unit 315 of the first embodiment, that is, an offset amount detection error is also the carrier phase θc. Varies depending on

  For example, an offset amount detection error Ifer with respect to the carrier phase θc can be obtained in advance by an experiment or the like in which the motor current is actually measured after changing the carrier phase θc.

  As a result, as shown in FIG. 13, a characteristic line 510 of the offset amount detection error Ifer with respect to the carrier phase θc is obtained in advance. In the characteristic line 510, the offset amount detection error Ifer also changes due to the change of the carrier phase θc, and when the carrier phase θc changes by 360 °, it becomes the same as the original carrier phase, so the offset amount detection error Ifer also has the same value. That is, the characteristic line 510 is a periodic function for every electrical angle of 360 °.

  FIG. 14 is a flowchart illustrating a control processing procedure for detection and correction of the offset amount of the motor current in the AC motor control according to the first modification of the present embodiment.

  Referring to FIG. 14, control device 30 obtains the current for the latest N cycles in each phase with respect to the current sampling values sequentially sampled every time current sampling is instructed by steps S <b> 100 and S <b> 110 similar to FIG. 9. Extract sampling values.

  Further, in step S115, the control device 30 refers to the carrier phase θc known by the carrier wave control unit 270 and refers to a map or the like created in advance according to the characteristic line 510 (FIG. 13) to detect the offset amount detection error due to the ripple current. Read Ifer. Preferably, characteristic line 510 is set independently for each phase, and offset amount detection error Ifer is also obtained independently for each phase. Note that the offset amount detection error Ifer may be defined not by the current value itself but by a ratio (ratio) to the current value.

  In step S120, the control device 30 averages the current sampling values for N periods extracted in step S110 in the same manner as in FIG. In step S130, the control device 30 subtracts the offset amount detection error Ifer obtained in step S115 from the average value of the current sampling values obtained in step S120 for each phase, thereby obtaining the current offset amount ΔIu, ΔIv and ΔIw are calculated. That is, in the first modification of the first embodiment, the processing by steps S115 and S130 is further executed by the offset amount calculation unit 317.

  Furthermore, the control apparatus 30 reflects the offset amount calculated | required by step S120 to control calculation by step S150 similar to FIG.

  In this way, the AC motor control apparatus according to the first modification of the embodiment can determine the current offset amount more accurately in consideration of the current detection error characteristic in the synchronous PWM. As a result, in addition to the effects of the AC motor control device according to the embodiment, the calculation accuracy of the offset amount reflected in the control calculation can be further increased.

[Modification 2]
In the second modification of the embodiment, a method for more accurately obtaining the current offset amount in correspondence with the change in the number k of synchronous pulses in the synchronous PWM will be described.

  In the synchronous PWM, the smaller the number of synchronous pulses k, the more effective the switching loss is reduced, but the controllability is relatively lowered. Therefore, it is preferable that the synchronization pulse number k is changed according to the state of the AC motor M1 with respect to the synchronization pulse number k when the synchronous PWM is applied.

  FIG. 15 is a conceptual diagram for explaining an example of selection of a control mode for the operating state of AC electric motor M1.

  Referring to FIG. 15, the selected control mode changes depending on, for example, the rotational speed and torque of AC electric motor M1. For example, normal PWM control is applied to the low speed regions 501 to 504, while overmodulation PWM control is applied to the medium speed region 505, and rectangular wave voltage control is applied to the high speed region 506.

  Among the low speed regions 501 to 504 to which the PWM control is applied, the asynchronous PWM is applied to the region 501 with a low rotational speed, while the synchronous PWM is applied to the regions 502 to 503 with a relatively high rotational speed. . Further, a synchronous PWM with k = 15 is applied in the region 502, a synchronous PWM with k = 9 is applied in the region 503, and k = in the region 504 so that the number k of synchronous pulses decreases as the rotational speed increases. Six synchronous PWMs are applied.

  Note that the selection of the control mode according to FIG. 15 is merely an example, and the setting of the number of synchronization pulses k in the synchronization PWM is described in terms of confirmation. In the second modification of the embodiment, the current offset amount is accurately obtained in correspondence with the variable setting of the synchronization pulse number k.

  Referring to FIG. 16, the characteristic of offset amount detection error Ifer with respect to carrier phase θc varies depending on the number of synchronization pulses k. Then, for each applied synchronization pulse number k, as in FIG. 13, the offset amount detection error Ifer for the carrier phase θc is obtained in advance by experiments or the like in which the motor current is measured after changing the carrier phase θc. Can do.

  As an example, FIG. 16 further shows a characteristic line 520 when k = 9 and a characteristic line 530 when k = 15 in addition to the characteristic line 510 when k = 6. It is understood that each of the characteristic lines 520 and 530 is a periodic function every electrical angle 360 °.

  FIG. 1 is a flowchart for explaining a control processing procedure for detection and correction of an offset amount of a motor current in AC motor control according to Modification 2 of the present embodiment.

  Referring to FIG. 17, control device 30 uses the same steps S100 and S110 as in FIGS. 9 and 14 to determine the latest N cycles in each phase for the current sampling values sequentially sampled every time current sampling is instructed. The current sampling value is extracted.

  Further, in step S115 #, control device 30 determines the ripple based on the carrier phase θc and synchronization pulse number k known by carrier wave control unit 270 by referring to a map or the like created in advance according to characteristic line 510 (FIG. 13). The offset amount detection error Ifer due to the current is read. Preferably, the characteristic lines 510, 520, and 530 are set independently for each phase, and the offset amount detection error Ifer is also obtained independently for each phase. Note that the offset amount detection error Ifer may be defined not by the current value itself but by a ratio (ratio) to the current value.

  In step S120, the control device 30 averages the current sampling values for N periods extracted in step S110 in the same manner as in FIGS. In step S130, control device 30 subtracts offset amount detection error Ifer obtained in step S115 # from the average value of the current sampling values obtained in step S120 for each phase, thereby obtaining current offset amount ΔIu. , ΔIv, ΔIw are calculated. That is, in the second modification of the first embodiment, the processing by steps S115 # and S130 is further executed by offset amount calculation section 317.

  Furthermore, the control apparatus 30 reflects the offset amount calculated | required by step S120 to control calculation by step S150 similar to FIG.

  As described above, the AC motor control apparatus according to the second modification of the embodiment can determine the current offset amount more accurately in consideration of the current detection error characteristic in the synchronous PWM. As a result, in addition to the effects of the AC motor control apparatus according to the embodiment, the calculation accuracy of the offset amount reflected in the control calculation can be further improved in the synchronous PWM control in which the number of synchronization pulses k is variable.

  In the present embodiment, the three-phase motor is exemplified as the AC motor M1, but the offset calculation of the motor current when the synchronous PWM is applied according to the present invention can be applied to an AC motor other than the three-phase motor. it can.

  In FIG. 1, as a preferred configuration example, a configuration is shown in which DC voltage generation unit 10 # of the motor drive system includes converter 12 so that the input voltage (system voltage VH) to inverter 14 can be variably controlled. DC voltage generation unit 10 # is not limited to the configuration exemplified in the present embodiment. That is, it is not essential that the inverter input voltage is variable, and the present invention is also applied to a configuration in which the output voltage of the DC power supply B is directly input to the inverter 14 (for example, a configuration in which the converter 12 is omitted). Is possible.

  Further, with regard to the AC motor serving as a load of the motor drive system, in the present embodiment, a permanent magnet motor mounted for driving a vehicle in an electric vehicle (hybrid vehicle, electric vehicle, etc.) is assumed. The present invention can also be applied to a configuration in which an arbitrary AC motor used in is used as a load.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  The present invention can be used for AC motor control to which synchronous PWM is applied.

  5 Ground wire, 6, 7 Power line, 10, 13 Voltage sensor, 10 # DC voltage generator, 11, 24 Current sensor, 12 Converter, 14 Inverter, 15 U phase upper and lower arm, 16 V phase upper and lower arm, 17 W phase upper and lower Arm, 25 rotation angle sensor, 30 control unit (ECU), 100 motor drive control system, 160 carrier wave, 170 voltage command, 180 pulse width modulation voltage, 200 PWM control unit, 210 current command generation unit, 220, 250 coordinate conversion unit , 240 Voltage command generation unit, 250 Coordinate conversion unit, 255 Addition unit, 260 PWM modulation unit, 270 Carrier wave control unit, 280 Mode selection unit, 290 Rotational frequency calculation unit, 300 Current sampling unit, 310 Current offset detection unit, 315 sampling Value extraction unit, 317 Offset amount calculation unit, 3 0 offset correction unit, 400 current sampling point, 501 low speed region (asynchronous PWM), 502 to 504 low speed region (synchronous PWM), 505 medium speed region (overmodulation PWM), 506 high speed region (rectangular wave voltage control), 510, 520, 530 Characteristic line (carrier phase-offset amount detection error), B DC power supply, C0, C1 smoothing capacitor, D1-D8 antiparallel diode, fc carrier frequency, I1-I13 current sampling value, Ib DC current, Id d-axis Current, Idcom d-axis current command value, Ifer offset amount detection error, Iof offset amount, Iq d-axis current, Iqcom d-axis current command value, iu, iv, iw Motor current, Iu, Iv, Iw Current detection value (each phase Motor current), k number of synchronous pulses, L1 reactor, M1 AC motor, D mode signal, Q1-Q8 power semiconductor switching element, SG, SG1-SG8 switching control signal, SR1, SR2 system relay, Trqcom torque command value, Vd # d-axis voltage command value, Vg gate voltage, VH DC voltage (system Voltage), VL DC voltage, Vq # q-axis voltage command value, Vu #, Vv #, Vw # phase command (before correction), Vu, Vv, Vw phase voltage command (final), ΔId d-axis current deviation , ΔIq q-axis current deviation, ΔIrp ripple component, ΔIu, ΔIv, ΔIw offset amount (motor current), ΔIu, ΔIv, ΔIw current offset amount, ΔVu, ΔVv, ΔVw offset correction amount, θc carrier phase, θ rotor rotation angle, ωe Rotational frequency (AC motor).

Claims (5)

An AC motor control device in which an applied voltage is controlled by an inverter,
A current detector configured to detect the current of the AC motor;
An electric motor control unit configured to control the inverter according to a control calculation using a current detection value detected by the current detector;
The motor controller is
A PWM modulator that controls a pulse width modulation voltage applied from the inverter to the AC motor based on a voltage comparison between a phase voltage command and a carrier wave;
A carrier wave control unit for controlling to generate the carrier wave at a frequency k times (k: natural number) the frequency of the phase voltage command;
When the frequency of the carrier wave is controlled by the carrier wave control unit to be k times the frequency of the phase voltage command, the current detector outputs the current detector by sampling the output of the current detector at a predetermined period according to the phase of the carrier wave. A current sampling unit configured to obtain a detection value;
When the frequency of the carrier wave is controlled to be k times the frequency of the phase voltage command by the carrier wave control unit, N (N: natural number) of the current is obtained from a plurality of sequential current sampling values by the current sampling unit. ) An extractor configured to extract the latest M current sampling values corresponding to the period;
And an offset amount calculation unit configured to calculate an offset amount of the current detection value according to an average value of the M current sampling values extracted by the extraction unit. The carrier wave control unit is further configured to control a phase difference of the carrier wave with respect to the phase voltage command,
The offset amount calculation unit is configured to calculate an offset amount of the current detection value based on an average value of the M current sampling values and a correction amount corresponding to the phase difference. 1. The control apparatus for an AC motor according to 1. The carrier wave control unit is further configured to variably control the k according to an operating state of the AC motor,
The offset amount calculation unit is configured to calculate an offset amount of the current detection value based on an average value of the M current sampling values and a correction amount corresponding to the phase difference and the k. The control apparatus for an AC motor according to claim 1.   The control apparatus for an AC motor according to claim 1, wherein N is 1. 5.   The electric motor control unit further includes an offset correction unit configured to correct the control calculation based on the offset amount calculated by the offset amount calculation unit. The control apparatus of the alternating current motor described in 1.

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