JP2010273518A - AC motor control device - Google Patents

Abstract

In an AC motor control configured to sample the outputs of a plurality of current detectors sequentially rather than in parallel, adverse effects on control due to current detection errors are suppressed.
Outputs of current sensors provided in a V phase and a W phase are sampled sequentially rather than in parallel with respect to a common sampling instruction, so that a delay of a predetermined time Tsp is inevitable in sampling timing between phases. Occurs. The current sampling order between the phases for each sampling instruction is not fixed. Specifically, the current sampling order is within one cycle (electrical angle 360 degrees) so that the phase is sampled first at the timing corresponding to the electrical angle at which the current of any phase becomes an extreme value. It is changed appropriately.
[Selection] Figure 10

Description

  The present invention relates to an AC motor control device, and more particularly, to motor control based on sampling values of current sensors provided in a plurality of phases.

  Conventionally, a drive system using an inverter has been adopted to control an AC motor. For example, in an electric vehicle such as an electric vehicle, a hybrid vehicle, and a fuel cell vehicle, the output torque of an AC motor for traveling is generally controlled by an inverter. In such control of an AC motor, a drive current of the motor is detected by a current sensor, and current control based on the detected value is generally applied.

  In such motor control based on current detection, it is important to remove current detection errors in order to ensure control accuracy. For example, Japanese Patent Laid-Open Publication No. 2001-29892 (Patent Document 1) has a control configuration for calculating an offset amount of a drive current when driving an AC motor and correcting a duty ratio of a drive signal based on the offset amount. Are listed.

  Japanese Patent Laid-Open No. 2008-278573 (Patent Document 2) describes a technique for accurately measuring an offset of a current detection means (current sensor). Specifically, in an inverter device that operates a three-phase motor, control is performed so that one of the three phases is non-conductive and current is supplied to two phases including current detection means, and current detection at that time is detected. It describes that the offset amount is obtained based on the value.

  JP-A-2004-32917 (Patent Document 3), JP-A-2004-120814 (Patent Document 4) and JP-A-2006-271045 (Patent Document 5) are represented by three-phase motors. In the multi-phase motor control, a method for improving the motor control by paying attention to variations between phases is described.

JP 2001-298990 A JP 2008-278573 A JP 2004-32917 A JP 2004-120814 A JP 2006-271045 A

  In the motor control based on the current detection as described above, the current sensor output is sampled every predetermined control period, and the sampled output value is subjected to analog / digital conversion (A / D conversion). A current detection value used for a representative control calculation is acquired.

  On the other hand, in a multiphase AC motor represented by a three-phase AC motor, it is necessary to arrange current sensors in a plurality of phases in order to acquire a current detection value of each phase. For example, in a three-phase AC motor, it is necessary to arrange current sensors in at least two phases. Furthermore, it is necessary to sample the output of a plurality of current sensors in each control cycle. In this case, it is theoretically preferable to sample a plurality of phases of current simultaneously.

  However, in order to realize such parallel sampling processing, it is necessary to provide input ports corresponding to a plurality of phases and process sampling in each phase in parallel. Realization of such a configuration requires advanced A / D converters and arithmetic processing units (CPUs), which increases the cost of the control device.

  For this reason, from the viewpoint of cost, it is preferable to obtain a current detection value of each phase by sequentially sampling the outputs of the current sensors of a plurality of phases in order. On the other hand, when the sampling is sequentially processed as described above, an error due to the difference in sampling timing is included between the current detection values of the respective phases corresponding to the same timing. For this reason, it is necessary to consider so as not to adversely affect the control due to such errors.

  The present invention was made to solve such problems, and the object of the present invention is to control AC motors configured to sequentially sample the outputs of a plurality of current detectors instead of in parallel. It is to suppress the adverse effect on the control due to the current detection error.

  An AC motor control device according to the present invention is an AC motor control device in which an applied voltage is controlled by an inverter, and includes a plurality of current detectors, an electric motor control unit, a sampling timing instruction unit, a sampling processing unit, A sampling order designating unit. The plurality of current detectors are respectively provided in a plurality of phases of the AC motor and configured to detect a plurality of phase currents. The sampling timing instruction unit is configured to generate a current sampling instruction according to a predetermined control cycle. The sampling processing unit is configured to sequentially sample the detection values of each of the plurality of current detectors one by one in response to the current sampling instruction. The electric motor control unit is configured to control the AC electric motor according to a control calculation using the phase current sampled by the sampling processing unit. When the timing of the current sampling instruction corresponds to the timing at which the phase current takes an extreme value in the first phase of the plurality of phases, the sampling order designating unit sets the first phase of the plurality of current detectors. The sampling processing unit is configured to designate the sampling order so that the detection value of the corresponding current detector is sampled first.

  According to the control apparatus for an AC motor described above, a configuration in which the output of a plurality of current detectors is sampled sequentially rather than in parallel, focusing on the point that the amplitude of the phase current is detected at the timing when the phase current takes an extreme value. The sampling order can be specified so that the current amplitude of each phase is sampled at an equal phase. Thereby, even when the ripple current is superimposed on the phase current, it is possible to prevent an offset between phases from occurring in the detected value of the current amplitude due to the difference in sampling timing between a plurality of phases by the sequential sampling process. As a result, even when the outputs of the multiple-phase current detectors are sequentially sampled one by one, it is possible to avoid adverse effects on the control due to current detection errors.

  Preferably, the sampling order designating unit is configured to designate the sampling order based on the electrical angle of the AC motor at the timing of the current sampling instruction.

  If it does in this way, based on the electrical angle of the AC motor which can typically be detected by the rotation angle sensor (resolver etc.) of the AC motor, the phase (the maximum value or the minimum value) in which the phase current becomes an extreme value (maximum value or minimum value). When there is one phase), the sampling order among a plurality of phases can be determined so that the current of the phase is sampled first.

  Preferably, the sampling order designating unit is configured to designate the sampling order in the current sampling instruction based on a comparison of sampling values of a plurality of phase currents in the past current sampling instruction.

  In this way, when there is a phase (first phase) in which the phase current becomes an extreme value (maximum value or minimum value) based on the past output of the current detector without using an electrical angle, The sampling order between the phases can be determined so that the current of the phase is sampled first.

  More preferably, the electric motor control unit includes a calculation unit, a PWM modulation unit, and a carrier wave control unit. The calculation unit is configured to generate a voltage command for each phase of the AC motor in accordance with the control calculation. The PWM modulation unit is configured to control the pulse width modulation voltage of each phase applied from the inverter to the AC motor based on the voltage comparison between the voltage command and the carrier wave. The carrier wave control unit generates a carrier wave at a frequency k times the frequency of the voltage command (k: an integer of 2 or more).

  In this way, in the first mode (synchronous PWM), in which the carrier frequency is low and the control cycle is long, the detection error of the amplitude of each phase current due to the current phase shift due to the sequential sampling process is reduced. The sampling order can be appropriately specified to suppress.

  More preferably, the carrier wave control unit generates a carrier wave at a frequency k times the voltage command when the first mode is selected, while the carrier frequency when the first mode is selected when the second mode is selected. Is configured to generate a carrier wave at a higher predetermined frequency. The sampling order specifying unit fixes the sampling order in the sampling processing unit to a predetermined order regardless of the timing of the current sampling instruction when the second mode is selected.

  In this way, in the second mode (asynchronous PWM) in which the carrier frequency is high and the control cycle is short, the sampling order is designated in consideration of the small current detection error (amplitude error) due to the sampling order. Arithmetic processing can be non-executed. As a result, the calculation load of the control device in the second mode with a short control cycle can be reduced.

  Preferably, the AC motor is a three-phase AC motor. The plurality of current detectors are arranged in three phases, respectively. Alternatively, the plurality of current detectors are respectively arranged in two phases of the three phases.

  In this way, even if the configuration is such that the outputs of the plurality of current detectors are sequentially sampled in the control of the three-phase AC motor in which a plurality of current detectors are arranged in each phase or two phases, phase current amplitude detection It is possible to prevent the control accuracy from being lowered due to the error.

  According to the present invention, in the AC motor control configured to sample the outputs of a plurality of phase current detectors in order, adverse control effects due to current detection errors can be suppressed.

1 is an overall configuration diagram of a motor drive control system to which an AC motor control device according to an embodiment of the present invention is applied. It is a wave form diagram explaining the basic operation of PWM control. It is a wave form diagram explaining the carrier wave in synchronous PWM. It is a functional block diagram explaining the structure of the PWM control by the control apparatus of the AC motor according to the embodiment of the present invention. FIG. 5 is a functional block diagram illustrating a configuration of a current sampling unit illustrated in FIG. 4. It is a 1st waveform diagram explaining the problem by a sequential sampling process. It is a 2nd waveform diagram explaining the problem by a sequential sampling process. It is a conceptual diagram explaining the 1st example of the current sampling process by the control apparatus of the alternating current motor according to the embodiment of the present invention. It is a flowchart which shows the process sequence of the 1st example of the current sampling process by this Embodiment. It is a conceptual diagram explaining the sequential sampling process by the control apparatus of the alternating current motor by this Embodiment. It is a conceptual diagram explaining the modification of the 1st example of the current sampling process by the control apparatus of the AC motor according to the embodiment of the present invention. It is a flowchart which shows the process sequence of the modification of the 1st example of the current sampling process by this Embodiment. It is a flowchart which shows the process sequence of the 2nd example of the current sampling process by this Embodiment. It is a conceptual diagram explaining the 2nd example of the current sampling process by the control apparatus of the alternating current motor according to the embodiment of the present invention. It is a flowchart which shows the process sequence of the modification of the 2nd example of the current sampling process by this Embodiment. It is a conceptual diagram explaining the modification of the 2nd example of the current sampling process by the control apparatus of the AC motor according to the embodiment of the present invention.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

  FIG. 1 is an overall configuration diagram of a motor drive control system to which an AC motor control device according to an embodiment of the present invention is applied.

  Referring to FIG. 1, motor drive control system 100 includes a DC voltage generation unit 10 #, a smoothing capacitor C0, an inverter 14, an AC motor M1, and a control device 30.

  For example, AC electric motor M1 generates torque for driving drive wheels of an electric vehicle (referred to as a vehicle that generates vehicle driving force by electric energy such as a hybrid vehicle, an electric vehicle, and a fuel cell vehicle). This is an electric motor for traveling. Alternatively, AC electric motor M1 may be configured to have a function of a generator driven by an engine, or may be configured to have both functions of an electric motor and a generator. Further, AC electric motor M1 may operate as an electric motor for the engine, and may be incorporated in a hybrid vehicle as one that can start the engine, for example. That is, in the present embodiment, the “AC motor” includes an AC drive motor, a generator, and a motor generator (motor generator).

  DC voltage generation unit 10 # includes a DC power supply B, system relays SR1 and SR2, a smoothing capacitor C1, and a converter 12.

  The DC power supply B is typically constituted by a rechargeable power storage device such as a secondary battery such as nickel metal hydride or lithium ion or an electric double layer capacitor. The DC voltage VL output from the DC power supply B and the input / output DC current Ib are detected by the voltage sensor 10 and the current sensor 11, respectively.

  System relay SR 1 is connected between the positive terminal of DC power supply B and power line 6, and system relay SR 1 is connected between the negative terminal of DC power supply B and ground line 5. System relays SR1 and SR2 are turned on / off by signal SE from control device 30.

  Converter 12 includes a reactor L1, power semiconductor switching elements Q1, Q2, and diodes D1, D2. Power semiconductor switching elements Q 1 and Q 2 are connected in series between power line 7 and ground line 5. On / off of power semiconductor switching elements Q1 and Q2 is controlled by switching control signals SG1 and SG2 from control device 30.

In the embodiment of the present invention, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, or a power bipolar transistor is used as a power semiconductor switching element (hereinafter simply referred to as “switching element”). Etc. can be used. Anti-parallel diodes D1, D2 are arranged for switching elements Q1, Q2. Reactor L1 is connected between a connection node of switching elements Q1 and Q2 and power line 6. Further, the smoothing capacitor C 0 is connected between the power line 7 and the ground line 5.

  Inverter 14 includes a U-phase upper and lower arm 15, a V-phase upper and lower arm 16, and a W-phase upper and lower arm 17 provided in parallel between power line 7 and ground line 5. Each phase upper and lower arm is constituted by a switching element connected in series between the power line 7 and the ground line 5. For example, the U-phase upper and lower arms 15 are composed of switching elements Q3 and Q4, the V-phase upper and lower arms 16 are composed of switching elements Q5 and Q6, and the W-phase upper and lower arms 17 are composed of switching elements Q7 and Q8. Further, antiparallel diodes D3 to D8 are connected to switching elements Q3 to Q8, respectively. Switching elements Q3 to Q8 are turned on / off by switching control signals SG3 to SG8 from control device 30.

  Typically, AC motor M1 is a three-phase permanent magnet type synchronous motor, and is configured by commonly connecting one end of three coils of U, V, and W phases to a neutral point. Furthermore, the other end of each phase coil is connected to the midpoint of the switching elements of the upper and lower arms 15 to 17 of each phase.

  Converter 12 is basically controlled such that switching elements Q1 and Q2 are turned on and off in a complementary manner in each switching period. During the boosting operation, converter 12 boosts DC voltage VL supplied from DC power supply B to DC voltage VH (this DC voltage corresponding to the input voltage to inverter 14 is hereinafter also referred to as “system voltage”). This step-up operation is performed by supplying the electromagnetic energy stored in reactor L1 during the ON period of switching element Q2 to power line 7 via switching element Q1 and antiparallel diode D1.

Converter 12 steps down DC voltage VH to DC voltage VL during the step-down operation. This step-down operation is performed by supplying the electromagnetic energy accumulated in the reactor L1 during the ON period of the switching element Q1 to the power line 6 via the switching element Q2 and the antiparallel diode D2. The voltage conversion ratio (the ratio of VH and VL) in these step-up or step-down operations is controlled by the on-period ratio (duty ratio) of the switching elements Q1 and Q2 with respect to the switching period. If switching elements Q1 and Q2 are fixed to ON and OFF, respectively, VH = VL (voltage conversion ratio = 1.0) can be obtained.

  Smoothing capacitor C 0 smoothes the DC voltage from converter 12 and supplies the smoothed DC voltage to inverter 14. The voltage sensor 13 detects the voltage across the smoothing capacitor C 0, that is, the system voltage VH, and outputs the detected value to the control device 30.

  When the torque command value of AC motor M1 is positive (Trqcom> 0), inverter 14 responds to switching control signals SG3 to SG8 from control device 30 when a DC voltage is supplied from smoothing capacitor C0. AC motor M1 is driven so as to output a positive torque by converting a DC voltage into an AC voltage by switching operation of elements Q3 to Q8. Further, when the torque command value of AC electric motor M1 is zero (Trqcom = 0), inverter 14 converts the DC voltage into the AC voltage by the switching operation in response to switching control signals SG3 to SG8, and the torque is zero. The AC motor M1 is driven so that Thus, AC electric motor M1 is driven to generate zero or positive torque designated by torque command value Trqcom.

  Further, during regenerative braking of an electric vehicle equipped with motor drive control system 100, torque command value Trqcom of AC electric motor M1 is set to a negative value (Trqcom <0). In this case, the inverter 14 converts the AC voltage generated by the AC motor M1 into a DC voltage by a switching operation in response to the switching control signals SG3 to SG8, and converts the converted DC voltage (system voltage) to the smoothing capacitor C0. To the converter 12 via The regenerative braking here refers to braking with regenerative power generation when the driver operating the electric vehicle performs a footbrake operation, or regenerative braking by turning off the accelerator pedal while driving, although the footbrake is not operated. This includes decelerating (or stopping acceleration) the vehicle while generating electricity.

  Current sensor 24 detects a current (phase current) flowing through AC electric motor M <b> 1 and outputs the detected value to control device 30. Since the sum of instantaneous values of the three-phase currents iu, iv, and iw is zero, the motor currents for two phases (for example, the V-phase current iv and the W-phase current iw) are detected as shown in FIG. You may arrange in.

  The rotation angle sensor (resolver) 25 detects the rotor rotation angle θ of the AC electric motor M 1 and sends the detected rotation angle θ to the control device 30. The control device 30 can calculate the rotation speed and the rotation frequency ωe of the AC motor M1 based on the rotation angle θ. Note that the rotation angle sensor 25 may be omitted by directly calculating the rotation angle θ from the motor voltage or current by the control device 30.

  The control device 30 is composed of an electronic control unit (ECU), and performs motor processing by executing software stored in advance by a CPU (Central Processing Unit) (not shown) and / or hardware processing by a dedicated electronic circuit. The operation of the drive control system 100 is controlled.

  As a typical function, the control device 30 includes the input torque command value Trqcom, the DC voltage VL detected by the voltage sensor 10, the DC current Ib detected by the current sensor 11, and the system voltage detected by the voltage sensor 13. Based on VH and motor currents iv and iw detected by current sensor 24, rotation angle θ from rotation angle sensor 25, etc., AC motor M1 outputs torque according to torque command value Trqcom by a control method described later. Next, the operation of the converter 12 and the inverter 14 is controlled. That is, switching control signals SG1 to SG8 for controlling converter 12 and inverter 14 as described above are generated and output to converter 12 and inverter 14.

  During the boosting operation of converter 12, control device 30 performs feedback control of system voltage VH and generates switching control signals SG1 and SG2 so that system voltage VH matches the voltage command value.

  In addition, when control device 30 receives signal RGE indicating that the electric vehicle has entered the regenerative braking mode from an external ECU, switching control signal SG <b> 3 to convert the AC voltage generated by AC motor M <b> 1 into a DC voltage. SG8 is generated and output to the inverter 14. Thereby, the inverter 14 converts the AC voltage generated by the AC motor M <b> 1 into a DC voltage and supplies it to the converter 12.

  Furthermore, when receiving a signal RGE indicating that the electric vehicle has entered the regenerative braking mode from the external ECU, control device 30 generates switching control signals SG1, SG2 so as to step down the DC voltage supplied from inverter 14. And output to the converter 12. As a result, the AC voltage generated by AC motor M1 is converted to a DC voltage, stepped down, and supplied to DC power supply B.

Next, the operation of PWM control will be described using FIG. 2 and FIG.
Referring to FIG. 2, in PWM control, each phase of AC motor M <b> 1 is controlled by controlling on / off of switching elements of each phase of inverter 14 based on voltage comparison between carrier wave 160 and voltage command 170. A pulse width modulation voltage 180 as a pseudo sine wave voltage is applied. The carrier wave 160 can be configured by a triangular wave or a sawtooth wave having a predetermined frequency. Below, a triangular wave is illustrated.

  In asynchronous PWM, the frequency of the carrier wave 160 (hereinafter referred to as the carrier wave frequency) does not change in synchronization with the rotational speed (rotational frequency) of the AC motor M1, and is fixed at a relatively high predetermined frequency at which electromagnetic noise is difficult to detect. Is done.

  On the other hand, as shown in FIG. 3, in synchronous PWM, the rotational frequency (rotational frequency) of AC electric motor M1 is synchronized with the rotational frequency (rotational frequency) of AC electric motor M1 so as to be k times (k: an integer of 2 or more). In addition, the carrier frequency is controlled. As a result, in the synchronous PWM, the number of carriers of the carrier wave 160 included in the electrical angle 360 degrees (one cycle) of the AC motor M1 is controlled to a constant value k. In the present embodiment, synchronous PWM is applied in distinction from so-called rectangular wave voltage control in which a rectangular wave voltage of one positive and negative pulse is applied in synchronization with the rotation frequency of AC electric motor M1, as described above. k ≧ 2.

Since phase voltage command 170 is also synchronized with the rotational frequency of AC electric motor M1, as a result, the frequency ratio between carrier wave 160 and phase voltage command 170 is also k: 1.

  In synchronous PWM, the positive / negative symmetry of the pulse width modulation voltage 180 (FIG. 2) can be ensured even if the number of carriers per electrical angle period (360 degrees) is reduced. For this reason, by applying synchronous PWM, the carrier frequency can be set lower than that of asynchronous PWM without impairing control stability, so by reducing the number of switching per unit time of each switching element of the inverter 14, Switching loss (power loss) can be reduced. Therefore, by applying synchronous PWM, it is possible to improve the fuel efficiency of the electric vehicle by improving the power conversion efficiency in the inverter 14 and to suppress the temperature rise of each switching element of the inverter 14.

  FIG. 4 is a functional block diagram illustrating a configuration of PWM control by the AC motor control device according to the embodiment of the present invention. Each functional block shown in FIG. 4 may be configured by a circuit (hardware) having a function corresponding to the block, or the control device 30 (ECU) performs software processing according to a preset program. You may implement | achieve by performing.

  Referring to FIG. 4, current sampling section 300 generates V-phase and W-phase current detection values Iv and Iw by sampling output values ivs and iws of V-phase and W-phase current sensors 24. The detailed configuration of the current sampling unit 300 will be described later.

  The PWM control unit 200 is configured to control the AC motor M1 according to a control calculation described later using current detection values Iv and Iw according to the phase current sampled by the current sampling unit 300.

  The PWM control unit 200 includes a current command generation unit 210, coordinate conversion units 220 and 250, a voltage command generation unit 240, a PWM modulation unit 260, a carrier wave control unit 270, a PWM mode selection unit 280, and a rotation frequency calculation. Part 290.

  Current command generation unit 210 generates a d-axis current command value Idcom and a q-axis current command value Iqcom according to torque command value Trqcom of AC electric motor M1 according to a table or the like created in advance.

  The coordinate conversion unit 220 is based on the current detection values Iv and Iw from the current sampling unit 300 by coordinate conversion (3 phase → 2 phase) using the rotation angle θ of the AC motor M1 detected by the rotation angle sensor 25. , D-axis current Id and q-axis current Iq are calculated.

  Deviation ΔId (ΔId = Idcom−Id) relative to the command value of the d-axis current and deviation ΔIq (ΔIq = Iqcom−Iq) relative to the command value of the q-axis current are input to the voltage command generation unit 240. The voltage command generation unit 240 obtains a control deviation by performing PI (proportional integration) calculation with a predetermined gain for each of the d-axis current deviation ΔId and the q-axis current deviation ΔIq, and a d-axis voltage command value corresponding to the control deviation. Vd # and q-axis voltage command value Vq # are generated.

  The coordinate conversion unit 250 converts the d-axis voltage command value Vd # and the q-axis voltage command value Vq # into the U-phase, V-phase, W-phase by coordinate conversion (2 phase → 3 phase) using the rotation angle θ of the AC motor M1. Each phase voltage command Vu, Vv, Vw of the phase is converted.

  PWM mode selection unit 280 selects one of synchronous PWM and asynchronous PWM based on the operating state (rotational speed, output torque, temperature, etc.) of AC electric motor M1 and / or the operating state of inverter 14 (temperature of the switching element, etc.). A mode signal MD indicating the selection result is generated.

  The rotation frequency calculation unit 290 calculates the rotation frequency ωe of the AC motor M1 based on the output (rotation angle θ) of the rotation angle sensor 25.

The carrier wave control unit 270 determines the carrier frequency fc based on the mode signal MD from the PWM mode selection unit 280, the rotation frequency ωe calculated by the rotation frequency calculation unit 290, and the number of carriers k in the preset synchronous PWM. Set. The number of carriers k may be a fixed value, but may be variably set according to the operating state of AC motor M1 and / or inverter 14.

  The carrier wave control unit 270 sets the carrier wave frequency fc to a predetermined frequency when asynchronous PWM is selected. As described above, the predetermined frequency is set to a frequency at which electromagnetic noise is relatively difficult to detect in consideration of an audible frequency band. Note that the carrier frequency fc may be changed according to the operating state of the AC motor M1. However, in such a change, the synchronization between the carrier wave and the rotation of the AC motor is not ensured.

  On the other hand, when synchronous PWM is selected, the carrier wave control unit 270 sets the carrier frequency fc based on the determined carrier number k and the calculated rotation frequency ωe. As described with reference to FIG. 3, the carrier frequency fc in the synchronous PWM is set to fc = k · ωe. The carrier number k is set such that the carrier frequency fc in the synchronous PWM is lower than the carrier frequency fc in the asynchronous PWM. When a three-phase motor is used as the AC motor M1, k is a multiple of 3, so that k = 6 is set as an example.

  The PWM modulation unit 260 generates a carrier wave 160 (FIGS. 2 and 3) according to the carrier frequency fc set by the carrier wave control unit 270, and each phase voltage command Vu, Vv, Vw (FIG. 2, FIG. 2) from the coordinate conversion unit 250. 3) and the switching control signals SG3 to SG8 of the inverter 14 are generated in accordance with a voltage comparison with the carrier wave 160. By controlling on / off of the upper and lower arm elements of each phase of the inverter 14 in accordance with the switching control signals SG3 to SG8, a pseudo sine wave voltage corresponding to the pulse width modulation voltage 180 of FIG. Applied.

  Note that the amplitude of the carrier wave 160 in PWM modulation corresponds to the input DC voltage (system voltage VH) of the inverter 14. However, if the amplitude of each phase voltage command Vu, Vv, Vw is converted to a value obtained by dividing the original amplitude value based on Vd #, Vq # by the system voltage VH, the amplitude of the carrier wave 160 used in the PWM modulator 260 is changed. Can be fixed.

Next, the configuration of the current sampling unit 300 will be described with reference to FIG.
Referring to FIG. 5, current sampling unit 300 includes a sampling timing specifying unit 310, a sampling order specifying unit 320, and a sampling processing unit 330.

  The sampling timing designating unit 310 generates a signal Fsp that designates the sampling timing by the sampling processing unit 330 according to the control period of the PWM control unit 200. That is, the signal Fsp corresponds to “current sampling instruction”.

  Sampling timing designating section 310 generates signal Fsp at different timings between synchronous PWM and asynchronous PWM in accordance with mode signal MD. When asynchronous PWM is selected, the sampling timing designating unit 310 generates the signal Fsp at a constant time interval corresponding to the control period in accordance with a clock signal (not shown) or the like.

  On the other hand, when the synchronous PWM is selected, the sampling timing designating unit 310 basically generates the signal Fsp in synchronization with the carrier wave 160. In synchronous PWM, since the synchronization number k is known in advance, a sampling period instruction can be issued at predetermined intervals of the rotation angle θ by associating it with the period (phase) of the carrier wave 160. Hereinafter, it is assumed that the current sampling instruction Fsp is generated every half cycle of the carrier wave 160. Alternatively, the signal Fsp instructing sampling may be generated according to the rotation angle θ detected by the rotation angle sensor 25.

  In response to the signal Fsp from the sampling timing designating unit 310, the sampling processing unit 330 sequentially samples the output values ivs and iws of the U-phase and W-phase current sensors 24 in accordance with the order designated by the sampling order designating unit 320. To do. That is, the sampling processing unit 330 is configured to sample the detection values by the current sensors 24 provided in the plurality of phases one by one in order. Therefore, the cost of the sampling processing unit 330 can be reduced as compared with the configuration in which the detection values of the plurality of phase current sensors 24 are sampled simultaneously in parallel.

  Furthermore, the sampling processing unit 330 generates current detection values Iv and Iw according to the sampled output values ivs and iws, and sends them to the coordinate conversion unit 220 of the PWM control unit 200. The coordinate conversion unit 220 can also obtain the u-phase current Iu according to Iu + Iv + Iw = 0.

  The sampling order designating unit 320 generates a signal For that indicates the sampling order between a plurality of phases. The signal For is sent to the sampling processing unit 330.

  In the example of FIG. 4, the signal For designates which of the V-phase and W-phase currents is sampled first. If the current sensor 24 is provided in each of the U-phase, V-phase, and W-phase in the configuration of FIG. 4, the current sampling order for these three phases is specified by the signal For.

  Here, with reference to FIGS. 6 and 7, problems that arise when sequentially sampling a plurality of phases of current will be described.

  FIG. 6 shows changes in the output values ivs and iws of the V-phase and W-phase current sensors 24. These output values ivs and iws have a waveform in which a high-frequency ripple current is superimposed on a sinusoidal current having an electrical angle of 360 degrees as one cycle. The ripple current is generated in accordance with the switching in the inverter 14 due to the inductance of the winding or the like of the AC motor M1.

  In FIG. 6, in response to a current sampling instruction (signal Fsp) generated every half cycle of the carrier wave 160, the V-phase current is fixedly sampled first (S1, S3, S5, S7), and thereafter The operation when the W-phase current is sampled (S2, S4, S6, S8) is shown.

  In the sequential sampling process, a fixed time difference Tsp occurs between the sampling timing of the V-phase current and the sampling timing of the W-phase current for the same current sampling instruction. In the example of FIG. 6, the sampling timing of the W-phase current is fixedly delayed by a predetermined time Tsp from the sampling timing of the V-phase current. The predetermined time Tsp corresponds to the time required for the output value capturing process and A / D conversion, and is thus determined by the capability of the control device 30.

  Sampling points S1 and S5 correspond to the timing when the V-phase current takes an extreme value, and sampling points S4 and S8 correspond to the timing when the W-phase current takes an extreme value. The timing at which each phase current takes an extreme value corresponds to the detection timing of the current amplitude.

  However, as can be understood from the comparison of the sampling points S1 and S4 and the comparison of the sampling points S5 and S8 in FIG. 6, the combination of the current change due to the ripple current and the fixed sampling time difference Tsp results in the current amplitude. There is a possibility that an offset current difference occurs between the V phase and the W phase for the corresponding current sampling value.

  Generally, current detection values (Iv, Iw) used for PWM control are generated in accordance with the current sampling value that has passed through the low-pass filter. Therefore, the detected current difference between the phases as described above gives an offset error to the current control by the PWM control.

Specifically, as shown in FIG. 7, the current feedback control by the PWM control unit 200 includes the current difference between the sampling points S1 and S4, and the sampling points S5 and S4.
It acts to eliminate the current difference between 8. As a result, the offset error Iof is superimposed on the W-phase current as a direct current component.

  Therefore, in the control apparatus for an AC motor according to the embodiment of the present invention, even if the output values from the multiple-phase current sensors are sequentially sampled, the problems as shown in FIGS. 6 and 7 do not occur. The current sampling order between the phases is appropriately changed.

  Note that the above problem becomes conspicuous at the time of applying synchronous PWM in which the number of times of control (that is, the number of times of current sampling) is reduced every 360 degrees of electrical angle of AC electric motor M1. For this reason, in the following, the current sampling process in the synchronous PWM with k = 6 will be exemplified.

  FIG. 8 is a conceptual diagram illustrating a first example of current sampling processing performed by the AC motor control device according to the embodiment of the present invention.

  Referring to FIG. 8, in k = 6 synchronous PWM, a current sampling instruction (signal Fsp in FIG. 5) is generated every half cycle of carrier wave 160, that is, every 30 electrical degrees. Therefore, in the synchronous PWM, the electrical angle corresponding to the current sampling instruction can be obtained from the count of the number of carriers of the carrier wave 160 and the phase. On the other hand, also in the asynchronous PWM, it is possible to obtain the electrical angle corresponding to the generated current sampling instruction based on the output (rotation angle θ) of the rotation angle sensor 25 or the like.

  As shown in FIG. 8, each phase current has a relationship in which the phase is shifted by 120 degrees in electrical angle, so it is determined at which electrical angle the extreme value (maximum value or minimum value) is obtained. Specifically, the W-phase current (Iw) has a minimum value and a maximum value at electrical angles of 150 degrees and 330 degrees, respectively, and the V-phase current (Iv) has a minimum value and an electrical angle of 30 degrees and 210 degrees, respectively. Take the maximum value. That is, by obtaining the electrical angle corresponding to the current sampling instruction, it is possible to determine the presence / absence of a phase in which the phase current has an extreme value at the sampling timing and to identify the phase in which the phase current has an extreme value. is there.

  In the current sampling processing according to the present embodiment, the current sampling order between a plurality of phases is not fixed, and the current phase of any phase is set to the timing corresponding to the electrical angle at which the current reaches a maximum value (maximum value or minimum value). The current sampling order is changed within one cycle (electrical angle 360 degrees) so that sampling is performed first.

  FIG. 9 is a flowchart showing a processing procedure of a first example of the current sampling processing according to the present embodiment. Control processing according to the flowchart shown in FIG. 9 is executed by control device 30 every time a current sampling instruction is generated.

  Referring to FIG. 9, control device 30 obtains an electrical angle in step S <b> 110 in response to generation of a current sampling instruction. As described above, the electrical angle can be obtained in both synchronous PWM and asynchronous PWM.

  Further, in step S120, control device 30 determines whether the electrical angle obtained in step S110 is an electrical angle at which the W-phase current has an extreme value, that is, whether the electrical angle is 150 degrees or 330 degrees. judge.

  When YES is determined in step S120, the control device 30 performs current sampling at a timing when the W-phase current reaches an extreme value, and thus proceeds to step S130 to execute the W-phase current (that is, the W-phase current sensor). 24 output values iws) are sampled first. Thereafter, in step S132, control device 30 samples the remaining V-phase current (that is, output value ivs of V-phase current sensor 24).

  When NO is determined in step S210, control device 30 proceeds to step S140 and samples the V-phase current first. Thereafter, control device 30 samples the remaining W-phase current in step S142. Thereby, at the sampling timing corresponding to the electrical angles of 30 degrees and 210 degrees, the V-phase current having the extreme value can be sampled first.

  It should be noted that the sampling order can be arbitrary for electrical angles (electrical angles other than 30 degrees, 150 degrees, 210 degrees, and 330 degrees) in which neither the V-phase current nor the W-phase current takes an extreme value. In FIG. 9, for simplification of the processing, the processing when the electrical angle is not 150 degrees or 330 degrees (when NO is determined in S120) is constant.

  FIG. 10 shows a current sampling process according to the processing procedure of FIG. 9 as compared with FIG.

  In FIG. 10, unlike FIG. 6, sampling points S <b> 4 and S <b> 8 at which the W-phase current takes an extreme value are provided prior to V-phase current sampling points S <b> 3 and S <b> 6 corresponding to the same current sampling instruction. As a result, the influence of the ripple current is made uniform between the V-phase current (maximum value) sampled at the sampling points S1 and S5 and the W-phase current (maximum value) sampled at the sampling points S4 and S8. be able to.

  Therefore, even if a time difference (Tsp) occurs in the sampling between the V phase and the W phase due to the sequential sampling process, the detection error between phases at the detection timing of the current amplitude as pointed out in FIG. 6 is prevented. it can. As a result, the problem that the current detection error is reflected in the current feedback control as shown in FIG. 7 can be avoided.

  That is, even when the current sensors 24 arranged in a plurality of phases are sequentially sampled, it is possible to avoid an adverse effect on the motor control due to the current detection error.

  4 illustrates the configuration in which the current sensors 24 are arranged in the V phase and the W phase, but the same current sampling is possible even in a configuration in which the current sensors 24 are arranged in each of the U phase, the V phase, and the W phase. Processing can be executed. The process at this time will be described with reference to FIGS. 11 and 12 as a modification of the current sampling process (first example) shown in FIGS.

  FIG. 11 further shows a U-phase current (Iu) in addition to the V-phase and W-phase currents shown in FIG. Since the current phase of each phase is shifted by 120 degrees in electrical angle, the U-phase current has a local maximum value at an electrical angle of 90 degrees and a local minimum value at an electrical angle of 270 degrees. The electrical angles at which the V-phase current and the W-phase current take extreme values are as described in FIG.

  Referring to FIG. 12, in response to the generation of the current sampling instruction, control device 30 obtains an electrical angle in step S110 as in FIG. 9, and in step S120, electric power at which the W-phase current takes an extreme value. It is determined whether the angle is 150 degrees or 330 degrees.

  At the time of YES determination in step S120, control device 30 samples W-phase current, which is the timing of the extreme value, first in step S130 similar to FIG. Further, control device 30 sequentially samples U-phase current and W-phase current after completion of sampling of W-phase current in steps S132 and S134. The order of steps S132 and S134 can be interchanged.

  When NO is determined in step S120, control device 30 determines in step S125 whether or not the V-phase current is an electrical angle (30 degrees or 210 degrees) at which an extreme value is obtained.

  At the time of YES determination in step S125, control device 30 samples V-phase current, which is the timing at which the extreme value is reached, in step S140 first. Further, control device 30 sequentially samples the W-phase current and the U-phase current after completing the sampling of the W-phase current in steps S142 and S144. The order of steps S142 and S144 can be interchanged.

  When NO is determined in step S125, control device 30 proceeds to step S150 and samples U-phase current first. Therefore, at the sampling timing corresponding to the electrical angles of 90 degrees and 270 degrees, the U-phase current taking the extreme value can be sampled first. Thereafter, control device 30 sequentially samples the remaining V-phase current and W-phase current in steps S152 and S154. The order of steps S152 and S154 can be interchanged.

  Also, the electrical angle at which none of the U-phase current, V-phase current, and W-phase current at which the current sensor 24 is disposed takes an extreme value (other than 30 degrees, 90 degrees, 150 degrees, 210 degrees, 270 degrees, and 330 degrees) In electrical angle, the sampling order can be arbitrary. In FIG. 12, for simplification of the processing, the processing when the electrical angle is not any of 30 °, 150 °, 210 °, and 330 ° (when NO is determined in S120 and S125) is constant.

  In this way, even in the configuration in which the current sensor 24 is arranged in each phase, current detection errors between phases can be suppressed even if there is a time difference in actual sampling timing between phases due to sequential sampling processing. That is, as described with reference to FIGS. 8 to 10, even when the current sensors 24 arranged in a plurality of phases are sequentially sampled, it is possible to avoid an adverse effect on the motor control due to the current detection error.

  In the above description, the description is focused on the synchronous PWM in which the influence of the current detection error is large. However, it is confirmed that the same current sampling process can be applied to the asynchronous PWM control based on the electrical angle. To do. Also, the PWM control unit of FIG. 4 has exemplified the configuration that can select synchronous PWM or asynchronous PWM. However, even when one of synchronous PWM or asynchronous PWM is fixedly used, current sampling processing according to the present embodiment Is applicable.

  Furthermore, in the configuration illustrated in FIG. 4 where the synchronous PWM and the asynchronous PWM can be selected, the above-described current sampling process is applied only when the synchronous PWM is applied, while the current sampling instruction timing ( The current sampling order may be fixed regardless of the electrical angle. In this way, the processing load of the control device 30 can be reduced in the asynchronous PWM in which the current detection error due to the sampling order is relatively small because the number of times of current sampling is large while the control cycle is relatively short.

  Next, as a second example of current sampling according to the present embodiment, a control processing procedure for specifying the same sampling order as in the first example based on only the detected current value without obtaining an electrical angle. explain.

  FIG. 13 is a flowchart showing a processing procedure of a second example of the current sampling processing according to the present embodiment. FIG. 13 shows a processing procedure in a configuration in which current sensors 24 are arranged in the V phase and the W phase. Control processing according to the flowchart shown in FIG. 13 is executed by control device 30 every time a current sampling instruction is generated.

  Referring to FIG. 13, in response to the generation of the current sampling instruction, control device 30 reads the V-phase current value and the W-phase current value at the previous current sampling from steps S210 and S212, respectively.

  In step S220, control device 30 compares the absolute values between W-phase current (Iw) and V-phase current (Iv). In step S220, it is determined whether | Iw |> | Iv | + α is satisfied. Therefore, when | Iw | = | Iv |, S220 is NO.

  When YES is determined in step S220, control device 30 proceeds to step S230 and samples W-phase current (that is, output value iws of W-phase current sensor 24) first. Thereafter, in step S232, control device 30 samples the remaining V-phase current (that is, output value ivs of V-phase current sensor 24).

  On the other hand, when NO is determined in step S220, control device 30 proceeds to step S240 and samples the V-phase current (that is, output value ivs of V-phase current sensor 24) first. After that, in step S242, control device 30 samples the remaining W-phase current (that is, output value iws of W-phase current sensor 24).

  In this way, the transition of the V-phase current and the W-phase current and the current sampling order of the V-phase and the W-phase are as shown in FIG.

  Referring to FIG. 14, the V-phase current and the W-phase current that are 120 degrees out of phase have the same absolute value at electrical angles of 0 degrees, 90 degrees, 180 degrees, 270 degrees, and 360 degrees. Since the margin value α is provided in the determination formula in step S220, step S220 is NO in electrical angles of 0 degrees, 90 degrees, 180 degrees, 270 degrees and 360 degrees.

  As a result, in the period of electrical angle 0 degree (360 degrees) to 90 degrees and the period of 180 degrees to 270 degrees, the V-phase current is sampled first, while the electrical angles are 120 degrees, 150 degrees, 300 degrees, and 330 degrees. In the case of the degree, the W-phase current is sampled first.

  As a result, the V-phase current can be sampled first at electrical angles of 30 degrees and 210 degrees at which the V-phase current takes extreme values. Further, the W-phase current can be sampled first at the electrical angles of 150 degrees and 330 degrees at which the W-phase current takes extreme values.

  Note that the sampling order can be arbitrary for electrical angles in which the current does not become an extreme value in any phase, but in FIG. 13, in order to simplify the processing, in these cases, steps S240 and S242 are commonly followed. It is assumed that the current is sampled in the same order.

  As described above, even if only the detection value by the current sensor 24 is used, as in the first example of the current sampling process described above, the electrical angle at which the current of any phase becomes an extreme value (maximum value or minimum value). At the timing corresponding to, the current sampling order can be determined within one cycle (electrical angle 360 degrees) so that the phase is sampled first.

  Next, a control processing procedure for applying the current sampling processing (second example) according to FIGS. 13 and 14 to the configuration in which the current sensor 24 is arranged in each of the U phase, the V phase, and the W phase, This will be described with reference to FIGS. 15 and 16.

  Referring to FIG. 15, similarly to FIG. 13, control device 30 reads the V-phase current value and the W-phase current value at the time of the previous current sampling in steps S <b> 210 and S <b> 212, respectively. Further, in step S214, control device 30 reads the U-phase current value at the previous current sampling.

  Then, similarly to FIG. 13, the control device 30 determines whether or not | Iw |> | Iv | + α in step S220. Further, similarly to FIG. 13, when the determination in step S220 is YES, control device 30 proceeds to step S230 and samples the W-phase current first. Thereafter, control device 30 sequentially samples the remaining U-phase current and V-phase current in steps S232 and S234. The order of steps S232 and S234 can be interchanged.

  On the other hand, when NO is determined in step S220, the control device 30 determines whether or not | Iv |> | Iu | + α in step S225. Then, when YES is determined in step S220, control device 30 proceeds to step S240 to sample the V-phase current first. Thereafter, control device 30 sequentially samples the remaining W-phase current and U-phase current in steps S242 and S244. The order of steps S242 and S244 can be interchanged.

  When NO is determined in step S225, control device 30 proceeds to step S250 and samples the U-phase current first. Thereafter, control device 30 sequentially samples the remaining V-phase current and W-phase current in steps S252 and S254. The order of steps S252 and S254 can be interchanged.

  Referring to FIG. 16, based on the comparison of | Iw | and | Iv | similar to FIG. 13, at the electrical angles of 150 degrees and 330 degrees at which the W-phase current takes extreme values, the processing of steps S230 to S234 is performed. The W-phase current is sampled first.

  Further, at electrical angles of 30 degrees and 210 degrees at which the V-phase current takes extreme values, | Iu | = 0 at the time of the previous current sampling (electrical angles of 0 degrees and 180 degrees), so step S225 is determined as YES. The Therefore, the V-phase current is sampled first by the processes of steps S240 to S244.

  Further, at the electrical angles of 90 degrees and 270 degrees at which the U-phase current has an extreme value, | Iu | = | Iv | at the previous current sampling (electrical angles of 60 degrees and 240 degrees), step S225 is NO. It is said. Therefore, the U-phase current is sampled first by the processes of steps S250 to S254.

  Thus, even in the configuration in which the current sensor 24 is disposed in each phase, the same current sampling process (second example) as in FIGS. 13 and 14 can be realized. That is, based on only the current detection value, at the timing corresponding to the electrical angle at which the current of any phase becomes an extreme value (maximum value or minimum value), the phase is sampled first. The current sampling order can be determined within the period (electrical angle 360 degrees).

  Note that the second example of the current sampling process described with reference to FIGS. 13 to 16 can also be applied to asynchronous PWM by appropriately setting the margin value α in steps S220 and S225. Further, in asynchronous PWM, steps S220 and S225 are performed based not only on the previous current sampling but also on the detection value at the time of current sampling earlier (or the average value or filter processing value of a plurality of current detection values). The comparison determination may be performed.

  Furthermore, in addition to the configuration in which synchronous PWM and asynchronous PWM can be selected as illustrated in FIG. 4, the second example of the current sampling process described above is also applied when one of synchronous PWM or asynchronous PWM is fixedly used. Describe what is possible.

  In the second example of the current sampling process, in the configuration in which the synchronous PWM and the asynchronous PWM can be selected, the above-described current sampling process is applied only when the synchronous PWM is applied. The current sampling order may be fixed regardless of the timing (electrical angle) of the sampling instruction.

  As described above, according to the current sampling process (each of the first and second examples) by the AC motor control device according to the present embodiment, the current sensors 24 arranged in a plurality of phases are sequentially sampled. However, it is possible to avoid an adverse effect on the motor control due to the current detection error.

  In the present embodiment, the three-phase motor is exemplified as the AC motor M1, but the current sampling process according to the present invention can be applied to a multiphase motor other than the three-phase motor.

  In FIG. 1, as a preferred configuration example, a configuration is shown in which DC voltage generation unit 10 # of the motor drive system includes converter 12 so that the input voltage (system voltage VH) to inverter 14 can be variably controlled. DC voltage generation unit 10 # is not limited to the configuration exemplified in the present embodiment. That is, it is not essential that the inverter input voltage is variable, and the present invention is also applied to a configuration in which the output voltage of the DC power supply B is directly input to the inverter 14 (for example, a configuration in which the converter 12 is omitted). Is possible.

  Further, with regard to the AC motor serving as a load of the motor drive system, in the present embodiment, a permanent magnet motor mounted for driving a vehicle in an electric vehicle (hybrid vehicle, electric vehicle, etc.) is assumed. The present invention can also be applied to a configuration in which an arbitrary AC motor used in is used as a load.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  The present invention can be applied to electric motor control based on sampling values of current sensors provided in a plurality of phases.

  5 Ground wire, 6, 7 Power line, 10, 13 Voltage sensor, 10 # DC voltage generator, 11, 24 Current sensor, 12 Converter, 14 Inverter, 15 U phase upper and lower arm, 16 V phase upper and lower arm, 17 W phase upper and lower Arm, 25 rotation angle sensor, 30 control unit (ECU), 100 motor drive control system, 160 carrier wave, 170 voltage command, 180 pulse width modulation voltage, 200 PWM control unit, 210 current command generation unit, 220, 250 coordinate conversion unit 240 voltage command generation unit, 260 PWM modulation unit, 270 carrier wave control unit, 280 mode selection unit, 290 rotation frequency calculation unit, 300 current sampling unit, 310 sampling timing designation unit, 320 sampling order designation unit, 330 sampling processing unit, B DC power supply, C0, C1 Capacitor, D1 to D8 Anti-parallel diode, fc carrier frequency, For signal (sampling order), Fsp signal (current sampling instruction), Ib DC current, Id d-axis current, Idcom d-axis current command value, Iq q-axis current, Iqcom q-axis current command value, iu, iv, iw Three-phase current (AC motor), Iv, Iw Current detection value, ivs, iws Current sensor output value, k number of carriers (synchronous PWM), L1 reactor, M1 AC motor, MD Mode signal (synchronous PWM / asynchronous PWM), Q1-Q8 power semiconductor switching element, S1-S8 current sampling point, SG1-SG8 switching control signal, SR1, SR2 system relay, Trqcom torque command value, Tsp sampling time difference (interphase) , Vd # d-axis voltage command value VH system voltage (direct current voltage), VL DC voltage, Vu, Vv. Vw Each phase voltage command, Vq # q-axis voltage command value, ΔId d-axis current deviation, ΔIq q-axis current deviation, θ rotation angle, ωe rotation frequency.

Claims (7)

An AC motor control device in which an applied voltage is controlled by an inverter,
A plurality of current detectors provided respectively in a plurality of phases of the AC motor and configured to detect phase currents of the plurality of phases;
A sampling timing instruction unit configured to generate a current sampling instruction according to a predetermined control period;
In response to the current sampling instruction, a sampling processing unit configured to sequentially sample detection values by each of the plurality of current detectors one by one;
An electric motor control unit configured to control the AC electric motor according to a control calculation using the phase current sampled by the sampling processing unit;
Corresponding to the first phase of the plurality of current detectors when the timing of the current sampling instruction corresponds to the timing at which the phase current takes an extreme value in the first phase of the plurality of phases A control apparatus for an AC motor, comprising: a sampling order designating unit configured to designate a sampling order in the sampling processing unit so that a detection value of a current detector to be sampled is sampled first.   The AC motor control apparatus according to claim 1, wherein the sampling order designating unit is configured to designate a sampling order based on an electrical angle of the AC motor at a timing of the current sampling instruction.   The sampling order specifying unit is configured to specify a sampling order in the current sampling instruction based on a comparison of sampling values of the phase currents of the plurality of phases in the current sampling instruction in the past. 1. The control apparatus for an AC motor according to 1. The motor controller is
A calculation unit configured to generate a voltage command for each phase of the AC motor according to the control calculation;
A PWM modulation unit configured to control a pulse width modulation voltage of each phase applied from the inverter to the AC electric motor based on a voltage comparison between the voltage command and a carrier wave;
The AC motor control according to any one of claims 1 to 3, further comprising: a carrier wave control unit for generating the carrier wave at a frequency of k times the frequency of the voltage command (k: an integer of 2 or more). apparatus. The carrier wave control unit generates the carrier wave at a frequency k times the voltage command when the first mode is selected, while the carrier frequency when the first mode is selected when the second mode is selected. Configured to generate the carrier at a predetermined frequency higher than
5. The control of an AC motor according to claim 4, wherein, when the second mode is selected, the sampling order designating unit fixes the sampling order in the sampling processing unit to a predetermined order regardless of the timing of the current sampling instruction. 6. apparatus. The AC motor is a three-phase AC motor,
The control device for an AC motor according to claim 1, wherein the plurality of current detectors are arranged in three phases, respectively. The AC motor is a three-phase AC motor,
The control device for an AC motor according to any one of claims 1 to 5, wherein the plurality of current detectors are respectively arranged in two phases of three phases.

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