HK1063895A - Multipath cdma receiver for reduced pilot - Google Patents

Description

Reduced pilot multipath CDMA receiver

The present application is a divisional application of the patent application entitled "pilot-reduced multipath code division multiple access receiver" filed on 27.1.1999, application number 99806092.5 (international application number PCT/US 99/01794).

Technical Field

The present invention relates generally to digital communications. More particularly, the present invention relates to a system and method for using a code division multiple access air interface that significantly reduces the signal power required for global and assigned pilots while improving operational performance by using Quadrature Phase Shift Keying (QPSK) traffic signals for a particular channel to perform channel estimation and carrier recovery.

Prior Art

Most advanced communication technologies today use digital spread spectrum modulation or Code Division Multiple Access (CDMA). Digital spread spectrum is a communication technique in which data is transmitted in a frequency band (spread spectrum) that is widened by modulating the data to be transmitted with a pseudo noise signal. CDMA can transmit data without being affected by signal distortion or interference frequencies in the transmission path.

A simplified CDMA communications system is shown in fig. 1 and includes a single communications channel of a given bandwidth which is mixed by a predetermined pattern of spreading codes generated by a repeating pseudo-noise (pn) sequence generator. Modulating the data signal with the pn sequence produces a digital spread spectrum signal. The carrier signal is then modulated with a digital spread spectrum signal, a forward link is established, and the carrier signal is transmitted. The receiver demodulates the transmitted carrier signal and extracts a digital spread spectrum signal. The transmitted data is regenerated after correlation with the matching pn sequence. The same process is repeated to establish a reverse link.

During terrestrial communications, the transmitted signal is subject to disturbances due to various terrain and environmental conditions, as well as man-made obstacle reflections. This produces multiple received signals at the receiver with different time delays. This effect is commonly referred to as multipath propagation. In addition, each path arrives at the receiver with a delayed signal with a unique amplitude and carrier phase.

In order to identify multiple components in multipath propagation, the relative delay as well as the amplitude and phase must be determined. This determination can be performed with a modulated data signal, but more accurate reproduction is generally obtained when compared to an unmodulated signal. In most digital spread spectrum systems, it is more efficient to use an unmodulated pilot signal that is scattered from the transmitted modulated data by assigning a separate pn sequence to the pilot signal. For systems that transmit many signals from one base station to multiple users, the global pilot signal is most valuable.

In the case of a base station transmitting many channels, the global pilot signal is provided by the same pilot sequence by multiple users serving that particular base station and is used for initial acquisition by an individual user and for users to obtain channel estimates for coherent reception and to combine multipath components. However, at the required signal strength, the global pilot signal may use up to ten percent of the forward air capacity.

The same multipath distortion affects the reverse link transmission of the user to the base station. Inserting a given pilot in each individual user return signal may consume up to twenty percent of the total reverse channel air capacity.

A typical prior art system is disclosed in EP0675606a1, which combines a remodulated signal with a baseband signal to obtain frequency offset components. This frequency offset component of the combined signal corrects for the frequency offset of the channel signal.

Without phase and amplitude estimation, non-coherent or differentially coherent reception techniques must be performed. Accordingly, there is a need for a coherent demodulation system that reduces the air capacity of the global pilot and designated pilot signals while maintaining desired air interface performance.

Summary of the invention

The present invention relates to a digital spread spectrum communication system that utilizes pilot-assisted coherent multipath demodulation to substantially reduce global pilot and assigned pilot overhead. The system and method uses a QPSK modulated data signal to remove the modulated data and use the recovered carrier for channel amplitude and phase estimation. The resulting signal is free of data modulation and is used as a pseudo pilot signal. In one embodiment of the invention, in combination with the pseudo pilot signals, a multiple input phase locked loop is used to further eliminate errors due to carrier offset caused by the use of multiple pseudo pilot signals. A pilot signal is still required to resolve the absolute phase ambiguity, but at a much reduced amplitude.

It is therefore an object of the present invention to provide a code division multiple access communication system which reduces the required global and dedicated pilot signal strengths.

It is another object of the invention to reduce the transmit levels of the global and designated pilot signals so that they consume negligible overhead in the air interface while providing the information needed for coherent demodulation.

Other objects and advantages of the systems and methods will become apparent to those skilled in the art upon a reading of the detailed description of the preferred embodiments.

Fig. 1 is a simplified block diagram of a typical CDMA communication system of the prior art;

FIG. 2 is a detailed block diagram of a B-CDMATM communication system;

FIG. 3A is a graph of an in-phase bit stream;

FIG. 3B is a graph of an orthogonal bit stream;

FIG. 3C is a graph of a pseudo-noise (pn) bit sequence;

FIG. 4 is a detailed block diagram of the present invention using a pseudo pilot signal with carrier offset correction performed at the chip level;

fig. 5 is a block diagram of a rake receiver;

FIG. 6 is a diagram of a received symbol p on a QPSK constellation showing hard decisions₀A schematic diagram of (a);

FIG. 7 is a schematic diagram of a correction angle corresponding to a given symbol;

FIG. 8 is a schematic illustration of the resulting symbol error after applying a correction corresponding to a given symbol;

FIG. 9 is a block diagram of a conventional phase locked loop;

FIG. 10 is a detailed block diagram of the present invention that performs carrier offset correction at the symbol level using a dummy pilot signal;

FIG. 11 is a detailed block diagram of the present invention for performing carrier offset correction at the chip level using a dummy pilot and MIPLL;

FIG. 12 is a block diagram of a multiple input phase-locked loop (MIPLL);

fig. 13 is a detailed block diagram of the present invention that performs carrier offset correction at the symbol level using a dummy pilot signal and MIPLLs.

The preferred embodiments are described below with reference to the attached drawings, wherein like reference numerals refer to like elements throughout.

A B-cdma tm communication system, as shown in fig. 2, includes a transmitter 27 and a receiver 29, which may be present in either a base station or a mobile subscriber receiver. Transmitter 27 includes a signal processor 31 that encodes voice or non-voice signals 33 into data at different data rates, for example, data rates of 8kbps, 16kbps, 32kbps, or 64 kbps. The signal processor 31 selects a data rate or responds to a set data rate depending on the type of signal.

By way of background, the generation of a transmitted signal in a multiple access environment involves two steps. First, input data 33, which may be considered a bi-phase modulated signal, is encoded with a forward error correction code (FEC) 35. For example, if an R-1/2 convolutional code is used, the single bi-phase modulated data signal becomes a bivariate or two bi-phase modulated signal. One signal is assigned to the in-phase channel I41 a. The other signal is assigned to the quadrature channel Q41 b. A plurality ofIs of the form a + bj, where a and b are real numbers, and j²Is-1. Bi-phase modulated I and Q signals are commonly referred to as Quadrature Phase Shift Keying (QPSK). In the preferred embodiment, the tap generator polynomial for the convolutional code rate of K-7 and R-1/2 is G₁＝171₈37 and G₂＝133₈39。

In a second step, the two bi-phase modulated data or symbols 41a, 41b are spread with a complex pseudo-noise (pn) sequence. The resulting I45 a and Q45 b spread signals are mixed 53 with other spread signals (channels) having different spreading codes, multiplied (mixed) with a carrier signal 51, and transmitted 55. The transmission 55 may comprise a plurality of separate channels having different data rates.

The receiver 29 includes demodulators 57a, 57b that down-mix the transmitted wideband signal 55 to an intermediate carrier frequency 59a, 59 b. The second down-conversion reduces the signal to baseband. The QPSK signal is then filtered 61 and mixed with a locally generated complex pn sequence 43a, 43b, which complex pn sequence 43a, 43b matches the conjugate of the transmitted complex code. Only the original waveform spread with the same code at the transmitter 27 is effectively despread. Other waveforms appear as noise to the receiver 29. The data 65a, 65b is then passed to a signal processor 67 where FEC decoding of the convolutionally encoded data is performed in the signal processor 67.

As shown in fig. 3A and 3B, one QPSK symbol is composed of one bit from each of in-phase (I) and quadrature (Q) signals. These bits represent an analog sample or a quantized version of digital data. The duration t of the symbol can be seen_sEqual to the duration of the bit.

The transmitted symbols are spread by multiplying the QPSK symbol stream by a unique complex pn sequence. Both the I and Q pn sequences are made up of a bit stream that is generated at a much higher rate, typically 100 to 200 times the symbol rate. One such pn sequence is shown in fig. 3C. The complex pn sequence is mixed with a complex symbol bit stream to produce a digital spread signal. The component of the spread signal is called a chip, which has a much smaller duration t_c。

The baseband signal is at the chip level when the signal is received and demodulated. Both the I and Q components of the signal are despread with the conjugate of the pn sequence used during spreading, returning the signal to the symbol level. However, due to the carrier offset, the phase deterioration suffered during transmission is manifested by distortion of the respective chip waveforms. If the carrier offset correction is done at the chip level, it can be seen that the overall accuracy is improved due to the inherent resolution of the chip level signal. Carrier offset correction can also be done at the symbol level, but with less overall accuracy. However, since the symbol rate is much smaller than the chip rate, less overall processing speed is required when correction is performed at the symbol level.

The following is a system configuration of a receiver in accordance with the system and method of the present invention that does not require a large amplitude pilot signal. The following system replaces the filtering, despreading and signal processing in fig. 2. The system has carrier offset correction at the chip and symbol level.

As shown in fig. 4, a receiver using the system 75 and method of the present invention is shown. A complex baseband digital spread spectrum signal 77 composed of in-phase and quadrature-phase components is input and filtered with an Adaptive Matched Filter (AMF)79 or other adaptive filtering means. The AMF79 is a transversal filter (finite impulse response filter) that uses filter coefficients 81 to superimpose delayed replicas of the received signal 77 on each other to provide a filtered signal 83 having an increased signal-to-noise ratio (SNR). Coupling the output 83 of the AMF79 to a plurality of channel despreaders 85₁，85₂，85_nAnd a pilot despreader 87. In a preferred embodiment, n-3. In the pn sequence 93 assigned to each user₁，93₂，93_nDe-spreading 85₁，85₂，85_nWhile transmitting data 77, a separate despreader 87 and pn sequence 91 are used to despread pilot signal 89. Despreading 85 in data channel₁，85₂，85_nThereafter, the data bit stream 95₁，95₂，95_nCoupled to viterbi(Viterbi) decoder, and output 99₁，99₂，99_n。

The filter coefficients 81, or weights, used in adjusting the AMF79 are obtained by demodulation of the individual multipath propagation paths. This operation is performed by a rake receiver 101. Compensation for multipath distortion with rake receiver 101 is well known to those familiar with communications technology.

As shown in fig. 5, rake receiver 101 is formed by a plurality of path demodulators ("fingers") 103 that demodulate a particular multipath component₀，103₁，103₂，103_nAre combined in parallel. The pilot sequence tracking loop for a particular demodulator is initialized with the timing estimate for a given path determined by the pn sequence 105. In the prior art, one pilot signal is used to despread each signal of the rake receiver. In this embodiment of the invention, the pn sequence 105 may belong to any channel 93 of the communication system₁. The channel with the largest received signal is typically used.

Each path demodulator includes a complex mixer 107₀，107₁，107₂，107_nAnd adder and latch 109₀，109₁，109₂，109_n. For each rake element, pn sequence 105 is delayed by τ 111₁，111₂，111_nOne chip and mixed 107 with the baseband spread spectrum signal 113₁，107₂，107_nThereby despreading each signal. Each product is input to an accumulator 109₀，109₁，109₂，109_nIt is added to the previous product in the accumulator and latched after the next symbol-clock cycle. The rake receiver 101 provides a relative path value to each multipath component. A plurality of n-dimensional outputs 115₀，115₁，115₂，115_nEstimates of the sampled channel impulse response are provided that contain a relative phase error of 0, 90, 180 or 270.

Referring back to fig. 4, the multiple outputs from the rake receiver are coupled to an n-dimensional complex mixer 117. Mixed with each rake receiver 101 output 115 is a correction signal to remove the relative phase error contained in the rake receiver output.

The pilot signal is also a complex QPSK signal, but with the quadrature component set to zero. The error correction signal 119 of the present invention is generated by first performing despreading of the despread signal 95₁From the despread channel 95 by hard decision 121 per symbol₁And (4) deriving. A hard decision processor 121 determines the QPSK constellation position that is closest to the despread symbol value.

As shown in fig. 6, the euclidean distance processor processes the received symbols p for channel 1₀And four QPSK constellation points x_1，1，x_-1，1，x_-1，-1，x_1，-1And (6) comparing. Due to multipath or noise and distortion corruption during the radio frequency transmission 55, each received symbol p needs to be examined₀. Hard decision processor 121 computes a slave received symbol p₀Four distances d to each quadrant₁，d_-2，d₃，d₄And selecting the shortest distance d_-2And specifies the symbol position x_-1，1. Discarding the original symbol coordinate p₀。

Referring back to fig. 4, after each hard symbol decision 121 is passed, the complex conjugate 123 of each symbol output 125 is determined. A complex conjugate is one of a pair of complex numbers with the same real part and imaginary parts that differ only in sign.

As shown in fig. 7, by first determining the designated symbol coordinate x_-1，1Demodulates or derotates one symbol to form a correction signal 119 for removing the relative phase error contained in the rake receiver output. Thus, the relative phase error is eliminated by effectively de-shuffling the rake receiver output by the angles associated with the hard decisions. This operation effectively provides a rake receiver driven by the pilot signal but without an absolute phase reference.

Referring back to fig. 4, the output 119 from complex conjugate 123 is coupled to a complex n-dimensional mixer 117 where each output of rake receiver 101 is mixed with correction signal 119. The resulting product 127 is the channel impulse response p as shown in fig. 8₁The noise estimate of (2). The error shown in fig. 8 is expressed in radians distance of pi/6 from the in-phase axis.

Referring back to fig. 4, the output 129 of the complex n-dimensional mixer 117 is coupled to an n-dimensional channel estimator 131. The channel estimator 131 is a plurality of low pass filters that filter each multipath component. The output of the n-dimensional mixer 117 is coupled to the AMF 79. These signals serve as filtering weights for the AMF 79. The AMF79 filters a baseband signal to compensate for channel distortion due to multipath transmission without a pilot signal of large amplitude.

The rake receiver 101 is used in conjunction with a phase-locked loop (PLL)133 circuit to remove carrier offset. Carrier offset is generated as a result of transmitter/receiver component mismatch and other RF distortions. The present invention 75 requires that a low level pilot signal 135 be generated by despreading 81 the pilot from the baseband signal 77 with a pilot pn sequence 91. The pilot signal is coupled to a single input PLL 133. PLL133 measures the phase difference between pilot signal 135 and a 0 reference phase. The despread pilot signal 135 is the actual error signal coupled to the PLL 133.

A conventional PLL133 is shown in fig. 9. The PLL133 includes an arctangent analyzer 136, a complex filter 137, an integrator 139, and a phase-to-complex converter 141. The pilot signal 135 is an error signal input to the PLL133 and is coupled to a complex filter 137. The complex filter 137 includes two gain stages, an integrator 145, and a summer 147. The output from the complex filter is coupled to an integrator 139. The integral of the frequency is the phase of the output 140 to the converter 141. The phase output 140 is coupled to a converter 141, which converter 141 converts the phase signal into a complex signal for mixing 151 with the baseband signal 77. Since upstream operations are interchangeable, the output 149 of the PLL133 is also fed back to the system 75.

By performing hard decision 121 and derotation 123 of data modulation, the process provides channel estimation without using large pilot signals. If an error occurs during the hard decision process and the quadrant receiving the data symbol is not correctly assigned, the process generates a phase error. However, the likelihood of phase error is reduced due to the increased signal-to-noise ratio of the traffic channel. Errors generated during the channel estimation and carrier recovery processes are filtered out. The traffic channel is approximately 6dB (2x) stronger than the despread pilot signal.

As previously mentioned, the invention can also be performed with carrier offset correction at the symbol level. An alternative embodiment 150 implemented at the symbol level is shown in fig. 10. The difference between chip and symbol level processing is generated in the case of combining the outputs of the conventional PLL 133. At the symbol level, the PLL output 140 is not subjected to chip conversion 141 and is weighted by another n-dimensional mixer 153 after rake receiver 101, introduced to AMF 97. Phase correction signal 140 feedback must also be communicated to channel despreader 85₁，85₂，85_nA plurality of outputs 95₁，95₂，95_nEach of the mixers 154₁，154₂，154_nAnd mixed 156 with the output 135 of pilot despreader 87.

Another alternative embodiment 193, as shown in fig. 11, uses a variation of the previous embodiment whereby hard decisions are made for each received symbol after despreading and derotation is performed by an amount of radians equal to the complex conjugate. As shown in FIG. 11, an alternative method 193 uses multiple channel despreaders 85₁，85₂，85_nAnd a pilot despreader 87 as input to a Multiple Input Phase Locked Loop (MIPLL) 157. Since each despreading channel 95₁，95₂，95_nContains an ambiguous representation of the pilot signal and therefore requires a small signal pilot signal 135 as an absolute reference. The despread symbols from all channels combined with the despread small-signal pilot signal are input to the MIPLL 157.

Referring to fig. 12, from each channel 95₁，95₂，95_nIs coupled to a hard decision/complex conjugate operation 159₁，159₂，159_n. Then, the derotated pseudo pilot signal 161 is transmitted₁，161₂，161_nMixing 163 with delayed symbols₁，163₂，163_nA complex voltage error is generated. Error 165 will be₁，165₂，165_nInput to converter 167₁，167₂，167_n，167_n+1The converter converts the complex arc tangent to a phase error 169₁，169₂，169_n，169_n+1. Each phase error 169₁，169₂，169_n，169_n+1The inputs are fed to a maximum likelihood combiner 171 which assigns different weights to the inputs and produces a sum output. Also included in the sum are despreading 135 and conversion 167_n+1Phase 169 of the small signal pilot signal 135_n+1. The weighting of the small pilot signal can be emphasized because its phase is unambiguous.

The output 173 of the combiner is an estimate of the carrier offset and is coupled to a complex filter 175 and to an integrator 177. All channels provide an absolute phase error reference for the estimation of the carrier offset frequency that is removed by the unambiguous pilot signal. The integrator accumulates the history of the summed signal over a number of samples. After integration, the phase error estimate 179 is output, converted to a complex voltage 183 and output.

Referring back to fig. 11, the output 183 of the MIPLL157 is coupled to a complex mixer 185 upstream of the rake receiver. This completes the error feedback of the MIPLL 157. The MIPLL157 architecture can be efficiently implemented and executed in a Digital Signal Processor (DSP), although this embodiment requires additional resources and increases complexity.

Referring now to an alternative embodiment 195 shown in fig. 13, this embodiment 195 mixes the output of MIPLL157 at the symbol level. The output of the MIPLL157 is mixed 197 with the output of the rake receiver 101. As described above, the output of rake receiver 101 is at the symbol level. Disabling symbols in MIPLL157 structureChip conversion 181. Since the output 183 of MIPLL157 is mixed with the output of rake receiver 101, which is only weighted by AMF79, phase correction of the carrier offset must be added to the receiver portion that processes the traffic data. Therefore, a despreader 85 is required at each channel₁，85₂，85_nDownstream multiple mixers 199₁，199₂，199_nAnd a mixer 201 downstream of pilot despreader 87 mixes the phase corrected output 183 (at the symbol level) as feedback to the system.

The present invention maintains the transmitted pilot signal at a low level to provide an absolute phase reference while reducing interference and increasing air capacity. The net effect is to effectively eliminate pilot overhead.

While particular embodiments of the present invention have been shown and described, many modifications may be made by one skilled in the art without departing from the spirit and scope of the invention. The foregoing description is by way of illustration only, and is not intended to be in any way limiting to the particular forms disclosed.

Claims (14)

1. A base station for recovering carrier offset during reception of a communication signal over an air interface using a plurality of channels and a pilot signal, said base station comprising:

an adaptive matched filter for receiving the demodulated communication signal and generating a filtered signal using the weighted signal;

a rake receiver for receiving the demodulated communication signal and the generated pseudo-noise signal for a selected channel and generating a filter weight signal;

means for defining filter weighting signals using correction signals for generating weighting signals used by said adaptive matched filter;

a channel despreader coupled to said selected channel at said output of said adaptive matched filter for despreading said filtered signal with a pseudo-noise signal generated for said selected channel to produce a despread channel signal for said selected channel;

a pilot channel despreader coupled to a pilot channel at an output of said adaptive matched filter for despreading said filtered signal with a pseudo-noise signal generated for said pilot channel to produce a despread pilot signal for said pilot channel;

a hard decision processor, associated with a complex conjugate processor, for receiving the despread channel signal for the selected channel and generating the correction signal; and

a phase locked loop which uses at least the despread pilot signal to generate a phase correction signal for use in generating the phase corrected channel signal.

2. The base station of claim 1, further comprising: a plurality of channel despreaders, each coupled to the output of the adaptive matched filter for despreading the filtered signal, each channel despreader producing a plurality of despread channel signals using an associated pseudo-noise signal generator.

3. The base station of claim 2 wherein said number of channel despreaders is three.

4. The base station of claim 2, wherein said phase locked loop phase correction signal is on-chip and is applied to said demodulated communication signal.

5. The base station of claim 2 wherein each of the plurality of channels is a complex, bi-phase modulated signal, said signal being comprised of a plurality of data-representing symbols including in-phase and quadrature components, said hard decision processor comparing each despread channel signal symbol to one of four possible quadrature constellation points and assigning each of said symbols to a closest constellation point, and said complex conjugate processor de-scrambling each of said symbols by determining a complex conjugate for each of said assigned points to produce said correction signal.

6. The base station of claim 2, wherein said phase locked loop further comprises a plurality of inputs corresponding to said plurality of channel despreaders.

7. The base station of claim 6, wherein said phase locked loop further comprises:

a hard decision processor associated with said complex conjugate processor with a local feedback loop for each of said corresponding channel despreader inputs to produce an error estimate signal for each channel signal;

each of said error estimation signals and said despreader pilot signal coupled to an arctan processor produces a corresponding phase correction signal; and

the respective channel phase correction signal and pilot phase correction signal are coupled to a maximum likelihood combiner for producing a combined correction signal coupled to an integrator to produce the phase correction signal.

8. The base station of claim 7 wherein said number of channel despreaders is three.

9. The base station of claim 1 wherein said phase locked loop phase correction signal is at the symbol level and is applied to said filter weighting signal and to the despread channel signal of said channel and pilot despreader.

10. The base station of claim 9 further comprising a plurality of channel despreaders, each channel despreader coupled to an output of said adaptive matched filter for despreading said filtered signal with an associated pseudo-noise signal generator to produce a plurality of said despread channel signals.

11. The base station of claim 10 wherein said number of channel despreaders is three.

12. The base station of claim 10, wherein said phase locked loop further comprises a plurality of inputs corresponding to a plurality of said channel despreaders.

13. The base station of claim 12, wherein said phase locked loop further comprises:

a hard decision processor associated with a complex conjugate processor having a local feedback loop for each of said plurality of signal inputs, each producing an error estimate for a respective channel signal;

outputting a channel phase correction signal for each of said channel error estimates and said despreader pilot signal coupled to an arctan processor; and

the channel and the pilot phase correction signal are coupled to a maximum likelihood combiner for generating a combined correction signal coupled to an integrator to generate the phase correction signal.

14. The base station of claim 13 wherein said number of channel despreaders is three.