HK1057429B - Cdma system which uses pre-rotation before transmission - Google Patents

Description

CDMA system using pre-rotation method before transmission

Background

The present invention relates generally to digital communications, and more particularly to a system and method for pre-rotating a digital spread spectrum signal prior to transmission, thereby increasing receiver accuracy and improving recovery of receiver phase and frequency information.

Many existing communication systems use digital spread spectrum modulation or Code Division Multiple Access (CDMA) techniques. Digital spread spectrum modulation is a communication technique in which data to be transmitted is modulated with a pseudo random noise signal and transmitted through a widened frequency band (spread spectrum). Data transmitted using code division multiple access techniques may be unaffected by signal distortion or interfering frequencies in the transmit path.

Fig. 1 is a simplified diagram of a code division multiple access communication system including a single communication channel having a given bandwidth. The channels are mixed by a spread spectrum code that repeats a predetermined pattern generated by a pseudo random noise (pn) sequence generator. The data signal is modulated with a pseudorandom noise sequence to generate a digital spread spectrum signal. The carrier signal is modulated with a digital spread spectrum signal for establishing a forward link and then transmitted. The receiver demodulates the transmitted content to extract the digital spread spectrum signal. The same is true when establishing the reverse link.

In terrestrial communications, the transmitted signal tends to be distributed via reflections due to varying terrain, environmental conditions, and man-made obstructions. The result is that a single transmitted signal is generated as multiple received signals with different delays in reaching the receiver. This effect is commonly referred to as multipath distortion. When multipath distortion occurs, signals arriving from different paths arrive at the receiver with different respective amplitude and carrier phase delays.

Us patent 5,659,573 proposes a system in which errors associated with multipath distortion are typically corrected by the receiver after the signal is associated with a matching pseudo-random noise sequence and the transmitted data is reproduced. Thus, the signal already contains errors while the correlation is complete. Similar multipath distortion also affects the reverse link transmission process.

French patent 2767238 proposes a system for estimating a received signal in which a predetermined function is used to estimate the phase shift of the received signal. The phase shift is used in a phase locked loop to bring the system to converge to zero error.

European patent No. 0818892 and us patent No. 5,499,236 propose a system in which an error signal is sent by a base station in the downlink transmission indicating that adjustments are to be made by the terminal station for reverse line transmission.

Therefore, there is a need for a system that can correct encountered signal errors during transmission.

Disclosure of Invention

The present invention relates to a digital spread spectrum communication system which calculates errors in phase and frequency of a received signal transmitted by a communicating entity in wireless communication and pre-corrects for these errors in signal phase and frequency prior to transmission to the entity.

Drawings

Fig. 1 is a simplified block diagram of a prior art Code Division Multiple Access (CDMA) communication system.

Fig. 2 is a detailed block diagram of the B-cdmat communication system.

Fig. 3A is a detailed block diagram of the use of pseudo-random pilot signals with carrier offset error correction at the chip level in accordance with the present invention.

Fig. 3B is a block diagram of a rake receiver.

FIG. 4 shows a received symbol p on a QPSK group₀A diagram showing a hard decision process.

Fig. 5 is a schematic diagram illustrating an error correction angle corresponding to a specific symbol.

Fig. 6 is a schematic diagram of a synthesized symbol after applying an error correction angle corresponding to a designated symbol.

Fig. 7 is a block diagram of a conventional phase-locked loop.

Fig. 8A is a simplified block diagram of a transmitter in accordance with a preferred embodiment of the present invention.

Fig. 8B is a simplified block diagram of a transmitter in accordance with another embodiment of the present invention.

Fig. 8C is a simplified block diagram of a transmitter in accordance with another embodiment of the present invention.

Detailed Description

The preferred embodiments of the present invention will now be described in detail with reference to the drawings, wherein like reference numerals designate like elements throughout.

As shown in fig. 2, the cdma communication system 25 includes a transmitter 27 and a receiver 29, which may be located at a base station or a mobile user receiver. Transmitter 27 includes a signal processor 31 that encodes audio signals and non-audio signals into data at different rates, such as a data rate of eight kilobytes per second, sixteen kilobytes per second, thirty-two kilobytes per second, or sixty-matched kilobytes per second. The signal processor 31 selects a specific data rate according to the kind of signal or a set data rate.

In the background art, the generation of a transmission signal in a multiple access system includes the following two steps. In a first step, input data 33 is encoded by Forward Error Correction (FEC) encoding 35. Where the input data 33 may be considered a bi-phase modulated signal. For example, if a convolutional code of r-1/2 is used, the single bi-phase modulated data signal becomes a bivariate or two bi-phase modulated signal. One signal is designated in the in-phase (I) channel 41 a. The other signal is designated in quadrature (Q) channel 41 b. The complex numbers take the form of a + bj, where a and b are real numbers, and j is²Is-1. Bi-phase modulated I and Q signals are often referred to as Quadrature Phase Shift Keying (QPSK). In the preferred embodiment, the constraint length k 7 and convolutional code rate r 1/2 have a topological generator polynomial G₁＝171₈37 and G₂＝133₈39。

In a second step, the two bi-phase modulated data or symbols 41a, 41b are spread by a complex pseudo-random noise (pn) sequence. The combined I45a and Q45b spread spectrum signals are combined 53 with other spread spectrum signals (channels) having different spreading codes and mixed with a carrier signal 51 and then transmitted 55. This transmission contains multiple individual channels with different data rates.

The receiver 29 includes demodulators 57a, 57b that downconvert the transmitted wideband signal 55 to intermediate frequency signals 59a, 59 b. The second down-conversion reduces the signal to baseband. The quadrature phase shift keying signal is then filtered 61 and mixed 63a, 63b with a complex pseudo-random noise (pn) sequence 43a, 43 b. Where a complex pseudo-random noise (pn) sequence 43a, 43b is matched to the conjugate of the transmitted complex code. Only the original waveform spread with the same code at the transmitter 27 is effectively despread. Other waveforms appear as noise at the receiver 29. The data 65a, 65b are then sent to a signal processor 67 where the convolutionally encoded data are subjected to forward error correction decoding.

When the signal is received and demodulated, the baseband signal is at the chip level. Both the I and Q components of the signal are despread by the conjugate of the pseudorandom noise sequence used in spreading and the signal is returned to the symbol level. However, due to carrier offset, the phase error occurring in the transmission is reflected by the change in the single chip waveform. If carrier offset correction is performed at the chip level, the overall accuracy is improved due to the inherent convolution of the chip-level signal. Carrier offset correction can also be done at the symbol level, but the overall accuracy is not as high. However, since the symbol rate is much less than the chip rate, the overall processing speed is lower when error correction is performed at the symbol level.

Fig. 3A illustrates a receiver used in accordance with the system 75 and method of the present invention. The complex baseband digital spread spectrum signal 77 constituting the in-phase and quadrature phase elements is input and filtered via an Adaptive Matched Filter (AMF) or other adaptive filtering method. Adaptive matched filter 79 is a transversal filter (finite impulse response) that employs filter coefficients 81 to overlay delayed copies of received signal 77 with one another to produce filtered signal output 83. The signal output has an increased signal-to-noise ratio (SNR). Output 83 generated by adaptive matched filter 79 and a plurality of channel despreaders 85₁、85₂、85_nAnd a pilot despreader 87. The pilot signal, which is a component of the signal output 83, is despread by the separate despreader 87 and pseudorandom noise sequence 91, while the transmit data 77 is assigned to the despread pseudorandom noise sequence 91₁、92₂、93_nRespectively despread 85₁、85₂、85_nThe channel of (2). Data channel completion despreading 85₁、85₂、85_nThen, the data bit stream 95₁、95₂、95_nAnd a Viterbi decoder 97₁、97₂、97_nConnected and output 99₁、99₂、99_n。

The filter coefficients 81 or weights used to adjust the adaptive matched filter 79 are obtained by demodulation of a plurality of individual multipath propagation paths. This operation is performed by rake receiver 101. Methods for compensating for multipath distortion using rake receiver 101 are well known to those skilled in the art of communications.

As shown in fig. 3B, rake receiver 101 includes parallel combined path demodulator "fingers" 103₀、103₁、103₂、103_nThese elements demodulate specific multiplexing elements. The pilot sequence tracking loop for a particular demodulator begins with a timing estimate for a given path determined by the pn sequence 105. In the prior art, the pseudo-random noise sequence 105 may belong to any channel 93 of the communication system₁. Typically using the channel with the largest received signal.

Each path demodulator includes a complex mixer 107₀、107₁、107₂、107_nAdder and latch 109₀、109₁、109₂、109_n. For each rake element, pseudorandom noise sequence 105 is subject to a chip delay of τ 111₁、111₂、111_nAnd spread signals 113 and 107 via baseband₁、107₂、107_nMixing is performed so that each signal is despread. Each multiplication product is input to the accumulator 109₀、109₁、109₂、109_nAnd adds and latches the original product after the next symbol clock cycle is completed. Rake receiver 101 provides relative path values for each multipath element. A plurality of n-dimensional outputs 115₀、115₁、115₂、115_nAn estimate of the sampled impulse channel response is provided that includes a relative phase error of 0 °, 90 °, 180 °, or 270 °.

Referring to fig. 3A above, the multiple outputs generated by the rake receiver are combined with an n-dimensional complex mixer 117. Output 115 is a correction to the relative phase error in the anti-fading output, after mixing with each rake receiver 101.

The pilot signal is a complex quadrature phase shift keyed signal but the quadrature element is set to zero. The error corrected 119 signal of the present invention is formed from the despread channel 95₁By applying a frequency band to each of the despread signals 95₁Is first derived by making a hard decision 121. The hard decision processor 121 determines the position of the qpsk constellation closest to the despread symbol value.

As shown in fig. 4, the hard decision processor 121 is an omega-several meter distance processor to convert the received symbols p of channel 1 into a number of symbols p₀Comparing four quadrature phase shift keying group points x_1，1，x_-1，1，x_-1，-1，x_1，-1. Since the transmission 55 process has noise and distortion effects in both multipath and radio frequency scenarios, it is necessary to transmit each received symbol p₀And (6) checking. The hard decision processor 121 computes the slave received symbols p₀Four distances d to each quadrant₁、d₂、d₃、d₄And selecting the closest distance d₂Specifying the symbol location x_-1，1. Original symbolic coordinate p₀Is discarded.

Referring to fig. 3A above, after each hard symbol decision 121, the complex conjugate 123 of each symbol output 125 is determined. The complex conjugate is a pair of complex numbers that have the same real part and the imaginary part only differs in sign. As shown in fig. 5, by first determining the designated symbol coordinate x_-1，-1Demodulates or derotates the symbols and forms an error correction signal 119 to remove the relative phase error in the anti-fading output. In this way, the anti-fading output is effectively de-rotated by the hard decision correlation angle and the relative phase error is removed. This operation effectively provides fading immunity driven by the pilot signal without the need for an absolute phase reference.

Referring to fig. 3A above, the output 119 generated by complex conjugate 123 is coupled to a complex n-dimensional mixer 117, and the output 115 of each rake receiver 101 is mixed with an error correction signal 119. As shown in fig. 6, the composite product is a noise estimate of the impulse channel response. The error shown in fig. 6 indicates the radian distance pi/6 on the in-phase axis.

Referring to FIG. 3A above, the output 129 of the complex n-dimensional channel mixer 117 is connected to an n-dimensional estimator 131. The channel estimator 131 is a plurality of low pass filters each provided for filtering by a multipath component. The output 81 of the n-dimensional estimator 131 is connected to an adaptive matched filter 79. These outputs 81 are used as adaptive matched filter 79 weights. The adaptive matched filter 79 filters the baseband signal to compensate for channel distortion due to the lack of significant use of the pilot signal in the multipath scenario.

The rake receiver 101 is used in conjunction with a phase-locked loop (PLL)133 circuit to remove carrier offset. Carrier offset is due to transmitter/receiver component mismatch or other radio frequency distortion. The present invention 75 uses a low level pilot signal 135 that is generated by despreading 87 the pilot signal of the baseband signal 77 using a pilot pseudo-random noise sequence 91. As shown in fig. 7, the pilot signal is connected to a single input phase locked loop 133. The phase locked loop 133 measures the phase difference between the pilot signal 135 and the reference phase 0. The despread pilot signal 135 is the actual error signal connected to the phase locked loop 133.

The phase locked loop 133 includes an arctangent analyzer 136, a complex filter 137, an integrator 139, and a phase-to-complex converter 141. The pilot signal 135 is the error signal input to the phase locked loop 133 and is connected to a complex filter 137. The complex filter 137 comprises two gain stages 143a, 143b, an integrator 145 and an adder 147. The output of the complex filter 137 is connected to an integrator 139. The integral of the frequency is the phase, which is output 140 to the converter 141. The phase output 140 is connected to a converter 141 which converts the phase signal into a complex signal for mixing 151 with the baseband signal 77. Since upstream operation is interchangeable, the output 149 of the pll 133 is also a feedback loop into the system 75.

As shown in fig. 8A, the error correction signal 119 of the complex conjugate 123 and the output signal 149 of the phase locked loop 133 are each connected to a mixer located in the transmitter 181 for the purpose of correcting the signal before transmission. The transmitter 181 of fig. 8A operates in a similar manner to the transmitter 27 of fig. 2, except that the signal to be transmitted is pre-rotated prior to transmission. Referring to FIG. 8A above, data 164₁、164₂、164₃The encoding is performed by using forward error correction coding (FEC) 35. The two bi-phase modulated data or symbols 41a, 41b are spread with a complex pseudo-random noise (pn) sequence, the resulting I45a and Q45b spread signals are mixed with the error correction signal 119, up-converted with the carrier signal 51, and combined 53 with other spread signals having different spreading codes. The resulting signal 55 is subjected to re-error correction by a signal 149 sent from the receiver phase locked loop 133. The signal 56 with the completed phase and frequency pre-correction is then transmitted. Thus, the present invention uses the signals 119, 149 generated by the system 75 of fig. 3A to pre-correct the transmitted signal to reduce errors in the phase and frequency of the resulting signal at the receiver.

Referring previously to fig. 8B, a transmitter 183 constructed in accordance with another embodiment of the present invention is shown. This embodiment is similar to the embodiment of fig. 8A, with the only difference that the error correction signal 119 is mixed with the baseband data signal by mixer 157. Thus, the baseband data is pre-corrected prior to encoding and spreading. Of course, those skilled in the art will recognize that other steps may be included before the error correction signal 119 is mixed with the data signal.

Referring previously to fig. 8C, a transmitter 188 made in accordance with another embodiment of the present invention is shown. In this embodiment, the error correction signal 119 and the carrier offset signal 149 are input to a combiner which combines the signals with a single pre-correction signal prior to transmission and mixes with the output of the adder 53 via a mixer 169.

Finally, note that carrier offset error correction and pre-rotation error correction are mutually independent error correction methods. Each may be used alone. For example, the system may only pre-correct for carrier offset error without using pre-rotation. Alternatively, the system performs pre-rotation without correcting for carrier offset errors.

While particular embodiments of the present invention have been illustrated and described, it will be obvious to those skilled in the art that various changes and modifications can be made without departing from the spirit and scope of the invention. The foregoing description is for the purpose of illustration only and is not intended to limit the invention in any way.

Claims (4)

1. A method of reducing transmission errors in a code division multiple access communication system having at least two communication units, wherein a first communication unit receives a code division multiple access communication signal transmitted from a second communication unit, said method characterized by having said first communication unit;

analyzing a phase error of the received signal at the first communication unit;

analyzing a frequency error of the received signal at the first communication unit;

generating an error correction signal at the first communication unit based on the analysis; and is

The information signal, including the audio or data signal, is corrected using the error correction signal before it is transmitted from the first communication unit to the second communication unit.

2. The method of claim 1, further comprising: correcting said received signal with said error correction signal.

3. A code division multiple access communication system for reducing transmission errors in communications between at least two communication units, each communication unit comprising:

a receiver for receiving a code division multiple access communication signal transmitted from another communication unit, said receiver comprising:

an adaptive matched filter for receiving the demodulated cdma communication signal and generating a filtered signal using the weighted signal;

a rake receiver for receiving the demodulated code division multiple access communication signal and a pseudo-random noise signal generated corresponding to the selected channel and generating a filter weighted signal; at least one despreader coupled to an output of said adaptive matched filter for despreading the filtered signal using a pseudo-random noise generator for said selected channel to produce a despread signal;

a hard decision processor for analyzing said received signal error and generating an error correction signal, an

An error correction unit having a channel mixer for performing error correction on the received signal using the error correction signal;

a composite signal transmitter for pre-correcting the composite signal using the error correction signal prior to transmission to another communication unit.

4. A communication system according to claim 3, wherein the system comprises a plurality of communication units, the plurality of communication units communicating with each other over a code division multiple access air interface using a plurality of channels and a pilot signal for carrier offset recovery in reception; each communication unit comprising one said receiver and one said transmitter;

the receiver further comprises:

means for mixing the filtered weighted signal with the error correction signal to produce a weighted signal for use by the adaptive matched filter; and is

The transmitter includes:

a data input device for providing an information signal;

at least one despreader for despreading said information signal;

a mixer for mixing the spread spectrum signal with the error correction signal.