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EP1573891B1 - Convertisseur a resonance et procede pour alimenter des charges variables - Google Patents

Convertisseur a resonance et procede pour alimenter des charges variables Download PDF

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Publication number
EP1573891B1
EP1573891B1 EP03782433A EP03782433A EP1573891B1 EP 1573891 B1 EP1573891 B1 EP 1573891B1 EP 03782433 A EP03782433 A EP 03782433A EP 03782433 A EP03782433 A EP 03782433A EP 1573891 B1 EP1573891 B1 EP 1573891B1
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EP
European Patent Office
Prior art keywords
input
switch
voltage
load
current
Prior art date
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EP03782433A
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German (de)
English (en)
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EP1573891A2 (fr
Inventor
Matthias Radecker
Gunther LÖHMANN
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Fraunhofer Gesellschaft zur Foerderung der Angewandten Forschung eV
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Fraunhofer Gesellschaft zur Foerderung der Angewandten Forschung eV
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices
    • H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a bridge converter in the final stage
    • H05B41/2828—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a bridge converter in the final stage using control circuits for the switching elements
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/36—Controlling
    • H05B41/38—Controlling the intensity of light
    • H05B41/39—Controlling the intensity of light continuously
    • H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor

Definitions

  • the present invention relates to a switching power supply for driving variable ohmic-capacitive or ohmic-inductive loads, comprising a resonant circuit, an electromechanical energy converter, a switch and a control device.
  • Switching power supplies with or without resonant circuit usually do not come without inductive electromagnetic components. To achieve a low-loss switching operation such circuits can be operated only up to a certain maximum frequency and only with resonant inductive elements or broadband transformers or inductors. Such components are volume-intensive and cause a significant share of the costs of the entire device.
  • a self-or externally excited half-bridge circuit which works with bipolar transistors, reverse-sequence diodes, a series resonant circuit and inductive base feedback.
  • An exemplary embodiment of such a half-bridge circuit is disclosed in the following document (1): Lowbridge, M. Maytum, K. Rutgers, "Electronic Ballasts for Fluorescent Lamps Using BUL 770/791 Transistors” (Texas Instruments, 1992 ).
  • the load circuit is predominantly inductive, whereby a low-loss switching in different load cases is possible.
  • This circuit can also be classified as a class D amplifier.
  • ZVS Zero Voltage Switching
  • This circuit also requires a resonant inductance in the load circuit, but achieves zero-voltage behavior (ZVS) in parallel with a sufficiently large capacitance.
  • ZVS zero-voltage behavior
  • the parallel capacitance to the switch is chosen to be as small as possible in order to achieve the zero-voltage behavior (ZVS) without problems by means of a resonance inductance
  • this parallel capacitance is made as large as possible to be the maximum voltage across the switch to be as small as possible during turn-off.
  • the capacitance is chosen too large, the voltage can not return to zero, and impermissible switch-on losses occur.
  • any desired transformation ratios can be realized.
  • these components usually do not provide predominantly inductive input behavior.
  • Such electromechanical converters are usually also very narrow-band and can transmit only sinusoidal oscillations with regard to their frequency behavior.
  • a hard-switching converter topology is therefore less suitable for their operation.
  • resonant operation favorably also in a resonant converter topology, must be chosen.
  • a capacitive input and output behavior is essentially predetermined by a piezoceramic material
  • such a converter can only replace conventional inductances or transformers if care is taken in the event of a desired inductive load circuit behavior for additional inductive shaping of the load circuit.
  • a half-bridge circuit such an inductive load circuit behavior is required to keep the switching losses small.
  • the simplest measure is an additional, albeit small conventional inductance in the load circuit insert. If the turn-on losses are small enough due to correspondingly low input voltage levels (eg extra-low voltages up to 24 V), a capacitive behavior of the electromagnetic converter in the half-bridge may also be acceptable.
  • the switching in a resonant case using a piezoelectric transformer can be designed so that the switching losses are minimized when a Umladezeit the relatively large input capacitances of the piezoelectric transformer by accurately observing required driving times by temporarily switching off both switches (dead times) bridged becomes.
  • a precisely adjustable high-side and low-side driver circuit is required, which further usually has an integrated circuit.
  • An embodiment of such a circuit is published in the following document (3): RL Lin, Lee FC, EM Baker, DY Chen, "Inductor-less Piezoelectric Transformer Electronic Ballast for Linear Fluorescent Lamps ", APEC2001, Anaheim, CA, USA, Proceedings, Vol. 2, pp. 664-669 ,
  • a class E resonant circuit the predominantly capacitive input behavior of a piezo transformer is beneficial in that the magnitude of the input capacitance can be adjusted to an electrically required value and thus not disturbing, as is the case with a half-bridge or other inductively-acting load circuit.
  • Such class E circuits with a piezoelectric transformer are already known from the following document (4): T.Abe, Sh. Jomura, T. Tamakai, "Discharge tube driving device and piezoelectric transformer," EP 0 665 600 B1 , European Patent of 21.07.1999.
  • such a circuit usually requires an additional input parallel capacitance if the input capacitance of the piezoelectric transformer is not sufficiently large. This is not the case in a transformer-off case where the input capacitance of some embodiments of piezoelectric transformers may be too large.
  • an input-side smoothing choke prevents a direct action of high-frequency current oscillations on an input or on a smoothing capacitor with respect to a non-input side smoothing or Resonanzinduktterrorism, so that an input-side smoothing reactor (hereinafter called inductor inductance) is preferable to other arrangements of inductance.
  • phase lock loop is a typical way of frequency tuning.
  • PLL phase lock loop
  • US 5,866,968 a possibility is described to adjust the phase shift between the output voltage and the drive signal of a circuit according to (4) so that a PLL circuit with a simple oscillator / driver IC can be realized.
  • This class E control circuit is well suited for piezotransformers with high transformation properties, since the maximum voltage at the output of the transformer is a significant point at the same time for the rated power. Because of the low current load in the high transformation, the frequency characteristic of the output voltage will almost correspond to an idle case, so that the transformation ratio between idle and nominal load changes little.
  • a phase control by sampling the zero crossings of output voltage and output current in a half-bridge circuit is again inaccurate because of the dispersion of Umlade réelle at the input of the piezoelectric transformer, so that there is an evaluation of the amplitude of the output current is required to adjust the rated power.
  • an inversion circuit for igniting a cold cathode arc tube by using a piezoelectric transformer includes a piezoelectric transformer for supplying a voltage signal to the arc tube, a driver for driving the piezoelectric transformer, and a voltage controlled oscillator for generating an oscillation pulse voltage signal having an oscillation frequency controlled by a control voltage.
  • a control circuit For generating the control voltage, a control circuit is used, which generates the control voltage based on a detected phase difference between a voltage at the output of the driver and a voltage at a connected to the output of the piezoelectric transformer voltage divider.
  • the object of the present invention is to provide a resonant converter and a method for efficiently driving variable loads.
  • the present invention is based on the finding that a piezotransformer can be used for driving variable loads in a rated load operation for the Abtransformationsfall by a switch for switching a voltage applied to the piezotransformer voltage signal is used whose switching frequency based on a phase shift between a switch current and a load current is controlled.
  • a negative switch current profile can and should occur by eg an antiparallel diode to the switch in all operating cases, whereby the zero voltage switching (ZVS) can always be guaranteed.
  • a voltage transformation ratio of the piezoelectric transformer is just chosen to match the load impedance, and an input capacitance of the piezotransformer is selected so that it can resonantly store the required reactive power component, so that neither the Switch voltage is exceeded, nor does the voltage return to zero.
  • the external capacitance shown there parallel to the switch is unnecessary, since the input capacitance of the piezoelectric transformer for a mains voltage application can be chosen sufficiently large, while their value is achieved in low-voltage applications of a piezoelectric transformer less well and below Circumstances is too small.
  • the circuit according to the invention requires comparatively to half-bridge circuits for mains voltage applications only a low-side driver and thus has a reasonable tax expense. This simplifies the control effort for the entire circuit and is comparable to a driving effort of a hard-switching DC-DC converter (flyback or boost arrangement).
  • the switch comes only for a short time, and comparable to the effect of a power source, in a reverse operation and therefore operates very low loss especially at a use of MOS transistors, but also when using IGBT with reverse diode even at high frequencies up to 100 kHz ,
  • the present invention makes it possible, within certain limits, to drive variable loads with low loss and with a simple driving effort at high frequencies, with only a minimal amount of circuitry including, for example, a switch (MOSFET or IGBT with a reverse diode), an input direct current choke (Drosselihdukttechnik) and an electromechanical Energy converter (piezoelectric transformer) is obtained.
  • a rectified mains voltage with certain fluctuations in the input voltage is just as useful as a constant input DC voltage.
  • the converter (resonant converter) generates a nearly sinusoidal output voltage due to a high quality of the electromechanical transformer, whereby the crest factor at downstream ohmic loads, such as gas discharge lamps, can be kept sufficiently small.
  • the electromechanical transducer piezoelectric transformer
  • the electromechanical transducer satisfies the requirement of up-transformation in an unloaded state, so that a low-pressure gas discharge lamp can be easily ignited.
  • a lamp Prior to ignition, such a lamp represents a very large resistance, which in an ignited state (combustion mode) changes into a defined load with a negative differential resistance, and can be approximately approximated with an ohmic resistance at an operating point.
  • the ignition circuit realized by further components in a conventional ballast can be realized in a ballast with a piezoelectric transformer exclusively by this transformer, whereby a further cost reduction is brought about.
  • a load-dependent phase shift between a load current and a switch current such a rated load point is set that it can be regulated via a phase locked loop (PLL).
  • PLL phase locked loop
  • a simple integrated drive circuit can be used.
  • a detection of an input or a lamp voltage is not required for an adjustment of an operating point, since a parameter dependence of a phase shift is small enough to adjust the lamp power alone via a setpoint adjustment of the phase shift.
  • the amplitude of the output current need not be sampled for approximate power adjustment because of the change in the transformation ratio If the load changes, the nominal power can be reproduced exactly enough on the phase shift of the current zero crossings of the switch and load current.
  • Fig. 1 is a rough representation of a resonance converter according to the invention is shown, which comprises a source 101, a switch 103, a piezo transformer 105, a variable load 107 and a control device 109.
  • a voltage supplied by the source 101 or a current supplied thereby is switched by means of the switch 103 at a switching frequency, whereby an input signal is applied to the piezo transformer 105, which is converted into an output signal having a frequency which is the switching frequency of the switch 103 depends.
  • This output signal serves to drive a load 107, for example a low-pressure gas discharge lamp, whose load characteristic is variable.
  • the switching frequency at which the switch 103 is switched is controlled by the controller 109 based on a phase shift between the current through the switch 103 and the load current through the load 107.
  • This phase shift can be determined from a plurality of signals which can be tapped, for example, before and after the piezo transformer 105 and before or after the switch 103.
  • Fig. 2 shows an embodiment of a resonance converter, wherein a control device for controlling the switching frequency is not shown.
  • the source 101 is coupled to a first terminal 2011 of an input choke 201.
  • a second terminal 2013 of the input choke 201 is coupled to a first input 1031 of the switch 103.
  • the first input 1031 of the switch 103 is coupled to a first terminal 1051 of an input port 1052 of the piezotransformer.
  • the source 101 is further coupled to a second input 1033 of the switch 103, which is further coupled to a second terminal 1053 of the input port 1052 of the piezo transformer 105.
  • the variable load 107 is connected between a first terminal 1055 of an output gate 1056 of the piezotransformer and a second terminal 1057 of the output gate 1056.
  • the switch 103 further has a control input 1035, to which a control signal can be applied, which controls the switching frequency of the switch 103.
  • the source 101 which may be a DC voltage source
  • an approximately constant or sawtooth DC current is fed via the input choke 201.
  • the switch 103 is operated with a relative on time D and an operating frequency f, so that a resonance of the converter 105 is reached, and an output signal, such as a voltage, the variable load 107, such as a gas discharge lamp or other ohmic-capacitive load, drives.
  • Fig. 3 is a detailed circuit diagram of a resonant converter including a class E amplifier.
  • the source 101 is first coupled to the first terminal 2011 of the input choke 201.
  • the second terminal of the inductor inductor is coupled to the first input 1031 of the switch 103, the first input 1031 being further coupled to the first terminal 1051 of the converter 105.
  • the source 101 is also coupled to the second input 1033 of the switch 103, the second input 1033 being further coupled to the second terminal 1053 of the converter 105.
  • the load 107 is arranged between the first terminal 1055 and the second terminal 1057 of the output port of the converter 105.
  • the switch 103 in this embodiment comprises a voltage-controlled power switch 1037 whose source or emitter is coupled to the first input 1031 of the switch and whose drain or collector is coupled to the second input 1033 of the switch 103 are.
  • the control input 1035 of the switch 103 is simultaneously implemented as a gate of the voltage-controlled circuit breaker 1037 in this embodiment.
  • a diode 1039 is connected in the flow direction.
  • FIG. 1 a simplified equivalent circuit diagram of a piezotransformer 105 is shown in FIG.
  • the equivalent circuit comprises an input capacitance 10501, which is connected between the first terminal 1051 and the second terminal 1053 of the input port of the piezo transformer 105 and is thus arranged parallel to the switch 103.
  • the equivalent circuit of the converter 105 comprises a resonant circuit consisting of a series circuit of a capacitance 10502, an inductor 10503 and a resistor 10504.
  • the resonant circuit which is further characterized by a high quality, consists of the capacitance 10502, the inductor 10503 and the resistor 10504, is connected between the first terminal 1051 of the converter 105 and another terminal 10506 a primary side of the transformer assembly 10505. Parallel to a secondary side of the transmission arrangement 10505, an output capacitance 10508 is arranged.
  • the piezo transformer 105 is characterized in that the transmission ratio ü is subject to change depending on the load 107.
  • the voltage controlled power switch 1037 may be, for example, a fast IGBT (eg, a field stop IGBT) or a MOS transistor (For example, a cool MOS transistor), which is used together with an anti-parallel reverse diode. In the following the operation of the circuit shown in Fig. 3 will be explained.
  • the voltage-controlled power switch 1037 When the voltage-controlled power switch 1037 is made conductive by application of a control signal to the control input 1035, a current flowing through the voltage-controlled power switch can not rise abruptly due to the input reactor 201. In addition, the input capacitance 10501 of the converter 105 is discharged. If the voltage-controlled power switch 1035 is switched off again by applying a corresponding control signal, that is to say in a blocking state, then a voltage across the voltage-controlled power switch only grows slowly, since the input capacitance 10501 charges up.
  • the voltage-controlled power switch at the gate 1035 can therefore be de-energized with respect to the collector / emitter or drain / source, so that no turn-on losses occur.
  • Such a current-controlled antiparallel diode is not necessarily designed as a fast diode, so that in this case also a low-cost slow diode can be used.
  • the switch 103 is now operated at a predetermined frequency, the resonant circuit consisting of the capacitance 10502, the inductor 10503 and the resistor 10504 is excited. If a resonant frequency of the resonant circuit is reached, the converter 105 reaches a maximum voltage transfer ratio u.
  • a piezoelectric transformer can be, for example, a voltage transfer function (at a defined input voltage 101 and a defined load 107) with respect to the frequency as described by a Gaussian function (bell curve), as illustrated for example in Fig. 3a.
  • the voltage transfer function reaches a maximum value in the load state.
  • the voltage transfer function decreases such that it follows a course of the Gaussian curve. For example, at a frequency f 1 above the resonant frequency, the voltage transfer function has assumed a value that is significantly less than the value of the voltage transfer function in the resonant case. If the frequency decreases again in the over-resonant mode, the voltage transfer ratio ü increases again.
  • the gas discharge lamp exemplarily connected to the secondary side of the transformation arrangement 10505 is characterized by a variable load characteristic.
  • a high voltage is applied, which enables the ignition of the gas discharge lamp 301. If the gas discharge lamp 301 is ignited in the no-load state, then the voltage applied to the gas discharge lamp 301 drops while the load current flowing through the gas discharge lamp increases.
  • Is a nominal load operation (load condition) achieved that is, the gas discharge lamp is converted into a combustion operation, so its load behavior, as already stated, be approximated by a variable resistance in each operating point, wherein a negative differential characteristic of the ohmic resistance occurs.
  • the voltage transfer function chosen by a suitable design of the electro-mechanical transducer 105 so broad that a deviation from the resonant frequency, a suitable reduction of the voltage transfer ratio occurs, it can be counteracted an increase in a voltage at the gas discharge lamp during load operation.
  • the piezoelectric transformer because of its capacitive output due to the capacitance 10508, acts as a class E converter with a predominantly capacitive output load.
  • the transmitted total power does not decrease to such a degree as if a constant ohmic resistance were operated as a load with the same frequency change.
  • the transmitted total power is divided into the reactive power conducted via the capacitor 10508 and the active power conducted via the load 107.
  • the transmitted total power can drop less with a deviation from the resonant frequency less than a constant ohmic load with the same converter, as a larger capacitive reactive power due to larger output voltage on the capacity 10508 is performed.
  • the piezoelectric transformer 105 is designed so that it generates at about a same or only a slightly different resonant frequency with respect to the load case (ie, burning operation of the gas discharge lamp) a load-less up-transformation of the output voltage, so that an ignition of the gas discharge lamp is made possible.
  • This attribute is achievable in piezoelectric transformers due to an undamped mechanical vibration in a load-free state in a simple and cost-effective manner by narrowing the resonance curve shown in Fig. 3a in the unloaded state, and the broadband resonant curve in the load state encloses the narrowband load-free curve.
  • phase angle ⁇ LT which is determined by the zero crossings of the switch current I S and the load current I L is not zero and relatively large in this exemplary diagram, since the load current I L has a larger capacitive component, which is equivalent to the fact that Gas discharge lamp has not yet been converted to its nominal operation (approximately ohmic resistance), where the phase angle ⁇ LT is smaller and may even become almost zero.
  • the reverse time t rev becomes ever smaller and can become almost zero, so that the negatively flowing reverse current through the diode 1039 disappears.
  • the frequency-dependent voltage transmission ratio of a piezoelectric transformer is used in the embodiment shown in FIG. 3 according to the invention to realize a frequency-dependent power transmission in response to a variable load, as has already been explained with reference to FIG. 3a. This will be explained in detail below with reference to the voltage transfer ratio of a piezoelectric transformer 105 shown in FIG. 5 depending on a load characteristic of a gas discharge lamp 301.
  • the resonant frequency in an unloaded operation is higher than the optimum one Frequency under load (for example, for maximum power or for maximum efficiency).
  • the resonant frequency of the electromechanical transducer is realized without load only slightly above the resonant frequency under load, which is technically easily possible by a suitable design of a piezoelectric transformer.
  • the rated frequency for the nominal load combustion mode is intended to coincide approximately with the resonant frequency in a no-load condition.
  • the converter is initially driven starting from the resonant frequency with a frequency which is preferably variable around the no-load resonance point, which increases slowly and / or slowly decreases again, and follows a curve of a voltage transfer ratio curve 501 shown in FIG ,
  • a phase angle ⁇ LT between the load current and the switch current is evaluated for controlling and regulating the converter constructed in this way in order, for example, to realize a superresonant control.
  • FIG. 5 Also illustrated in FIG. 5 is an example waveform of the phase angle ⁇ LT at rated load versus frequency (curve 507) together with the voltage transfer functions in a no-load condition (ignition) and a load condition (rated load).
  • the phase angle ⁇ LT steadily decreases until a maximum power transmission is reached, while it increases in the direction of load-free operation.
  • the load changes so that below f OPT the rated load or an even larger load (small voltage transfer ratio ü) occurs, and above f OPT a smaller load (greater voltage transfer ratio ü) up to the no-load ignition characteristic in association with the function of the phase angle ⁇ LT 507 occurs.
  • the supersorant range above a frequency f opt can be used for controlling or regulating the gas discharge lamp power. Accordingly, it is therefore not necessary to detect a maximum value of the gas discharge lamp current to perform the control or the regulation of the converter. It is sufficient to sample the phase angle ⁇ LT between the switch and the load current and set it to a nominal value. If the frequency becomes smaller, the active power transfer increases in an over-resonant operation up to its maximum at the resonance frequency.
  • the switch current embodies approximately the input current of the piezo transformer 105, which is distributed via the transformation ratio to the load (gas discharge lamp) and to the output capacitance 10508 of the converter 105.
  • FIG. 5a shows a dependence of the output power transmission on the input voltage at a constant output impedance.
  • the power can be increased under rated load by increasing the input voltage from a minimum rated input voltage 505 'to a higher input voltage 503' to a maximum load characteristic 501 '.
  • the output power can not be increased significantly, this being dependent on the volume of the piezoelectric transformer used. A smaller volume allows only a smaller maximum load. It is therefore important to ensure that the piezotransformer is designed at least for a load slightly greater than the rated load, so that the control circuit of FIG. 8 remains functional beyond the rated load.
  • phase angle ⁇ LT The course of the phase angle ⁇ LT at a constant frequency is shown once again in FIG. 6 as a function of an input voltage U in applied to the gas discharge lamp.
  • U in the phase angle ⁇ LT decreases, since in this case more active power is transmitted to the gas discharge lamp, see, for example, Fig. 5a, überresonanter operation.
  • This has the consequence that the active component of the lamp current increases. From this example it is clear that variations in the input voltage U in reflected also in the size of the phase angle ⁇ LT.
  • such fluctuations of the voltage U in can be compensated by more power is passed to the gas discharge lamp by lowering the frequency at a falling input voltage U in in the over-resonant mode of operation.
  • phase angle ⁇ LT can be set by setting a different transmission ratio u as a function of the input voltage. This phase angle is at constant output voltage due to the parallel circuit of the approximately constant capacitance 10508 and the load 107 in load operation, a measure of the size of the load current, and thus for the output power.
  • FIG. 7 shows an exemplary embodiment of a resonance converter according to the invention for low-pressure gas discharge lamps, including switching frequency control. Since this embodiment is based on the embodiment shown in Fig. 3, the functionalities will not be described again with the same reference numerals in the following.
  • the embodiment shown in FIG. 7 initially comprises an input rectifier 701 having a first mains terminal 70101 and a second mains terminal 70103. Between an output 7015 and an input 7017 of the input rectifier 701 is a capacitance 703, e.g. a charge capacitor can be coupled. In addition to the capacitor 703, a drive part 705 is further coupled together with a resistor 70501. The output 7015 of the input rectifier 701 is further coupled to the first terminal 2011 of the input inductor 201. The drive part 705 further has a control output 7051, which according to the present invention is coupled to the control input 1035 of the switch 103, which in this embodiment comprises the current-controlled power switch 1037.
  • a capacitance 703 e.g. a charge capacitor can be coupled.
  • a drive part 705 is further coupled together with a resistor 70501.
  • the output 7015 of the input rectifier 701 is further coupled to the first terminal 2011 of the input inductor 201.
  • the drive part 705 further has
  • the drive part 705 further has a first one Input 7053 and a second input 7055 on.
  • the first input 7053 is coupled to the second input 1033 of the switch.
  • a sense resistor 707 is further arranged between the first input 7053 of the drive part 705 and the input 7017 of the input rectifier 701.
  • a second sense resistor 709 is arranged between the load 107 and the second terminal 1057 of the converter 105.
  • the second input 7055 of the driver 705 is coupled between the load 107 and the second sense resistor 709.
  • the drive part 705 further has a power supply input 7057, which is coupled to the input 7017 of the input rectifier 701 via a capacitor 70111, which may be designed, for example, as a blocking capacitor. Between the second terminal 1053 of the converter 105 and the power supply input 1057 of the drive part 705, a first diode 70131 is coupled in the flow direction. Between the input 7017 of the input rectifier 701 and the first terminal 1051 of the input port of the converter 105, there is further coupled a parallel circuit consisting of an external capacitance 70151 and a diode 70171 operated in the forward direction.
  • a parallel circuit consisting of an external capacitance 70151 and a diode 70171 operated in the forward direction.
  • the task of the control part 705 is to detect the indicated in Fig. 7 with an arrow switch current I S and the load current I L suitable to determine a phase difference between the two currents, and so at the control output 7051, a control signal for controlling the Switching frequency of the switch 103 output.
  • a variable dependent on the switch current I S is initially generated, which can be applied to the first input 7053 of the control part 705.
  • the switch current I s at the first sense resistor 707 is converted to a voltage applied to the first input 7053.
  • the size dependent on the switch current can be generated by any functionality, such as a current mirror or a current-controlled voltage source.
  • the piezotransformer 105 with a voltage transfer ratio ü drives a gas discharge lamp with the load resistor 107, through which the load current I L flows.
  • a second sense resistor 709 is used in the exemplary embodiment shown in FIG. 7, so that the load current I L generates a voltage across the resistor 709 which is applied to the second input 7055 of the drive part 705 is applied.
  • the phase difference between the switch current I S and the load current I L is first determined in the drive part 705 and, as has already been described above, a control signal is output which controls the switching frequency of the switch 103.
  • the power supply of the drive part 701 is realized via a primary-side connection of the piezoelectric transformer 105 via a pumping circuit with the diodes 70131 and 70171, and via the external capacitor 70151, while the capacitor 70111 (blocking capacitor) Supply voltage of the control part 701 smoothes.
  • a simple power supply device contains no special requirements for electromagnetic compatibility and without further options for dimming or power factor correction, only three capacitances 703, 70111 and 70151, which are designed for example as capacitors, an input rectifier 701 (mains rectifier), an input choke 201, a piezo transformer 105, for example a fast IGBT 1037 with a reverse diode 1039, a possibly integrated control section 705, two diodes 70131 and 70171 as well as some miniature resistors.
  • the ballast thus obtained can thus be accommodated in a compact design in the smallest space, for example, a height of 10 mm is easily accessible.
  • a size EF 13 up to a power of 18 watts is sufficient for the input choke 201 (choke inductance).
  • a cylindrical design with a height of 9mm and a diameter of 20mm may also be considered sufficient for 18W.
  • the transistor 1037 for example, implemented as a fieldstop IGBT, can be housed in a small SOT package, and the drive IC (IC) for the driver 705 can be packaged in a standard 8-pin package.
  • a complete integration of the 1039 reverse diode, such as a Fieldstop IGBT 1037 and a drive IC is also cost-effective in an 8-pin package as a multi-chip solution.
  • FIG. 8 shows an exemplary embodiment of the control device 109 according to the invention together with the switch 103 and the load resistor 107.
  • the control device 109 comprises first a device 801 for detecting a dependent of the switch current I S size, means 803 for detecting a dependent of the load current I L size and a phase locked loop 805.
  • the phase locked loop 805 comprises in this embodiment means 807 for determining the Phase shift between switch current and load current from the quantities detected by device 801 and device 803.
  • the device 807 has a first input 8071, a second input 8073 and an output 8075.
  • the output 8075 of the device 807 is via a Resistor 8091 and a capacitor 8093 coupled to a reference potential, such as ground.
  • Means 805 further includes a voltage controlled oscillator 811 (VCO) and a gate driver 813.
  • VCO voltage controlled oscillator
  • An input 81101 of the VCO 811 is coupled between the resistor 8091 and the capacitor 8093.
  • An output 81103 of the VCO is coupled to an input of the gate driver 813 whose output is coupled to the control input 1035 of the switch 103.
  • the device 801 has a comparator 8011 with a first input 80111, a second input 80112 and an output 80113.
  • the first input 80111 of the comparator 8011 is coupled to the second input 1033 of the switch 103.
  • the second input 80112 is coupled to the output 80131 of a reference source 8013.
  • the output 80113 of the comparator 8011 is coupled to the first input 8071 of the device 107.
  • the device 803 comprises a comparator 8031 having a first input 80311 and a second input 80312 and an output 80313.
  • the first input 80311 of the comparator 80131 is coupled between the resistors 107 and 709.
  • the second input 80312 of the comparator 8031 is coupled to the output 8031 of the reference source 8013.
  • the output 80313 of the comparator 8031 is further coupled to the second input 8073 of the device 807.
  • the switch current at the sense resistor 707 is converted into a voltage applied to the first input 80111 of the comparator 8011.
  • a reference signal supplied from the reference source 8013 is applied.
  • the comparator 8011 thus samples the zero crossings of the switch current I S by a comparison between the one falling on the sense resistor 707 Voltage and the reference signal near zero.
  • an output signal is thus output whose instantaneous phase results from the comparison between the signals present at the inputs 80111 and 80112 and which in this embodiment represents a variable dependent on the switch current I S.
  • a symmetrical arrangement is located on the load side.
  • the load current I L is converted into a voltage applied to the first input 80311.
  • the second input 80312 is also the reference signal 80131, which is supplied by the reference source 8013.
  • the load current is sampled via the sense resistor 709 and the comparator 8031 outputs at its output 80313 an output signal representing a magnitude dependent on the switch current I L.
  • the two second inputs 80112 and 80312 are coupled to the same output 80131 of the reference source 8013.
  • This reference source is designed in this embodiment as a DC voltage source.
  • reference source 8013 may be any source, such as an AC source, or other arrangement such as a current or voltage controlled voltage source that provides a predetermined, eg, time-dependent reference signal.
  • the switch current and the load current at the two sense resistors 707 and 709 are sampled.
  • the switch current and the load current can be sampled using any functionality, such.
  • Means 807 is for detecting the phase shift between the switch current and the load current from the detected quantities applied to the two inputs 8071 and 8073.
  • means 807 is implemented as a phase detector which is part of phase locked loop 805.
  • the phase difference signal determined by the phase detector 807 which further depends on its frequency dependency on whether there is an over-resonant or an under-resonant operation, is integrated by the integrator 809, in this embodiment as a filter consisting of a resistor and a capacitor.
  • the filter output signal is applied to the input 81101 of the VCO 811, which generates a suitable frequency f and an associated duty cycle D f from the filter output signal. This output signal is forwarded to the control input 8035 of the switch 103.
  • the integrator device 809 which in this example is particularly cost-effective, can also be implemented in other ways, for example by a suitably connected operational amplifier, or another time-delaying circuit.
  • the switch 103 includes a voltage controlled power switch 1037.
  • the output of the VCO 813 is supplied to the gate driver 813, the output of which is passed to a gate of, for example, a field stop IGBT or MOSFET as possible embodiments of the voltage controlled power switch.
  • the device 807 is implemented as a phase detector for detecting the phase difference and generating a difference signal. This has the advantage that for controlling the switch 103, the phase locked loop 805 can be used, which can be realized inexpensively.
  • Fig. 8 works as follows: When the output load increases (smaller ohmic resistance), Thus, a smaller transmission ratio will occur according to FIG. 5. At the same time, however, the phase angle ⁇ LT decreases in the over-resonant case, so that the voltage Up which is output at the output of the device 809 increases. Via the VCO 811, a larger frequency is set by the filter 809 at the gate of the fieldstop IGBT 1037, for example, which causes a reduction of the transmitted power. This leads to the over-resonant characteristic of FIG. 5 to a smaller transmission ratio ü. However, since the output load has a negative differential resistance when it is a gas discharge lamp (fluorescent lamp), the output voltage increases and the lamp current decreases disproportionately.
  • a gas discharge lamp fluorescent lamp
  • the VCO 811 may therefore have a means, not shown, to store the angelege frequency at the time of ignition.
  • the ignition frequency in überresonanten branch of the load curve in Fig. 3a This ignition frequency is then not targeted during control under load, or only by an amount defined by the parameters of the piezo transformer 105 falls below, so that even a disproportionate change in the phase voltage U p to smaller values no frequency reduction in the sub-resonant operation permits.
  • the VCO 811 can continue to have a device, not shown, to ensure a minimum, lower limit frequency in load operation, which ensures this behavior.
  • the VCO 811 is further characterized by a duty cycle D f , which is adjustable.
  • D f a duty cycle
  • these switch-on times are supplied by the voltage-controlled oscillator 811 in such a way that the current in the switch only rises during the switch-on time, as illustrated in FIG. 4 by the profile of I S in an interval marked by t on .
  • the VCO 811 is therefore designed so that it provides a duty cycle D f necessary for this purpose. This can be realized, for example, by means not shown in FIG. 8 for setting a predetermined duty cycle of the output signal of the oscillator 811.
  • a resonant inverter consisting of a self-excited or a foreign-excited class E amplifier with a mode tuned to resonance frequency at a high frequency using a high load circuit electromechanical power converter, high efficiency, and limited load variation and input voltage swing, by using a dynamically fast switch with at least about three times reverse bias voltage versus the maximum DC input voltage. Since only one switch and a relatively simple drive circuit are required, the circuit can be realized as a one-chip solution (eg in a SMART-POWER technology) or in a known cost-effective multi-chip design without a need a bridge-capable high-voltage technology for the drive circuit.
  • a resonance inductance is thus no longer required, just as little as a high-side driver device, which is not the case for comparable half-bridge solutions with narrow-band energy converters, or only with restrictions with regard to the control accuracy.
  • no reactive components capacitor, inductors
  • the inverse voltage transfer ratio 1 / ⁇ (input voltage / output voltage) of the electromechanical transducer is selected to be 1.5: 1 to 5: 1 in terms of sinusoidal transmission at resonant frequency to accommodate typical grid applications for discharge lamps (eg, low pressure lamps).
  • the input mains voltage can be between 80 and, for example, 260 volts AC.
  • From the electrical filter behavior of the electronic-mechanical transducer eg piezoelectric transformer
  • a load voltage burning voltage
  • a load adjustment with optimal switch voltage limit for the described network applications can not be achieved, which is why the correct dimensioned gear ratio in rated load operation is an essential idea of the inventive solution.
  • the input capacitance of the electromechanical transducer is to be chosen so that in addition to the parallel to the converter input switched semiconductor switch no further parallel capacitance is needed.
  • the value of this input capacitance will be between 100 pF and 1 nF at a frequency of typically 100 kHz and a power of 10 to 20 watts, depending on the input voltage.
  • the value of the capacitance should be about 500 pF to 1 nF, for a large input voltage (160 to 260 V AC), this value should be selected as 100 pF to 500 pF.
  • the parallel acting capacity of the switch is on the order of less than 200 pF.
  • the value of the input capacitance shifts up (higher power) or down (smaller power).
  • a construction of a piezoelectric transformer Preferably, a circular or a laterally oscillating piezoelectric transformer is used here.
  • a piezoelectric transformer which works on the basis of a thickness vibration or a rose type transformer is less suitable for this application because they do not allow a corresponding transformation ratio in the given power range and the required input capacitance at a sufficient efficiency. It should be noted, however, that these two types of piezotransformers can be used according to the invention.
  • the negative differential resistance of a gas discharge lamp in the burning operation helps to stabilize the class E zero-voltage circuit, and as a load in conjunction with a narrow-band electromechanical converter, this is better suited than a constant ohmic resistance.
  • the piezoelectric transformer is designed so that its voltage transfer function has a sufficient bandwidth, which, as already mentioned, follows in frequency about a Gaussian function, and is chosen so wide that at a deviation from the resonance frequency Reduction of the voltage transfer ratio ü occurs, which counteracts the increase in the voltage at the gas discharge lamp.
  • a control or regulation for gas discharge lamps via detection of the lamp current can be technically reliably implemented if the frequency bandwidth is up to the drop to half a power at least about 5 to 10% of the nominal frequency.
  • the behavior changes
  • the class-E circuit with respect to the zero voltage circuit and the switch current load hardly, so that no significant changes in the switch maximum current, the switch reverse current and the switch maximum voltage occur at approximately constant relative on-time. This is due to the fact that the output capacitance of the piezoelectric transformer is large enough (eg a few nanofarads) to take enough reactive power even in the no-load condition, and resonantly fed back to the input when the load becomes smaller than its nominal value.
  • the class E converter responds to an increased capacitive or less resistive output load with an increase in reactive current component without violating the zero voltage condition.
  • the inherent output capacitance of the piezoelectric transformer has a stabilizing effect in this sense.
  • the switch voltage continues to return to zero even if the gas discharge lamp has been extinguished or removed again.
  • This only increases the proportion of reverse current in the switch.
  • the maximum reverse current is equal to the maximum inrush current of the switch (ignition mode).
  • detection of end-of-life effects or load circuit interruption can be done by sampling the reverse current in the switch without having to monitor the lamp voltage.
  • the transformation ratio of the electromechanical transducer changes little, so that the power changes approximately with the square of the input voltage. If the DC input voltage of the converter decreases, the active current and the reactive current in the load circuit decrease accordingly, and the switch reverse current decreases. If the reverse bias of the switch is large, the input capacitance of the piezoelectric transformer can be reduced to achieve zero voltage switching (ZVS) to lower input voltages. If, however, the reverse voltage reserve of the switch is small, the input voltage must not fall below a certain minimum value.
  • a switch a component whose maximum permitted voltage in any possible operating state of a gas discharge lamp with electromechanical transducer (piezoelectric transformer) is exceeded. Therefore, a non-avalanche-proof switch (MOSFET or IGBT) is well suited for this application, since the output capacitance of the converter, which acts on the input, has a compensating effect with decreasing resistive load, and a maximum transmittable power can not be exceeded.
  • MOSFET or IGBT non-avalanche-proof switch
  • non-avalanche-resistant components in particular field stop IGBT as a switch
  • the present application is cheaper by no protective element against overvoltages must be used on the switch, since the output circuit already ensures the protection of the switch by its electromechanical and thus electrical properties.
  • the phase angle between the load current and the switch current can be evaluated for controlling and regulating a converter constructed in this way.
  • the switch current is superimposed only by the DC component of the input inductor, which changes the phase shift by a fixed amount, and therefore not or only little depends on the power or input voltage. If the input throttle of the converter is chosen so small that the inductor current can decay to zero or less than zero, one can significantly reduce the proportion of superimposed DC flow from the throttle or almost make it to zero, because then the inductor current is typically in the moment switching on the switch reaches about a zero crossing. Even if the input choke is selected to be larger, a phase detection for power control is possible and only slightly adapted to the respective value of the input choke, since the effective input current in this application is much smaller than the load current.
  • the fluctuation of the input voltage can be compensated via the phase detection and a corresponding frequency change, since the capacitive component of the output current in the converter increases, if due to decreasing input voltage, the active power is smaller.
  • the electromechanical transducer usually has the property to transmit at a decreasing input voltage with this square decreasing power.
  • the converter can only react with an increase in the output voltage even if a small input voltage is applied.
  • the transformation ratio shifts towards larger values and the transformer internal losses increase slightly.
  • the inherent output capacitance of the converter is subjected to a larger voltage, whereby the capacitive current component increases and the ohmic current component decreases.
  • the enlargement of the transformation ratio can be adjusted by a design of the electromechanical transducer so that the output voltage from maximum load (minimum load resistance) to smaller loads (larger load resistance) increases so that the resulting equivalent resistance with respect to the input remains approximately constant or changes little.
  • the size of the input choke can also be used to adjust the power within certain limits at a given frequency. If the input choke is made larger, the transmission power increases, because at the same frequency, because of the electrical characteristics of the class E converter, the effective stored energy in the input choke increases, which is passed to the load circuit.
  • the adjustment of the power through the input inductor is possible only within smaller limits because of the limited bandwidth of the electromechanical transducer and will affect the overall performance insignificantly within the usual tolerances of inductive components.
  • the adjustment of the input choke can be used to adjust the operating point, if another adjustment should not be made. An advantage of the finite design of the input choke is thus the possibility to adjust the lamp power.
  • the input choke is made too large, it can cause an improved smoothing of the current harmonics to the network (noise voltage), but also causes a necessary adjustment of the input capacitance of the converter to smaller values at a power increase and at a constant transformation ratio or a smaller step-down ratio and more consistent or larger input capacitance of the converter.
  • the values for the input choke required for a typical implementation of the invention are to be chosen between 3mH and 20mH at a typical frequency of 100kHz ,
  • a PLL control loop is put into operation in time after the detection of the ignition, in which the zero crossings are sampled by the switch and load current and passed on to a phase detector. Furthermore, this phase difference is passed to a filter which generates a smoothed output voltage. This is switched to a suitable VCO (voltage controlled oscillator), which should be adjusted to a desired value (setpoint comparison) and has a suitable gain.
  • VCO voltage controlled oscillator
  • the output signal of the VCO is returned as a frequency signal with associated duty cycle according to the invention (constant or slightly variable within said range) via a driver to the switch (gate of an IGBT or MOSFET). In this case, the duty cycle may increase slightly with decreasing frequency and slightly decrease with increasing frequency or it is kept constant.
  • the phase difference signal is positively turned on, thereby producing an approximately constant power.
  • load change load reduction
  • the converter will keep the power approximately constant by first "determining" that the phase angle has become larger due to sinking load.
  • the resulting phase voltage Up has become smaller, according to the diagram. If it is given positive to the VCO, then the frequency drops slightly and the power is increased again. As a result, the phase angle decreases again, and the phase difference voltage increases again according to the diagram.
  • the regulation comes eventually to a new actual value of the lamp power, but not very different from the initial value.

Landscapes

  • Circuit Arrangements For Discharge Lamps (AREA)
  • Inverter Devices (AREA)
  • Dc-Dc Converters (AREA)

Claims (20)

  1. Convertisseur à résonance pour alimenter des charges variables, aux caractéristiques suivantes:
    un transformateur piézoélectrique (105) avec une porte d'entrée et une porte de sortie pour fournir un signal de sortie, pour alimenter la charge variable (107);
    un interrupteur (103) destiné à fournir un signal d'entrée d'une source (101) à la porte d'entrée du transformateur piézoélectrique (105);
    un moyen de régulation (109) destiné à réguler une fréquence de commutation de l'interrupteur;
    caractérisé par le fait que le transformateur piézoélectrique (105) est dimensionné et disposé de sorte que, en cas de fourniture d'une puissance nominale à la charge variable (107), un rapport de transformation en abaissement entre le signal d'entrée et le signal de sortie soit de 1,5:1 à 5:1; et
    que le moyen de régulation (109) destiné à réguler la fréquence de commutation (103) est conçu sur base d'un déphasage entre le courant d'interrupteur et le courant de charge à charge variable et/ou à tension d'entrée variable.
  2. Convertisseur à résonance selon la revendication 1, avec une porte d'entrée qui est reliée à une source (101) fournissant une tension d'entrée à partir de laquelle l'interrupteur (103) génère le signal d'entrée pour la porte d'entrée du transformateur piézoélectrique (105), une capacité d'entrée du transformateur piézoélectrique (105) étant réglée de manière fixe en fonction de la grandeur de la tension d'entrée et de la puissance de sortie nominale.
  3. Convertisseur à résonance selon la revendication 2, dans lequel la capacité d'entrée du transformateur piézoélectrique (105) est réglée de manière fixe à une valeur comprise entre 100 pF et 1 nF.
  4. Convertisseur à résonance selon la revendication 2 ou 3, dans lequel la tension d'entrée est comprise entre 80 et 160 volts et la capacité d'entrée du transformateur piézoélectrique (105) est comprise entre 500 pF et 1 nF.
  5. Convertisseur à résonance selon la revendication 2 ou 3, dans lequel la tension d'entrée est comprise entre 160 et 260 volts et la capacité d'entrée du transformateur piézoélectrique (105) est comprise entre 100 pF et 500 pF.
  6. Convertisseur à résonance selon l'une des revendications 1 à 5, qui présente une bobine d'induction (201) qui est connectée entre la source (101) et l'interrupteur (103) disposé en parallèle avec le transformateur piézoélectrique (105).
  7. Convertisseur à résonance selon la revendication 6, dans lequel l'inductance de la bobine d'induction (201) présente une valeur de 3 mH à 20 mH.
  8. Convertisseur à résonance selon l'une des revendications 1 à 7, dans lequel l'interrupteur (103) comporte un disjoncteur commandé par la tension (1037), qui peut être réalisé sous forme de 'Fieldstop IGBT' ou de transistor ,Cool MOS', avec une première entrée, une deuxième entrée et une entrée de régulation à laquelle peut être appliqué un signal de régulation.
  9. Convertisseur à résonance selon la revendication 8, entre la deuxième entrée et la première entrée du disjoncteur commandé par la tension étant connectée une diode (1039).
  10. Convertisseur à résonance selon l'une des revendications 1 à 9, le moyen de régulation (109) comportant un moyen pour détecter (801) une grandeur fonction du courant d'interrupteur, un moyen (803) pour détecter une grandeur fonction du courant de charge et un moyen (807) pour détecter le déphasage entre le courant d'interrupteur et le courant de charge à partir des grandeurs détectées.
  11. Convertisseur à résonance selon la revendication 10, dans lequel le moyen de régulation (109) présente une boucle de réglage de phase (805) destinée à régler un déphasage moyen à une valeur de consigne constante.
  12. Convertisseur à résonance selon la revendication 10 ou 11, dans lequel le moyen de régulation (109) comporte, par ailleurs, un oscillateur (811) dont la fréquence peut être réglée en fonction du déphasage moyen et dont le signal de sortie sert à la régulation de l'interrupteur (103).
  13. Convertisseur à résonance selon la revendication 12, présentant, par ailleurs, un moyen de réglage d'un rapport de balayage prédéterminé du signal de sortie de l'oscillateur (811).
  14. Convertisseur à résonance selon l'une des revendications 1 à 13, dans lequel la charge variable comporte une lampe à décharge de gaz.
  15. Procédé pour alimenter des charges variables par un convertisseur à résonance qui contient un transformateur piézoélectrique (105) avec une porte d'entrée, le transformateur piézoélectrique (105) étant dimensionné de sorte que, en cas de fourniture d'une puissance nominale à la charge variable (107), le rapport de transformation en abaissement de la tension est compris entre 1,5:1 et 5:1, qui comprend, par ailleurs, un interrupteur (103) et un moyen de régulation (109), aux étapes suivantes consistant à:
    réguler une fréquence de commutation de l'interrupteur (103) par le moyen de régulation (109) sur base d'un déphasage entre le courant d'interrupteur et le courant de charge à charge variable et/ou à tension d'entrée variable, pour appliquer un signal d'entrée à la porte d'entrée du transformateur piézoélectrique (105) et générer, de ce fait, un signal de sortie pour alimenter la charge variable.
  16. Procédé selon la revendication 15, aux étapes suivantes consistant à:
    détecter une grandeur fonction du courant d'interrupteur;
    détecter une grandeur fonction du courant de charge;
    déterminer le déphasage entre le courant d'interrupteur et le courant de charge à partir des grandeurs détectées.
  17. Procédé selon la revendication 16 avec une étape de réglage d'un déphasage moyen à une valeur nominale à l'aide d'une boucle de réglage de phase (805).
  18. Procédé selon la revendication 16 ou la revendication 17, aux étapes suivantes consistant à:
    réguler un oscillateur commandé par la tension (811) sur base du déphasage moyen; et
    utiliser le signal de sortie de l'oscillateur comme signal de commande pour l'interrupteur (103).
  19. Procédé selon la revendication 18, avec une étape de réglage d'un rapport de balayage prédéterminée du signal de sortie de l'oscillateur (811).
  20. Procédé selon l'une des revendications 14 à 19, dans lequel est utilisée, comme charge variable, une lampe à décharge de gaz.
EP03782433A 2002-12-17 2003-12-17 Convertisseur a resonance et procede pour alimenter des charges variables Expired - Lifetime EP1573891B1 (fr)

Applications Claiming Priority (3)

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DE2002159069 DE10259069B4 (de) 2002-12-17 2002-12-17 Resonanzkonverter und Verfahren zum Treiben von veränderlichen Lasten
DE10259069 2002-12-17
PCT/EP2003/014427 WO2004055961A2 (fr) 2002-12-17 2003-12-17 Convertisseur de resonance et procede pour faire fonctionner des charges variables

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EP1573891A2 EP1573891A2 (fr) 2005-09-14
EP1573891B1 true EP1573891B1 (fr) 2007-11-07

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CN (1) CN100468941C (fr)
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DE10259069B4 (de) 2007-01-25
WO2004055961A2 (fr) 2004-07-01
CN1726633A (zh) 2006-01-25
WO2004055961A3 (fr) 2004-12-09
DE10259069A1 (de) 2004-07-22
DE50308562D1 (de) 2007-12-20
EP1573891A2 (fr) 2005-09-14
CN100468941C (zh) 2009-03-11

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