DE3921650A1 - CAPACITIVE SENSOR AND ELECTRONIC CIRCUIT FOR CONTACTLESS DISTANCE MEASUREMENT - Google Patents

Description

This invention relates generally to non-contact electronic distance measuring systems and the electronic circuit used for this. In particular This invention relates to a novel one capacitive electronic transducers for gap widths and on a new circuit called Double-T resonance circuit, which in connection with the Transducer can be used.

So far there has always been a lack of useful and accurate devices for measuring the width of columns or spaces without touching the interfaces. For example, in measurement technology have long been Difficulty in measuring the distance of two Known plates. The simple known method at the calibrated feeler blades of just the right thickness have the disadvantage that the Touch is inaccurate that the measurement is time consuming and that no electrical output signal is obtained. Checking the alignment of a car door requires therefore often a caliper measurement of lumps of clay, that were compressed in the gap.

The state of the art is electronic Devices are known for such non-contact gap width measurements to execute. With these previous measuring systems often become electromagnetic induction phenomena exploited to change distances between a transducer and a metallic object measure up. Such measuring systems can with a wide variety of applications, especially when it is used is impossible or undesirable, the object that the distance to be measured determined to touch physically. Other applications are pressure transducers, accelerometers, electronic micrometer, dimensional Comparators, drilling gauges, limit gauges and liquid metal Level sensor.

With the previous electromagnetic induction measuring systems  wasn't the one for quick and accurate Distance measurements required accuracy and Stability achieved. They have certain limitations Development of this state of the art Systems inhibited, such as the difficulty sufficient sensitivity and resolution over the to obtain the actual measuring range of the system. These The limitation is due to the fact that the state systems corresponding to the technology do not exist between the distinguish magnetic properties of the object can and can not compensate for these properties. Another limitation is that caused by temperature changes caused errors. Temperature changes cause impedance changes of the object and the Components for inductive distance measurement of the system, and these changes in impedance are expressed as changes in distance, although such distance changes are in reality have not occurred. Another problem with that Systems corresponding to the state of the art are non-linearity.

U.S. Patent No. 4,160,204 issued to the assignee of this patent has been assigned and incorporated by reference into the present patent refers to an improved, non-contact Distance measuring system, which is high sensitivity and Resolution over the actual measuring range of the system exhibits, practically insensitive to temperature changes is, and an almost linear relationship between the delivered output signal and the to offers measuring distance. That in Patent No. 41 60 204 specified measuring system generally comprises a radio frequency signal Source, an inductive transducer and a reference impedance (both in a signal phase Network and are connected to the source) and a means for comparing the signals from transducers and Reference impedance, whereby an output signal is obtained,  that in relation to the distance between the transducer and the object stands. Is parallel to the transducer a component, such as a capacitor, placed to the sensitivity and resolution of the System increase to errors due to temperature changes the transducer and the measured object significantly reduce or completely eliminate and for an almost linear relationship between the delivered output signal and the measured distance receive.

The inductance sensor specified in patent 41 60 204 Measuring system is for the intended purpose well suited, but has the disadvantage that the Measurement of the material of the interfaces of the gap can be influenced. This sensitivity to the material of the interfaces of the gap Adversely affect the accuracy of the measuring system. There that corresponding to the state of the art in the patent 41 60 204 specified system for inductive measurement based, it cannot with non-metallic interfaces (such as plastic interfaces) will.

The above discussed and other disadvantages of state of the art devices in the non-contact distance measuring device present invention eliminated or reduced. in the Contrary to the inductive measuring system according to the patent 41 60 204 is in the present invention Capacity used to measure the gap width. This has the consequence that the according to the present invention proposed distance measurement regardless of the Materials of the interfaces is and therefore for measurement the gap width for non-metals, such as Plastics, can be used.

An important feature of the present invention is a new type of thin, capacitive transducer with  three connections, the one in the gap to be measured is arranged. This transducer consists of a isolated unit with three parallel ones senior levels, of which the first and the second Level a single electrode and the third level an electrode pair of two at a certain distance electrodes arranged from each other. Each Electrode preferably consists of a thin copper layer, on an insulating substrate, for example made of an epoxy glass composite. Between each electrode of the pair of electrodes in the third level and the single electrode in the second An air gap is provided on the level.

Another important feature of the present Invention is a novel electronic circuit for Use in connection with the new capacitive Three-port transducer. This circuit can create an "artificial resonance" in which a small Capacity (such as that of the invention capacitive sensor with three connections) behaves like this like at or near series resonance with a artificial large inductance. This is through use an artificial network (which is a variant a known double-T circuit) is in the feedback loop a high-gain amplifier reached, the normal lock cut into a resonance-like peak is reversed, namely at a frequency resulting from the following equation results in:

Here means:

C = input arm capacity of the first T branch
C ₂ = input arm capacity of the second T branch
C ₃ = output arm capacity of the second T branch
L = inductance of the middle leg of the second T branch

The novel resonance circuit according to the invention can be used to produce an artificial resonance with other than capacitive measuring devices be used. For example, with inductive Sensors of the type specified in patent 41 60 204 be used.

The features discussed above and other features and Advantages of the present invention will become apparent to those skilled in the art from the following detailed description and the Drawings can be seen.

The following refers to the drawings taken in the same parts in the different Figures have the same reference numbers:

Fig. 1A and 1B are diagrams representing the dielectric field of the prior art and the present invention a corresponding capacitive transducer;

Fig. 2A is a perspective view of the capacitive transducer of the present invention;

Figure 2B is an electrical schematic of the transducer of Figure 2A;

Fig. 3 is another electrical circuit diagram of the transducer of Fig. 2A;

Figure 4 is a graph showing capacitance versus distance for the transducer of Figure 2A;

Fig. 5 is an electrical circuit diagram of a known RLC circuit;

Fig. 6 is a diagram showing the natural series resonance for the RCL circuit of Fig. 5;

Fig. 7 is an electrical circuit diagram of a known double T inductive circuit;

Fig. 8 is an electrical circuit diagram of a known double-T bridge circuit;

Fig. 9 is an electrical schematic of a double T circuit arranged in the feedback loop of a high gain amplifier;

Fig. 10 is an electrical schematic of the resonant circuit according to the invention for producing an artificial resonance;

Fig. 11 is a diagram showing the artificial resonance for the circuit of Fig. 10;

Fig. 12 is an electrical schematic of the network that makes up the inductive leg;

Fig. 13 is a graph _showing L _päq and R _päq as a function of temperature.

Capacitive sensors are known components that respond to the change in the dielectric field between electrodes of different potential. In the conventional capacitive sensors according to the prior art, two parallel plates are used, as is shown at 10 in FIG. 1A. According to an important feature of this invention, however, two capacitor plates 12 and 14 are used for the sensor according to the invention, which are arranged at a certain distance from one another in the same plane and generate an arcuate electric field 16 . It has been found that when the electrodes are arranged according to FIG. 1B, a maximum change in the dielectric field between the electrodes is obtained, which enables more precise and reliable measurements.

We now proceed to FIGS. 2A and 2B, in which the novel capacitive sensor or transducer of the present invention is generally designated by the reference number 18 . The sensor 18 has a pair of electrodes consisting of two electrodes 12 and 14 arranged at a certain distance from one another, both of which lie in the same plane. A coupling plate 20 is arranged at a certain distance below the electrodes 12, 14 in a plane parallel to the said electrodes. Suitable spacers 22 made of an insulating, dielectric material, such as an epoxy-glass composite, are arranged between the coupling plate 20 and the electrodes 12, 14 in order to define air spaces 24 and 26 and a central channel 28 . In addition, a layer 30 of an insulating, dielectric material is applied on the opposite side of the plate 20 with respect to the electrodes 12 and 14 . The structure of the electrodes 12, 14 , the coupling plate 20 and the insulating layers 22 and 30 defines an insulating / coupling structure which is attached to a flat base plate 32 which is at zero potential. It is recommended to place the base plate 32 under the isolation / coupling structure during the measurement in order to have a reference point for the physical contact with one side or interface of the gap to be measured. Finally, two coaxial cables 34 and 36 are connected to the base plate 32 and the electrode 12 or to the base plate 32 and the electrode 14 . In this way, a capacitive T network is obtained which corresponds to the "Pi" network of a capacitor with three connections and for which the following equation applies:

The above equation (1) is illustrated in FIG. 3, in which an equivalent electrical circuit diagram of the sensor of FIGS. 2A and 2B is shown.

The capacitive sensor of FIGS. 2A, 2B and 3 has various essential features and advantages which can be attributed to the novel coupling structure consisting of the coupling plate 20 and the electrodes 12 and 14 . For example, in accordance with the present invention, the sensor geometry can be scaled without changing the sensor capacitance ( C _eq ). This is because the ratio of C _B to C _{A is determined} by the relative position of the coupling plate 20 between the electrodes 12, 14 and the base plate 32 (as well as the type and dimensions of the insulating material selected).

Yet another important feature of the capacitive sensor of this invention is that the temperature coefficient of the sensor capacitance ( C _eq ) can be made zero even if the dielectric constant of the insulating material 22 has a positive (or negative) temperature coefficient. Depending on the ratio of C _B to C _A , their relative temperature coefficients can be set so that C _eq remains constant. This is most easily achieved by lowering the temperature coefficient of C _A , for which purpose the area of the air gap 24, 26 is brought into the correct ratio to the area of the insulating material 22 .

In a preferred embodiment of the present invention, the coaxial cables 34 and 36 of the sensor are connected to a three-port capacitance bridge with a known capacitance (such as the Model 2500, 1 kHz automatic capacitance bridge manufactured by Andeen-Hagerling of Chagrin Falls, Ohio ) or preferably connected to the novel circuit for generating an artificial resonance described below with reference to FIG. 10. Indeed, both circuits do not measure parallel capacitance (and cable capacitance) to ground and only measure the value of C _eq .

The sensor 18 is preferably made of copper-coated epoxy glass printed circuit board material which is dimensioned such that C _B is 15 pF and C _A is 5 pF. The sensor capacitance C _eq is then 1 pF. If C _{A has} dimensions such that the correct air gap area to epoxy glass area ratio is obtained, the high positive temperature coefficient of C _B (denominator) can be compensated for by the lower coefficient of C _A so that a ratio of one is obtained.

We now refer to FIG. 4, in which the change in capacitance for the transmitter 18 is shown when the distance from a metal plate connected to ground is changed. It can be seen that the sensor capacitance of 1 pF decreases as the metal plate is approached, in fact the arcuate dielectric field between the electrodes is weakened (and both C _A 's are reduced).

In FIG. 2A, although only a single pair of electrodes of the coplanar electrodes 12 and 14, but it is obvious that in the present invention, the use is contemplated of a plurality of coplanar electrode pairs with interleaved structure.

We now proceed to Figures 5-13 and describe the novel resonant circuit of the present invention. As mentioned above, this novel circuit is the preferred electronic circuit for the novel capacitive transmitter of FIG. 2A. However, it is obvious that the resonance circuit according to the invention for generating an artificial resonance can also be used in a large number of other applications.

In Fig. 5 and 6, a simple known RCL is - network play, which offers a great potential for development for a wide range of physical measurements. If a physical quantity causes a change in one or more of these electrical proportionality constants, the series resonance network offers extremely interesting possibilities for conversion. If L or C is the variable, a very sensitive change in the phase of E ₀ is detectable (with the sensitivity of R being determined). If R is the variable, the amplitude of E ₀ is the detectable change. It is therefore possible to produce either an amplitude modulation or a phase modulation using the three basic electrical parameters. The limits of application of this relatively simple network usually result from the fact that it is impossible to combine practical values in such a way that the following resonance equations are satisfied:

If L or C is necessarily small due to the nature of the physical measurement, f may become too large, impractical, or, more commonly, R may become too large (or thermally too unstable) to provide stable sensitivity.

It is therefore highly desirable to find a means to overcome the natural limitations on the device values required in the simple network of Figures 5 and 6. A successful way is to build a network whose electrical behavior corresponds to the behavior of a series resonance circuit, but which uses simple, inexpensive components to obtain the properties that cannot be obtained with a real induction coil and a small capacitor. In the 1940s, a double-T network was described in the measurement technology literature, in which a small inductive leg was used and with which it was possible to calibrate exactly to zero. The mathematical analysis showed that the network at zero corresponded to a very large induction coil with a small capacitor placed in parallel. The real small parallel inductor did indeed have the same electrical behavior as an inductor that was several hundred thousand times larger. At zero, however, there is no usable signal voltage for detection or further processing. However, if this null network is placed in the loop of a high-gain amplifier intended for negative feedback, the null signal can be reversed into a signal peak (elimination of the negative feedback). It can be shown that the transfer function of this "signal selective" circuit corresponds to the transfer function of a network for generating a natural resonance (which would require an inductance which cannot be realized).

Transfer function:

With
Φ = phase angle between the voltages E ₀ and E _i
ω = 2 π f

The conventional inductive double-T network (also known as an inductive parallel T network) was implemented (as shown in FIG. 7) with two variable capacitors C ₁ and C ₀ in order to obtain a zero-voltage transmission. As can be seen from FIG. 8, after mathematical conversion of the parallel T networks into equivalent “pi” networks, the essential properties for the generation of an artificial resonance can be seen more clearly.

The equivalent "Pi" transformation gives:

In the balanced state, the actual C lies parallel to an equivalent resonance inductance, which is equal to L , multiplied by the large ratio:

However, it can be seen that various features of this double T must be adapted in order to obtain a practical resonant circuit for the generation of an artificial resonance. First, the zero voltage must be reversed into an amplitude peak. Referring to Fig. 9, it is noted that when a double T network is placed in the feedback loop of a high gain amplifier, the network transfer function is reversed by the feedback equation. Since the double-T is normally operated with a zero signal adjusted to zero, an infinite resonance peak is neither possible nor desirable. In order for the Doppel-T to act as a feedback network, the zero signal is attenuated to approximately -80 dB and set to a frequency slightly lower than the desired resonance frequency. Since the zero signal equations only apply exactly in the real, balanced state, they are used as approximations in the resonance loop, with "Q" and the resonance phase angle of -90 ° in practice with the aid of at least two adjustable double-T components be determined. The closed loop transfer function for the feedback equation of Figure 9 is:

Second, the classic double T needs to be changed to maintain stability against vibrations. It is necessary not just the transfer function to adjust so that a phase angle at the zero signal of + 90 ° is obtained, but also the phase lead at higher frequencies under + 180 ° to keep a to avoid positive feedback. (A preferred one Exemplary embodiment with a capacitive sensor from FIG. 1 Picofarad is inFig. 10 reproduced.) Note thatR _s (5 ohms) andR _p (35 ohms) were provided to the Q the induction coilL and their coil resistanceR _i to Reduce. Otherwise, wouldC.₂ andL at 360 kHz one  Series resonance circuit with highQ form what a excessively large, positive phase angle and possible Interfering vibrations. The entire loop gain is a function of all resistances (R₂ /R₁, R₄ /R₃,R _p,R _s,R _i,R ) and fromC.₁. A simple setting fromQ at the resonance can therefore over the PotentiometerR respectively. The resonance frequency is in primarily a function ofC, L, C₂ andC.₃ (at a certain interaction with the gain determining Variables).R _p andR _s also have an important function for temperature compensation of the coil resistanceR _i. TheFig. 12 shows the network with the inductive leg and the equivalentL _päq andR _päq. TheFig. 13 is a Diagram in which the behavior ofL _päq andR _päq at a change in temperature ofR _i is shown. ThereL _päq  is a variable of the resonance frequency, its value and their rate of change through the ratio of temperature stableR _s toR _i to be controlled. Furthermore becomesR _p selected so that the rate of change fromR _päq the loop gain adapts enough to the Resonance frequency with temperature changes constantly hold.

The example of FIGS. 10 and 11 shows that a 1 picofarad sensor (such as the sensor shown in FIG. 2A) has the equivalent transfer function of a 30 kHz series resonance with an artificial inductance of 27.9 H (or 600,000 times) actual inductance). In this case the sensor has a capacitance change of 0.1 (10%) over its range of displacement (which results in a phase angle change of 10 °).

In the example shown in FIGS. 10 and 11:

Although it has been shown that the resonance circuit according to the invention for generating an artificial resonance can be very sensitive to small changes in capacitance, it should be noted that a similar configuration is advantageous for inductive sensors (eddy current sensors). When using an inductive sensor as the L- leg of the double-T, the equivalent series inductance is usually one million times as large as L , which makes this configuration operate as a resonant network at a frequency much lower than that normally used with such small C - and L Values would be possible. Using a varactor for C ₃ and a phase detector, the double-T can also be phase locked (or forced to match) at resonance, thereby obtaining a control loop output voltage, with the advantages that a closed loop system has over a transducer has an open loop.

Claims (27)

1. Capacitive sensor, characterized in that it consists of
at least one pair of electrodes made of first electrodes which are at a certain distance from one another and are arranged coplanar in a first plane;
a coupling plate located at a certain distance from said pair of electrodes in a second plane, said second plane being parallel to said first plane;
a base plate spaced from a third plane from said coupling plate, said third plane being parallel to said first and second planes, and said second plane being arranged between said first and third planes;
a first dielectric material disposed between said first electrodes and said coupling plate;
a second dielectric material disposed between said coupling plate and said base plate;
a first gap between one of said first electrodes and said coupling plate and
a second gap between the other of said first electrodes and said coupling plate. 2. Sensor according to claim 1, characterized in that
said first dielectric material includes first and second pairs of individual elements, each electrode of said first pair of electrodes abutting a pair of said elements;
said first gap is located between said first pair of elements and
said second gap is located between said second pair of elements. 3. Sensor according to claim 1, characterized in that said first and second gaps are air gaps. 4. Sensor according to claim 2, characterized in that that said first and second gaps are air gaps. 5. Sensor according to claim 1, characterized in that said first and second dielectric material consists of an epoxy glass composite. 6. Sensor according to claim 2, characterized in that said first and second dielectric material consists of an epoxy glass composite. 7. Sensor according to claim 1, characterized in that that said first electrodes that said Coupling plate and said base plate copper contain. 8. Sensor according to claim 1, characterized in that that between those at a certain distance from each other a channel is provided in the arranged first electrodes, that by said first dielectric material and the said coupling plate is defined. 9. Sensor according to claim 2, characterized in that that between those at a certain distance from each other a channel is provided in the arranged first electrodes, that by said first dielectric material and the said coupling plate is defined. 10. Sensor according to claim 1, characterized in that it comprises
a first coaxial cable connected to one of said first electrodes and to said base plate;
a second coaxial cable connected to the other one of said first electrodes and said base plate. 11. Sensor according to claim 10, characterized in that that he had one with said first and second Having coaxial cable connected electronic circuit,  to measure the distance between two interfaces. 12. Electronic circuit for generating an artificial resonance, characterized in that it consists of
an inductive parallel T network and
a high gain amplifier, said parallel parallel T network being electrically connected to the feedback loop of said high gain amplifier to reverse the cutout of said parallel parallel T network into a resonance peak occurring at a predetermined frequency. 13. Circuit according to claim 12, characterized in that said high gain amplifier has a first amplifier with a second Amplifier is connected in series. 14. Circuit according to claim 12, characterized in that said inductive parallel T network consists of
a first T-branch having a first input arm, a first output arm and a first middle leg between said first input arm and said first output arm, having a first input capacitance C in said first input arm, a first output resistance R in said first output arm and has a first mass variable capacitance C ₁ in said middle leg;
a second T-arm with a second input arm, a second output arm and a second middle leg between said second input arm and said second output arm, there being a second input capacitance C ₂ in said second input arm, a second output capacitance C ₃ in said second output arm and an inductance L in said middle leg; and
a loop of a resistor R _p , which is electrically connected to a variable capacitance C ₀, said loop being electrically connected between the capacitance C ₂ of the second input arm and the capacitance C ₃ of the second output arm. 15. A circuit according to claim 14, characterized in that said predetermined frequency is derived from the formula results. 16. Circuit according to claim 14, characterized in that said high gain amplifier has a first amplifier with a second Amplifier is connected in series. 17. The circuit according to claim 16, characterized in that it has a first resistor R ₂, a second resistor R ₃ and a third resistor R ₄, all of which are connected in series and influence the sensitivity of said first and second amplifiers. 18. A circuit according to claim 13, characterized in that it has a first resistor R ₂, a second resistor R ₃ and a third resistor R ₄, all of which are connected in series and influence the sensitivity of said first and second amplifiers. 19. A circuit according to claim 12, characterized in that said inductive parallel T network consists of
a first T-branch having a first input arm, a first output arm and a first middle leg between said first input arm and said first output arm, having a first variable input capacitance C in said first input arm, a first variable output resistance R in said has a first output arm and a first capacitance C ₁ in said middle leg;
a second T-branch with a second input arm, a second output arm and a second middle leg between said second input arm and said second output arm, there being a second input capacitance C ₂ in said second input arm, a second output capacitance C ₃ in said second output arm and an inductor L connected in series to a resistor R _i and a resistor R _s , and said inductor L , said resistor R _i and said resistor R _{s are} in said middle leg;
a resistor R _p electrically connected between said second input arm and said second middle leg resistor R _s . 20. Circuit according to claim 19, characterized in that said high gain amplifier has a first amplifier with a second Amplifier is connected in series. 21. A circuit according to claim 20, characterized in that it has a first resistor R ₂, a second resistor R ₃ and a third resistor R ₄, all of which are connected in series and influence the sensitivity of said first and second amplifiers. 22. A circuit according to claim 19, characterized in that said predetermined frequency is derived from the formula results. 23. A circuit according to claim 19, characterized in that said variable capacitance C consists of a capacitive sensor. 24. Circuit according to claim 23, characterized in that the capacitive sensor consists of
at least one pair of electrodes made of first electrodes which are at a certain distance from one another and are arranged coplanar in a first plane;
a coupling plate located at a certain distance from said pair of electrodes in a second plane, said second plane being parallel to said first plane;
a base plate spaced from a third plane from said coupling plate, said third plane being parallel to said first and second planes, and said second plane being arranged between said first and third planes;
a first dielectric material disposed between said first electrodes and said coupling plate;
a second dielectric material disposed between said coupling plate and said base plate;
a first gap between one of said first electrodes and said coupling plate and
a second gap between the other of said first electrodes and said coupling plate. 25. Sensor according to claim 24, characterized in that
said first dielectric material comprises first and second pairs of individual elements, each of said first electrodes abutting a pair of said elements;
said first gap is located between said first pair of elements and
said second gap is located between said second pair of elements. 26. Sensor according to claim 24, characterized in that between those at a certain distance a channel is provided in the arranged first electrodes, that by said first dielectric material and the said coupling plate is defined. 27. Sensor according to claim 24, characterized in that the
a first coaxial cable connected to one of said first electrodes and to said base plate;
a second coaxial cable connected to the other one of said first electrodes and said base plate.