DE1204265B - Electrical switchgear, in particular for controlling a multi-phase converter circuit - Google Patents

Description

FEDERAL REPUBLIC OF GERMANY

GERMAN

PATENT OFFICE

EDITORIAL

Int. α .:

H03k

German KL: 21 al-36/18

Number: 1204 265

File number: E 22681 VIII a / 21 al

Filing date: April 5, 1962

Opening day: November 4, 1965

The invention relates to electrical switchgear, in particular for controlling a multi-phase converter circuit, with a series of electronic switches, each of which has a control electrode and brought into the conductive state by a control pulse applied to this control electrode while it is by reducing the DC voltage applied to the switch to or below a predetermined value is converted into the non-conductive state and at which this electronic Switches are connected to a common supply voltage source and so interconnected are that they are sequentially by cyclic control signals applied to their control electrodes in the conductive state are transferred and in this way generate a sequence of output signals and with means which come into effect when a switch is switched to the conductive state, which the DC voltage applied to the switch preceding in the sequence to the specified Value or less and so the switch is switched to the non-conductive state and where normally only one switch is conductive at a time.

The electronic switches can be mercury arc rectifiers or preferably controlled semiconductor (e.g. silicon) rectifiers.

The mentioned connection of the electronic switches preferably comprises capacitive coupling means between the anodes of the rectifier; the output devices are connected to the cathodes. Alternatively, the output devices with the anodes of the rectifiers and the capacitive coupling means be connected to their cathodes.

In such a switchgear, it is possible that, as a result of external influences, a fault condition occurs in which 2 or more of the said electronic switches are conducting at the same time instead of one after the other. Unless all electronic switches are conducting at the same time, the fault state normally balances itself out automatically as soon as the next non-conducting electronic switch is brought into the conducting state in the course of the normal operating sequence. If, however, all electronic switches are in the conductive state at the same time, this disturbance state does not normally equalize itself; However, it has been found that the persistence of this last-mentioned disturbance state can be avoided in a simple manner by using electrical switchgear, in particular for
Control of a multi-phase converter circuit

Applicant:

The English Electric Company Limited, London

Representative:

Dipl.-Ing. C. Wallach, patent attorney,

Munich 2, Kaufingerstr. 8th

Named as inventor:

Eric Wistow,

Michael Brown, Stafford (Great Britain)

Claimed priority:

Great Britain April 5, 1961 (12183)

that temporarily the DC potential normally supplied to the electronic switches interrupts.

Based on this knowledge, the invention proposes an electrical switchgear of the above Kind before, which is characterized in that a detector device is provided which at Occurrence of a fault condition in which all of the electronic switches in the series are conductive at the same time are, responds, and that an interrupter device is provided which upon detection of a such disturbance state by the detector device temporarily the turn of electronic switches lying DC voltage is reduced to the specified value or below and thus temporarily non-conductive power.

Such a switchgear according to the invention can be used as the output stage of an electrical switching system for the supply of control signals to the control electrodes of the converter elements, such as controlled semiconductor z. B. silicon) converters or mercury arc converters a static power converter system (i.e. rectifier or inverter system). With such a

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term system, the switchgear according to the invention can be used to create a cyclical sequence short pulses which are fed to the control electrodes of the switchgear from a control circuit are converted into so-called "long impulses", which are the control electrodes of the converter elements are fed. "Long impulses" are understood to mean impulses whose duration is im is substantially equal to the ignition periods of the converter elements; in the case of a three-phase converter these ignition periods (and thus the duration of the control electrode pulses) amount to 120 electrical Grade.

The control circuit can be of such a type that the cyclically successive short pulses can be changed in their phase position with respect to a reference phase; for this purpose a AC voltage of essentially sinusoidal waveform with a variable DC bias voltage are combined, the amount of which does not exceed the peak value of the alternating voltage. The waveform preferably has the aforementioned alternating voltage at its wave peaks and valleys pointed extensions (extensions, elevations) to ensure that the alternating voltage is the The amount of DC bias over a small fraction of the period also exceeds when the magnitude of the DC bias is substantially equal to the peak value of the AC voltage is.

The “peak value” is to be understood as the maximum value which the alternating voltage without the mentioned processes.

The alternating voltage can, for example, have a “sawtooth” waveform, preferably it has however, a substantially sinusoidal waveform.

The invention therefore also relates, in combination, to an electrical switching system in which a AC voltage is combined with a variable DC bias voltage, the amount of which denotes the Does not exceed the peak value of the alternating voltage, the waveform of the alternating voltage being peaked Has extensions on their wave crests and valleys; with such an electrical switching system an electrical switchgear of the type mentioned above, which preferably includes detector devices, which respond to a state in which all electronic switches are at the same time are in the conductive state, as well as having interrupter devices that respond to said Responding detector devices and are arranged so that when such a condition occurs the direct current potential supplied to the electronic switches during normal operation for a period of time, which is sufficient to bring the electronic switch into the non-conductive state, interrupted and then restored so that the electronic switch is normal, cyclical successive mode of action can resume.

Further advantages and details of the invention emerge from the following description of a Embodiment in the last-mentioned combination; forms the switchgear according to the invention while the output stage of a control circuit of the type mentioned assigned to it, together with the a switching system for controlling a six-phase mercury arc converter forms; the description of the embodiment is based on the drawings; in this shows

Fig. 1 is a block diagram of the switchgear according to the invention and the control circuit existing switching system,

Fig. 2 is a circuit diagram for three phases of the switchgear together with the power supply and an interference suppression circuit,

ίο Fig. 3 and 4 circuit diagrams of a first or a second stage for one phase of the control circuit,

5 shows a circuit diagram of the device for generating the (hereinafter referred to as "phase switching pulse") pointed extensions for two opposite ones Phases,

6 shows the input and output signals of a Phase of the first stage of the control circuit for a certain value of the DC bias voltage,

Fig. 7 shows the input and output signals of a phase of the first stage of the control circuit for a other value of the DC bias voltage,

8 shows the input signal and an intermediate signal of the phase switching pulse circuit according to FIG. 5, F i g. 9 the output signal of the phase switching pulse circuit mentioned,

10 is a graph showing the relationship between control current and converter output voltage;

11 is a graph showing the relationship between control voltage and converter output voltage;

FIG. 12 is a circuit diagram of that in FIG. 2 in block form interference suppression circuit shown,

13 shows a circuit diagram for 3 phases of a Alternative version to the switchgear shown in Fig. 2, but omitting the power supply and the ducking circuit.

The switching system shown in the block diagram in Fig. 1 consists of the control circuit 1, whose six-phase output of the switchgear 2 is supplied, which is the output stage of the switching system forms; the output of switchgear 2 becomes the cathode and the six control grids of the through fed to the switching system controlled converter. The control circuit 1 consists essentially of a first stage 3 and a second stage 4. The first stage has 3 inputs, namely for one Six-phase sinusoidal alternating voltage source 5, a source 5 a for six-phase switching pulses and one DC power source 6 for the DC bias voltage; this forms the control signal for changing the Phase position of an output signal from circuit 1 with respect to a reference phase, as will be discussed further below will be described in detail.

The first stage 3 generates a six-phase output variable in the form of square-wave pulses, which are shown in the second stage 4 are shaped so that the output of the second stage 4 is a six-phase sequence receives sharp unipolar pulses.

In Fig. 2 only 3 phases are shown since the Circuit for the other three phases is identical; the power supply of the system and the interference suppression circuit (which will be described below) are common for all 6 phases.

The circuit of the system consists essentially of a controlled silicon rectifier for each phase 30, 31, 32, a relatively large

ßen cathode resistor 33, 34, 35, a proportionate small anode resistor 36, 37, 38, one to the cathode of the controlled silicon rectifier connected resistor 39, 40, 41 and one in parallel with the controlled silicon rectifier and the diode 42, 43, 44 lying to the anode resistance. The cathode resistance is in each case bridged by a capacitor 45,46,47.

The circuit is made from a three-phase AC power source 51 fed in star connection, via a rectifier bridge circuit 52 and resistors 53, 54. The neutral star point of the star connection power source 51 forms the connection for the cathode of the mercury arc converter; the center tap between two parallel to the DC output of the rectifier-resistor combination 52, 53, 54 lying storage capacitors 55, 56 is connected to this cathode connection via a further resistor 57. The grid resistors 39, 40, 41 are also connected to this cathode line via grid capacitors 58, 59, 60 tied together.

The anodes of the controlled silicon rectifiers are connected via three anode capacitors 61, 62, 63 in Delta or delta connection connected to one another.

The interference suppression circuit 48 is shown in detail in FIG. 12; it basically consists of a blocking part with a controlled silicon rectifier 70, from a switch part with a controlled silicon rectifier 71 as well as from a part determining the disturbance state a unipolar transistor 72.

The circuit associated with the silicon controlled rectifier 70 consists of a resistor 73, a capacitor 74 and a diode 75. Those associated with the controlled silicon rectifier 71 Circuit consists of resistors 76 to 79 and a capacitor 80. The one with the unipolar transistor The circuit connected to 72 consists of resistors 81 and 82 (the latter of which is changeable) and from a capacitor 83. In the positive lead is on the input side a current limiter resistor 84 is provided; the output of the interference suppression circuit is on a capacitor 85 removed.

Fig. 3 shows the circuit for one phase of the first stage 3 of the control circuit 1, however the circuit 5a (shown separately in FIG. 5) for supplying the phase switching pulses for clarity is omitted for the sake of The circuit of FIG. 3 consists essentially of a p-n-p transistor 7, whose collector-emitter circuit is connected to a 14 volt DC power supply, from a collector load resistor 8 and a diode 9 for limiting the base-emitter reverse voltage of the transistor 7; the DC voltage control signal is applied via a diode 12 to a low input resistance 10 applied; this input resistance is in series with one phase of the secondary winding a transformer 4, the primary winding of which is supplied with a sinusoidal alternating supply voltage; the Secondary winding is connected to the base of transistor 7 via a resistor 11; the diode 12 is intended to prevent the resistor 10 from being bridged by the impedance of the control signal source.

As will become apparent from the following description, the feed of the control signal in FIG Way on the resistor 10 the advantage that a control signal zero a fully delayed output pulse, d. H. a state of complete inversion. That means a decrease of the control signal, the converter grid pulses in the direction of a reduction in the converter current shifts.

Alternatively, the control signal could also be applied at AB (where the resistor 10 would be omitted or selected with a large value in order to obtain a small bias voltage at the beginning); in this case, a control signal zero leads to an output pulse lying in the middle between maximum phase delay and maximum phase lead.

As a further alternative, a variable direct current control signal could be applied between point B and a further resistor (not shown) connected to the base of transistor 7, and zv / ar in addition to a direct voltage signal of fixed magnitude applied between AB.

By suitable selection of the size of the DC voltage signal supplied at AB , the output pulse for a control signal zero can therefore have any given phase position between the limits of full delay and full lead. The size of the DC voltage signal applied at AB then determines the horizontal position of the control characteristics shown in FIGS.

The output of this first stage of the control circuit appears at resistor 8 and becomes the in Fig. 4 is supplied to the second stage shown; this essentially consists of a p-n-p transistor 13, to which a diode 14 is connected to limit the applied base-emitter reverse voltage is, from an output coupling transformer 15, from one connected to the collector of the transistor 13 Series connection of a resistor 16 and a capacitor 17 for positive feedback to the base of the transistor 7 of the first stage and from a parallel arrangement of a capacitor 18 and a resistor 19 in the emitter circuit of the transistor 13. Via the secondary winding of the Transformer 15 is a diode 15 a.

For reasons that will be given in the later description of the mode of operation of the overall arrangement 6 through 9 is understood in FIG. 5 the phase switching pulse circuit for two exactly opposite phases of the six-phase converter shown. This circuit consists essentially of a transformer 20, its secondary winding has a center tap; also from current limiting resistors 21, 22, Zener diodes 23, 24, as well as signal differentiating capacitors and resistors 25, 26, 27, 28. The circuit is with the Bases of the transistors 7 (see FIG. 3) of the respective first stages of two opposite phases of the Six-phase connected. For the other two pairs of opposite phases of the arrangement are corresponding phase switching pulse circuits intended.

In the following, the mode of operation of the arrangement according to FIGS. 1 to 5 and 12 will now be described with reference to FIGS. 6 to 11. First, the control circuit (Figs. 1, 3, 4 and 5) will be discussed.
Fig. 6 shows a negative bias at the base of the transistor 7 based on its emitter potential, such that the base potential is only over a small fraction of the period of the sine change.

selstromes becomes positive with respect to the emitter potential; during that small fraction of the period will the transistor? thus blocked, so that its collector current drops to zero in the manner shown. This negative bias is derived from the fixed 14 volt DC potential shown in FIGS here.

As the base negative bias potential decreases to zero and then to increases in the positive direction in that the control signal fed in via resistor 10 increases, the switch-off point moves downwards along the sine wave until the one shown in FIG. 7 state shown is reached; under these conditions the transistor 7 is non-conductive for most of the period, and the collector current only flows during the small fraction of the period during which the Base potential is negative with respect to the emitter potential.

If, as in the following with reference to FIG is explained in more detail, the moment in which the collector current drops to zero, in the second stage the arrangement is used to generate the desired switching pulse, so results from a comparison 6 and 7 indicate that a change in the magnitude of the DC bias signal is a Caused phase shift of this moment compared to a reference phase. As can be seen in this Way, a maximum phase shift of this moment of slightly less than 180 ° is reached will.

The in F i g. 4, the second stage of the control circuit shown has the task of generating the square-wave pulse output the first stage, d. H. the collector current of transistor 7, into a sharp positive pulse to convert, which is fed to the switchgear 2, which forms the output of the overall system.

The in F i g. 4 works in the following way: If the transistor 7 is switched off, then it falls its collector potential to the potential of the negative feed line, and this negative potential becomes also fed to the base of transistor 13 of the second stage. The transistor 13 is therefore in the conductive Brought state, and the capacitor 18 charges quickly across the primary winding of the pulse transformer 15 on. The initially high collector current of transistor 13 falls as it increases Charging of the capacitor 18 and then remains on a determined by the resistor 19 low stationary value. The provided across the resistor 16 and the capacitor 17 positive Feedback from the collector of transistor 13 to the base of transistor 7 ensures that the Switching process runs very quickly and that with it a rapid increase in current in the pulse transformer 15 is reached. This rapid switching also reduces the power loss in the transistor 13.

Turning on the transistor? causes the base of transistor 13 to become positive and the Transistor 13 is thus blocked; the small current flowing in the pulse transformer 15 becomes therefore interrupted, as a result of which a negative pulse per period is generated in the output of transistor 15 will. Normally only the positive pulse caused by the switching on of the transistor 13 becomes needed, and the mentioned negative pulse is generated by means of the over the secondary winding of the transformer 15 lying diode 15 a suppressed.

The disconnection of the transistor 13 also leads to a discharge of the capacitor 18 through the resistor 19, whereby the readiness for the next cycle is restored.
From the representation of the phase switching pulse circuit in FIG. 5 and the waveform representations in FIG. 8 and 9 it can be seen that the sinusoidal output voltage of the transformer 20 (shown in dashed lines in FIG. 8) with a center tap through

ίο the Zener diodes 23, 24 is cut so that the voltage occurring at the capacitors 25, 26 has the curve shown in full in FIG. 8. As a result of the characteristics of the Zener diodes 23, 24, the approximately rectangular waveform of the voltage appearing on the capacitors 25, 26 is asymmetrical, since the Zener voltage is about 2 volts in the forward direction, but about 10 volts in the reverse direction.
This square wave is differentiated by the capacitors 25, 26, together with the base-emitter resistors of the transistors 7 of two exactly opposite phases of the six-phase arrangement; the resulting signals are then sent to the bases of the transistors 7 via resistors 27,

as 28 in parallel with the sinusoidal alternating voltage source fed.

The waveform of the output of the phase switch pulse circuit is shown in FIG. 9 shown; together with the sinusoidal alternating supply voltage, the result shown in FIGS. 6th and Fig. 7 shows a sharp waveform for the base potential.

The values of the resistors 21, 22, the diodes 23, 24 and the capacitors 25, 26 are chosen so that there is a smooth transition between the sinusoidal parts and the phase switching pulse parts the waveform results.

As can be seen, between the output variables of transformers 4 (Fig. 3) and 20 (Fig. 5) a phase shift of 90 ° is required; this comes from a 60 ° phase shift between the two transformers (using suitable star-delta connections) together with a 30 ° phase shift comes about, which is achieved by a (not shown) capacitor-resistor network is effected on the input side of the transformer 4. This capacitor-resistor circuit reduces any possible distortion in the sinusoidal AC supply voltage to a negligible one Value that can hardly give rise to faulty operation. You can see that through the phase switching pulses at the otherwise essentially sinusoidal alternating supply voltage pose a risk it is reduced that no output signal is given by the control circuit because the DC bias coincidentally exceeds approximately the sine peak value of the AC supply voltage. The phase switching pulses also enable, as has already been described with reference to FIGS. 6 and 7, an enlargement the range of permissible phase changes of the output signal sequence of the arrangement compared to a Reference phase to practically 180 °, while so far an extension of this range over about 165 ° leads to uncertainties led.

In the following, the mode of operation will be explained with reference to FIG the switchgear according to the invention, which forms the output stage of the switching system, described:

It is assumed that the silicon rectifier 30 is in its non-conductive state; the No. 1 grid of the mercury arc converter is held negative via the cathode resistor 33 and the grid resistor 39. Used by the transformer 15 (see. Fig. 4) of the second stage 4 of the control circuit 1 is a sharp positive Pulse fed to the control electrode of the silicon rectifier 30, this is switched on and its cathode potential rises to almost the full positive value, thereby making electrode No. 1 of the Mercury arc rectifier is supplied with a positive potential.

This positive potential remains on this electrode until the silicon rectifier 30 again is blocked, which occurs simultaneously with the switching on of the silicon rectifier 31 in the following manner takes place.

If the silicon rectifier 30 is in the conductive state, the potential at points a, d and e is almost equal to the full positive value, and the potential at point b has the full negative value. If the silicon rectifier 31 is now switched on (by the control circuit 1, in the same way as described above for the silicon rectifier 30), the potential of point e immediately falls to that of point b (ά. H full negative value) and then increases exponentially as the capacitor 46 charges through the resistor 37 and the silicon rectifier 31. When the potential at point e drops from the positive to the negative value, the potential at point d is reduced by a similar amount due to the connection via capacitor 62. The potential at point d then rises exponentially to the full positive Value, to the extent that the capacitor 62 charges through the resistor 36. As soon as the potential of the point d drops to the full negative value, the voltage applied to the silicon rectifier 30 is reversed, since the point is initially held by the capacitor 45 at a potential which is close to the full positive value. This reverse bias voltage is maintained at the silicon rectifier 30 for a period of time which is greater than the switch-off period of the rectifier, to be precise by a suitable choice of the time constants of the resistor-capacitor combinations 36, 62; 36, 45 and 33.45.

From the symmetry of the FIG. 2 it can be seen that in the same way as the silicon rectifier 30 is blocked as soon as the rectifier 31 is switched on, as above described, the silicon rectifier 31 is also blocked as soon as the rectifier 32 is switched on by applying a pulse to its control electrode, as was done above for the rectifier 30 was described.

As already mentioned, in the case of the more common six-phase mercury arc converter circuit, namely a double-star connection with inter-phase reactance, two three-phase switchgear of the type shown in FIG. 2 (but with a common DC power supply 51 to 54 and common interference suppression circuit 48) applied, the control pulses from the relevant Three-phase elements of the control circuit 1 are 60 ° out of phase. The principle of The switchgear assembly according to FIG. 2 can alternatively be used for any number of phases; so For example, a ring of six controlled silicon rectifiers could be used to drive a six-phase Siro converter system to serve.

The output pulses of the switchgear 2 can be output by any series and / or parallel connection multiple controlled silicon rectifiers per phase to any desired Value can be set.

In the event of malfunctions in the mercury arc converter, high positive Surge voltages occur. If they are not suppressed, these surge voltages would be applied to the Cathode of the respective silicon rectifier occur, which leads to a reverse bias on this Consequence would have. To avoid this, the diodes 42, 43 and 44 are provided, which the structure of this ao prevent reverse voltage by allowing the current to pass directly into the storage capacitors 55, 56, with the result that the surge voltages are only applied to the relevant grid resistor (39, 40 or 41) occur.

In the following, the mode of operation will be shown with reference to FIG the interference suppression circuit 48 (see FIG. F i g. 2) are described: If this circuit initially the DC supply voltage is supplied, the controlled silicon rectifier 70 is blocked and thus keeps the voltage from the silicon rectifiers 30, 31 and 32 of the output stage (cf. Fig. 2) remote; the capacitor 74 starts over to charge the resistor 73 until the control electrode of the silicon rectifier 70 is sufficient Receives power to turn on the rectifier. The capacitor 85 then charges through the Resistor 84 on, and the DC voltage is fed to the output stage (Fig. 2) and the silicon controlled rectifier 71 supplied. The voltage across the silicon rectifier 70 then approximately drops to zero and capacitor 74 discharges.

As long as the silicon rectifiers 30, 31 and 32 of the output stage (FIG. 2) fire normally (ie sequentially), the voltage across the variable resistor 82, which is proportional to the current in the negative DC current line, is not sufficient to fire the unipolar -Transistor 72 off, so that the silicon rectifier 71 remains non-conductive. However, if silicon rectifiers 30, 31 and 32 were to conduct simultaneously, the current in the negative DC line would increase to three times its normal value, causing the voltage across resistor 82 to increase to a value at which the emitter-base circuit would be the unipolar transistor 72 is forward biased; This has the consequence that the capacitor 83 discharges into the control electrode of the controlled silicon rectifier 71, which is switched on as a result. As soon as this rectifier is switched on and when the resistors 78 and 79 are of the same size, the potential of the point g falls approximately to the voltage zero and thereby takes the anode potential of the silicon rectifier 70 with it. The potential in point / remains approximately at the full positive voltage, so that the anode of the silicon rectifier 70 becomes negative with respect to its cathode for a short time, which is determined by the time constant of the capacitor 85. For this purpose

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it is essential that capacitor 80 be larger than capacitor 85.

The short time mentioned is selected so that it exceeds the recovery time of the silicon rectifier 70. This is blocked, and the capacitor 85 discharges through the resistors 79, 78 and the silicon rectifier 71; this lowers the output voltage of the interference suppression circuit to zero and the silicon rectifiers 30, 31 and 32 (cf. FIG. 2) and the silicon rectifier 71 can resume their blocking state. After the silicon rectifier 70 is blocked, the voltage builds up across it as a result of the charging of the capacitor 80 and the discharge of the capacitor 85 again. This leads to the capacitor 74 charges up to the ignition point and triggers the silicon rectifier 70 again, thereby restoring the normal DC voltage supply to the output stage (Fig. 2) and normal operation of the switching system can be resumed.

FIG. 13 shows an alternative embodiment of the switchgear shown in FIG. 2; same Parts are shown in FIG. 13 with the same reference numerals as in FIG. 2 provided.

A comparison of the two circuit diagrams shows the main simplification of the arrangement according to Fig. 2 that the capacitors 61 to 63 and replace the three separate anode resistors 36 to 38 in FIG. 2 a common Anode resistance 36 'occurs for three of the six phases (as in the previous case, there are only three Phases shown).

The connection of the circuit to the power supply and the connection with the one to be controlled Mercury arc converters are not shown in detail. The power supply can be similar as in Fig. 2 with the inclusion of an interference suppression circuit 48 (FIGS. 2 and 12) and as in the previous case, the cathode of the mercury arc converter can be with the star point connected to the AC power source. The mercury arc converter control grid are connected to the switchgear via suitable resistor-capacitor-rectifier circuits (not shown) tied together.

The switchgear shown in Fig. 13 works as follows: As soon as the controlled silicon rectifier 30 is turned on, its cathode capacitor 45 discharges; This allows the potential to be transferred to the The anodes of the silicon rectifiers 30 to 32 drop immediately to the potential of the negative line. Since the resistance of the anode resistors, such as 36, is small compared to is the resistance of the cathode resistors such as 33, the cathode capacitor charges in another phase in which the silicon rectifier is currently switched on is, almost to the full line potential. Therefore the cathode of the controlled silicon rectifier lies in this last-mentioned phase at a positive potential as soon as its anode potential as a result of the switching on of the silicon rectifier 30 has been made negative. As a result the silicon rectifier is switched off in the latter phase. In the same way will the silicon rectifier 30 then by switching on the next silicon rectifier switched off within the cycle, and the cyclical operation of all silicon rectifiers follows in the same way.

From Figs. 10 and 11 it can be seen that about the normal control range is the ratio between the output voltage of one of the operated Control circuit controlled converter on the one hand and the current or the voltage of the control circuit supplied DC, bias or control signal is, on the other hand, essentially linear, provided that the device at 4 (Fig. 1) AC voltage supplied has a sine waveform. The phase switching pulses make these Characteristic curves reshaped at the extreme ends of the control range.

Claims (3)

Patent claims: 1. Electrical switchgear, especially for controlling a multi-phase converter circuit, with a series of electronic switches, each of which has a control electrode and brought into the conductive state by a control pulse applied to this control electrode is, while by reducing the DC voltage applied to the switch to or is converted into the non-conductive state below a predetermined value, and when these electronic switches are connected to a common supply voltage source and are interconnected in such a way that they are successively connected to their control electrodes by cyclic / applied control signals are transferred into the conductive state and in this way a Generate sequence of output signals, and with the transition of a switch to the conductive State effective means, which are connected to the switch preceding in the order DC voltage is reduced to the specified value or below, and so the switch transferred to the non-conductive state and normally only one switch at a time is conductive, characterized in that a detector device is provided which when a fault condition occurs in which all of the electronic switches (30, 31, 32, F i g. 2 and 13) are simultaneously conductive, responds, and that an interrupter device is provided, which when such a disturbance state is detected by the detector device temporarily the turn of the electronic switches (30, 31, 32, Fig. 2 and 13) DC voltage is reduced to the specified value or below and thus temporarily non-conductive power. 2. Electrical switchgear according to claim 1, characterized in that the detector device a signal generated by the strength of the current flowing from the common supply current source to the series of electronic switches (30, 31, 32, Fig. 2 and 13) flows, depends, and reached a predetermined value when all electronic switches (30, 31, 32, Fig. 2 and 13) of the series are simultaneously conductive, and that the interrupter device is a switching device which comes into operation when the signal from the detector reaches its specified value, and those at all electronic switches 10 (30, 31, 32, Fig. 2 and 13) lying in the row DC voltage reduced to a predetermined value or below, as well as delay devices that when the switching device is actuated are put into action and after a predetermined delay unit the the electronic switches (30, 31, 32, Fig. 2 and 13) to bring the DC voltage back to normal. 3. Electrical switchgear according to claim 2, characterized in that the switching device the interrupter device has a further electronic switch (70, Fig. 12), between the supply current source and the series of electronic switches (30, 31, 32, Fig.2 and 13) is connected and is normally conductive, so that the DC voltage of the supply current source on the series of electronic switches (30, 31, 32, Fi g. 2 and 13), and that also on the of the detector device generated signal responsive devices are provided that the reduces the DC voltage applied to the said further electronic switch (70, Fig. 12), when the detector signal reaches the specified value and this electronic switch (70, Fig. 12) blocks, whereby the DC voltage for the series of electronic switches (30, 31, 32, Fig. 2 and 13) is switched off, and that the delay device has a first Contains capacitor (85, Fig. 12), which is connected to the further electronic switch (70, Fig. 12) lying DC voltage after a delay time that is smaller than the specified Delay time is, turns on again, as well as a second capacitor (74, Fig. 12), the after the predetermined delay time, a control signal of the control electrode of the aforesaid further electronic switch (70, Fig. 12) and makes it conductive again. Considered publications:
German interpretation document No. 1082625. For this purpose 2 sheets of drawings 509 720/383 10.65 ® Bundesdruckerei Berlin