CN1472991A - High frequency heating device - Google Patents

Abstract

The present invention provides a high-frequency heating apparatus, comprising: the filter inductor is connected with a positive end of a direct current power supply; the middle tap transformer comprises a middle tap end, a first end and a second end, wherein the middle tap end is connected with the other end of the filter inductor; one end of the filter capacitor is connected with the first end of the middle tap transformer, and the other end of the filter capacitor is connected with a negative end of the direct current power supply; a first switch connected in series with the second terminal of the center-tapped transformer and also connected to the negative terminal of the DC power supply; the series circuit comprises a second switch and a second capacitor which are connected in series and is connected with the middle tap transformer; a first capacitor connected to the middle tap transformer; a rectifier connected to a secondary side coil of the center-tapped transformer; and a magnetron connected to the rectifying device, wherein the first capacitor, the second capacitor and the center-tapped transformer form a resonant circuit.

Description

High frequency heating device

(1) Technical field

The invention relates to a high-frequency heating device applied to a magnetron.

(2) Background technology

FIG. 1 is a schematic diagram of a conventional magnetron circuit. As shown in Figure 1, a magnetron is a vacuum tube used to generate microwaves. The condition for its normal operation is: when the cathode temperature exceeds 2100K (absolute temperature), a negative high voltage is applied between the cathode and the anode (thousands of volts). However, different magnetrons have different operating voltages, but their voltage and current characteristic curves are basically similar, as shown in Figure 2. When the voltage between the cathode and the anode reaches the operating voltage, the magnetron generates a microwave, and the voltage between the cathode and the anode is clamped near the operating voltage. At this time, the characteristics of the magnetron are quite In a regulator tube.

FIG. 3 is a schematic circuit diagram of a conventional clamped forward-flyback converter. As shown in FIG. 3 , the working principle of the clamping forward (forward)-flyback (flyback) converter 100 is as follows: the driving signal of a main switch 101 and an auxiliary switch 102 is a complementary signal, and the circuit converter A capacitor 103 is used to clamp and control the primary side voltage of a transformer 104 , and also to reset the magnetic field of the transformer 104 .

Please refer to FIG. 4 , which is a schematic diagram of circuit waveforms of a conventional clamp-type forward-flyback converter. Wherein, V _GS1 is the driving signal of the main switch 101 , V _GS2 is the driving signal of the auxiliary switch 102 , I ₁ represents the conduction current of the main switch 101 , and I ₂ represents the conduction current of the auxiliary switch 102 . The advantages are: both the main switch 101 and the auxiliary switch 102 are turned on at zero voltage (ZVS); the rectifier diode on the secondary side is turned off at zero current (ZCS), and there is no reverse recovery problem. The disadvantages of the known clamping forward (forward)-flyback (flyback) converter are: (1) because the capacitance value of the filter capacitor 105 is small, in order to reduce the current ripple of a filter inductor 106 ( current ripple), the inductance value of the filter inductor 106 must be increased. (2) There is a large DC bias in the magnetic flux of a high-voltage transformer. In order to prevent the transformer from being saturated, the air gap of the transformer core must be increased, thus increasing the loss of the transformer.

In order to clarify the problem of the DC bias of the transformer, the description is as follows: FIG. 5 is a transformer equivalent circuit diagram of a conventional clamped forward-flyback converter. 107 is the magnetizing inductance corresponding to the primary side of the transformer 104 . Since the capacitors 108 and 109 cannot have a DC component flowing through them, no DC component flows through the secondary side of the transformer 104, the RMS current in the exciting inductor 106 is equal to I _in , and the peak value of the exciting current is I _m . Assuming the power factor of this power supply is 1, then:

(1)

i _in =I _m sin ωt $P_{\mathrm{in}}=V_{\mathrm{in}}{I}_{\mathrm{in}}=P_{\mathrm{out}}/\mathrm{\η}---\left(2\right)$ $I_m=\sqrt{2}{I}_{\mathrm{in}}=\sqrt{2}{P}_{\mathrm{out}}/V_{\mathrm{in}}\mathrm{\η}---\left(3\right)$ $I_{m\max }=\sqrt{2}{I}_{\mathrm{in}\max }=\sqrt{2}{P}_{\mathrm{out}\max }/V_{\mathrm{in}\min }\mathrm{\η}---\left(4\right)$

Among them, i _in represents the input current, P _in represents the average input power, V _in represents the root mean square value of the input voltage, I _in represents the root mean square value of the input current, P _out represents the average output power, and η represents the efficiency of the transformer.

Also, the peak value of the DC bias value of the magnetomotive force in the transformer core is:

U _{dc max} = N _{Im max} (5)

Among them, N represents the number of turns of the primary side winding.

However, the DC bias peak value of the magnetomotive force will be very large at full load and low input voltage, resulting in low utilization of the transformer core, so the transformer core must have a large air gap, thus increasing the losses of the transformer.

(3) Contents of the invention

The main purpose of the present invention is to provide a magnetron (magnetron) high-frequency heating device to reduce the DC bias in the magnetic flux of a high-voltage transformer and prevent the transformer from being saturated.

Another object of the present invention is to provide a magnetron high-frequency heating device to solve the problem of input current ripple in the circuit and the bias value of the transformer, and improve the power factor (PowerFactor) and efficiency.

Another object of the present invention is to provide a magnetron high-frequency heating device to improve the utilization rate of the high-voltage transformer magnetic core in the high-frequency heating device.

Another object of the present invention is to provide a magnetron high-frequency heating device, the output rectifier diode of the high-frequency heating device can realize zero current switching (ZCS), to eliminate the reverse recovery problem of the diode, so that the device can obtain higher efficiency and power density.

According to the above idea, the high-frequency heating device of the present invention includes: a filter inductor connected to a positive terminal of a DC power supply; a center tapped transformer including a center tap terminal, a first terminal and a second terminal, the The center tap end is connected to the other end of the filter inductor; a filter capacitor, one end of which is connected to the first end of the center tap transformer, and the other end is connected to a negative end of the DC power supply; a first switch is connected in series to the center tap transformer The second end of the DC power supply is also connected to the negative end of the DC power supply; a series circuit includes a second switch and a second capacitor connected in series, connected to the center tap transformer; a first capacitor connected to the center tap transformer; A rectifying device connected to a secondary side coil of the center-tapped transformer; and a magnetron connected to the rectifying device, wherein the first capacitor, the second capacitor and the center-tapped transformer form a resonant circuit.

According to the above idea, wherein the first capacitor is connected in parallel with the center-tapped transformer.

According to the above concept, the first capacitor is connected in parallel with the first terminal and the second terminal of the center-tapped transformer.

According to the above idea, wherein the first capacitor is connected in series with the center-tapped transformer, and at the same time is connected in parallel with the first switch.

According to the above concept, wherein the first capacitor is connected in series with the second terminal of the center-tapped transformer.

According to the above idea, wherein the series circuit is connected in parallel with the center-tapped transformer.

According to the above idea, wherein the series circuit is connected in parallel with the first terminal and the second terminal of the center-tapped transformer.

According to the above concept, wherein the series circuit is connected in series with the center-tapped transformers.

According to the above concept, wherein the series circuit is connected in series with the second end of the center-tapped transformer.

According to the above concept, the rectification device is one of the following devices: full wave voltage doubler rectification; half wave voltage doubler rectification; full wave rectification ( full wave rectification); full bridge rectification device (full bridge rectification).

According to the above idea, the transformer is a transformer with leakage inductance.

According to the above concept, the first capacitor is a bulk capacitor of the first switch.

In order to further illustrate the purpose, structural features and effects of the present invention, the present invention will be described in detail below in conjunction with the accompanying drawings.

(4) Description of drawings

Fig. 1 is the circuit diagram of conventional magnetron (magnetron);

Fig. 2 is a schematic diagram of voltage-current characteristics of a conventional magnetron;

FIG. 3 is a schematic circuit diagram of a conventional clamped forward-flyback converter;

Fig. 4 is a circuit waveform diagram of a conventional clamped forward-flyback converter;

FIG. 5 is a transformer equivalent circuit of a conventional clamped forward-flyback converter;

6 is a schematic circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) in the first preferred embodiment of the present invention;

7 is a schematic diagram of an equivalent circuit of a current-mode regulating transformer DC-DC converter (DC/DC Converter) in the first preferred embodiment of the present invention;

FIG. 8 is a schematic diagram of an equivalent circuit of a secondary-side rectification circuit of the transformer in FIG. 7;

Fig. 9 is a schematic diagram of an equivalent circuit simplified according to Fig. 7 and Fig. 8;

10 is a schematic diagram of a circuit waveform of a current-mode regulating transformer DC-DC converter according to a first preferred embodiment of the present invention;

Fig. 11 (a) ~ (g) is the schematic diagram of the action of the DC-DC converter circuit of the current-mode regulating transformer of the first preferred embodiment of the present invention;

Fig. 12 is an equivalent circuit of a current-mode regulating transformer DC-DC converter transformer in the first preferred embodiment of the present invention;

Fig. 13 is the equivalent analysis circuit of the first preferred embodiment of the present invention;

14 is a schematic diagram of the node N1 voltage and the filter capacitor voltage V _c1 voltage waveform of the current-mode regulating transformer DC-DC converter in the first preferred embodiment of the present invention;

Fig. 15 is a schematic circuit diagram of the inverter part and the rectification part of the current-mode regulating transformer DC-DC converter according to the first preferred embodiment of the present invention;

16 is a partial circuit schematic diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to a second preferred embodiment of the present invention;

17 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to a third preferred embodiment of the present invention;

18 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) in a fourth preferred embodiment of the present invention;

19 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to a fifth preferred embodiment of the present invention;

Fig. 20 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to a sixth preferred embodiment of the present invention;

21 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to a seventh preferred embodiment of the present invention; and

FIG. 22 is a partial circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to an eighth preferred embodiment of the present invention.

(5) specific implementation

Please refer to FIG. 6 , which is a schematic circuit diagram of a current-mode regulating transformer DC-DC converter (DC/DC Converter) in the first preferred embodiment of the present invention, that is, a CTT (Current Tapping Transformer) DC/DC converter. As shown in Figure 6, a high-frequency heating device 200 includes: a filter inductor 201, a center tapped transformer 202, a filter capacitor 203, a first switch 204, a series circuit including a second switch 205 connected in series and a second capacitor 206 , a first capacitor 207 , a rectifier 208 and a magnetron 209 . The filter inductor 201 is connected to a positive terminal (+) of a DC power supply V _dc . The center tap transformer 202 includes a center tap end, a first end and a second end, and the center tap end is connected to the other end of the filter inductor 201 . One end of the filter capacitor 203 is connected to the first end of the center-tapped transformer 202, and the other end is connected to a negative terminal (-) of the DC power supply V _dc . The first switch 204 is connected in series with the second terminal of the center-tapped transformer 202 and also connected with the negative terminal (-) of the DC power supply V _dc . The series circuit is connected in parallel with the center-tapped transformer 202 . The first capacitor 203 is connected in parallel with the center-tapped transformer 202 . The rectifying device is connected to a secondary side coil of the center-tapped transformer. And, the magnetron 209 is connected to the rectifying device 208 , wherein the first capacitor 207 , the second capacitor 206 and the center-tap transformer 202 form a resonant circuit.

The rectification device 208 can be a full wave voltage doubler rectification device (full wave voltage doubler rectification). The full-wave voltage doubler rectifier is composed of two diodes 210, 211 and two capacitors 212, 213.

For the power supply of microwave ovens, the DC-DC converter with current mode output has no reverse recovery problem of the rectifier diode and is suitable for high voltage output. The present invention applies this circuit structure to a current-type output DC-DC converter. The DC-DC converter has all the advantages of the circuit in Figure 3, and also solves the problem of input current ripple and transformer bias in the circuit in Figure 3. It can be proved that its power factor (PowerFactor) and efficiency are higher than the former.

Please refer to FIG. 7 , which is a schematic diagram of an equivalent circuit of a current-mode regulating transformer DC-DC converter (DC/DC Converter) according to the first preferred embodiment of the present invention. As shown in Figure 7, in order to facilitate the analysis of the working principle of the circuit, the circuit is simplified. In one switching cycle, the following assumptions can be made: (1) because the filter inductor 201 is relatively large, it can be equivalent to a current source 214; (2) because the clamping capacitor 206 is relatively large, it can be equivalent to a voltage source V _C2 ; (3) When the magnetron is in operation, its characteristic is equivalent to a voltage source V _m ; (4) In the transformer 202, since the DC component cannot flow through the winding n1, the secondary winding has no DC component , so the input direct current all flows through the winding n2, and the direct current component can be equivalent to a current source I _m2 whose size is I _in ; (5) the power of the cathode heating part of the magnetron and the working power of the magnetron It is relatively small, so it is ignored in the analysis, and only the secondary side winding n3 is analyzed. Wherein L _S1 and L _S2 are the leakage inductances of the transformer winding _n1 and winding _n2 respectively, L _m1 and L _m2 are the excitation inductances of the transformer winding _n1 and winding _n2 respectively; the first capacitor 207 can be equivalent to It is connected in parallel with both ends of the main switch 204; the main switch 204 and the auxiliary switch 205 have parasitic diodes D ₁ and D ₂ respectively. The transformer 202 is a high-voltage transformer. In order to ensure good insulation, the winding method of the winding is generally to separate the primary side from the secondary side, resulting in a large leakage inductance, but the coupling between the primary side and the secondary side windings is better. Leakage inductance is ignored.

The equivalent circuit shown in FIG. 7 is further simplified, and the simplification of the rectification circuit on the secondary side of the transformer 202 is shown in FIG. 8 . Fig. 8A shows the working process when the current in winding n3 is in different directions, and the result is equivalent to the circuit in Fig. 8B.

Combining the schematic diagram of the equivalent circuit shown in FIG. 8 , the schematic diagram of the equivalent circuit shown in FIG. 9 can be obtained after simplified processing.

Please refer to FIG. 10 , which is a schematic diagram of a circuit waveform of a current-mode regulating transformer DC-DC converter (DC/DC Converter) in the first preferred embodiment of the present invention, wherein V _p1 is the terminal voltage of the primary side winding n ₁ , V _p2 is the terminal voltage of the primary winding n ₂ , i _LM1 is the excitation current of the primary winding n ₁ , i _LM2 is the excitation current of the primary winding n ₂ , V _DS1 is the cross voltage of the main switch 101, V _DS2 is the For the voltage across the auxiliary switch 102, i _DS1 is the current of the main switch 101, i _DS2 is the current of the auxiliary switch 102, i _S is the current of the secondary winding, and V _S is the terminal voltage of the secondary winding. As shown in FIG. 10 , the main switch 204 and the auxiliary switch 205 are cross-complementarily conducted, and the DC-DC converter can be divided into 7 operation modes in one working cycle.

First, a steady-state analysis of the circuit is performed. For the loop: DC power supply V _dc (+)-filter inductance 105-primary side winding n ₁ -filter capacitor 203-DC power supply V _dc (-), because there can be no DC voltage on the filter inductance 105 and the primary side winding n ₁ component, so the DC voltage V _C1 on the filter capacitor 203 is equal to the input voltage V _dc (the rectified voltage is a half-sine wave of 120 Hz). The capacitance value of the filter capacitor 203 is relatively small, so V _C1 is actually a half sine wave with a frequency of 120 Hz. Since a high-frequency inverter part is connected behind it, it has a relatively large voltage ripple.

For the loop: DC power supply V _dc (+)-filter inductor 105-secondary side winding n ₂ -main switch 204-DC power supply V _dc (-), assuming that the duty ratio (dutyratio) of the main switch 204 is D _Q1 , Since the volt-second (Volt-Sec) on the magnetic element filter inductance 105-secondary side winding n ₂ needs to be balanced, the voltage of the main switch 204 during the cut-off period is the relationship between the voltage V _C2 on the second capacitor 206 and the input voltage It is the relationship between the output voltage and the input voltage in a boost circuit (boost), that is: ${\mathrm{<figure-callout\; id="2"\; label="V\; C"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">V</figure-callout>}}_C=\frac{V_{\mathrm{dc}}}{1-{\mathrm{D.}}_{Q1}}---\left(6\right)$ Analyzing the node N1, it can be concluded that the DC component I _m2 of the transformer is equal to I _in . Because the two windings of n1 and n2 are wound in the same magnetic circuit, and the voltages of the two windings are in phase. so:

I _Lm1 =I _Lm2 -I _m2 (7)

I _n1 =I _n2 (8)

Please refer to FIG. 11(a)~(g) which are schematic diagrams of the circuit operation of the DC-DC converter circuit of the current-source regulating transformer in the first preferred embodiment of the present invention. Its main working principle is described as follows:

Mode 1 (t ₀ -t ₁ ): as shown in Fig. 11(a)A, the main switch 204 is turned on, the auxiliary switch 205 is turned off, and the energy in the filter capacitor 203 starts to transfer to the secondary side (ie i _Ls >I _in ). The input current I _in is stored in the transformer as magnetic energy (which lays the foundation for continuing to transfer energy to the secondary side after the main switch 204 is turned off). The equivalent circuit at this time is shown in Figure 11(a)B, and the following equation can be obtained after analysis:

i _Ls ≥ I _m2 = I _in (9) $i_{\mathrm{L\; m}1}=i_{\mathrm{L\; m}1t0}+\frac{{\&Integral;}_{t0}^{t1}{\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c\mathrm{<figure-callout\; id="1"\; label="dt\; L\; m"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">dt</figure-callout>}}{L_m}---\left(10\right)$ ${\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c=u_{c1t0}-\frac{{\&Integral;}_{t0}^{t1}(i_{\mathrm{the\; s}}^{\&prime;}+i_{\mathrm{L\; m}1})\mathrm{dt}}{C_1}---\left(11\right)$ $i_{\mathrm{the\; s}}^{\&prime;}=i_{\mathrm{st}0}^{\&prime;}+\frac{(u_{c1t0}-u_{(c5+c6)t0}^{\&prime;})}{\sqrt{\frac{L_{\mathrm{the\; <figure-callout\; id="1"\; label="s\; C"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">s</figure-callout>}}}{C}}}\sin {\mathrm{\&omega;}}_{0}t---\left(12\right)$ ${\mathrm{\&omega;}}_0=1/2\mathrm{\&pi;}\sqrt{L_{\mathrm{the\; s}}(\mathrm{<figure-callout\; id="1"\; label="C"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">C</figure-callout>}1//{(C5+C6)}^{\&prime;})}---\left(13\right)$

Among them, C ₁ is the capacitance value of the filter capacitor 203, C ₅ is the capacitance value of the capacitor 212, C ₆ is the capacitance value of the capacitor 213, u _c1 is the terminal voltage of the filter capacitor 203, and i' _s is the conversion from the secondary side to the primary side current (namely: the difference between the current flowing through winding n1 and the current i _Lm1 ), (C5+C6)' is the capacitance value converted from the secondary side capacitors 212 and 213 to the primary side of the transformer, C1//(C5+C6 )' is the capacitance value of the filter capacitor and the parallel connection of capacitors 212 and 213, u' _(C5+C6) is the voltage value converted from the secondary side voltage of the transformer to the primary side, and L _s is the sum of leakage inductance L _s1 and L _s2 .

Mode 2 (t ₁ -t ₂ ): as shown in FIG. 11(b)A, the main switch 204 is turned off, and the auxiliary switch 205 is also turned off. Since the current in the inductor Ls cannot change abruptly, it continues to flow to the first capacitor 207 Charging until the voltage on the first capacitor 207 reaches the value of the clamping voltage V _c2 . In this mode of operation, the primary side continues to transfer energy to the secondary side. The magnetic energy stored in the transformer reaches a maximum. In this mode of operation, the time is very short so it can be assumed that: the excitation current i _Lm (=i _Lm1 +i _Lm2 ) does not change, the voltage of the filter capacitor 203, the secondary side capacitors 212 and 213 (C5+C6)' does not change (Because the values of the two capacitors are larger than the value of the first capacitor 207, so this assumption is reasonable), the voltage on the first capacitor 207 changes from zero to positive V _c2 +u _c1t1 , it can be assumed that its effect on the current i The effect of _s is equivalent to (V _c2 +u _c1t1 )/2 The equivalent circuit at this time is shown in Figure 11(b)B, and the following equation can be obtained. Right now:

                       

i _Lm1t1 = i _Lm1t2

(15)

u _c1 =u _c1t1 $i_{\mathrm{the\; s}}^{\&prime;}=i_{\mathrm{<figure-callout\; id="1"\; label="st"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">st</figure-callout>}1}^{\&prime;}-\frac{(u_{(C5+C6)\mathrm{<figure-callout\; id="1"\; label="t"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">t</figure-callout>}1}^{\&prime;}+\frac{1}{2}{V}_{c2}-\frac{1}{2}{u}_{c1t1})t}{L_{\mathrm{the\; s}}}---\left(16\right)$ $T_{12}\&ap;\frac{({\mathrm{<figure-callout\; id="2"\; label="V\; c"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">V</figure-callout>}}_c+u_{c1t1})C_3}{I_{m2}+\frac{i_{\mathrm{<figure-callout\; id="1"\; label="st"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">st</figure-callout>}1}^{\&prime;}+i_{\mathrm{<figure-callout\; id="2"\; label="st"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">st</figure-callout>}2}^{\&prime;}}{2}}---\left(17\right)$

Mode 3 (t ₂ -t ₃ ): as shown in FIG. 11(c)A, when the first capacitor 207 is charged to a certain value, the parasitic diode of the main switch 204 is turned on, which is the ZVS of the auxiliary switch 205 Conduction creates the conditions. Because the energy in the leakage inductance is large (the current in the inductor _L5 is still greater than the excitation current at this time), the energy is still transmitted to the secondary side. Since this time period is short, it can be assumed that the voltage of the capacitor (212+213) does not change. At this time, its equivalent circuit is shown in Fig. 11(c)B. The following equation can be obtained: $i_{\mathrm{L\; m}1}=i_{\mathrm{L\; m}1t2}-\frac{V_{C2}\mathrm{<figure-callout\; id="1"\; label="t\; L\; m"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">t</figure-callout>}}{L_m}---\left(18\right)$ ${\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c=u_{c1t2}-\frac{I_{m2}t}{C_1}---\left(19\right)$ $i_{\mathrm{the\; s}}^{\&prime;}\&ap;i_{\mathrm{<figure-callout\; id="2"\; label="st"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">st</figure-callout>}2}^{\&prime;}\cos {\mathrm{\&omega;}}_{1}t+\frac{V_{C2}-u_{(c5+c6)}^{\&prime;}}{\sqrt{L_{\mathrm{the\; s}}/{(C5+C6)}^{\&prime;}}}\sin {\mathrm{\&omega;}}_{1}t---\left(20\right)$ ${\mathrm{\&omega;}}_1=\frac{1}{2\mathrm{\&pi;}\sqrt{L_{\mathrm{the\; s}}{(C5+C6)}^{\&prime;}}}---\left(\mathrm{twenty\; one}\right)$

Mode 4 (t ₃ -t ₄ ): As shown in Figure 11(d), at time t ₃ the current in the inductor L _s is less than the excitation current, and the secondary side current decreases to zero, so the cut-off of the secondary side diode is ZCS cut off. After the commutation is completed, the energy stored in the inductor L _s continues to provide energy to the second capacitor 206 . The equivalent circuit in this mode of operation is shown in Fig. 11(d)B. The following equations can be obtained: $i_{\mathrm{lm}1}=i_{\mathrm{L\; m}1t3}-\frac{V_{C2}\mathrm{<figure-callout\; id="1"\; label="t\; L\; m"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">t</figure-callout>}}{L_m}---\left(\mathrm{twenty\; two}\right)$ ${\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c=u_{c1t3}+\frac{I_{m}t}{C_1}---\left(\mathrm{twenty\; three}\right)$ $i_{\mathrm{the\; s}}^{\&prime;}=\frac{{(C5+C6)}^{\&prime;}}{L_{\mathrm{the\; s}}}{V^2}_{c2}\sin {\mathrm{\&omega;}}_{1}t---\left(\mathrm{twenty\; four}\right)$

Mode 5 (t ₄ -t ₅ ): As shown in Figure 11(e)A, the auxiliary switch 205 is turned off, the current in the inductor L _s cannot change abruptly, and resonates with the first capacitor 207, and starts to discharge the filter capacitor 203 , and its equivalent circuit is shown in Fig. 11(e)B. Therefore, the operating time of this mode is shorter, similar to mode two, the following assumptions can be made: the current i _Lm is constant; The first capacitor 207 is relatively large, so this assumption is reasonable), the voltage on the first capacitor 207 changes from positive V _c2 +u _c1t1 to zero. It can be assumed that its effect on the current i _s is equivalent to -(V _c2 +u _c1t1 )/2. The following equations can be obtained:

(25)

i _Lm1t4 = i _Lm1t5

(26)

u _c1 =u _c1t4 $i_{\mathrm{the\; s}}^{\&prime;}=i_{\mathrm{st}4}^{\&prime;}-\frac{(u_{(C5+C6)}^{\&prime;}-\frac{1}{2}{\mathrm{<figure-callout\; id="2"\; label="V\; c"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">V</figure-callout>}}_c+\frac{1}{2}{u}_{c1t4})t}{L_{\mathrm{the\; s}}}---\left(27\right)$ $T_{45}\&ap;\frac{({\mathrm{<figure-callout\; id="2"\; label="V\; c"\; filenames="A0212784000392.png"\; state="\{\{state\}\}">V</figure-callout>}}_c+u_{c1t4})C_3}{I_{m2}+\frac{i_{\mathrm{st}4}^{\&prime;}+i_{\mathrm{st}5}^{\&prime;}}{2}}---\left(28\right)$

Mode 6 (t ₆ -t ₇ ): As shown in FIG. 11(f), the body diode of the main switch 204 conducts, creating conditions for it to realize ZVS conduction. The current of the inductor Ls is still greater than the excitation current, so energy is still transferred to the secondary side. At this point the following equations can be obtained: $i_{\mathrm{L\; m}1}=i_{\mathrm{L\; m}1t5}+\frac{{\&Integral;}_{t5}^{t6}{\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c\mathrm{<figure-callout\; id="1"\; label="dt\; L\; m"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">dt</figure-callout>}}{L_m}---\left(29\right)$ ${\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c=u_{c1t5}+\frac{{\&Integral;}_{t5}^{t6}(i_{\mathrm{the\; s}}^{\&prime;}-i_{\mathrm{L\; m}1})\mathrm{dt}}{C_1}---\left(30\right)$ $i_{\mathrm{the\; s}}^{\&prime;}\&ap;i_{\mathrm{st}5}^{\&prime;}\cos {\mathrm{\&omega;}}_{0}t-\frac{V_{C2}-u_{(C5+C6)}^{\&prime;}}{\sqrt{L_{\mathrm{the\; s}}/{\mathrm{<figure-callout\; id="1"\; label="C"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">C</figure-callout>}}_1//{(C5+C6)}^{\&prime;}}}\sin {\mathrm{\&omega;}}_{0}t---\left(31\right)$

Mode 7 (t ₆ -t ₇ ): As shown in Figure 11(g)A, at time t ₆ , the current in the inductor L _s is less than the excitation current, and the secondary side current decreases to zero, so the secondary side diode The cutoff is the ZCS cutoff. After the commutation is completed, the energy stored in the inductor L _s continues to provide energy to the second capacitor 206 . The equivalent circuit in this mode of operation is shown in Fig. 11(g)B. The following equations can be obtained: $i_{\mathrm{L\; m}1}=i_{\mathrm{L\; m}1t6}+\frac{{\&Integral;}_{t6}^{t7}{\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c\mathrm{<figure-callout\; id="1"\; label="dt\; L\; m"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">dt</figure-callout>}}{L_m}---\left(32\right)$ ${\mathrm{<figure-callout\; id="1"\; label="u\; c"\; filenames="A0212784000381.png"\; state="\{\{state\}\}">u</figure-callout>}}_c=u_{c1t6}+{\&Integral;}_{t6}^{t7}(i_{\mathrm{the\; s}}^{\&prime;}+i_{\mathrm{L\; m}1})\mathrm{dt}---\left(33\right)$ $i_{\mathrm{the\; s}}^{\&prime;}=\frac{{(C5+C6)}^{\&prime;}}{L_{\mathrm{the\; s}}}{V^2}_{c2}\sin {\mathrm{\&omega;}}_{1}t---\left(34\right)$ ${\mathrm{\&omega;}}_1=\frac{1}{2\mathrm{\&pi;}\sqrt{L_{\mathrm{the\; s}}{(C5+C6)}^{\&prime;}}}---\left(35\right)$

After mode 7 ends, the circuit returns to mode 1 again.

The following analysis for DC magnetic bias is as follows:

In this circuit, among the two windings of the primary side and the secondary side of the transformer, the winding n1 has no DC magnetic bias, but there is a DC magnetic bias in the winding n2. For the convenience of analysis, the analysis model of transformer 202 is established as shown in FIG. 12 . Wherein L _m1 and L _m2 respectively correspond to the excitation inductance of the primary side windings n ₁ and n ₂ of the transformer 202 . Because capacitors C _a and C _b cannot have DC current components, the DC current component in L _m2 is equal to the input DC current component. Assuming that the power factor of the power supply is 1, then:

i _in =I _m sinωt (36) $P_{\mathrm{in}}=V_{\mathrm{in}}{I}_{\mathrm{in}}=P_{\mathrm{out}}/\mathrm{\&eta;}---\left(37\right)$ $I_m=\sqrt{2}{I}_{\mathrm{in}}=\sqrt{2}{P}_{\mathrm{out}}/V_{\mathrm{in}}\mathrm{\&eta;}---\left(38\right)$ $I_{m\max }=\sqrt{2}{I}_{\mathrm{in}\max }=\sqrt{2}{P}_{\mathrm{out}\max }/V_{\mathrm{in}\min }\mathrm{\&eta;}---\left(39\right)$ The peak DC bias value of the magnetomotive force in the transformer core is:

U _{dc max} = n2I _{m max} (40)

The peak value of the DC bias value of the magnetomotive force in the transformer core of the circuit shown in Figure 3 is:

U _{dc max} =NI _{m max} =(n2+n1)I _{m max} (41)

Compared with the DC bias peak value of the magnetomotive force in the two transformer magnetic cores, the present invention is smaller (determined by design), which improves the magnetic core utilization rate of the transformer, so the air gap of the transformer magnetic core can be reduced, thereby reducing Reduce the loss of the transformer.

The analysis of the input current ripple is as follows:

In order to facilitate the analysis, the analysis model shown in Figure 13 is established. Wherein the voltage source V1 is the voltage on the winding n1 of the transformer. According to the previous analysis of the magnetic circuit, it is known that when the main switch 204 is turned on, the voltage of the node N1 is equivalent to adding a negative V _c1 on the basis of the voltage of the filter capacitor 203. When the main switch 204 is turned off, the node N1 The voltage of N1 is equivalent to superimposing a positive V _c1 on the basis of the voltage of the filter capacitor 203 , as shown in FIG. 13 . It can be seen from Figure 14 that if the winding n1 is correctly selected, a bimodal voltage ripple waveform can be obtained at the node N1, and its effect is equivalent to doubling the frequency of the subsequent high-frequency inverter. Thus greatly reducing the input current ripple. The input power factor of the power supply is improved.

According to the above analysis, it can be known that the present invention has the following advantages:

(1) The input current is a continuous conduction mode, and since the filter inductor is connected to the filter capacitor through the n1 winding, the current ripple is smaller than that of the circuit shown in Figure 3 (under the same ripple condition, the input filter inductor value can be reduced), so the power factor (PF) is higher.

(2) There is no DC bias in the winding n1, and the DC component only passes through the winding n2, so the bias magnetomotive force of the magnetic core is lower than that in Figure 3, which improves the utilization rate of the high-voltage transformer magnetic core.

(3) Both the main power element and the auxiliary power element can realize zero-voltage switching (ZVS) when they are turned on, and are buffered by the first capacitor 207 when they are turned off, so that the switching loss is small. The output rectifier diode is capable of zero-current switching (ZCS), which eliminates the diode's reverse recovery problem, allowing the device to achieve high efficiency and power density.

However, the above-mentioned analyzes all take the circuit diagram shown in Figure 6 as an example, and the examples of equal changes are as follows. For the convenience of explanation, the circuit diagram shown in Figure 6 is divided into two parts, as shown in Figure 15 : The first part is the inverter part; the second part is the rectification part.

(1) Equal change embodiment of the first part:

Second preferred embodiment: the first capacitor 207 is connected in parallel to the primary side of the transformer, which is equivalent to connecting the first capacitor 207 in parallel to both ends of the main switch 204 or using the bulk capacitance of the main switch 204 to replace the capacitor. As shown in Figure 16.

Third preferred embodiment: the series circuit of the second capacitor 206 and the auxiliary switch 205 is connected in parallel to the primary side of the transformer for current absorption and resetting of the transformer, which is equivalent to connecting the second capacitor 206 with the auxiliary switch The series circuit of 205 is connected in parallel with both ends of the main switch 204 . As shown in Figure 17. The auxiliary switch 205 can be driven by a common ground if a P-channel IGBT or MOS is used.

The fourth preferred embodiment: the above two equivalent principles are combined: the first capacitor 207 is connected in parallel to both ends of the main switch 204 or the bulk capacitance of the main switch 204 is used to replace the capacitor; the second capacitor 206 A series circuit with the auxiliary switch 205 is connected across the main switch 204 in parallel. As shown in Figure 18.

(2) Equal change embodiment of the second part

The fifth preferred embodiment: the second part shown in Figure 16 is a full-wave voltage doubler rectifier, if the second part is replaced by a half-wave voltage doubler rectifier, it is also an all-change embodiment of the present invention, as shown in Figure 19 Show.

The sixth preferred embodiment: the second part shown in FIG. 16 is a full-wave voltage doubler rectifier. If the second part is replaced by a full-bridge rectifier, it is also an equivalent embodiment of the present invention, as shown in FIG. 20 .

The seventh preferred embodiment: the second part shown in Figure 16 is full-wave voltage doubler rectification, if the second part is replaced by full-wave rectification, it is also an equivalent embodiment of the present invention, as shown in Figure 21.

The eighth preferred embodiment: the second part shown in Figure 16 is a full-wave voltage doubler rectification, if another half-wave voltage doubler rectifier replaces the second part, it is also an equal change of the present invention

Embodiment, as shown in Figure 22.

Based on the above, the present invention can provide a magnetron (magnetron) high-frequency heating device, which reduces the DC bias in the magnetic flux of the high-voltage transformer and prevents the transformer from being saturated.

Of course, those of ordinary skill in the art should recognize that the above embodiments are only used to illustrate the present invention, rather than as a limitation to the present invention, as long as within the scope of the spirit of the present invention, the implementation of the above Changes and modifications of the examples will fall within the scope of the claims of the present invention.

Claims (12)

1. a thermatron is characterized in that, comprising:

One filter inductance connects an anode of a direct current power supply;

One interphase reactor transformer comprises a centre tap end, one first end and one second end, and this centre tap end connects the other end of this filter inductance;

One filter capacitor, one end connect this first end of this interphase reactor transformer, and the other end connects a negative terminal of this DC power supply;

One first switch, this second end of this interphase reactor transformer that is connected in series also connects this negative terminal of this DC power supply;

One series connection circuit comprises a second switch and one second electric capacity of serial connection, connects this interphase reactor transformer;

One first electric capacity connects this interphase reactor transformer;

One rectifying device connects a second siding ring of this interphase reactor transformer; And

One magnetron connects this rectifying device, and wherein this first electric capacity, this second electric capacity and this interphase reactor transformer form a resonance circuit.

2. thermatron as claimed in claim 1 is characterized in that, this first electric capacity is this interphase reactor transformer that is connected in parallel.

3. thermatron as claimed in claim 2 is characterized in that, this first electric capacity is be connected in parallel this first end and this second end of this interphase reactor transformer.

4. thermatron as claimed in claim 1 is characterized in that, this first electric capacity is this interphase reactor transformer that is connected in series, and this first switch simultaneously is connected in parallel.

5. thermatron as claimed in claim 4 is characterized in that, this first electric capacity is this second end of this interphase reactor transformer of being connected in series.

6. thermatron as claimed in claim 4 is characterized in that, this first electric capacity is the body capacitance for this first switch.

7. thermatron as claimed in claim 1 is characterized in that, this series circuit is this interphase reactor transformer that is connected in parallel.

8. thermatron as claimed in claim 7 is characterized in that, this series circuit is be connected in parallel this first end and this second end of this interphase reactor transformer.

9. thermatron as claimed in claim 1 is characterized in that, this series circuit is this interphase reactor transformer that is connected in series.

10. thermatron as claimed in claim 9 is characterized in that, this series circuit is this second end of this interphase reactor transformer of being connected in series.

11. thermatron as claimed in claim 1 is characterized in that, this rectifying device is to be one of following apparatus:

(1) full wave and voltage doubling rectifying device;

(2) half-wave voltage multiplying rectifier device;

(3) full-wave fairing attachment;

(4) full-bridge rectification device.

12. thermatron as claimed in claim 1 is characterized in that, this transformer is the transformer that has leakage inductance for.

Images