MXPA99001860A - Dual loop frequency and power control - Google Patents

Abstract

The invention is an improved phacoemulsification probe drive circuit for supplying electrical power to an ultrasonic transducer. The drive circuit has a power control loop (1412) and a frequency control loop (1413). The power control loop (1412) has a variable gain amplifier (1416) whose output is an input to a power amplifier (1417). After the power amplifier amplifies power, power is delivered to atransformer (1436) and, thereafter, to a transducer (1439). The voltage and current applied to the primary of the transformer are sensed to generate a signal proportional to the power (real or apparent), and the result is compared against a power command originating from a foot pedal. Once compared, the result of this comparison is sent to a first controller which acts upon the information by sending a corrective signal to the variable gain amplifier.

Description

FREQUENCY CONTROL AND DOUBLE LOOP POWER Field of the Invention This invention relates to phacoemulsification devices and, more particularly, to a method for controlling a phacoemulsification device.

Previous Technique Ultrasonic probes have traditionally been used for phacoemulsification, namely, for breaking cataracts in the eye and for aspirating pieces of altered tissue. These ultrasonic probes must be activated carefully for proper operation. The operation of the ultrasonic probe at its resonant frequency takes advantage of the resonant characteristics of the ultrasonic transducer. Resonance is defined as the phenomenon where a system is driven in or near one of its natural modes. Accordingly, the prior art has focused on how to determine the resonant frequency of a transducer. Theoretically, this problem has been solved. A typical way of determining the resonant frequency of an ultrasonic transducer is to compare the phase angle between the shape of the wave of the voltage applied to the ultrasonic transducer and the shape of the wave of the current emitted by the transducer. When voltage is applied to a circuit, current will flow through the circuit. When the voltage and current waveforms are observed for a particular circuit, the shape of the current wave will be delayed behind the waveform of the voltage if the circuit is inductive, and the shape of the voltage wave will be delayed behind the wave shape of the current if the circuit 'is capacitive. The time difference between the points when the shape of the wave of the current and the shape of the stress wave intersect the zero axis is measured in terms of trigonometry by the phase angle f. For purely resistive circuits, f is equal to zero and the voltage in the waveform of the current is said to be in phase. For purely inductive circuits, f equals 90 ° and for purely capacitive circuits, f equals -90 ° and the voltage in the waveform of the current is said to be out of phase. The presence of an inductive or capacitive reactance component in a load impedance will decrease the power supply efficiency of the system, since only the resistive components can actually dissipate the power.

For circuits that contain the three elements, resistors, inductors and capacitors, there will be some frequencies where the total impedance of the circuit will appear purely resistive, although the circuit contains reactive elements, that is, the resistive elements plus the imaginary component caused by the presence of inductive and capacitive elements. These frequencies are at or near the resonance and / or anti-resonance frequencies. Therefore, in theory, a method to determine the resonant frequencies of certain types of complex circuits is to apply an alternating voltage to the circuit and to vary the frequency until the phase angle f between voltage and current is zero. The frequencies where this condition occurs are the actual resonance frequencies of that particular circuit. The resonant frequency is that frequency or frequencies at which the circuit response (ie, admittance) is locally a maximum, and the anti-resonant frequency is that frequency or frequencies at which the response reaches a local minimum. When a circuit that has both resistive and reactive components is operated, it is important to know the value of the phase angle f since the power supplied to a load is given by the following equation: Power = VI eos (f) where V is the voltage drop across the load impedance; I is the series current flowing through the load impedance; and cosine p i is the power factor of the circuit. Clearly, for a pass angle equal to zero, cosine (0) is equal to 1 and the power transfer from the source to the circuit is at the maximum. This situation exists where there is a purely resistive load. When these theoretical principles have been applied, problems have been found. Specifically, as the environmental conditions such as temperature, time, etc., change, the characteristics of the probe change. These changes are reflected as changes in the values of the various resistive and reactive components of the electric ultrasonic probe model of FIG. 1. In other words, since the environmental factors change, the mechanical resonant frequency of the ultrasonic probe also changes. To solve this problem, there has been an address in the prior art to provide a phase locked circuit to ensure that the system phase angle, f, is zero, such as for example in U.S. Patent Nos. 5,446 .416; 5,210,509; 5,097,219; 5,072,195; 4,973,876; 4,484,154; and 4,114,110. However, the load on the transducer will have a dampening effect on the vibrations of the transducer. In other words, the load can dampen the vibrations of the transducer. When this condition occurs, the resonant frequency may change and the phase angle f will be larger than zero and the power transfer will no longer be optimal. Therefore, unless provisions are made in the circuit to alter the phase angle f, the optimal power transfer can not be achieved. Accordingly, methods other than the phase angle lock f have been explored such as using a tunable inductor in a control system to cancel the capacitive reactants of the load impedance presented by the ultrasonic transducer, as described in the Patent Nos. United States No. 4,970,656; and 4,954,960. Alternatively, the use of ultrasonic transducer admittance as the tuning parameter instead of the phase angle has also been explored in U.S. Patent No. 5,431,664. The approach to this problem from a purely power output point of view has also been explored in U.S. Patent No. 5,331,951, in which the actual electrical power supplied to the drive circuit is examined and the voltage of supply is varied after comparing the electric power supplied with the desired power level of the transducer. Tangentially, this patent also addresses a way to substantially reduce to the minimum the power consumption of the power amplifier by providing a servo regulator to supply voltage to the amplifier. In yet another method, phase-regulated power and frequency control are used, such as in U.S. Patent No. 4,849. 872. In it the initial resonance frequency of the ultrasonic transducer is determined and a capacitive phase angle between the waveform of the voltage and the waveform of the current is introduced and maintained so that, by control phase of the phase control circuit, the oscillator drive frequency is reduced with respect to the series resonance frequency of the transducer. The phase angle is typically maintained as a non-zero constant. Similarly, in U.S. Patent No. 4,888,565, a power control feedback loop is used to monitor the output signal and a frequency control feedback circuit to provide maximum current. This procedure is based on maintaining the current of the network constant. An electrical model of a phacoemulsification probe • Ultrasonic in the proximity of resonance is provided in figure 1. The model has a voltage source 1401 connected to a capacitor 1130 of picofarad 1402 connected in parallel to an RLC circuit of series 1403, where the resistor is 220 ohms, the inductor is 1708 henris, and the capacitor is 18 picofarads. When the apparent power resulting from the electric model is examined, the graphs of Figures 2 and 3 are obtained. As seen in these figures, the apparent power is' maximum at 28.661 kHz with a phase angle of about -42 degrees. This is expected due to the parallel capacitance in the RLC 1403 circuit. When the real power resulting from the electric model is examined, the graphs of figures 4 and 5 are obtained. As shown in the figures, the real power is maximally correct 28.7 kHz, but the phase angle is approximately -24.5 degrees. When a compensation inductor with a calculated value of 27.21 millihenris is placed in a phantom block 1404 of FIG. 1 to cancel the reactive component of FIG. 1 and the resulting apparent power and actual power information are obtained as in FIG. Figures 6 and 7, the apparent power and the real power are both now maximally correctly at 28.7 kHz with a phase of about -0.5 degrees. Therefore, it can be seen that the inductor in the phantom block 1404 compensates for the parallel capacitance 1402 and makes the circuit look resistive (zero phase) in resonance. From these graphs, it is evident that the actual power provides a more reliable view of the resonance frequency, unless the compensation inductor is added close to resonance. Therefore, the resonant frequency is defined here as the frequency at which the real power reaches a maximum (local). However, the apparent power can be used to determine the resonance frequency if the parallel capacitance is compensated at resonance. The apparent power provides a resonant frequency approximation (frequency at which the local maximum occurs) if a compensation inductor compensates for the parallel capacitance 1402 close to the resonance. Therefore, there is a need in the art to maximize the power output to an ultrasonic transducer that is sensitive to both environmental changes and changes in load, and yet does not necessarily require a fixed phase angle or a constant current.

Description of the invention In view of the above problems, the present invention was developed. The invention is an improved phacoemulsification probe drive circuit for supplying electrical power to an ultrasonic transducer. The drive circuit has a power control loop and a frequency control loop. The power control loop has a variable gain amplifier, whose output is an input to a power amplifier. After the power amplifier amplifies the power, the power is supplied to a transformer and then to a transducer. The voltage and current applied to the primary of the transformer are detected to generate a signal proportional to the power (real or apparent) and the result is compared with a power command that originates from a pedal. Once compared, the result of this comparison is sent to a first controller that acts on the information by sending a corrective signal to the variable gain amplifier. In addition, the phase of the waveforms of the voltage and current applied to the primary of the transformer are detected by a phase detector. The phase angle is derived and then compared with a phase command that is determined from the initial calibration of the system. The summing / differential block sends its resulting comparison to a second controller that sends a control signal to the voltage controlled oscillator (VCl). The VCO receives the signal and sends a specific frequency to a fixed voltage to the variable gain amplifier. Before operation, the phacoemulsification probe is calibrated by applying a constant voltage to the probe and sweeping the drive circuit through a series of frequencies. Then, a different voltage is selected and another frequency sweep is performed. This process is repeated for one or more voltage levels and the information on the power and phase with respect to the frequency is stored in memory, so that the optimum phase angle at the resonance associated with a certain power requirement can be determined easily, although the phase angle can be relatively constant over a range of power levels. Additionally, when the power and phase information is stored in memory, a range of frequencies around a certain resonant frequency is used to create a window beyond which certain frequencies can not be used.

In operation, a pedal is lowered by providing a power command that is compared to the existing power. The difference between these two levels is transmitted to the power loop controller. Acting on the information stored in the memory, the power loop controller selects the appropriate voltage level necessary to correct the difference between the power and the power command and sends this information to the control input of the variable gain amplifier. The variable gain amplifier sends its output to a 'power amplifier. The output of the power amplifier is applied to the transformer and simultaneously to both the power monitor and the phase detector. The power is then calculated and compared to the power command signal received from the pedal control and the power loop starts again. The phase detector sends its phase information to an adder / differential block that compares the actual phase with a calculated phase command. The difference between the phase command and the existing phase is then sent to the frequency circuit controller which communicates a signal to the voltage controlled oscillator to emit a certain frequency at the input of the variable gain amplifier which completes the frequency loop. The phase command is determined from the information taken at the calibration time and from the current power command. The additional features and advantages of the present invention, as well as the structure and operation of various embodiments of the present invention, are described below in detail with reference to the accompanying drawings.

Brief Description of the Drawings The accompanying drawings, which are incorporated and form part of the specification, illustrate the embodiments of the present invention and together with the description, serve to explain the principles of the invention. In the drawings: Figure 1 illustrates a block diagram of an electrical model of an ultrasonic phacoemulsification probe operating close to the resonance frequency.

Figure 2 is a graph of apparent power according to the electrical model of Figure 1. Figure 3 is a graph of the phase angle between the wave forms of voltage and current with respect to the graph of apparent power of Figure 2 and resulting from the electric model of Figure 1. Figure 4 is a graph of actual power according to the electric model of Figure 1. Figure 5 is a graph of the phase angle between the shapes of the wave of voltage and current with respect to the actual power graph of Figure 4 and resulting from the electric model of Figure 1. Figure 6 is a graph of apparent power and phase angle with the addition of an inductor of compensation to the electrical model of Figure 5. Figure 7 is a graph of actual power and phase angle with the addition of a compensation inductor to the electric model of Figure 5. Figure 8 illustrates a block diagram of the power system of Figure 5. s phacoemulsification wave of the present invention. Figure 9 illustrates a more detailed apparent power block diagram of the power monitor block in Figure 8. Figure 10 illustrates a more detailed real power block diagram of the power monitor block in Figure 8.

Figures 11, 12, 13, 14 and 15 illustrate a hardware implemented embodiment of the present invention which represents a coprocessor and an electronically programmable logic device. Figures 16, 17, 18 and 19 illustrate an implementation implemented with hardware of the present invention that represents the memory for the coprocessor and a reset circuit. Figures 20, 21 and 22 illustrate an embodiment implemented with hardware of the present invention that represents a transceiver, and an integrated circuit chip of neurons. Figures 23, 24, 25 and 26 illustrate a hardware implemented embodiment of the present invention that represents a servo regulator, a voltage controlled oscillator, a digital-to-analog multiplication converter, a variable gain amplifier, an amplifier power, a first coupling capacitor, an isolating transformer, a second coupling capacitor, a compensation inductor, and an ultrasonic transducer. FIGS. 27 and 28 illustrate a hardware implemented embodiment of the present invention representing DC voltage and current converters RMS, and a medium power detector. Figures 29 and 30 illustrate a hardware implemented embodiment of the present invention representing several minor hardware aspects; and Figures 31 and 32 illustrate an embodiment implemented with hardware of the present invention representing several minor hardware aspects.

Mode (s) for Carrying Out the Invention With reference to the accompanying drawings, in which similar reference numerals indicate similar elements, Figure 8 shows the phacoemulsification probe system, generally shown at 1411, of the present invention. The phacoemulsification probe system 1411 comprises the power loop generally shown at 1412, the frequency loop generally shown at 1413, and the isolated transducer circuit generally shown at 1414. As shown in FIG. 8, the power loop 1412 comprises the power loop controller 1415, the variable gain amplifier 1416, the power amplifier 1417, the first coupling capacitor 1418, the secondary of the transformer 1436, the power monitor 1419, the first summing / differential block 1425, and the power command signal input 1426. The power loop controller 1415 has an output to the variable gain amplifier 1416. The function of the power loop controller 1415 is twofold: (1) perform a square root operation ( the power is proportional to the square of the voltage); and (2) ensure the stability of the loop and ensure the desired response characteristics of the system. Optionally, the power loop controller 1415 may store peak power information in the memory, although this may also be handled by a combination of coprocessor and coprocessor memory. The power amplifier 1417 receives an input from the output of the variable gain amplifier 1416. The output of the power amplifier 1417 continues through the coupling capacitor 1418 which compensates for the leakage inductance as well as blocks any current "Direct from the power amplifier 1417. The power is then supplied to the primary transformer 1436 and from there to the isolated transducer circuit 1414. Additionally, the voltage and current applied to the isolated transducer circuit 1414 are detected by the power monitor 1419. The 1419 power monitor generates a signal proportional to the power (real or apparent). As shown in Figure 9, the power monitor 1419 can be an apparent power monitor comprising a converter 1420 of the square root of the mean of the instantaneous squares (RMS) in DC 1420, a converter 1421 of the RMS current in DC, and a multiplier 1422. A DC signal is produced which provides an apparent power value which is then communicated to a first summing / differential block 1425. Alternatively, the power monitor 1419 may be a real power monitor comprising a voltage and current multiplier 1423 connected to low pass filter 1424. An actual power value is produced which is then communicated to the first summing / differential block 1425. The first summing / differential block 1425 compares the power level detected by the monitor of power 1419 and the power command provided at the signal input of the power command 1426. In hardware, any summing / differential block described herein it can be incorporated as a differential amplifier, and in software it is commonly referred to as a "subtraction" operation. The results of the comparison are communicated to the power loop controller 1415. A calculation is made on the required correction amount, and the power loop controller 1415 sends a new signal to the voltage gain amplifier 1416 based on the calculation. The calculation can be performed by the power loop controller 1415, or any other component associated with the power loop controller 1415, such as a coprocessor and coprocessor memory. This completes a cycle of the power loop 1412. The frequency loop 1413 comprises the frequency loop controller 1430 which communicates a signal to the voltage controlled oscillator 1431 which itself provides an input to a variable gain amplifier 1416., from there to the power amplifier 1417, through the coupling capacitor 1418, to the isolated transducer circuit 1414. The phase of the voltage and current waveforms applied to the isolated transducer circuit 1414 are detected by the detector of phase 1432 and then communicated to the second adder / differential block 1433. A phase command that is determined from the initial calibration of the system and possibly from the subsequent calculation is also communicated to the input of the phase 1434 command of the second block adder / differential 1433. After this, the second adder / differential block 1433 communicates an error signal based on the phase difference between the actual phase and the phase command to the frequency loop controller 1430. A calculation of the correction amount required, and the frequency loop controller 1430 sends a new signal to the voltage controlled oscillator 1431 based on the calculation The calculation can be performed by the frequency loop controller 1430, or any other component associated with the power loop controller 1430 such as a coprocessor and the coprocessor memory. This completes an iteration of 1413 frequency loop. Returning back to the isolated transducer circuit 1414, the isolated transducer circuit 1414 comprises the insulating secondary transformer 1436, the second coupling capacitor 1437, the compensation inductor 1438, and the ultrasonic transducer 1439. More specifically, the parallel combination of the ultrasonic transducer 1439 and the compensation inductor 1438 is connected in series with the secondary of the transformer 1436 and the coupling capacitor 1437. The function of the second coupling capacitor 1437 is to compensate for any leakage inductance from the secondary isolation transformer 1436. The value of the compensation inductor 1438 is selected such that the magnitude of its reactance is equal to the magnitude of the reactance (C) of the parallel capacitance of the ultrasonic transducer 1439. If F represents the resonant frequency of the ultrasonic transducer, then the appropriate value of inductance to compensate the ultrasonic transducer is one divided by the square footage of l a quantity of two per pi for F, final quantity, for C, final quantity. In calculating the value of the compensation inductor 1438, it is commonly known that the values for the ultrasonic transducer 1439 experience some amount of variation. Accordingly, a sampling of parts of the ultrasonic transducer 1439 can be performed to derive the mean value of the parallel capacitance and, therefore, to calculate the value of the compensation inductor 1438. Since the compensation inductor 1438 is a fixed value, it is known that this circuit is designed to provide a relatively reliable inductor value to make the phacoemulsification probe system 1401 appear purely resistive with the parallel combination of the ultrasonic transducer 1439 and the compensation inductor 1438, with a small degree of error that it is compensated using the power loop 1412 and the frequency loop 1413 in combination. The phacoemulsification probe system 1411 has two separate and distinct modes. One mode is calibration in which the control loops are open, and the adder / differential blocks, 1425, 1433 respectively, are removed, and the other is an operation, in which the control loops are closed so that they can be a response to a power command from a pedal. Returning to the operation of the phacoemulsification probe system 1411, prior to actual surgical use, a calibration of the entire 1411 system must first be provided. The purpose of the calibration step is to initialize a window of operating system voltage and operating frequencies. phacoemulsification probe 1411. Briefly, the purpose of the calibration is to find a window of operation of voltages and frequencies' by successively iterating a series of frequencies at a constant voltage (sweeping the frequency), and then possibly repeating this for different voltages to derive the resonant frequency at various power levels. This information is stored in memory and then used to determine the phase commands in the control of the dual-loop phacoemulsification probe system 1411. Calibration is initiated by a request from the user. As an overview, the calibration consists of one or more frequency sweeps. The frequency is swept from a lower start frequency to a higher final frequency by the frequency loop controller 1430. During this frequency sweep, the excitation level is kept constant by the power loop controller 1415. At a further level In detail, a command signal to calibrate is received by the power loop controller 1415 and the frequency loop controller 1427. The power loop controller 1415 then sends a command signal to the variable gain amplifier 1416 so that the variable gain amplifier will emit a 'fixed voltage. In a similar way, the frequency loop controller 1430 sends a signal to the voltage controlled oscillator 1431. After the reception of the signal from the frequency loop controller 1430, the voltage controlled oscillator 1431 transmits a frequency sweep to the variable gain amplifier 1416. The variable gain amplifier 1416 applies a voltage gain to the frequency sweep voltage to produce an output voltage. This output voltage is communicated as an input voltage to the power amplifier 1417. The power amplifier 1417 amplifies the power and supplies the power to the secondary isolation transformer 1436 via the coupling capacitor 1418 (whose operation was previously described). The power monitor 1419 determines the frequency at which the peak power reaches a local maximum, and the phase detector 1432 determines the frequency where the phase passes through zero. A window of operating frequencies is then determined on this critical frequency. The rear end of the window is determined first by determining where the -frequency reaches a local maximum peak power as well as a proximity to a zero phase angle crossing. From this area, the frequency sweep is examined at lower frequencies to determine the frequency at which a crossing of the previous zero phase angle occurs. From the frequency of this above zero phase angle crossing, a fixed frequency amount is added to establish the rear end of the operating frequency window. The front end of the operating frequency window can be set in a similar way. Alternatively, a fixed frequency band such as 1 kHz can be set back and in front of the critical frequency. The purpose of establishing an operating frequency window is to ensure that the resonant frequency will occur within the operating frequency window, without encountering other zero phase crossings. Information on peak power, peak power phase, operating power level, and frequency window can be stored in memory. It should be noted that it may be preferable to conduct a wider frequency sweep to identify the general areas of interest, and then carry out a finer frequency sweep to focus on the area of general interest. In this way, the memory requirements are minimized, since the sweep information stored in the memory is larger, but it exists only temporarily to allow the derivation of the window information, while the window information is relatively more permanent, but smaller in memory space requirements. After the frequency sweep, the power loop controller 1415 changes the voltage gain and a different voltage (power / excitation level) is used to sweep the frequency, the phase and power information that results therefrom being stored in the memory. . This data taken during the calibration allows the determination of varying the phase angles so that a phase command can be determined during the subsequent operation based on the error signals derived from the first summing / differential block 1425 and the second adder / differential block 1433 during the operation of the phacoemulsification probe system 1411. After the calibration of the phacoemulsification probe system 1411 has been completed, a process that can take between four and six seconds, can begin the actual operation of the phacoemulsification probe system 1411 as a manual piece of phacoemulsification. In operation, the surgeon presses a pedal (not shown) that sends a power command to the power command signal input 1426 of the first summing / differential block 1425. Based on the difference between the new power command and the level of the system's existing power, the first summing / differential block 1425 sends an error signal to the power loop controller 1415. The power loop controller 1415 calculates a new voltage requirement and sends a signal to the variable gain amplifier. Similarly, a phase command signal determined from the power command and information stored during calibration inputs the input of the phase command signal 1434 to the second summing / differential block 1433. The second summing / differential block 1433 generates an error signal and communicates this signal to the voltage controlled oscillator 1431. The voltage controlled oscillator 1431 outputs a frequency changed to an input of the variable gain amplifier 1416. Now having two inputs, the variable gain amplifier 1416 emits a voltage to the power amplifier 1417 which then supplies power to the insulating secondary transformer 1436. The secondary insulating transformer 1436 supplies power through the second coupling capacitor 1437 and the compensation inductor 1438 to the ultrasonic transducer 1439. Simultaneously with the power supply from the 1417 power amplifier to the circuit or isolated transducer 1414, the voltage and current waveforms are communicated (in parallel with the secondary isolation transformer 1436) to the power monitor 1419 and the phase detector 1432. The average DC power signal is received by the first adder block / differential 1425 and compared to the existing power command provided at the power command signal input 1426. An error signal is then communicated from the first summing / differential block 1425 to the power loop controller 1415. Similarly, the phase angle of the phase detector 1432 is communicated to the second adder / differential block 1433 and compared to the input of the phase command signal 1434 and communicated to the frequency loop controller 1430. The power loop controllers and frequency loop 1415 and 1430, respectively, then send corrective signals to the variable gain amplifier 1416, and to the variable controlled oscillator 1431, as described above.

It should be noted that the phase command signal is more likely than not a non-zero phase command, since the final phacoemulsification probe system 1411 is very likely not exactly a purely resistive circuit. The reason that it is very likely that the system 1411 is not a circuit is purely because the compensation inductor 1438 has a fixed value that has a slight tolerance variation from unit to unit, and because the parallel capacitance of the transducer 1439 may vary from manual part to manual part, and due to environmental factors that could cause the resonant frequency of the ultrasonic transducer 1439 to change. For this reason, it is very likely that the optimum phase angle f for a particular power level is also no zero. When the phase angle f is zero, the circuit is purely resistive. If there is an imbalance in the circuit, the phase angle can not be zero, since the circuit is not purely resistive. However, on average it is estimated that the optimum phase angle will generally be within at least twenty degrees of zero. Figures 11-32 are first intended to simply allow the reader to prepare the detailed circuit diagrams of the blocking diagram shown in Figure 8, and secondly to reveal the best way to practice the invention. The power loop controller functions 1415 and the frequency loop controller 1430 are physically combined in hardware within a coprocessor 1441 shown in FIGS. 11 and 12. The coprocessor 1441 of FIGS. 11 and 12 is connected to the control oscillator. of voltage 1431 shown in FIG. 23, a sign wave generator, which passes its signal to the variable gain amplifier 1416, incorporated in LF labeled 412 in combination with the digital-to-analog multiplication converter (MDAC), indicated by the number reference 1444. MDAC 1444 is a two-channel DAC that passes a signal to the servo-regulator circuit shown in FIG. 24 that provides the power supply and deviation voltage to the power amplifier 1417 shown in the LM12 labeled operational amplifier block. in figure 25. It should be noted that the servo-regulator circuit is not necessary to realize the present invention. A servo regulator circuit is simply a different means for providing the voltage supply to the power amplifier 1417, and the use of the circuit requires an additional controller to calculate the required servo voltage output and to send a servo command to the servo regulator. The output of the power amplifier 1417 is passed through the coupling capacitor 1418 and thence to the isolating transformer 1436. As shown in the far right of the figure 25, the conductor cable of the current monitor 1446 and the cable voltage monitor conductor 1447 are provided to detect the current and voltage supplied to the isolating transformer 1436. The monitor lead wires, 1446 and 1447 respectively, of FIG. 25 are continued on FIG. 27, where the monitor signals are scaled by operation amplifier blocks labeled LF412, with numerical reference numbers 1448 and 1449. After scaling, the power monitor 1419 detects the power supplied to the first transformer (primary transformer) 1436. Specifically, the voltage converter RMS to DC 1420 is shown in block AD536 and the current converter RMS to DC 1421 is shown as simulated ilar AD536. After this, the outputs communicate with an analog to digital converter shown in block MAX182 (with number 1450) which converts the sine wave signal into DC and from there to coprocessor 1441 shown in figure 11. In figure 27, after the voltage and current monitors, 1446-1447 are scaled, they also communicate with the phase detector 1432 which is composed of two parts: (1) the zero-crossing detector operation amplifier shown in the blocks labeled LM319 ( with numbers 1451 and 1452) in figure 27; and (2) from there to Figure 13 to the Electronic Programmable Logic Device (EPLD) 1453 shown in the PLSI1032 block. After leaving the EPLD 1453 in FIG. 13, the output is transmitted to FIG. 27, the delay low pass filter of the lead wire shown in block LF412 and thence to analog to digital converter 1450 shown in block MAX182, and then to coprocessor 1441 shown in Fig. 11. Returning to Fig. 20, a NEURON chip 1454 (NEURON is a registered trademark) is shown in block U25. This chip has the following function. When the surgeon presses the pedal control, a communication is transmitted from the pedal control to the transceiver 1455 shown in block U23 of figure 22. Once the communication of the power command is received by the transceiver 1455, it is sent to the chip 1454 NEURON in block U25, and then compressor 1441 shown in figure 11. In view of the foregoing, it will be noted that various objects of the invention are achieved and other advantages are achieved. The embodiments were chosen and described in order to better explain the principles of the invention and their practical application in order to allow other technicians in the field to better use the invention in various embodiments and with various modifications that are adapted to the use of the invention. particular contemplated. Since various modifications can be made to the constructions and methods described and illustrated herein without departing from the scope of the invention, it is intended that all matter contained in the preceding description or shown in the accompanying drawings be construed as illustrative rather than as an illustration. limitative. For example, the hardware implementation of the present invention can be changed by consolidation or expansion with other hardware, or it can be replaced by software. In another example, the power amplifier can obtain an additional input from a servo regulator that can provide a power supply and a voltage without departing from the spirit of the present invention. Specifically, after receipt of an error signal in the compared power values, the power loop controller can send a signal to the third controller which then applies an input to the servo controller, whose output is an input to the power amplifier. . Therefore, the breadth and scope of the present invention should not be limited to some of the exemplary embodiments described above, but should be defined only in accordance with the following appended claims and their equivalents.

Claims (16)

Claims

1. A method of driving a phacoemulsification device comprising the steps of: (a) receiving a command signal for power; (b) comparing with an adder the command signal with an existing power signal; (c) sending an error signal from said adder to a power loop controller; (d) calculate a new vge requirement; (e) sending a signal from said power loop controller to a variable gain amplifier based on the new vge requirement; (f) applying a vge gain controlled by the signal from said power loop controller to an input vge with said variable gain amplifier to produce an output vge; (g) amplifying the output vge from said variable gain amplifier as an input vge to a power amplifier; (h) using the input vge of the power amplifier to emit increased power; (i) supplying said power to a transformer; (j) detect the power supplied to said transformer with a power monitor; (k) detecting the phase difference between the vge and the current supplied to said transformer with a phase detector; (1) sending a signal indicating the phase difference of said phase detector to a phase difference adder; (m) comparing with said phase difference adder the signal indicating the phase difference with a phase command signal; (n) sending an error signal from said phase difference adder to a frequency loop controller; (o) calculate a new frequency requirement; (p) sending a control signal from said frequency loop controller to a vge controlled oscillator based on the new frequency requirement; and (q) sending a fixed output vge to the frequency calculated in step (o) from said vge controlled oscillator to the input of said variable gain amplifier.

2. The method of driving a phacoemulsification device according to claim 1, further comprising the steps of: (r) continuously repeating steps from (a) to (q)

3. The method of driving a phacoemulsification device according to claim 1, further comprising the steps of, instead of the step (h) (s) sending a signal from said power loop controller to a servo controller controller; (t) calculate the servo vge output required with the controller, - (u) send a servo command from said servo controller to a servo regulator; (v) sending an output from said servo regulator to said power amplifier; and (w) using the input vge of the power amplifier and the output of the servo regulator in said power amplifier to produce increased power.

4. The method of driving a phacoemulsification device according to claim 1, further comprising the step of: between steps (h) and (i), placing a coupling capacitor after said power amplifier and before said transformer to compensate for the leakage inductance of said transformer and to block any DC current from said power amplifier.

5. The method of driving a phacoemulsification device according to claim 1, wherein - the command signal for power in step (a) is derived from a spatial displacement of a pedal.

6. The method of driving a phacoemulsification device according to claim 1, further comprising the steps of: (x) on the output side of said transformer, providing a second coupling capacitor in series, and an inductor in parallel with an ultrasonic transducer, said capacitor being for compensating any leakage inductance of said transformer to improve the power and phase detection on the primary side of said transformer, and said inductor for compensating the inherent parallel capacitance of said ultrasonic transducer to make its impedance appears to be substantially resistive to resonance.

7. A method of calibrating a phacoemulsification device comprising the steps of: (a) receiving a command signal to calibrate by a power loop controller and a frequency loop controller; (b) sending a signal from said power loop controller to a variable gain amplifier; (c) sending a signal from a frequency loop controller to a voltage controlled oscillator; (d) transmitting a voltage frequency sweep from said voltage controlled oscillator to said variable gain amplifier; (e) applying a voltage gain to the frequency sweep voltage with said voltage gain amplifier to produce an output voltage; (f) using the output voltage from said voltage gain amplifier as an input voltage to a power amplifier; (g) using said power amplifier to produce the increased power; (h) supplying said power to a transformer; (i) detect the power supplied to said transformer with a power monitor and communicate the value of the power level to a power adder; (j) comparing with said power adder the existing power signal with zero; (k) sending an error signal from said power adder to said power loop controller for memory storage; (1) detecting the phase difference between the voltage and current supplied to said transformer with a phase detector; (m) sending a signal indicating the phase difference from said phase detector to a phase difference adder; (n) comparing with said phase difference adder the signal indicating the phase difference with zero; (o) sending an error signal from said phase difference adder to a frequency loop controller for memory storage; and (p) sending a signal from said power loop controller to said variable gain amplifier to change the voltage gain.

8. The calibration method of a phacoemulsification device according to claim 7, further comprising the steps of: (q) repeating steps (d) to (p) a plurality of times,

9. The calibration method of a phacoemulsification device according to claim 7, further comprising the step of: (r) determining the maximum power at certain voltages based on the information stored in the memory, on the frequency, phase and power to allow the calculation of new frequency, phase and power parameters based on error signals derived from said power and phase adders.

10. The calibration method of a phacoemulsification device according to claim 8, further comprising the step of: (s) defining an operable frequency range that avoids zero phase points other than those close to the resonant frequency .

11. The calibration method of a phacoemulsification device according to claim 7, further comprising the step of: between steps (g) and (h), placing a coupling capacitor after said power amplifier and before said transformer to compensate for the leakage inductance of said transformer and to block any DC current from said power amplifier.

12. The calibration method of a phacoemulsification device according to claim 7, further comprising the steps of: (s) on the output side of said transformer, providing a second coupling capacitor in series and an inductor in parallel with a ultrasonic transducer to supply power to said phacoemulsification device.

13. The calibration method of a phacoemulsification device according to claim 7, further comprising the steps of, instead of (g): (t) sending a signal from said power loop controller to a servo regulator; (u) sending an output from said servo regulator to said power amplifier; and (v) using the input voltage of the power amplifier and the output of the servo regulator in said power amplifier to produce increased power.

14. A circuit for a "phacoemulsification" probe system comprising: a frequency loop controller having an output connected to an input of a voltage controlled oscillator; a power loop controller having an output connected to an input of a variable gain amplifier; said voltage controlled oscillator having an output connected to an input of said variable gain amplifier; a power amplifier connected to an output of said variable gain amplifier; said power amplifier supplying a power to an isolated power transformer; a power monitor connected and detecting said power supplied to said isolated power transformer; a power adder having an input connected to an output of said power monitor, and having an output connected to said power loop controller; a phase detector connected and detecting said waveforms of the voltage and current supplied to said isolated power transformer; and a phase adder having an input connected to an output of said phase detector, and having an output connected to said frequency circuit controller.

15. A circuit for a phacoemulsification probe system according to claim 14, wherein said power monitor comprises: a DC RMS voltage converter connected and detecting the waveform of the voltage supplied to said isolated power transformer; a DC RMS current converter connected and detecting the waveform of the current supplied to said isolated power transformer, a multiplier having an input connected to said RMS voltage converter in DC and another input connected to said converter of RMS current in DC, and having an output connected to said power adder.

16. A circuit for a phacoemulsification probe system according to claim 14, wherein said power monitor comprises: a multiplier having an input connected and detecting the waveform of the voltage supplied to said isolated power transformer and another connected input and which senses the shape of the wave of the current supplied to said isolated transducer circuit; and a low pass filter having an input connected to an output of said multiplier, and having an output connected to said phase adder.