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WO2020224200A1 - 一种开关变换器及其控制方法 - Google Patents

一种开关变换器及其控制方法 Download PDF

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Publication number
WO2020224200A1
WO2020224200A1 PCT/CN2019/113677 CN2019113677W WO2020224200A1 WO 2020224200 A1 WO2020224200 A1 WO 2020224200A1 CN 2019113677 W CN2019113677 W CN 2019113677W WO 2020224200 A1 WO2020224200 A1 WO 2020224200A1
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Prior art keywords
inductor
switch
voltage
current
switching tube
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French (fr)
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卢鹏飞
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Mornsun Guangzhou Science and Technology Ltd
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Mornsun Guangzhou Science and Technology Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a switching power supply, in particular to a switching converter circuit and a control method thereof.
  • Figure 1 is a traditional step-down circuit.
  • the effective value of the current is relatively large when the circuit works in discontinuous mode.
  • the MOS tube Q1 is hard-switched, and the conduction loss of the diode D1 is large.
  • Figure 2 is a step-down circuit with synchronous rectification function. Compared with the traditional step-down circuit, the conduction loss in the freewheeling phase is reduced. In the FCCM mode, the ZVS of the MOS transistor Q1 can also be turned on, due to the ZVS of the MOS transistor Q1 The range of turn-on and high-efficiency work is relatively narrow, so in a wide voltage input range, there is a problem of low overall efficiency in the full load range.
  • Figure 3 is a drawing of the abstract of the US patent application number US13/794,588.
  • the core idea of the patent is to clamp the reverse current of the inductor in parallel with a unidirectional switch at both ends of the inductor, which can be realized under a larger input voltage range and load range.
  • the ZVS of the main MOS transistor Q1 is turned on.
  • this patent does not solve the problem of large current effective value and large conduction loss when the step-down circuit works in discontinuous mode; and when the ratio of the intermittent mode to the input and output voltage is greater than 3, the MOS transistor Q1 If the ton time is too small and the freewheeling time of the MOS transistor Q2 is too long and high frequency, it is difficult to achieve a compromise between large current output and high efficiency operation.
  • the present invention proposes a switching converter and its control method to solve the problem that the effective value of the current of the step-down circuit is relatively large when the step-down circuit works in discontinuous mode.
  • the problem of high loss is a switching converter and its control method to solve the problem that the effective value of the current of the step-down circuit is relatively large when the step-down circuit works in discontinuous mode. The problem of high loss.
  • a switching converter including input power supply positive, output voltage positive, power supply common ground, switching tube Q1, switching tube Q2, switching tube Q3, switching tube Q4, inductor L1 and capacitor C1; the drain of switching tube Q1 and the switch
  • the drain of the tube Q3 is connected to the input power supply
  • the source of the switching tube Q1 and the drain of the switching tube Q2 are connected to one end of the inductor L1
  • the source of the switching tube Q3 and the drain of the switching tube Q4 are connected to the inductor L1
  • the source of the switch Q4 is connected to one end of the capacitor C1
  • the source of the switch Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
  • the ratio of input to output voltage is greater than 3.
  • the switching tube Q1, the switching tube Q2, the switching tube Q3, and the switching tube Q4 are MOS tubes, triodes or IGBTs.
  • the first control method of the above switching converter includes the following steps:
  • Stage t0 ⁇ t1 At t0, the switch Q4 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the switch Q1 is turned off at the time t1;
  • Stage t1 ⁇ t2 After the switch Q1 is turned off, the current IL of the inductor L1 charges the output capacitor Coss1 of the switch Q1 and discharges the output capacitor Coss2 of the switch Q2. At t2, the voltage at one end of the inductor L1 drops by Vin At 0V, the switch Q2 realizes ZVS opening;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitance Coss4 of the switching tube Q4 and discharges the output capacitance Coss3 of the switching tube Q3. At t4, the voltage at the other end of the inductor L1 rises from Vo to Vin, and the switching tube Q3 Realize the opening of ZVS;
  • the current IL of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1.
  • the voltage at one end of the inductor L1 rises from 0V to Vin, and the switch Q1 realizes ZVS opened;
  • Stage t7 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss3 of the switch Q3 and discharges the output capacitor Coss4 of the switch Q4. At t0+Tx, the voltage at the other end of the inductor L1 drops from Vin to Vo , The switch Q4 realizes ZVS opening;
  • the stages t0 to t1 when the load is reduced to a certain value, the stages t0 to t1, the stages t2 to t3 and the stages t4 to t5 begin to decrease, and the stages t6 to t7 become longer.
  • the second control method of the above switching converter includes the following steps:
  • Stage t0 ⁇ t1 At t0, the switch Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the switch Q1 is turned off at the time t1;
  • Stage t1 ⁇ t2 After the switch Q1 is turned off, the current IL of the inductor L1 charges the output capacitor Coss1 of the switch Q1 and discharges the output capacitor Coss2 of the switch Q2. At t2, the voltage at one end of the inductor L1 drops by Vin At 0V, the switch Q2 realizes ZVS opening;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitance Coss4 of the switching tube Q4 and discharges the output capacitance Coss3 of the switching tube Q3. At t4, the voltage at the other end of the inductor L1 rises from Vo to Vin, and the switching tube Q3 Realize the opening of ZVS;
  • Stage t7 ⁇ t0+Tx The current of the inductor L1 discharges the output capacitor Coss1 of the switch Q1 and charges the output capacitor Coss2 of the switch Q2. At t0+Tx, the voltage at one end of the inductor L1 rises from zero to Vin, and the switch Manage Q1 to realize ZVS opening;
  • the stages t0 to t1 when the load is reduced to a certain value, the stages t0 to t1, the stages t2 to t3 and the stages t4 to t5 begin to decrease, and the stages t6 to t7 become longer.
  • the present invention also provides another switching converter with the same inventive concept.
  • the technical solution is as follows:
  • a switching converter including input power supply positive, output voltage positive, power supply common ground, switching tube Q1, switching tube Q2, switching tube Q4, diode D1, inductor L1 and capacitor C1; the drain of switching tube Q1 and diode D1
  • the cathode of the switch Q1 is connected to the input power supply, the source of the switch Q1 and the drain of the switch Q2 are connected to one end of the inductor L1, the anode of the diode D1 and the drain of the switch Q4 are connected to the other end of the inductor L1, and the switch
  • the source of the tube Q4 is connected to one end of the capacitor C1, and the source of the switching tube Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
  • the ratio of input to output voltage is greater than 3.
  • the switching tube Q1, the switching tube Q2, and the switching tube Q4 are MOS tubes, triodes or IGBTs.
  • the control method of the above switching converter includes the following steps:
  • Stage t0 ⁇ t1 At t0, the switch Q1 is turned on, the voltage across the inductor L1 is Vin-Vo, and the inductor L1 is excited, the current IL of the inductor L1 rises, and the switch Q1 is turned off at the time t1;
  • Stage t1 ⁇ t2 After the switch Q1 is turned off, the current IL of the inductor L1 charges the output capacitor Coss1 of the switch Q1 and discharges the output capacitor Coss2 of the switch Q2. At t2, the voltage at one end of the inductor L1 drops by Vin At 0V, the switch Q2 realizes ZVS opening;
  • the current IL of the inductor L1 charges the output capacitor Coss4 of the switch Q4, the voltage at the other end of the inductor L1 rises from Vo to Vin, and the voltage across the inductor L1 is clamped by Vin.
  • the current IL drops to At zero hour, the current IL reverses and the output capacitor Coss4 of the switching tube Q4 begins to discharge.
  • the voltage at the other end of the inductor L1 drops from Vin to Vo, and the switching tube Q4 realizes ZVS opening;
  • Stage t5 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss2 of the switch Q2 and discharges the output capacitor Coss1 of the switch Q1. At t6, the voltage at one end of the inductor L1 rises from 0V to Vin, and the switch Q1 realizes ZVS opening at t0+Tx;
  • the period t0 to t1 when the load is reduced to a certain value, the period t0 to t1, the period t2 to t3 and the period t4 to t5 begin to decrease, and the period t6 to t7 becomes longer.
  • the drain of the switching tube For MOS tube, it refers to the drain, for the triode, it is the collector, and for the IGBT, it is the drain.
  • Other switching tubes can correspond to the knowledge of those skilled in the art, not one by one. Enumerate
  • the source of the switching tube For MOS tube, it refers to the source, for triode, it is the emitter, and for IGBT, it is the source. Other switching tubes can correspond to each other according to the knowledge of those skilled in the art. Enumerate.
  • the present invention has the following beneficial effects:
  • Figure 1 is a schematic diagram of a traditional step-down circuit
  • Figure 2 is a schematic diagram of a step-down circuit with synchronous rectification function
  • Figure 3 is a schematic diagram of the step-down patented circuit with application number US13/794,588;
  • FIG. 4 is a schematic diagram of the circuit of the first embodiment of the present invention.
  • Figure 5 shows the relationship between the ratio of input and output voltage and the switching frequency
  • Fig. 6 is a first working sequence diagram of the first embodiment of the present invention.
  • Figure 7 is a second working sequence diagram of the first embodiment of the present invention.
  • Fig. 8 is a schematic circuit diagram of the second embodiment of the present invention.
  • Fig. 9 is a working sequence diagram of the second embodiment of the present invention.
  • Fig. 4 is a schematic circuit diagram of the first embodiment of the present invention. Including input power supply positive Vin, output voltage positive Vo, power supply common ground GND, MOS tube Q1, MOS tube Q2, MOS tube Q3, MOS tube Q4, inductor L1 and capacitor C1; the drain of MOS tube Q1 and MOS tube Q3 The drain is connected to the input power positive Vin, the source of MOS transistor Q1 and the drain of MOS transistor Q2 are connected to one end of inductor L1, and the source of MOS transistor Q3 and the drain of MOS transistor Q4 are connected to the other of inductor L1. At one end, the source of the MOS transistor Q4 is connected to one end of the capacitor C1, and the source of the MOS transistor Q2 and the other end of the capacitor C1 are connected to the power supply common ground GND.
  • Coss1, Coss2, Coss3, and Coss4 in Figure 4 are the output capacitors of MOS transistors Q1, MOS transistors Q2, MOS transistors Q3 and MOS transistors Q4, respectively.
  • MOS transistors Q1, MOS transistors Q2 and MOS are also shown in Figure 4 The body diode of the tube Q3 and MOS tube Q4.
  • MOS transistors Q1, MOS transistors Q2, MOS transistors Q3, and MOS transistors Q4 are replaced with other types of switching transistors such as transistors and IGBTs.
  • Figure 5 shows the waveform of the inductor L1 current IL and the output current Io when the buck circuit works in discontinuous mode.
  • the rising slope of current IL is The decreasing slope of current IL is Therefore, the current IL rises and falls in the same time, and the corresponding duty cycle is T1.
  • the rising slope of current IL is The decreasing slope of current IL is Change the inductance of inductor L1 so that the rising slope of current IL and
  • the falling time of current IL is When the current IL falls time twice, the corresponding duty cycle is T2, which is greater than T1.
  • the rising slope of current IL is The decreasing slope of current IL is Change the inductance of inductor L1 so that the rising slope of current IL and
  • the current IL falling time is When the current IL falls 4 times, the duty cycle is T3, which is greater than T2.
  • the ratio of input to output voltage is greater than 3, it can be ensured that the present invention can achieve better comparison. Good beneficial effect.
  • Figure 6 shows the first working sequence of the first embodiment, which is specifically as follows:
  • Stage t7 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss3 of the MOS transistor Q3, and the output capacitor Coss4 of Q4 discharges.
  • the voltage at the node SW2 (the other end of the inductor L1) is changed by Vin drops to Vo, MOS tube Q4 realizes ZVS opening;
  • Tx in the above t0+Tx represents a time length of X cycles.
  • FIG 7 shows the second working sequence of the first embodiment, which is specifically as follows:
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitance Coss4 of the MOS transistor Q4 and discharges the output capacitance Coss3 of the MOS transistor Q3. At t4, the voltage of the node SW2 rises from Vo to Vin, and the MOS transistor Q3 realizes ZVS Open
  • Stage t7 ⁇ t0+Tx The current of the inductor L1 discharges the output capacitor Coss1 of the MOS transistor Q1 and charges the output capacitor Coss2 of the MOS transistor Q2. At t0+Tx, the voltage of the node SW1 rises from zero to Vin, the MOS transistor Q1 realizes the opening of ZVS;
  • the waveform of the current IL of the inductor L1 is also quadrilateral, which also achieves the purpose of the invention.
  • mode switching can be used to improve the efficiency of the circuit at light load and improve methods. as follows:
  • Fig. 8 is a schematic circuit diagram of the second embodiment of the present invention.
  • the MOS transistor Q3 is replaced by a diode D1.
  • the cathode of the diode D1 is connected to the drain of the MOS transistor Q1 and the input power supply positive Vin, and the anode of the diode D1 is connected to the drain and the inductor of the MOS transistor Q4.
  • the time for the diode D1 to flow through the current is relatively small. Compared with the MOS tube solution, the conduction loss will not increase too much, but it does not need a floating drive, which reduces the drive loss, simplifies the drive circuit, and is suitable for small and medium current output. Scenes.
  • the ratio of input to output voltage is greater than 3 to obtain better implementation effects.
  • Figure 9 shows The working sequence of the second embodiment is specifically as follows:
  • Phase t2 ⁇ t3 The voltage across the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL drops, and the MOS transistor Q4 is turned off at t3;
  • Stage t3 ⁇ t4 The current IL of the inductor L1 charges the output capacitor Coss4 of the MOS transistor Q4, the voltage of the node SW2 rises from Vo to Vin, and the voltage across the inductor L1 is clamped by Vin.
  • the current IL drops to zero
  • the current IL reverses, the output capacitor Coss4 of the MOS transistor Q4 begins to discharge, the voltage of the node SW2 drops from Vin to Vo at t4, and the MOS transistor Q4 realizes the ZVS opening;
  • Phase t5 ⁇ t0+Tx The current IL of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2 and discharges the output capacitor Coss1 of the MOS transistor Q1. At t6, the voltage of the node SW1 rises from 0V to Vin, and the MOS transistor Q1 Realize ZVS opening at t0+Tx;
  • the waveform of the current IL of the inductor L1 is also quadrilateral, which also achieves the purpose of the invention.
  • switching converters with other parameters can also be selected, and the efficiency of the circuit can also be improved through the above-mentioned mode switching, which will not be repeated here.

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  • Power Engineering (AREA)
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Abstract

一种开关变换器及其控制方法，开关变换器包括输入电源正Vin、输出电压正Vo、电源公共地GND、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正Vin，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地GND，该开关变换器在输入输出电压之比较大的条件下能实现电感器L1快速去磁，并将电感器L1的电流波形从三角形变为四边形，实现开关变换器高频高效工作。

Description

一种开关变换器及其控制方法 技术领域
本发明涉及开关电源，特别涉及开关变换器电路及其控制方法。
背景技术
图1为传统的降压电路，电路工作在断续模式下电流有效值较大，MOS管Q1为硬开关，二极管D1的导通损耗大。
图2为具有同步整流功能的降压电路，相比传统的降压电路，降低了续流阶段的导通损耗，在FCCM模式还可以实现MOS管Q1管的ZVS开通，由于MOS管Q1的ZVS开通及高效率工作的范围比较窄，所以在宽电压输入范围，全负载范围存在综合效率偏低的问题。
图3为申请号为US13/794,588的美国专利摘要附图，该专利的核心思想是电感两端并联单向开关对电感的反向电流进行钳位，在较大输入电压范围和负载范围下实现主MOS管Q1的ZVS开通。但是此专利并没有解决降压电路工作在断续模式下电流有效值较大，导通损耗偏大的问题；并且该专利在断续模式和输入输出电压之比值大于3时，MOS管Q1的ton时间太小和MOS管Q2的续流时间太长且高频化，大电流输出和高效率工作难以实现折中。
发明内容
鉴于现有降压电路及其改进型专利电路和控制方式的技术缺陷，本发明提出一种开关变换器及其控制方法，解决降压电路工作在断续模式下电流有效值较大，导通损耗偏大的问题。
为了实现上述发明目的，本发明采用以下技术方案：
一种开关变换器，包括输入电源正、输出电压正、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
优选地，输入输出电压之比大于3。
优选地，所述的开关管Q1、开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
上述开关变换器的第一种控制方法包括如下步骤：
t0～t1阶段：在t0时刻开关管Q4导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t4时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
t4～t5阶段：电感器L1的电流IL存在一次换相，由正转负，在t5时刻关断开关管Q2；
t5～t6阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t6时刻电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；
t6～t7阶段：电感器L1两端的电压均为Vin，电压差为零，所以电感器L1的电流IL保持不变，t7时刻关断开关管Q3；
t7～t0+Tx阶段：电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，在t0+Tx时刻电感器L1另一端的电压由Vin降到Vo，开关管Q4实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述第一种控制方法的改进，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
上述开关变换器的第二种控制方法包括如下步骤：
t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t4时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
t4～t5阶段：电感器L1的电流IL在t5时刻下降到零，此时关断开关管Q3；
t5～t6阶段：开关管Q3的输出电容Coss3充电，开关管Q4的输出电容Coss4放电，电感器L1的电流IL从零下降为负向电流，在t6时刻电感器L1另一端的电压由Vin下降到Vo，开关管Q4实现ZVS开通；
t6～t7阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t7时刻关断开关管Q2；
t7～t0+Tx阶段：电感器L1的电流给开关管Q1的输出电容Coss1放电，给开关管Q2的输出电容Coss2充电，在t0+Tx时刻电感器L1一端的电压由零上升到Vin，开关管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述第二种控制方法的改进，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
本发明还提供另外一种相同发明构思的开关变换器，技术方案如下：
一种开关变换器，包括输入电源正、输出电压正、电源公共地、开关管Q1、开关管Q2、开关管Q4、二极管D1、电感器L1和电容器C1；开关管Q1的漏极和二极管D1的阴极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，二极管D1的阳极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
优选地，输入输出电压之比大于3。
优选地，所述的开关管Q1、开关管Q2和开关管Q4为MOS管、三极管或者IGBT。
上述开关变换器的控制方法包括如下步骤：
t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，电感器L1另一端的电压从Vo上升到Vin，电感器L1两端的电压被Vin钳位，当电流IL下降到零时，电流IL反向，开关管Q4的输出电容Coss4开始放电，在t4时刻电感器L1另一端的电压从Vin下降到Vo，开关管Q4实现ZVS开通；
t4～t5阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t5时刻关断开关管Q2；
t5～t0+Tx阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t6时刻电感器L1一端的电压从0V上升到Vin，开关管Q1在t0+Tx时刻实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
作为上述控制方法的改进，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
术语含义说明：
开关管的漏极：对于MOS管指的是漏极、对于三极管指的是集电极、对于IGBT指的是漏极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举；
开关管的源极：对于MOS管指的是源极、对于三极管指的是发射极、对于IGBT指的是源极，其它开关管依据本领域的技术人员的知识可以自行对应，不再一一列举。
与现有技术相比，本发明具有如下有益效果：
1)电路工作在断续模式，实现了所有MOS管的ZVS开通；
2)电感电流波形从三角形变为四边形，在相同输出功率下电感电流的有效值降低，导通损耗减小，效率提高，容易实现大电流输出；
3)输入输出电压之比较大时，MOS管Q4的关断大大缩短了MOS管Q2的续流时间，实现了高频化，而高频化使电感感值和电容容值降低，从而减小了电源尺寸，降低了成本。
附图说明
图1为传统的降压电路原理图；
图2为具有同步整流功能的降压电路原理图；
图3为申请号为US13/794,588的降压专利电路原理图；
图4为本发明第一实施例电路原理图；
图5为输入输出电压之比和开关频率的关系图；
图6为本发明第一实施例第一种工作时序图；
图7为本发明第一实施例第二种工作时序图；
图8为本发明第二实施例电路原理图；
图9为本发明第二实施例工作时序图。
具体实施方式
第一实施例
图4为本发明的第一实施例的电路原理图。包括输入电源正Vin、输出电压正Vo、电源公共地GND、MOS管Q1、MOS管Q2、MOS管Q3、MOS管Q4、电感器L1和电容器C1；MOS管Q1的漏极和MOS管Q3的漏极连接到输入电源正Vin，MOS管Q1的源极和MOS管Q2的漏极连接到电感器L1的一端，MOS管Q3的源极和MOS管Q4的漏极连接到电感器L1的另一端，MOS管Q4的源极连接到电容器C1的一端，MOS管Q2的源极和电容器C1的另一端连接到电源公共地GND。
图4中的Coss1、Coss2、Coss3和Coss4分别为MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4的输出电容，此外，图4中还画出了MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4的体二极管。
需要说明的是：将MOS管Q1、MOS管Q2、MOS管Q3和MOS管Q4替换为三 极管和IGBT等其它类型的开关管为本领域技术人员的惯用手段。
图5为降压电路工作在断续模式时，电感器L1电流IL的波形和输出电流Io，
Figure PCTCN2019113677-appb-000001
时，根据公式
Figure PCTCN2019113677-appb-000002
电流IL的上升斜率为
Figure PCTCN2019113677-appb-000003
电流IL的下降斜率为
Figure PCTCN2019113677-appb-000004
所以电流IL上升和下降的时间相同，对应的工作周期为T1。
当
Figure PCTCN2019113677-appb-000005
时，电流IL的上升斜率为
Figure PCTCN2019113677-appb-000006
电流IL的下降斜率为
Figure PCTCN2019113677-appb-000007
改变电感器L1的感量，使电流IL的上升斜率和
Figure PCTCN2019113677-appb-000008
时的上升斜率相同，则电流IL的下降时间为
Figure PCTCN2019113677-appb-000009
时电流IL下降时间的2倍，对应的工作周期为T2，大于T1。
当
Figure PCTCN2019113677-appb-000010
时，电流IL的上升斜率为
Figure PCTCN2019113677-appb-000011
电流IL的下降斜率为
Figure PCTCN2019113677-appb-000012
改变电感器L1的感量，使电流IL的上升斜率和
Figure PCTCN2019113677-appb-000013
时的上升斜率相同，则电流IL下降时间为
Figure PCTCN2019113677-appb-000014
时电流IL下降时间的4倍，工作周期为T3，大于T2。
因此在断续模式下，输入输出电压之比越大，对应的开关周期越大，频率越小，越难以实现高频化，输入输出电压之比的选择大于3时，才能保证本发明获得较好的有益效果。
针对Vin电压为380V，Vo电压为48V，电感器L1为1uH，输出电流为20A的开关变换器，图6所示为第一实施例第一种工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q4导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q1；
t1～t2阶段：MOS管Q1关断后，电感器L1的电流IL给MOS管Q1的输出电容Coss1充电，给MOS管Q2的输出电容Coss2放电，在t2时刻电路结点SW1(即电感器L1的一端)的电压由Vin降为0V，MOS管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q4；
t3～t4阶段：电感器L1的电流IL给MOS管Q4的输出电容Coss4充电， 给MOS管Q3的输出电容Coss3放电，在t4时刻结点SW2(即电感器L1的另一端)的电压从Vo上升到Vin，MOS管Q3实现ZVS开通；
t4～t5阶段：电感器L1的电流IL存在一次换相，由正转负，在t5时刻关断MOS管Q2；
t5～t6阶段：电感器L1的电流IL给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t6时刻结点SW1(即电感器L1的一端)的电压从0V上升到Vin，MOS管Q1实现ZVS开通；
t6～t7阶段：电感器L1两端的电压均为Vin，电压差为零，所以电感器L1的电流IL保持不变，t7时刻关断MOS管Q3；
t7～t0+Tx阶段：电感器L1的电流IL给MOS管Q3的输出电容Coss3充电，Q4的输出电容Coss4放电，在t0+Tx时刻结点SW2(即电感器L1的另一端)的电压由Vin降到Vo，MOS管Q4实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
由于电路为周期性的工作，上述t0+Tx中的Tx代表的含义为X个周期的时间长度。
从图6中可以看出电感器L1的电流IL的波形为四边形，在相同输出功率下，较现有技术的三角形波形而言，电感电流的峰值降低，有效值降低，所以导通损耗减小，效率提高，由公式
Figure PCTCN2019113677-appb-000015
得L*di＝N*dB*Ae，电流峰值降低使di减小，在电感器感量L，圈数N和磁芯dB不变的条件下，电感器磁芯的有效截面积Ae减小，则磁芯尺寸变小；在相同输出纹波要求下di减小，则所需的滤波电容器容值减小，电容尺寸变小；MOS管Q4的关断大大缩短了MOS管Q2的续流时间，实现了高频化，而高频化使电感感值和电容容值进一步降低；使电源尺寸减小，降低了成本。
图7所示为第一实施例第二种工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q1；
t1～t2阶段：MOS管Q1关断后，电感器L1的电流IL给MOS管Q1的输出电容Coss1充电，给MOS管Q2的输出电容Coss2放电，在t2时刻电路结点SW1的电压由Vin降为0V，MOS管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q4；
t3～t4阶段：电感器L1的电流IL给MOS管Q4的输出电容Coss4充电，给MOS管Q3的输出电容Coss3放电，在t4时刻结点SW2的电压从Vo上升到Vin，MOS管Q3实现ZVS开通；
t4～t5阶段：电感器L1的电流IL在t5时刻下降到零，此时关断MOS管Q3；
t5～t6阶段：MOS管Q3的输出电容Coss3充电，MOS管Q4的输出电容Coss4放电，电感器L1的电流IL从零下降为负向电流，在t6时刻结点SW2的电压由Vin下降到Vo，MOS管Q4实现ZVS开通；
t6～t7阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t7时刻关断MOS管Q2；
t7～t0+Tx阶段：电感器L1的电流给MOS管Q1的输出电容Coss1放电，给MOS管Q2的输出电容Coss2充电，在t0+Tx时刻结点SW1的电压由零上升到Vin，MOS管Q1实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
从图7中可以看出电感器L1的电流IL的波形也为四边形，同样实现发明目的。
需要说明的是，除了Vin电压为380V，Vo电压为48V，电感器L1为1uH，输出电流为20A的开关变换器，选择其他参数的开关变换器也具有类同的工作时序图，电感器L1的电流IL的波形也为四边形，只是各时间点的幅值有所区别。
另外，上述两种工作时序都是针对负载为满载时的应用场景，在实际的应用场合中，经常有轻载的情况出现，这时可以通过模式切换来改善轻载时电路的效率，改善方法如下：
1.当负载降低到一定值时(即输出电流减少到一定值时)，关闭Q3的驱动，减小驱动损耗，提高效率；
2.当负载进一步降低到一定值时(即输出电流进一步减小到一定值时)，t0～t1阶段，t2～t3阶段和t4～t5阶段减小较大，t6～t7阶段变长较多，使 总的开关周期基本维持不变。但有效的电流IL时间减小使得电流IL的有效值偏大，导通损耗偏大，这时使Q4处于持续导通状态，进一步降低驱动损耗，电路变为同步整流降压电路，电感器L1的电流IL的波形从四边形加较长的t6～t7阶段变为普通的三角形，在同等输出电流下电流有效值降低，效率提升。
第二实施例
图8为本发明的第二实施例的电路原理图。在第一实施例的基础上，将MOS管Q3换为二极管D1，二极管D1的阴极连接到MOS管Q1的漏极和输入电源正Vin，二极管D1的阳极连接到MOS管Q4的漏极和电感器L1的另一端。
二极管D1流过电流的时间相对较小，和MOS管方案相比，导通损耗不会增加太多，却省去了一路浮地驱动，降低了驱动损耗，简化了驱动电路，适合中小电流输出场景。
本实施例输入输出电压之比大于3同样能获得较佳的实施效果，针对Vin电压为380V，Vo电压为48V，电感器L1为1uH，输出电流为20A的开关变换器，图9所示为第二实施例工作时序，具体如下：
t0～t1阶段：在t0时刻MOS管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断MOS管Q1；
t1～t2阶段：MOS管Q1关断后，电感器L1的电流IL给MOS管Q1的输出电容Coss1充电，给MOS管Q2的输出电容Coss2放电，在t2时刻电路结点SW1的电压由Vin降为0V，MOS管Q2实现ZVS开通；
t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断MOS管Q4；
t3～t4阶段：电感器L1的电流IL给MOS管Q4的输出电容Coss4充电，结点SW2的电压从Vo上升到Vin，电感器L1两端的电压被Vin钳位，当电流IL下降到零时，电流IL反向，MOS管Q4的输出电容Coss4开始放电，在t4时刻结点SW2的电压从Vin下降到Vo，MOS管Q4实现ZVS开通；
t4～t5阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t5时刻关断MOS管Q2；
t5～t0+Tx阶段：电感器L1的电流IL给MOS管Q2的输出电容Coss2充电，给MOS管Q1的输出电容Coss1放电，在t6时刻结点SW1的电压从0V上升到 Vin，MOS管Q1在t0+Tx时刻实现ZVS开通；
本周期结束，下一个工作周期开始，重复上面的阶段。
从图9中可以看出电感器L1的电流IL的波形也为四边形，同样实现发明目的。
本实施例同样可以选择其他参数的开关变换器，也可以通过上述的模式切换来改善电路的效率，在此不赘述。
上述实施方式不应视为对本发明的限制，本发明的保护范围应当以权利要求所限定的范围为准。对于本技术领域的普通技术人员来说，在不脱离本发明的精神和范围内，还可以做出若干等同替换、改进和润饰，如根据应用场合的不同，通过器件的简单串并联等手段对电路微调，这些改进和润饰也应视为本发明的保护范围。

Claims (12)

  1. 一种开关变换器，其特征在于：包括输入电源正、输出电压正、电源公共地、开关管Q1、开关管Q2、开关管Q3、开关管Q4、电感器L1和电容器C1；开关管Q1的漏极和开关管Q3的漏极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，开关管Q3的源极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
  2. 根据权利要求1所述的开关变换器，其特征在于：输入输出电压之比大于3。
  3. 根据权利要求1所述的开关变换器，其特征在于：开关管Q1、开关管Q2、开关管Q3和开关管Q4为MOS管、三极管或者IGBT。
  4. 一种权利要求1至3任一项开关变换器的控制方法，其特征在于包括如下步骤：
    t0～t1阶段：在t0时刻开关管Q4导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
    t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
    t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
    t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t4时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
    t4～t5阶段：电感器L1的电流IL存在一次换相，由正转负，在t5时刻关断开关管Q2；
    t5～t6阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电，给开关管Q1的输出电容Coss1放电，在t6时刻电感器L1一端的电压从0V上升到Vin，开关管Q1实现ZVS开通；
    t6～t7阶段：电感器L1两端的电压均为Vin，电压差为零，所以电感器 L1的电流IL保持不变，t7时刻关断开关管Q3；
    t7～t0+Tx阶段：电感器L1的电流IL给开关管Q3的输出电容Coss3充电，给开关管Q4的输出电容Coss4放电，在t0+Tx时刻电感器L1另一端的电压由Vin降到Vo，开关管Q4实现ZVS开通；
    本周期结束，下一个工作周期开始，重复上面的阶段。
  5. 根据权利要求4所述的控制方法，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
  6. 一种权利要求1至3任一项开关变换器的控制方法，其特征在于包括如下步骤：
    t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
    t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
    t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
    t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，给开关管Q3的输出电容Coss3放电，在t4时刻电感器L1另一端的电压从Vo上升到Vin，开关管Q3实现ZVS开通；
    t4～t5阶段：电感器L1的电流IL在t5时刻下降到零，此时关断开关管Q3；
    t5～t6阶段：开关管Q3的输出电容Coss3充电，开关管Q4的输出电容Coss4放电，电感器L1的电流IL从零下降为负向电流，在t6时刻电感器L1另一端的电压由Vin下降到Vo，开关管Q4实现ZVS开通；
    t6～t7阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t7时刻关断开关管Q2；
    t7～t0+Tx阶段：电感器L1的电流给开关管Q1的输出电容Coss1放电，给开关管Q2的输出电容Coss2充电，在t0+Tx时刻电感器L1一端的电压由零上升到Vin，开关管Q1实现ZVS开通；
    本周期结束，下一个工作周期开始，重复上面的阶段。
  7. 根据权利要求6所述的控制方法，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
  8. 一种开关变换器，其特征在于：包括输入电源正、输出电压正、电源公共地、开关管Q1、开关管Q2、开关管Q4、二极管D1、电感器L1和电容器C1；开关管Q1的漏极和二极管D1的阴极连接到输入电源正，开关管Q1的源极和开关管Q2的漏极连接到电感器L1的一端，二极管D1的阳极和开关管Q4的漏极连接到电感器L1的另一端，开关管Q4的源极连接到电容器C1的一端，开关管Q2的源极和电容器C1的另一端连接到电源公共地。
  9. 根据权利要求8所述的开关变换器，其特征在于：输入输出电压之比大于3。
  10. 根据权利要求8所述的开关变换器，其特征在于：开关管Q1、开关管Q2和开关管Q4为MOS管、三极管或者IGBT。
  11. 权利要求8至10任一项开关变换器的控制方法，其特征在于包括如下步骤：
    t0～t1阶段：在t0时刻开关管Q1导通，电感器L1两端的电压为Vin-Vo，对电感器L1励磁，电感器L1的电流IL上升，在t1时刻关断开关管Q1；
    t1～t2阶段：开关管Q1关断后，电感器L1的电流IL给开关管Q1的输出电容Coss1充电，给开关管Q2的输出电容Coss2放电，在t2时刻电感器L1一端的电压由Vin降为0V，开关管Q2实现ZVS开通；
    t2～t3阶段：电感器L1两端的电压为Vo，对电感器L1去磁，电流IL下降，在t3时刻关断开关管Q4；
    t3～t4阶段：电感器L1的电流IL给开关管Q4的输出电容Coss4充电，电感器L1另一端的电压从Vo上升到Vin，电感器L1两端的电压被Vin钳位，当电流IL下降到零时，电流IL反向，开关管Q4的输出电容Coss4开始放电，在t4时刻电感器L1另一端的电压从Vin下降到Vo，开关管Q4实现ZVS开通；
    t4～t5阶段：电感器L1两端的电压为Vo，Vo对电感器L1反向励磁，在t5时刻关断开关管Q2；
    t5～t0+Tx阶段：电感器L1的电流IL给开关管Q2的输出电容Coss2充电， 给开关管Q1的输出电容Coss1放电，在t6时刻电感器L1一端的电压从0V上升到Vin，开关管Q1在t0+Tx时刻实现ZVS开通；
    本周期结束，下一个工作周期开始，重复上面的阶段。
  12. 根据权利要求11所述的控制方法，其特征在于：当负载降低到一定值时，t0～t1阶段，t2～t3阶段和t4～t5阶段开始减小，t6～t7阶段变长。
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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114499168A (zh) * 2021-12-28 2022-05-13 亚瑞源科技（深圳）有限公司 一种多相式升压转换装置

Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110086342B (zh) * 2019-05-23 2020-11-06 广州金升阳科技有限公司 一种开关变换器及其控制方法
CN110504835B (zh) * 2019-08-26 2021-09-14 广州金升阳科技有限公司 一种开关变换器及其控制方法
CN110474534B (zh) * 2019-09-05 2024-05-07 深圳市依崇微电子科技有限公司 一种改进型非隔离式电压变换器
CN110719026A (zh) * 2019-09-11 2020-01-21 广州金升阳科技有限公司 一种升压变换器及其控制方法

Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20060164051A1 (en) * 2002-07-26 2006-07-27 Thomas Durbaum Dc/dc down converter
CN101011537A (zh) * 2006-12-17 2007-08-08 杨绪民 专治失眠健忘使人聪明的状元醒灵丹
CN102751876A (zh) * 2012-07-18 2012-10-24 浙江大学 一种隔离型基于三端口功率变换器的新能源供电设备
CN103023322A (zh) * 2012-12-18 2013-04-03 黄文辉 一种软开关的降压式变换电路及其控制策略
CN103997828A (zh) * 2014-05-12 2014-08-20 上海大学 一种基于Buck电路的无电解电容LED驱动电源
CN107493638A (zh) * 2017-09-25 2017-12-19 江苏大学 基于boost‑buck变换器交错并联无电解电容led驱动电源及切换方法
CN107769541A (zh) * 2017-11-20 2018-03-06 哈尔滨工业大学 Lcc谐振缓冲型双向直流变换器及其调频软启动控制方法

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5307004A (en) * 1992-07-06 1994-04-26 Carsten Bruce W Soft switching boost and buck regulators
CN100508342C (zh) * 2006-09-15 2009-07-01 力博特公司 一种具有电压箝位功能的arcp软开关电路
CN106787735A (zh) * 2017-03-02 2017-05-31 成都优博创通信技术股份有限公司 H桥tec控制电路
CN107017772B (zh) * 2017-06-02 2019-02-19 哈尔滨工业大学 一种基于交错并联结构的高升压比双向dc/dc变换器

Patent Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20060164051A1 (en) * 2002-07-26 2006-07-27 Thomas Durbaum Dc/dc down converter
CN101011537A (zh) * 2006-12-17 2007-08-08 杨绪民 专治失眠健忘使人聪明的状元醒灵丹
CN102751876A (zh) * 2012-07-18 2012-10-24 浙江大学 一种隔离型基于三端口功率变换器的新能源供电设备
CN103023322A (zh) * 2012-12-18 2013-04-03 黄文辉 一种软开关的降压式变换电路及其控制策略
CN103997828A (zh) * 2014-05-12 2014-08-20 上海大学 一种基于Buck电路的无电解电容LED驱动电源
CN107493638A (zh) * 2017-09-25 2017-12-19 江苏大学 基于boost‑buck变换器交错并联无电解电容led驱动电源及切换方法
CN107769541A (zh) * 2017-11-20 2018-03-06 哈尔滨工业大学 Lcc谐振缓冲型双向直流变换器及其调频软启动控制方法

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114499168A (zh) * 2021-12-28 2022-05-13 亚瑞源科技（深圳）有限公司 一种多相式升压转换装置

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