[go: up one dir, main page]

WO2017098762A1 - Dispositif de conversion de courant, dispositif d'alimentation électrique, et leur procédé de commande - Google Patents

Dispositif de conversion de courant, dispositif d'alimentation électrique, et leur procédé de commande Download PDF

Info

Publication number
WO2017098762A1
WO2017098762A1 PCT/JP2016/076154 JP2016076154W WO2017098762A1 WO 2017098762 A1 WO2017098762 A1 WO 2017098762A1 JP 2016076154 W JP2016076154 W JP 2016076154W WO 2017098762 A1 WO2017098762 A1 WO 2017098762A1
Authority
WO
WIPO (PCT)
Prior art keywords
power
transformer
voltage
circuit
output
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/JP2016/076154
Other languages
English (en)
Japanese (ja)
Inventor
泰明 乗松
叶田　玲彦
馬淵　雄一
尊衛 嶋田
充弘 門田
祐樹 河口
輝 米川
瑞紀 中原
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Hitachi Ltd
Original Assignee
Hitachi Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Hitachi Ltd filed Critical Hitachi Ltd
Publication of WO2017098762A1 publication Critical patent/WO2017098762A1/fr
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode

Definitions

  • the present invention relates to a power conversion device, a power supply device, and a control method thereof for DC power and AC power.
  • isolation transformers are used in the power system, but it is difficult to reduce the size and weight because it is driven at a low frequency of several tens of Hz (50/60 Hz in Japan). There was a problem.
  • the SST Solid State Transformer: hereinafter simply referred to as SST
  • SST Solid State Transformer
  • the SST is composed of a high-frequency transformer and a power conversion circuit, and the output or input of the SST is AC power having the same frequency as the conventional one.
  • a high frequency is generated by a power conversion circuit (DC / DC converter or inverter) and a high frequency transformer is driven to connect an input or output to an AC power system having the same frequency as the conventional one. It replaces the isolation transformer.
  • the high-frequency transformer can be driven at a high frequency of several tens to several hundreds of kHz, so that a significant reduction in size and weight can be realized as compared with a conventional insulation transformer alone.
  • PCS Power Conditioning System
  • FIG. 2 shows a multi-level high-voltage inverter using multiple transformers as insulation transformers.
  • the multilevel in FIG. 2 shows an example of a one-stage configuration.
  • the electric power of the main circuit inverter 11 is input from the three-phase AC multiple transformer 12 and is output from the main circuit inverter 11 as a single-phase AC.
  • the main circuit inverter 11 is composed of a rectifier section R, a smoothing capacitor C2, and an inverter section In2.
  • the gate power supply 13 for supplying power for starting the semiconductor elements constituting the inverter unit In2 in the main circuit inverter 11 is provided for each of the single-phase transformer 4, the single-phase bridge rectifier circuit 16, the DC reactor 14, and the semiconductor elements.
  • the power supply circuit 17 is configured. The starting power is supplied by extracting a single phase from the output of the three-phase AC multiple transformer 12.
  • the multi-level high-voltage inverter shown in FIG. 2 insulation between the power system (several kV to several tens of kV) is achieved by the above-described connection configuration using the multiple transformer 12, the main circuit inverter 11, and the gate power supply 13. Is ensured by the multiple transformer 12. For this reason, the single-phase transformer 15 for the gate power supply 13 can be reduced to a voltage (about several hundred volts) required for one multilevel stage, and a control circuit for controlling voltage conversion can be eliminated.
  • Patent Document 1 discloses a high-voltage inverter power supply apparatus basically having the same configuration as that shown in FIG. 2, although there is a difference in that the multiple transformer 12 is a single phase.
  • the single-phase transformer 15 used in Patent Document 1 is driven at 50/60 Hz which is the same as the frequency of the power system, it is particularly difficult to reduce the size of the transformer core. Further, since the conversion to the DC voltage for the gate power supply 13 is performed by the single-phase bridge rectifier circuit 16, the voltage fluctuation is increased by the load of the control circuit, and a DC / DC converter or voltage fluctuation corresponding to the voltage fluctuation is obtained. It is necessary to add a DC reactor 14 for suppressing the power supply as necessary, and the power supply for the gate power supply 13 is enlarged.
  • the main circuit inverter 11 As a configuration of the main circuit inverter 11, a control power source configuration in which the DC voltage is stepped down by a DC / DC converter is conceivable. In this case as well, insulation between the power system can be ensured by the multiple transformer 12. Further, the transformer and the reactor necessary for the DC / DC converter are driven at the driving frequency of the DC / DC converter, so that the size can be reduced. However, in this configuration, since the DC voltage of the main circuit inverter 11 is several hundred volts, the application corresponding to the several hundred volts used for the main circuit inverter 11 is also applied to the DC / DC converter for the gate power supply 13.
  • an object of the present invention is to provide a power conversion device, a power supply device, and a control method thereof that can reduce the size of an insulating transformer and its peripheral circuits.
  • a first inverter unit that obtains a DC input and gives a high-frequency output
  • an LLC transformer that converts the high-frequency output of the first inverter unit, and an output of the LLC transformer
  • a rectifier unit for conversion a second inverter unit for converting the direct current output of the rectifier unit to alternating current, and a control circuit power supply transformer connected in parallel to the secondary circuit of the LLC transformer through the rectifier circuit
  • a power supply circuit that obtains the gate power of the semiconductor element constituting the inverter unit and the power of the control circuit that supplies the gate signal of the semiconductor element, and divides the DC voltage of the rectifier circuit and introduces it to the control circuit as the DC output of the rectifier unit.
  • This is a power conversion device and a power supply device.
  • a control method for a power conversion device or a power supply device in which the resonance frequency in the LLC transformer and the drive frequency in the first inverter unit are an operation mode in which the drive frequency and the resonance frequency are equal, and the drive frequency is greater than the resonance frequency
  • the control method is characterized in that the first inverter unit is controlled by switching between a higher operation mode and an operation mode in which the dynamic frequency is lower than the resonance frequency.
  • the present invention it is possible to provide a power conversion device, a power supply device, and a control method thereof that can reduce the size of the peripheral circuit including the isolation transformer.
  • the volume of the control circuit power transformer is reduced by increasing the frequency inputted to the control circuit power transformer, and the control circuit for voltage conversion becomes unnecessary.
  • a low-voltage element can be used in addition to the power transformer for the control circuit. As a result, the entire control circuit can be reduced in size and weight, so that the high-voltage power converter can be reduced in size and weight.
  • the figure which shows the structure of the power converter device which concerns on Example 1 of this invention The circuit diagram which shows the structure of the conventional multilevel high voltage inverter.
  • the figure which shows a voltage and electric current waveform when LLC resonant frequency is equal to a drive frequency.
  • FIG. 10 is a diagram illustrating a specific circuit configuration example of the gate power supply according to the third embodiment.
  • FIG. FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a fifth embodiment.
  • FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a fifth embodiment.
  • FIG. 10 is a diagram illustrating a specific circuit configuration example of a gate power supply according to a sixth embodiment.
  • a power conversion apparatus for a PCS of several hundred kW to several MW class whose input is DC 600 to 1000 V and whose output is connected to a high voltage system of 6.6 kV system. Assumed.
  • the main circuit 100 is configured by a series circuit of a full-bridge type LLC resonant converter 1 and a single-phase inverter 2.
  • the full-bridge type LLC resonant converter 1 is connected to a DC power source by, for example, photovoltaic power generation at an input terminal Ti, and provides an AC output to an output terminal To of the single-phase inverter 2.
  • the full-bridge type LLC resonant converter 1 includes a smoothing capacitor C1, a first inverter unit In1, an LLC transformer 3, a rectifier unit R, and a smoothing capacitor C2 from the input side.
  • the LLC transformer 3 is a transformer primary winding side of a first reactor 31, a second reactor 32 as a primary winding, and a capacitor 33 arranged in series. It is a vessel.
  • the single-phase inverter 2 includes a second inverter unit In2.
  • the first inverter unit In1 and the second inverter unit In2 have a single-phase full bridge configuration, and these inverter units In1 and In2 have their semiconductor elements configured by MOS-FETs. An example is shown.
  • a direct current input by photovoltaic power generation applied to the input terminal Ti is converted into a high frequency alternating current and adjusted to an arbitrary voltage in the LLC transformer 3.
  • the second inverter section In2 converts to a frequency (50/60 Hz) that can be connected to, for example, a commercial AC power source and outputs it to the output terminal To.
  • the power supply circuit 17 obtains power from the secondary winding 34 of the LLC transformer 3 via the control circuit power supply transformer 4.
  • the control circuit power transformer 4 includes a primary winding 41 and a secondary winding 42 electromagnetically coupled to the primary winding 41.
  • the secondary winding 42 supplies DC power to the power supply circuit 17 via the single-phase bridge rectifier circuit 16.
  • the DC power of the power supply circuit 17 is used as gate power given to the gates of the semiconductor elements Q1, Q2, Q3, and Q4 of the second inverter unit In2, or as control circuit power.
  • voltage dividing resistors R1 and R2 are connected in series on the output side of the single-phase bridge rectifier circuit 16, and the connection point potential is used as a detection signal for the DC voltage Vdc of the main circuit.
  • the LLC transformer 3 operates at a high frequency
  • the device configuration can be reduced in size.
  • the control circuit power transformer 4 also operates at a high frequency, it can be miniaturized.
  • the voltage dividing circuit for detecting the main circuit DC voltage Vdc is also lower in potential than the case where it is installed in the main circuit and is insulated, so that the circuit is miniaturized.
  • the power supply circuit 17 supplies the gate power of the MOS-FET semiconductor elements Q1, Q2, Q3, Q4 constituting the second inverter section In2 and the power supply for the gate control circuit.
  • the power supply circuit for one inverter unit In1 is not shown.
  • a power supply circuit for the first inverter unit In1 is separately installed but is not described here. The reason for this is that the voltage level to be insulated (usually about 100 volts AC) is low on the first inverter unit In1 side, so that a high degree of insulation measure that is applied to the power supply circuit 17 is not required. This is because it can be handled by technology.
  • the configuration of the main circuit 100 in FIG. 1 is a configuration in which the DC output after full-bridge diode rectification in the full-bridge type LLC resonant converter 1 is AC-output to the power system via the single-phase inverter 2.
  • a single-phase inverter 2 is assumed, but a three-level inverter configuration may be applied.
  • FIG. 1 shows a single-phase output, but three-phase is required for connection to an AC system. Therefore, practical applications suitable for the three-phase configuration of FIG. 3 are made.
  • FIG. 3 is a diagram illustrating the configuration of the three-phase power supply device using the power conversion device according to the first embodiment. In this application, the output side of the single-phase output main circuit 100 is connected in multiple stages for each of the three phases U, V, and W.
  • FIG. 1 shows a detailed configuration of the power converter
  • FIG. 3 shows the configuration of a three-phase power supply device as an example of deployment to a three-phase power system.
  • a series multi-level configuration connected in series can support high-voltage output.
  • the input voltage of each main circuit 100 is 600 to 1000 VDC, 6.6 kV can be obtained as the voltage between the three-phase lines.
  • relatively low-voltage semiconductor elements Q1, Q2, Q3, and Q4 such as 1700V, 1200V, and 650V can be used when viewed as a single-phase inverter 2.
  • the terminal voltage Vdc of the second capacitor C2 is also a voltage according to the semiconductor elements Q1, Q2, Q3, and Q4, it means that the DC capacitor C2 can use a capacitor having a lower voltage than the power system voltage. is doing.
  • the specifications and functions of the components constituting the power conversion device or power supply device shown in FIGS. 1 and 3 may be as follows, and the following effects can be exhibited.
  • the full-bridge type LLC resonant converter 1 has a DC voltage of 1000 V or less, it is assumed that a MOS-FET suitable for high-frequency driving is applied as the first inverter unit In1.
  • the switching frequency of the semiconductor element of the first inverter unit In1 is assumed to be several tens kHz to several hundreds kHz.
  • a SiC MOS-FET suitable for high withstand voltage / high frequency switching may be applied, and any other one having a similar function may be used.
  • the secondary side rectifier section R of the LLC resonant converter 1 is assumed to be smoothed by a diode.
  • Si-type diodes Si-type Schottky barrier diodes or SiC Schottky barrier diodes may be applied to reduce conduction loss, and loss can be reduced by using SiC MOS-FETs in synchronization. It does not matter if it has other similar functions.
  • the LLC transformer 3 has an insulating function from the power system voltage, and the first reactor 31 on the transformer primary winding side, the second reactor 32 as the primary winding, and the capacitor 33 are arranged in series. It is an insulation transformer called as follows. Of these circuit components, the first reactor 31 defines the leakage inductance Lr, the second reactor 32 as the primary winding defines the exciting inductance Lm of the high-frequency transformer, and the capacitor 33 defines the resonant capacitor capacitance Cr. is doing.
  • the LLC resonant transformer 3 has a configuration in which a leakage inductance Lr and a resonant capacitor capacitance Cr are connected to resonate with the exciting inductance Lm of the high-frequency transformer to achieve LLC resonance.
  • the leakage inductance Lr may be integrated in the high frequency transformer as a structure that can adjust the constant of the leakage magnetic flux in the high frequency transformer.
  • the resonance capacitor capacity Cr is assumed to use a film capacitor, but may have any similar function.
  • FIG. 1 shows an example in which the semiconductor element of the second inverter unit In2 is composed of a MOS-FET, this may be an IGBT.
  • the second inverter unit In2 may employ an IGBT because the switching frequency of the series multiplex PWM is as low as several kHz or less as a whole compared to the drive frequency of the LLC resonant converter 1.
  • the number of single-phase inverters 2 per phase is assumed to be about 8 to 6 stages in series.
  • a Y-connection configuration is assumed, but a ⁇ -connection configuration is also possible.
  • the phase voltage is 1 / ⁇ 3 with respect to the line voltage of 6.6 kV, and the DC voltage of the entire phase is based on ⁇ 2 times, so the second capacitor C2 in the case of eight stages
  • the terminal voltage Vdc is about 600 to 700V.
  • these specifications and functions may be as follows, and the following effects can be exhibited.
  • the power transformer 4 for the control circuit shown in FIG. 1 is configured to be connected to the secondary side of the LLC transformer 3, and is driven at several tens to several hundreds of kHz by the LLC control, so that several hundreds of volts on the primary side is secondary.
  • the side can be converted to several tens to several V.
  • the excitation inductance of the power transformer 4 for the control circuit has a sufficiently large value with respect to the LLC transformer 3.
  • the excitation inductance Lm of the LLC transformer 3 is several hundred ⁇ H
  • the excitation inductance of the power transformer 4 for the control circuit is set to be a value of several hundred mH or more. It is not limited.
  • control circuit power transformer 4 is driven by the LLC control, a controller for controlling voltage conversion becomes unnecessary.
  • the power transformer 4 for the control circuit can be reduced to an insulation function of about several hundred volts that is equal to or higher than the terminal voltage Vdc of the second capacitor C2.
  • the voltage dividing resistors R1 and R2 installed as the detection circuit for the DC voltage Vdc of the main circuit on the output side of the single-phase bridge rectifier circuit 16 can be applied to a low voltage circuit. The size can be greatly reduced compared to the case where it is provided.
  • each semiconductor element Q1, Q2, Q3, Q4 supplies isolated output to power supply circuits 17a, 17b, 17c, 17d for the gate and a controller (such as a microcomputer) for controlling the gate. is doing.
  • a controller such as a microcomputer
  • a configuration assuming that the gates of the semiconductor elements Q1, Q2, Q3, and Q4 have a voltage stabilizing circuit (such as a linear regulator) and a driving photoMOS, and the controller is controlled by driving a light emitting diode of the photoMOS. It is.
  • the power required for each output is assumed to be about several tens to several hundreds mW, but is not limited thereto.
  • an LLC transformer 3 is employed in the LLC resonant converter 1.
  • a control method peculiar to the LLC resonant converter 1 including the LLC transformer 3 and its effect will be described.
  • the resonance frequency is determined by the values of the excitation inductance Lm, leakage inductance Lr, and resonance capacitor capacitance Cr of the high-frequency transformer described above for the first inverter unit In1, and the resonance frequency is set to several tens to several hundreds kHz. Assumed.
  • the control of the LLC resonant converter 1 is executed by a control circuit that controls on / off of four sets of semiconductor elements constituting the full-bridge first inverter unit In1.
  • the control circuit itself has a circuit configuration that is usually performed, and a specific circuit configuration is omitted.
  • the semiconductor element is controlled as follows.
  • the upper right and lower left semiconductor elements Q9 and Q8 in FIG. 1 are the first set
  • the lower right and upper left semiconductor elements Q10 and Q7 are the second set.
  • the first conductive state in which the first set is ON and the second set is OFF, and the second conductive state in which the second set is ON and the first set is OFF are alternately formed.
  • the ON / OFF control is performed with a drive frequency having a period from the first conduction state to the first conduction state again through the second conduction state.
  • the horizontal axis represents time
  • the vertical axis represents the magnitude of voltage and current during one cycle.
  • FIG. 4 shows voltage and current waveforms when the LLC resonance frequency is equal to the drive frequency.
  • the drive frequency is controlled by a control circuit that controls on / off of the four sets of semiconductor elements constituting the full-bridge first inverter unit In1, and the drive frequency is a high-frequency transformer. This coincides with the resonance frequency determined by the values of the excitation inductance Lm, leakage inductance Lr, and resonance capacitor capacitance Cr of a certain LLC transformer 3.
  • FIG. 5 shows voltage and current waveforms when the drive frequency is lowered below the LLC resonance frequency and the boost operation is performed.
  • the secondary side voltage of the LLC transformer 3 becomes a rectangular wave voltage such as VTout.
  • This rectangular wave voltage is a waveform with a slight delay between rising and falling with respect to the power transformer 4 for the control circuit, but basically a rectangular wave is input. For this reason, since the voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary side output of the power transformer 4 for the control circuit, the voltage fluctuation is the same as in the LLC resonant converter 1. There is no need for a DC reactor.
  • FIG. 6 shows voltage and current waveforms when the drive frequency is raised above the LLC resonance frequency and the step-down operation is performed.
  • the MOS-FET In the state where the drive frequency is higher than the LLC resonance frequency, as shown in the waveform of the primary side current ITin of the LLC transformer 3, when the MOS-FET is ON, the current flowing through the MOS-FET is equal to that of the MOS-FET. Since it flows in the reverse direction through the body diode, it becomes ZVS (zero volt switching) and no switching loss occurs when it is ON. Since the current flowing through the MOS-FET does not decrease at the time of OFF, the switching loss at the time of OFF increases. However, in the case of the control operation by the PCS, the reduction in efficiency can be minimized by limiting the voltage range for the step-down control.
  • LLC resonance frequency drive frequency, LLC resonance frequency> drive frequency, LLC resonance frequency ⁇ drive frequency
  • FIG. 7 is a diagram illustrating a case example of LLC control when applied to solar power generation.
  • FIG. 7 shows the relationship between the terminal voltage Vdc of the second capacitor C2, the photovoltaic power generation output, and the driving frequency with respect to the input voltage Vin by photovoltaic power generation.
  • the magnitude of the input voltage Vin and the photovoltaic power generation output by the photovoltaic power generation is variable depending on the weather (fine weather, cloudy weather).
  • the range of the voltage input to the PCS (that is, the input voltage Vin by solar power generation) is divided into a first region D1, a second region D2, and a third region D3.
  • the drive frequency is made lower than the LLC resonance frequency by LLC control so that the terminal voltage Vdc of the second capacitor C2 is boosted so as not to be lower than the lower limit value.
  • the drive frequency is made higher than the LLC resonance frequency by LLC control so that the terminal voltage Vdc of the second capacitor C2 is stepped down so as not to exceed the upper limit value.
  • the drive frequency is made equal to the LLC resonance frequency by the LLC control and is made constant, thereby enabling high-efficiency operation.
  • the region may be set based on the upper and lower limits of the terminal voltage Vdc of the second capacitor C2.
  • the second threshold value Vin2 that defines the upper limit region is set to be equal to or higher than the maximum output point voltage in normal sunny weather. It is good. By doing so, the probability of the step-down operation in the third region is reduced.
  • the voltage on the secondary side of the LLC transformer 3 by this LLC control is VTout. Since a rectangular wave is input to the power transformer 4 for the control circuit, a voltage corresponding to the turn ratio with respect to the terminal voltage Vdc of the second capacitor C2 is output to the secondary output, so that the same as the LLC converter There is no voltage fluctuation and no DC reactor is required.
  • connecting the power transformer 4 for the control circuit to the secondary side of the LLC transformer 3 reduces the dielectric breakdown voltage, eliminates the need for voltage conversion drive control, suppresses voltage fluctuations due to the load, and reduces the size of the transformer due to higher frequencies. It becomes possible to satisfy the requirements, and the entire power supply device can be reduced in size and weight.
  • Example 3 shows a specific circuit configuration example of the gate power supply 13 of FIG.
  • a gate power supply circuit 17a, 17b, 17c, 17d and a single-phase bridge rectifier circuit are provided for each of the semiconductor elements Q1, Q2, Q3, Q4 of the second inverter unit In2.
  • 16a, 16b, 16c, and 16d are provided, and a control power supply circuit 17j and a single-phase bridge rectifier circuit 16j are provided.
  • the control circuit power transformer 4 is configured so that the secondary side of the control circuit transformer 4 has multiple outputs by secondary windings 42a, 42b, 42c, 42d, and 42j, so that each single-phase bridge rectifier circuit 16a, 16b, 16c, 16d and 16j.
  • the control circuit transformer 4 shows an example of a one-to-n connection method in which the primary side is a single winding and the secondary side is a plurality of windings.
  • voltage dividing resistors R1 and R2 are connected in series to the output side of the single-phase bridge rectifier circuit 16j, and the connection point potential is introduced into the control power supply circuit 17j as a detection signal of the DC voltage Vdc of the main circuit and used for control. ing.
  • the configuration of the third embodiment in FIG. 8 uses the secondary side of the control circuit transformer 4 as multiple outputs, so that it can be used for the voltage detection of the DC voltage Vdc and the microcomputer in the control power supply circuit 17j shown in the first embodiment.
  • the power source for driving the gates of the semiconductor elements Q1, Q2, Q3, and Q4 for driving the inverter is generated.
  • the gates of the semiconductor elements Q1, Q2, Q3, and Q4 have a voltage stabilizing circuit (such as a linear regulator) and a driving photo MOS, and the microcomputer controls the driving element by driving the light emitting diode of the photo MOS.
  • a voltage stabilizing circuit such as a linear regulator
  • the microcomputer controls the driving element by driving the light emitting diode of the photo MOS.
  • the inverter driving semiconductor elements Q1, Q2, Q3, and Q4 may use IGBTs instead of MOS FETs. Since the inverter section In2 for output has a switching frequency of the serial multiplex PWM as low as several kHz or less as a whole compared with the drive frequency of the LLC resonant converter, problems such as heat generation are small even when the IGBT is applied.
  • FIG. 10 shows a configuration when the control circuit power transformer 4 has a single output. It is assumed that the present invention is applied to a case where a plurality of control circuit power transformers 4a, 4b, 4c, 4d, and 4j can be individually arranged on the secondary side of the LLC transformer 3 so that downsizing in a distributed arrangement becomes possible. is doing.
  • FIG. 11 shows a configuration when the control circuit power transformer 4 is divided into four outputs for the gate power circuits 17a, 17b, 17c and 17d and a single output for the control power circuit 17j.
  • the turns ratio of the four outputs for the gate power supply circuits 17a, 17b, 17c and 17d is the same, and the turn ratio of one output for the control power supply circuit 17j is the four outputs for the gate power supply circuits 17a, 17b, 17c and 17d.
  • the configuration of FIG. 11 is a configuration in which the voltage detection of the DC voltage Vdc is divided on the side of the gate power supply circuit 17d and supplied to the control power supply circuit 17j. It is assumed that an error such as an influence of a voltage drop of the rectifying element of the control circuit power transformer 1 is reduced by dividing the voltage from a voltage higher than that on the control power circuit 17j side to improve detection accuracy.
  • the control method in the third embodiment is basically the same as that described above.
  • the third embodiment is also the control of the LLC resonant converter 1 and the duty control of 50% duty with a dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.
  • Example 3 the waveform when the voltage is changed is the same as in FIGS. 4, 5, and 6. Even if the load is changed, VTout is a rectangular wave, so the Vdc turns ratio is independent of the load. A control voltage can be generated.
  • control circuit transformer 4 by connecting the control circuit transformer 4 to the secondary side of the LLC transformer 3, downsizing due to a reduction in dielectric strength, no need for voltage conversion drive control, suppression of voltage fluctuation due to load, downsizing of the transformer due to higher frequency It is possible to satisfy all of the above, and the power supply as a whole can be reduced in size and weight.
  • FIG. 12 shows the configuration of the power conversion apparatus according to the fourth embodiment.
  • the configuration of the third embodiment is a configuration in which the number of semiconductor elements constituting the rectifier unit R of the LLC resonant converter 1 is halved.
  • the voltage width input to the primary-side LLC transformer 3 is 1 ⁇ 2 of the full-bridge configuration of FIG. 1, but can be similarly adjusted by the turn ratio of the LLC transformer 3.
  • the control method of Example 3 is a frequency control of Duty 50% performed by controlling the LLC resonant converter 1 and giving a dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.
  • the waveforms when the voltage is changed are the same as those in FIGS. 4, 5, and 6, and even if the load is changed, VTout is a rectangular wave. Therefore, the second capacitor C2 does not depend on the load.
  • the control voltage can be generated with the turn ratio of the terminal voltage Vdc.
  • control circuit transformer 1 is connected to the secondary side of the LLC transformer 2 to reduce the size due to a decrease in the withstand voltage, no need for voltage conversion drive control, the suppression of voltage fluctuation due to the load, and the downsizing of the transformer due to the higher frequency. It is possible to satisfy all of the above, and the power supply as a whole can be reduced in size and weight.
  • FIG. 13 and 14 show the configuration of the power conversion apparatus according to the fifth embodiment.
  • the configuration of the fifth embodiment assumes a system in which output is connected to three phases of U, V, and W phases in a single-stage configuration, and a motor and a pump are driven to output.
  • the output of the power transformer 3 for the control circuit is assumed to be a total of seven power circuits, including six for driving the semiconductor elements and one for the controller in accordance with the three-phase driving of the U, V, and W phases.
  • the number is not limited as long as the configuration has the same function.
  • FIG. 14 shows a configuration in which the number of outputs of the control circuit transformer 4 is reduced. Since the source voltages of the semiconductor elements Q2, Q4, and Q6 are the same, the number of outputs of the control circuit transformer 1 can be reduced by using the same output.
  • the control method of Example 5 is frequency control of Duty 50% performed by control of the LLC resonant converter 1 and provided with dead time. It is assumed that the resonance frequency is determined by the values of the excitation inductance Lm, the leakage inductance Lr, and the resonance capacitor capacitance Cr, and is set to several tens to several hundreds kHz.
  • the waveform when the voltage is changed is the same as that of FIGS. 4, 5, and 6. Even if the load is changed, VTout is a rectangular wave, so the second capacitor C2 does not depend on the load.
  • the control voltage can be generated with the turn ratio of the terminal voltage Vdc.
  • FIG. 15 shows a configuration in which the control circuit power transformer 4 of FIG. 1 is divided and distributed in series.
  • the primary winding and the secondary winding are configured to 1: n.
  • n 5 controls are provided as the control circuit power transformer 4 on the secondary side of the LLC transformer 3.
  • Circuit power transformers 4a, 4b, 4c, 4d, and 4j are connected in series. This connection form is applied when the control circuit power transformer 4 is divided into power supply circuits 17a, 17b, 17c, 17d and dispersed in series, and can be miniaturized in a distributed arrangement. Is assumed.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Dc-Dc Converters (AREA)

Abstract

L'invention porte sur un dispositif de conversion de courant dans lequel la taille d'un transformateur d'isolation et de circuits périphériques de ce dernier peut être réduite, sur un dispositif d'alimentation électrique, et sur leur procédé de commande. L'invention concerne un dispositif de conversion de courant et un dispositif d'alimentation électrique, comprenant : une première unité d'onduleur qui obtient une entrée de courant continu et fournit une sortie haute fréquence; un transformateur LLC qui convertit la tension de la sortie haute fréquence de la première unité d'onduleur; une unité de redresseur qui convertit la sortie du transformateur LLC en un courant continu; une seconde unité d'onduleur qui convertit la sortie de courant continu de l'unité de redresseur en un courant alternatif; et un circuit d'alimentation électrique qui obtient, par l'intermédiaire d'un circuit de redressement d'un transformateur d'alimentation de circuit de commande connecté en parallèle avec le circuit secondaire du transformateur LLC, de l'énergie électrique pour un circuit de commande qui fournit une énergie électrique de grille pour des éléments à semi-conducteurs constituant la seconde unité d'onduleur et un signal de grille pour lesdits éléments à semi-conducteurs. Une tension continue du circuit de redressement est divisée et est introduite dans le circuit de commande à titre de sortie de courant continu de l'unité de redresseur.
PCT/JP2016/076154 2015-12-10 2016-09-06 Dispositif de conversion de courant, dispositif d'alimentation électrique, et leur procédé de commande Ceased WO2017098762A1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
JP2015-240964 2015-12-10
JP2015240964A JP2019041427A (ja) 2015-12-10 2015-12-10 電力変換装置、電源装置およびその制御方法

Publications (1)

Publication Number Publication Date
WO2017098762A1 true WO2017098762A1 (fr) 2017-06-15

Family

ID=59012986

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/JP2016/076154 Ceased WO2017098762A1 (fr) 2015-12-10 2016-09-06 Dispositif de conversion de courant, dispositif d'alimentation électrique, et leur procédé de commande

Country Status (2)

Country Link
JP (1) JP2019041427A (fr)
WO (1) WO2017098762A1 (fr)

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2021527386A (ja) * 2018-06-15 2021-10-11 ルノー エス．ア．エス．Ｒｅｎａｕｌｔ Ｓ．Ａ．Ｓ． Ｄｃ／ｄｃ変換器の入力電圧周波数を制御するための方法

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR102349935B1 (ko) * 2020-11-27 2022-01-12 주식회사 팩테크 반도체 변압기 전원공급장치 및 그의 동작 방법

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2007151224A (ja) * 2005-11-24 2007-06-14 Shindengen Electric Mfg Co Ltd インバータ電源装置
JP2013172466A (ja) * 2012-02-17 2013-09-02 Fuji Electric Co Ltd 電力変換装置及びこれを用いた系統連系システム

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2007151224A (ja) * 2005-11-24 2007-06-14 Shindengen Electric Mfg Co Ltd インバータ電源装置
JP2013172466A (ja) * 2012-02-17 2013-09-02 Fuji Electric Co Ltd 電力変換装置及びこれを用いた系統連系システム

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2021527386A (ja) * 2018-06-15 2021-10-11 ルノー エス．ア．エス．Ｒｅｎａｕｌｔ Ｓ．Ａ．Ｓ． Ｄｃ／ｄｃ変換器の入力電圧周波数を制御するための方法
JP7258054B2 (ja) 2018-06-15 2023-04-14 ルノー エス．ア．エス． Ｄｃ／ｄｃ変換器の入力電圧周波数を制御するための方法

Also Published As

Publication number Publication date
JP2019041427A (ja) 2019-03-14

Similar Documents

Publication Publication Date Title
JP2017070047A (ja) 電源装置
US9887638B2 (en) Power conversion apparatus with frequency operation change based on input voltage
JP6172277B2 (ja) 双方向ｄｃ／ｄｃコンバータ
US7333348B2 (en) DC-DC converter
JP4995277B2 (ja) 双方向ｄｃ／ｄｃコンバータ
KR101811153B1 (ko) 직류 전원 장치 및 냉동 사이클 기기
JP6289618B2 (ja) 電力変換装置
JP5575235B2 (ja) 電力変換装置
JP6496608B2 (ja) 電力変換装置及びその電力変換制御手段
JP2016220482A (ja) 電源装置
US10193464B2 (en) DC-DC converter
JP7121971B2 (ja) 三相ａｃ－ｄｃコンバータ
US20070069581A1 (en) AC to DC converter circuit
WO2017098763A1 (fr) Dispositif d'alimentation électrique et procédé de commande de charge initiale de ce dernier
JP2016046973A (ja) 共振型ｄｃ−ｄｃコンバータ
JP2017147812A (ja) 電源装置およびその初充電制御方法
WO2017098762A1 (fr) Dispositif de conversion de courant, dispositif d'alimentation électrique, et leur procédé de commande
KR20170064076A (ko) 자속 공유형 대전압용 전원 공급 장치
WO2010098486A1 (fr) Convertisseur cc-cc
JP3934654B2 (ja) Ｄｃ−ｄｃコンバータ
Alshammari et al. Optimization of the Operating Frequency of a Bidirectional Synchronous H6 Inverter
JP4811786B2 (ja) 分散電源装置の電力変換装置

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 16872662

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 16872662

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: JP