WO2014045223A2

WO2014045223A2 - Method and device for measuring signals generated by a particle and/or ionizing radiations detector

Info

Publication number: WO2014045223A2
Application number: PCT/IB2013/058670
Authority: WO
Inventors: Francesca ZOCCA; Alberto PULLIA
Original assignee: Universita degli Studi di Milano; Instituto Nazionale di Fisica Nucleare INFN
Current assignee: Universita degli Studi di Milano; Instituto Nazionale di Fisica Nucleare INFN
Priority date: 2012-09-19
Filing date: 2013-09-19
Publication date: 2014-03-27
Anticipated expiration: 2015-03-19
Also published as: WO2014045223A3; ITMI20121560A1

Description

METHOD AND DEVICE FOR MEASURING SIGNALS GENERATED BY A PARTICLE AND/OR IONIZING RADIATIONS DETECTOR

DESCRIPTION

Field of the invention

The present invention relates to a method and a device for measuring signals generated by a detector of ionizing particles and/or radiations.

Background art

Circuits for measuring signals generated by a detector of ionizing particles and radiations are already known in the art, which generally consist of circuits that provide a measurement of the quantity of charge received at their input from the detector, while operating within a linear region of their input/output characteristic. As the quantity of charge received from the detector increases, said circuits may however reach the saturation region of their input/output characteristic. In such conditions, the quantity of charge received in saturation conditions, or "excess charge", is not measured and is lost and/or stays at the output electrode of the detector, thus altering the total measurement.

Circuits are also known which are adapted to remove the "excess" physical charge released by the detector, but these circuits are not equipped with a system for measuring such charge and cannot therefore be used for measuring signals that saturate the preamplifier. One example of such circuits is described in [1]: article by V. Radeka, "Overload Recovery Circuit for Charge Amplifiers', IEEE Trans. Nucl. Sci., vol. NS-17, no. 1 , pp. 269-275, 1970.

A technique and an algorithm are also known which are adapted to measure, with high resolution, signals exceeding a predetermined amplitude by means of an amplitude-time conversion device made out of discrete components, as described in [2]: article by A. PULLIA ET AL: " Active reset of digitized preamplifiers for ionizing-radiation sensors", IEEE TRANS. NUCL. SCIENCE, vol. 51. no. 3. I June 2004 (2004-06-01), pp. 831-835, or article by F. Zocca, A. Pullia, D. Bazzacco, G. Pascovici, "A Time-Over-Threshold Technique for Wide Dynamic Range Gamma-Ray Spectroscopy with the AGATA Detector", IEEE Trans. Nucl. Sci., vol. 56, n. 4, 2009. It should however be noted that (i) the device does not remove the "excess" physical charge released by the detector and cannot measure it, (ii) the first stage of the circuit (integrator) must necessarily be made out of discrete components because it uses relatively high power voltages (e.g. +-12V) as a strategy for providing a wide dynamic range, (iii) the data analysis algorithm requires two external "flash" ADC converters and a Personal Computer for each measurement channel. Due to the reasons (ii) and (iii), the device is unsuitable for highly integrated multichannel implementations (e.g. CMOS).

Integrated circuits are also known which are adapted to remove the "excess" physical charge released by the detector, wherein said charge is measured by means of an external apparatus, as described, for example, in [3]: article by A. Pullia, F. Zocca, "Extending the dynamic range of a charge-preamplifier far beyond its saturation limit: a 0.35um CMOS preamplifier for germanium detectors", 2009 IEEE Nucl. Sci. Symp., DOI: 10.1109/NSSMIC.2009.5402152. However, such circuits require an external apparatus for converting the signal into numerical form and processing it, typically a "flash" ADC converter and a Personal Computer for each measurement channel, and are therefore unsuitable for highly integrated multichannel implementations. Moreover, a simplified algorithm is used for measuring the charge, which produces significant (overestimate) errors due to tails of previous signals. They are not therefore suitable for high-resolution spectroscopy of signals generated, for example, by natural radionuclides.

Other circuits are also known which are equipped with an amplitude-time or amplitude-frequency converter for measuring the charge released by photodetectors and photomultipliers typically coupled to scintillators, as described, for example, in [4]: DE 10 2009 049304 A1 , and [5]: WO 2010/109347 A1. Said circuits, however, which have essentially been conceived for medical imaging applications, are not suitable for high-resolution spectroscopic measurements with ionizing-radiation detectors, e.g. gamma spectroscopy, which typically requires a line resolution of the order of 0.1 %. It is well known in this field that, in fact, electronic components such as the resistor "R1 " in [4] or the bias current generator "312" in [5], which are permanently connected to the input node of each circuit, introduce electronic noise that considerably degrades the circuit's performance, thus making high-resolution spectroscopy impossible.

Summary of the invention

The present invention aims at providing a method and a device for measuring signals generated by a detector of ionizing particles and/or radiations, which are adapted to overcome all the above-mentioned drawbacks.

The invention provides a technique and a device suitable for highly integrated multichannel implementations, i.e. with many independent replicas of the same circuit, hence using VLSI (Very Large Scale Integration) or higher technology, for measuring signals from detectors of ionizing particles and radiations, which allow measuring events with high resolution and linearity even when the preamplifier reaches or goes beyond the saturation condition. The dynamic range of spectroscopic measurements is thus hugely increased (by more than one order of magnitude), and events become measurable which would otherwise be lost. The detectors' whole intrinsic spectroscopic range can thus be used. The device operates automatically for signals which are higher than a predetermined threshold value. Under-threshold or low-energy events are still measured in the conventional manner, with no performance losses.

The present invention uses the known concept according to which the physical signal charge released by the detector is not lost when the preamplifier saturates, and can be measured by removing it at a constant rate and by measuring the time required for removing it, as described in [3]. The invention makes use of a processing circuit block that reconstructs the charge measurement by taking into account the presence of any tails from previous events, and provides a signal whose amplitude is proportional to the physical signal charge. The measurement can then be completed by using a simple PHA (Pulse Height Analyzer). The device is suitable for highly integrated implementations (e.g. CMOS).

The circuit of the invention allows to simultaneously remove and measure the physical signal generated by the detector, and is also effective when the preamplifier is strongly saturated. It also directly handles the physical charge generated by the detector (by removing it physically and by measuring the time required for removing it), as opposed to handling a secondary electric quantity dependent thereon (which might be subject to errors); moreover, it carries out a processing that takes into account the possible presence of a tail due to a previous event, correcting the measurement result accordingly.

The present invention aims at improving the measurement range of known spectrometers of ionizing radiations and particles, particularly those using highly integrated, multichannel front-ends, thus making it possible to measure the signals of the detectors even when they bring the preamplifier into saturation. In such conditions, the preamplifier signal no longer grows with the signal released by the detector (which is an electric charge at the electrode). Furthermore, the electric pulse is significantly distorted because the preamplifier exits the linear region, and conventional spectrometers either stop working or generate false lines and artefacts in the generated spectra. It is a common thought that no useful results can be obtained in these operating conditions, and that any attempts will have to deal with a very non-linear system.

The present invention allows to overcome this technical prejudice by introducing a technique that produces an output signal whose amplitude continues to grow linearly with the signal released by the detector even when the preamplifier enters a (deep) saturation state, thus ensuring a linear relationship between the detector signal and its measurement even in such conditions.

The dynamic range of spectroscopic measurements is thus hugely increased (by more than one order of magnitude), and events become measurable which would otherwise be lost. The detectors' whole intrinsic spectroscopic range can thus be used. At the same time, low-energy events under a predetermined threshold value can still be measured in a conventional manner, with no resolution losses. The advantage offered by this technique is particularly apparent when the electronic front-end is integrated and the output voltage swing is intrinsically limited by the scaling-down phenomenon.

From the measurement of the output signal of the detector, one can then go back to the measurement of the energy associated with the particles or radiations incident on the detectors.

The present invention is also applicable to measurements of output signals of capacitive feedback amplifiers, whenever a wide dynamic range and a high measurement resolution are required.

It is one object of the present invention to provide a method for measuring signals generated by a detector of ionizing particles and/or radiations, said signals being made available as electric charges at the outputs of the detector, the method being adapted for use within a multichannel environment with VLSI or higher technology and comprising the steps of: removing all said electric charges at a constant rate from said outputs of the detector, by using a type of preamplification in both linear and saturation conditions; measuring the time necessary for removing all said electric charges; converting said time into a measurement of the quantity of said electric charges; defining an event of a succession of events as characterized by all said electric charges, said measurement being referred to said event; eliminating from said measurement a contribution relating to electric charges of all previous events.

It is another object of the present invention to provide a device adapted to measure signals generated by a detector of ionizing particles and/or radiations, said signals being made available as electric charges at the outputs of the detector, said device being adapted for use in a multichannel device with VLSI or higher technology and comprising: means for removing all said electric charges at a constant rate from said outputs of the detector, all said electric charges being referred to an event of a succession of events, said means for removing comprising a preamplifier operating in both linear and saturation conditions; means for measuring the time necessary for removing all said electric charges and for converting said time into a measurement of the quantity of said electric charges; means for eliminating from said measurement a contribution relating to electric charges of all previous events.

It is a particular object of the present invention to provide a method and a device for measuring signals generated by a detector of ionizing particles and/or radiations as set out in the appended claims, which are an integral part of the present description.

Brief description of the drawings

Further objects and advantages of the present invention will become apparent from the following detailed description of a preferred embodiment (and variants) thereof and from the annexed drawings, which are only supplied by way of non- limiting example, wherein:

Figure 1 shows a circuit block diagram of a device according to the prior art;

Figure 2 shows the circuit block diagram of the device according to the present invention;

Figure 3 shows a more detailed circuit diagram relating to one example of embodiment of the device of Figure 2;

Figure 4 shows, by way of example, the various signals generated by the device of the present invention in response to a signal that brings the preamplifier into saturation;

Figures 5, 6, 7 and 8 are graphs relating to some examples of time trends of signals produced by the device of the present invention and to one example of high-resolution spectroscopic measurement;

Figure 9 shows a variant of an output equivalent circuit diagram for a capacitive feedback amplifier.

In the drawings, the same reference numerals and letters identify the same items or components.

Detailed description of some embodiments of the invention

With reference to Figs. 1 and 2, the dashed line indicates the equivalent output circuit RU of a detector of ionizing particles and/or radiations, which is of a per se known type and is therefore not shown in the drawing.

Particle detectors are known, e.g. of the silicon or germanium semiconductor type, or of the gas-ionization type. The particles may be, for example, X or alpha or beta or gamma rays, or light or heavy ions.

When the ionizing particle or radiation hits the detector, it interacts with the detector's material, which releases electric charges, electrons or holes (in the case of a semiconductor-type detector), or ions (in the case of an ionizing-radiation detector). Through the effect of the electric fields which are generated in the detector's material, these charges move and reach the output electrodes of the detector itself. Therefore, the measurement of the particles' energy is deduced from the quantity of produced charge that reaches the electrodes.

The equivalent output circuit RU of the detector of ionizing particles and/or radiations thus comprises an ideal current generator l_Q, which represents the accumulated charge, and a capacitive output impedance C_DE_T, which is generally dependent on the detector's geometry.

The current generator is of the impulsive type, and represents movements of electric charges that generate narrow pulses whose area Q5(t) defines the energy associated with the charges.

The output signal, i.e. an electric charge Q, is brought to a charge preamplifier stage (within the dashed lines PA), which acts as a negative-feedback active integrator. Therefore, the preamplifier essentially comprises an operational amplifier in negative feedback configuration OP, to which a capacity CF is applied across the output terminal and the negative input. A high-value resistor R_F, the function of which will be described below, is preferably arranged in parallel to the capacity CF; the negative input also receives the output signal Q of the detector. In linear conditions, the amplifier outputs a voltage which is proportional to the quantity of charge available at the input. Beyond the linear range, i.e. for higher levels of input charge quantity, the amplifier would enter a saturated state, in which the output voltage would tend to become constant as the charge increases, thus altering the measurement result. This problem is particularly important for integrated circuits, which are characterized by low levels of output voltage dynamics.

It is however known that the electric charge Q released by the detector in correspondence to an event is not lost even when the preamplifier saturates [3]. The excess charge Q', i.e. the charge exceeding the value that brings the preamplifier into saturation, stays at the electrode U of the detector without scattering because there is no DC path to ground where it can flow. The excess charge Q' may produce a voltage jump at the electrode U in accordance with the capacitor law applied to the detector itself (VDE_T=Q/CDET), but it will not be lost. The primary information thus remains intact despite the loss of preamplifier's linearity. A known circuit as described in [3], capable of measuring this charge (prior art), is shown in Fig. 1 . It comprises a dual-threshold Schmitt trigger comparator (T) and a controlled current generator l_REs.

By way of example, let us assume that the charge Q released by the detector is negative (electrons), e.g. the output is taken at the detector's anode.

The preamplifier PA will respond with a positive signal at its output U_PA. As soon as the preamplifier PA enters saturation (the output signal tends to become constant), the Schmitt trigger T switches, thereby generating a control signal that activates the controlled current generator l_REs- Then the generator starts draining electrons away from the electrode (incoming current corresponding to outgoing electrons) with a constant current value, since it is an ideal current generator. It is essential that the average number of electrons removed per time unit (i.e. the mean current) is constant, reproducible and stable against variations of environmental parameters such as temperature, humidity, etc. This is attained by using per se known design techniques to stabilize the current of the controlled current generator l_REs-

As the excess charge Q' is removed from the electrode U of the detector, at some point the preamplifier exits the saturation condition, re-entering a linear operating state. The charge removal process stops automatically when the signal of the preamplifier UPA returns to zero (baseline), i.e. when the total charge Q is depleted. At this instant, the Schmitt trigger T switches back and deactivates the controlled current generator l_REs-

It is now possible to obtain a measurement of the total charge to be removed Q, i.e. the signal generated by the detector, from the time ΔΤ taken for removing it. These two quantities are proportional: the higher the source charge, the longer it takes to remove it all (at a constant removal rate), in accordance with the very definition of the electric current concept. The time measurement ΔΤ is therefore an indirect but reliable measurement of the charge generated by the event (signal charge) and there is excellent linearity between the time ΔΤ and the charge Q, despite the fact that during most of the process the preamplifier operates in conditions of light or heavy non-linearity. The total time ΔΤ refers to the sum of the contributions given by the operation of the preamplifier PA in both the saturation region and the linear region. The time ΔΤ, and hence the signal charge, are measured by sending the comparator signal V_TRG to a fast flash-type ADC and by analyzing said signal, thus obtaining its time duration, by means of a computer or a DSP (Digital Signal Processor).

However, this known solution described in [3] has two drawbacks:

(i) it requires a complex external electronic block for each readout channel used for measuring ΔΤ, e.g. a fast ADC and a DSP, and therefore it is not compatible with highly integrated multichannel implementations,

(ii) it does not take into account the fact that the measurement of the total charge at the electrode U is not yet a correct measurement of the total charge associated with the event. As a matter of fact, prior to the event being measured a residual charge might already have been present at the electrode, associated with a previous event having a certain amplitude. Without an appropriate correction of this effect, it is not possible to perform high-resolution spectroscopic measurements. A processing algorithm that might solve this problem is described in [2], but the device proposed therein is substantially different, in that it does not remove the physical charge of the detector and, for several reasons already mentioned above, it is unsuitable for highly integrated multichannel implementations.

The present invention overcomes these problems by means of a device capable of generating a pulse whose height represents (regardless of the sign) the correct measurement of the total charge associated with the event, wherein the contribution due to the tail of previous events is eliminated without needing either a flash ADC or a DSP. The circuit bock diagram of the device and an example of generated signals are shown in Figures 2 and 4.

The circuit block diagram according to the present invention is shown in Fig. 2. It comprises two operational amplifiers OP1 and OP2, a Schmitt trigger T, two constant current generators I_RE_S and IT_AC, a low-pass filter PB, and some passive components. The measurement is completed with the help of a pulse height analyzer PHA_H, which may be external to the system. The figure also shows a traditional measurement chain consisting of a forming amplifier SH and a pulse height analyzer PHAL, which can be used for measuring low-energy signals (under the threshold of the comparator T). In correspondence to an above-threshold signal from the detector that brings the preamplifier into saturation, the device not only removes the charge Q as already described, but it also constructs a signal V₀ whose height represents the correct signal measurement, which takes into account the presence of a residual charge associated with a previous event.

The algorithm used for measuring the signal, including the correction term that takes into account the tail of a previous event, is of the following type:

y = h^*AT-m(V1-V2)+ c (alg) where y indicates the corrected final measurement, ΔΤ is the time width of the comparator signal, V1 and V2 are baseline voltage measurements immediately before the beginning and immediately after the end of the signal (i.e. immediately before the comparator switching instant and immediately after the reverse switching of the same comparator); h and m are constant positive parameters, and c is an offset term. Hereafter it will be shown that the absolute value of the peak voltage of Vo is proportional to y.

It can be demonstrated that when the algorithm (alg) is calibrated, h and m are bound to each other by the relation

where l_REs is the current used for removing the excess charge and CF is the value of the capacity in the preamplifier PA.

The value of y in the relation (alg) contains a contribution which is proportional to the time ΔΤ, and a corrective contribution which is given by the second term of the relation itself.

Through suitable conversion factors, y may also express the time value or even the charge quantity Q or the energy associated with the particles or radiations incident on the detector.

A further parameter of the algorithm is the averaging operation that may be carried out while measuring V1 and V2. It has been observed that, by mediating these measurements over a predefined time interval, e.g. of the order of 200ns, it is possible to significantly improve the quality of the measurement.

With reference to Figure 3, there is described one example of embodiment of that part of the circuit of Figure 2 which comprises the Schmitt trigger comparator T, the controlled current generator l_REs, with CMOS technology. For completeness' sake, the drawing also show the further portion of the circuit of Fig. 2 that comprises the block PA and further downstream circuitry, which can similarly be implemented by using CMOS technology. For example, the switch S may be implemented by means of a CMOS transistor.

The controlled current generator l_REs consists of a constant current generator IR, and a current switch D controlled by the comparator T. The switch D1 sends the current to the input node Q of the charge preamplifier PA (in conditions of measurement activation) or to ground (in idle conditions). The choice of using a current switch allows to keep the current generator always on, both during the measurement and in idle conditions, thereby promoting its thermal stabilization and improving its stability and reproducibility.

The current generated by the generator l_REs strictly depends on the resistors Ri and R₂ according to the relation:

IRI = V_cc (1-a)/Ri

where a is the resistive partition factor

The Schmitt trigger comparator T is implemented by means of an operational amplifier stage.

The threshold voltage V_th of the comparator T is:

th = Vcc RTI I/(RTH ⁺ RT12)

Therefore, it can be easily sized by appropriately choosing the values of the resistors R-m, RTI2-

Referring back to Figure 2, the circuit part that generates the pulse V₀ employs a current generator ITAC controlled by the comparator T and implemented by using a technique similar to that already described in regard to the generator l_REs- The current generator ITAC supplies current to the capacitive negative reaction amplifier OP2, which is implemented in a manner similar to the preamplifier PA. The amplifier OP2 receives the signal UPA through a low-pass filter PB and a capacity of C/K value.

There is also a switch S that can short-circuit the feedback capacity C of the amplifier OP2; the switch S is controlled by the signal of the comparator T.

The signal of the comparator T is such that the switch S is normally closed, thus short-circuiting the capacity C. When the comparator is turned on, i.e. when an above-threshold signal is detected, the current ITAC is deviated towards the virtual mass of the amplifier OP2. At the same time, the switch S opens and the current ITAC will accumulate charge on the capacitor C, thus generating a negative voltage ramp at the output of OP2.

The behaviour of the circuit in response to this input is similar to that of a single- ramp Wilkinson converter, i.e.:

y = -t ^*l_TAc/C + os O where y is the output voltage due to this contribution, os represents an offset term. In order to make it not very likely for the amplifier OP2 to enter saturation during the active phase, it is preferable to set:

ITAC/C « IRES/CF (oo)

The overall output signal of the amplifier OP2 also depends, however, on the signal of the preamplifier PA, which propagates through the block PB and reaches the amplifier OP2 through the capacity C/K. The amplifier OP2, as observed from this input, is an inverting amplifier, the gain of which, as is known, is given by the ratio between the capacities C/K and C. Assuming for simplicity that the comparator's switching time is t=0, we will obtain:

y = -1/K ^* [ V_pre_fiit(t)- V_pre_fiit(0) ] (**) where V_pre_fiit(t) is the signal of the preamplifier PA at the output of the block PB. At time 0 the signal must be null because the switch S was closed for t<0.

According to the effects superposition principle, the overall output signal of OP2 will then be given by the sum of the signals (^*) and (*^*). At the final time, i.e. t=AT, we will thus have:

Υτοτ(ΔΤ) = -ΔΤ ITAC/C - 1/K [ V_pre_fiit(At)- V_pre_f_ilt(0) ] + os (x) The relation (x) corresponds, regardless of the sign, to the algorithm (alg). The function of the block PB for filtering the voltage levels inputted to OP2 helps reduce the noise of the algorithm, as aforementioned.

It will then be sufficient to evaluate the absolute value of the height of y-τοτ, i.e. the total voltage detected at the output of the block OP2, at the comparator switching time t = At to obtain a measurement inclusive of tail correction. Considering that (o) must be verified in (alg), in (x) we should have

from which, considering (oo), we will obtain the following sizing rule

K = (IRES / C_F) / (ITAC / C) » 1

The amplitude of the output signal of the amplifier OP2 is then measured, whose peak value, taken as a modulus, provides the measurement to be considered, which may possibly be converted into a time or a charge quantity value by using suitable conversion factors. For example, the signal may be measured by means of a known analyzer called PHA (Pulse Height Analyzer).

The following remark should be made in regard to the saturation condition of the preamplifier PA. This condition occurs suddenly at the beginning of the measurement phase, caused by the large quantity of charge Q, de facto cancelling the time of linear behavior. Towards the end of the measurement procedure, instead, as the charge Q decreases through the effect of the intervention of the current generator l_REs, the preamplifier PA returns to linear operation in a time that, in this case, is added to the time during which PA was saturated, and contributes to the measurement of the total time, which is determined by the activation time of the generator current, which stays active until the end of this phase. Both contributions are therefore measured.

The circuit may be implemented by means of a 0.35pm, 3.3V/5.5V BiCMOS chip with Austria Micro System C35 technology.

A typical sizing value for K is 20, which means having a voltage range for Υ_ΤΟτ which is 20 times broader than that of the preamplifier PA. In some cases it might be useful to choose even higher values of K. By way of example, the following is a set of parameter values compatible with a CMOS implementation of the circuit: C_F = 0.4pF; IRES = 1 UA; C = 10pF; K=20; C/K = 0.5pF; l_TAc = 1.25uA

The circuit solution proposed herein is particularly simple, does not require the use of a fast flash-type ADC or a DSP or a PC, can be easily provided in monolithic form, and is therefore suitable for highly integrated multichannel implementations. Figure 5 shows a set of some of the signals produced by a circuit prototype in correspondence to events with increasing amplitude, which is useful to understand the operation of the device according to the invention. It should be noted that in this example the signs of all current generators indicated in Fig. 2 (e.g. detector signal current and current l_REs) are inverted. This makes the device suited to treating signals consisting of holes as opposed to electrons, i.e. signals generated by detector's "cathode" electrodes instead of "anode" electrodes.

The graph in the upper part represents the output U_PA of the preamplifier PA. The graph in the lower part represents the logic signal of the Schmitt trigger (T), which activates the charge removal current in the presence of above-threshold signals. The signals (1) to (4) have a low value, and the circuit responds in the conventional manner with step-like waves, meaning that in these cases the trigger T does not switch and the preamplifier PA remains in the linear operation region. In these cases, OP2 does not go on and takes no measurements. The output signal Up_A can thus be taken from the auxiliary output AUX and supplied to a known measurement circuit, e.g. of the "Spectroscopy Amplifier" type, which can take measurements of charge quantities in a per se known manner.

The signals of the group (5), (6),...,(23) have high values and cause the triggers T1 or T2 to switch. In these cases, the preamplifier PA saturates, thereby activating the measurement by 0P2. The circuit of the present invention thus removes the excess charge at a constant rate (i.e. constant current).

All the signals (7), (8), ...(23) have a level far beyond the amplitude saturation point of PA; in fact, they all "flatten out" at about -2V. Actually, also the signals (5), (6) are above the threshold, i.e. their amplitude is such that the trigger T will switch, though for a shorter time.

It should however be observed that in all these cases they regularly stretch out over time in a way proportional to the amplitude of the event. It should also be noted that the triggers' switching start condition occurs very rapidly and PA saturates right away, whereas the condition of end of saturation and return into the linear region shows a slower progress. At any rate, the triggers will switch back at the end of the total time inclusive of the saturation and linear region times, thus allowing to measure the total duration of the switching time, which will then be taken into account for measuring the total quantity of charge Q, which de facto is the time ΔΤ of the relation (alg).

According to one aspect of the invention, the time width of the triggers' switching signals, i.e. the width of the rectangular signals shown in the lower part of Figure 5, (the logic signals that activate the charge removal current generator) is then measured.

It is thus also possible to measure all wide signals, regardless of whether saturation is entered and of the extent thereof.

Figure 6 shows the experimental relation between event amplitude (i.e. the signal charge quantity supplied by the detector) and the time width of the above- mentioned signals.

This width is referred to in the drawing as "reset time".

Note the linearity of this relation (error smaller than 1/1 ,000) in a very high range of values.

The abscissa indicates signals with up to 100MeV of energy deposited in the detector (a huge value compared with the typical limit value of just a few MeV's), which bring the preamplifier into very deep saturation. In spite of this, the relation between amplitude and "reset time" stays perfectly linear.

It must also be pointed out that the technique of the invention implies clear advantages in terms of a significant reduction of the so-called "dead time", i.e. the delay with which a new measurement can take place, which depends on the time necessary for the charge of the ongoing measurement to be depleted. For example, Fig. 6 shows that, even in the presence of a considerable quantity of charge Q (in the case of the signal (23), a charge of 5.5 pC equivalent to an energy deposit of 100 MeV in a germanium detector; see also the axis of abscissas in Fig. 6), the dead time is very short and does not exceed 9 ps. For under-threshold signals, instead, the complete discharge of the input charge Q occurs in a very long time, e.g. not less than 1 ms for the signals (1) to (4). This is because this charge cannot be drained quickly, since the current generator l_RES, is not active, so that it stays "hung" on the input of the preamplifier PA. Discharge can take place through the resistor R_F (Fig. 2), which has a very high value (e.g. 1 GQ). This resistor in parallel to the capacity C_F has a high value to avoid altering the integration function of the amplifier OP.

Figure 7 shows a spectroscopic measurement of detector signals (Counts) simulated through the use of a pulser. The detector is simulated by a 15pF capacitor. Note that the x axis (signal charge amplitude or equivalent energy for germanium detectors) is logarithmic. The measurement was taken by using conventional techniques up to an equivalent energy of approx. 10 MeV (under- threshold signals) and by using the technique of the present invention for equivalent energies beyond 10 MeV (above-threshold signals). This latter measurement region is indicated in the drawing as "Fast reset mode". The figure also shows an enlargement of the low-energy signal region, which highlights a 1keV fwhm line span, compatible with the specifications of high-resolution gamma spectroscopy. It should be noted that the overall linear range of high-resolution spectroscopic measurements is exceptionally wide, from 5 keV to 0.7GeV, i.e. 103dB, and that the additional range corresponding to the "Fast reset mode" region provides an improvement of the overall dynamic range by more than one order of magnitude.

Figure 8 shows the sequence of three close signals having a very different amplitude. The first one (71) corresponds to a 40MeV equivalent energy deposit in germanium detectors (e.g. accelerated alpha particles), while the second one (72) and the third one (73) correspond to 1MeV and 2 MeV energy signals, respectively, which are typically found in nuclear gamma spectroscopy. From the above description, it can be understood that the system can also resolve 1 and 2 MeV signals, even though they are preceded by a much wider signal. In fact, the high-value signal (71) of Fig. 8 causes the trigger T to switch and is "resolved" in a short time (approx. 3 ps), thus allowing the next "under-threshold" signals (72) (73) to be treated without altering the measurement thereof.

If conventional techniques only were used, the second and third events could not be observed, due to the persisting saturation of the preamplifier caused by the first event.

On the other hand, this example also shows that a correction of the second term of the relation (alg) is required in order to take into account the still existing under- threshold signals, e.g. (73), which would alter the next measurement because of a very long discharge time.

The above-described example of embodiment may be subject to variations without departing from the protection scope of the present invention, including all equivalent designs known to a man skilled in the art.

The circuit of the present invention is also applicable to measurements of output signals or capacitive feedback amplifiers, whenever a wide dynamic range and a high measurement resolution are required. With reference to Figure 9, there is shown an equivalent output circuit of a per se known capacitive feedback amplifier. It comprises a step-type ideal voltage generator VQ and a capacitive series output impedance C_DET, generally dependent on the type of amplifier. The quantity of output charge Q is given by Q = VQ * C_DET

This charge Q is measurable, although its value is high, and is such as to bring the preamplifier PA (Fig. 2) into saturation conditions.

The advantages deriving from the application of the present invention are apparent.

The device of the invention, which is suitable for highly integrated implementations, allows to substantially increase (by more than one order of magnitude) the dynamic range of spectroscopic measurements, thus allowing events to be measured which would otherwise be lost. The detectors' whole intrinsic spectroscopic range can thus be used. At the same time, low-energy events can be measured in a conventional way with no performance losses.

The device finds application, for example, in the following fields: Nuclear Physics Experiments with germanium detectors (gamma spectroscopy and measurement of charged particles)

Subnuclear Physics Experiments (search for rare decays, WIMPs particles, measurement of high-energy cosmic radiations)

Astrophysical applications (X spectroscopy, gamma spectroscopy, cosmic radiations, WIMPs particles)

Applications in the field of spectroscopy and biomedical imaging with wide- dynamic range detectors (e.g. high-resolution X spectroscopy and gamma spectroscopy with a single detector)

Monitoring ionizing radiations in nuclear plants

Monitoring ionizing radiations in hospitals

Monitoring ionizing radiations for anti-terrorism purposes

From the above description, those skilled in the art will be able to produce the object of the invention without introducing any further construction details.

Claims

1. A method for measuring signals generated by a detector of ionizing particles and/or radiations, said signals being made available as electric charges at the outputs of the detector, the method being adapted for use within a multichannel environment with VLSI or higher technology and comprising the steps of:

- removing all said electric charges at a constant rate from said outputs of the detector, by using a type of preamplification in both linear and saturation conditions;

- measuring the time necessary for removing all said electric charges;

- converting said time into a measurement of the quantity of said electric charges;

- defining an event of a succession of events as characterized by all said electric charges, said measurement being referred to said event;

- eliminating from said measurement a contribution relating to electric charges of all previous events.

2. A device adapted to measure signals generated by a detector of ionizing particles and/or radiations, said signals being made available as electric charges at the outputs of the detector, said device being adapted for use in a multichannel apparatus with VLSI or higher technology and comprising:

- means for removing all said electric charges at a constant rate from said outputs of the detector, all said electric charges being referred to an event of a succession of events, said means for removing comprising a preamplifier operating in both linear and saturation conditions;

- means for measuring the time necessary for removing all said electric changes and for converting said time into a measurement of the quantity of said electric charges;

- means for eliminating from said measurement a contribution relating to electric charges of all previous events.

3. A device according to claim 2, wherein said means for removing all said electric charges at a constant rate from said outputs of the detector are so configured as to comprise:

- an input amplifier (PA) adapted to be connected to said outputs of the detector in order to receive said electric charges, said input preamplifier being characterized by a linear operating region and a saturation operating region; - means for generating constant current (IRES, T), adapted to supply said constant current to the input of said input amplifier (PA) in said conditions of saturation and linear region, for all the time necessary for removing all said electric charges.

4. A device according to claim 4, wherein said means for generating constant current (l_REs, T) comprise:

- at least one generator of uninterrupted constant current (IR);

- comparison means (T) adapted to switch upon detecting said condition of saturation at the output of the preamplifier (PA);

- at least one switch (D) controlled by said comparison means in such a way as to shunt the current generated by said at least one generator of constant current towards the input of said input amplifier (PA) in said conditions of saturation and linear region.

5. A device according to claim 3, wherein said means for measuring the time necessary for removing all said electric charges and for converting said time into a measurement of the quantity of said electric charges comprise means adapted to calculate the following relation:

y = h*AT- - m(V1 - V2) + c

where y indicates said measurement as an output voltage value; ΔΤ is said time necessary for removing all said electric charges, V1 and V2 are voltage measurements at the beginning and at the end of said time ΔΤ; h, m and c are constant parameters.