WO2008062146A1

WO2008062146A1 - Position sensor

Info

Publication number: WO2008062146A1
Application number: PCT/GB2006/004384
Authority: WO
Inventors: Victor Evgenievich Zhitomirsky
Original assignee: Sagentia Ltd
Current assignee: Sagentia Ltd
Priority date: 2006-11-23
Filing date: 2006-11-23
Publication date: 2008-05-29
Anticipated expiration: 2009-05-23

Abstract

A capacitive position sensor is provided having a plurality of electrodes spaced along a measurement path. Excitation circuitry is provided to generate and apply excitation signals to some of those electrodes and detection circuitry is provided to receive and process signals obtained from others of the electrodes. The detection circuitry includes circuitry to subtract some of the detection signals from each other to remove common mode signals.

Description

POSITION SENSOR

The present invention relates to capacitive position sensors and to parts thereof. The invention has particular, although not exclusive relevance to capacitive position sensors that can be used to sense the level of a liquid within a container and to transducers for use in such sensors.

It is known to provide capacitive liquid level sensors for sensing the level of a liquid within a container. In one such prior system, as disclosed in, for example, US4021707 by Ehref and Hoyer, the capacitive sensor uses two vertically spaced conductors and measures the change in capacitance between the conductors as the space between them fills with liquid. This type of capacitive sensor has the disadvantage that the capacitance of the conductors is not only dependent on the level of the liquid, but also to variations in the dielectric constant of the liquid. Consequently, if the composition of the liquid varies, which can be the case with automotive fuels for example, the liquid level sensor can become inaccurate. In order to compensate for the uncertainty in the dielectric constant of the liquid, Ehref and Hoyer teaches to use a reference capacitor that is fully immersed in the liquid. The change in the capacitance of the reference capacitor from its dry value is used to calibrate dielectric parameters of the liquid. Figure 1 schematically illustrates the form of electrodes used in this prior art type of sensor. As shown, the sensor includes three electrodes 9, 11 and 12 which are connected to excitation and processing circuitry 22 via a connection interface 20. The electrodes 9 and 12 provide the main sensor signal that will vary with the level of the liquid 103 in the container (not shown) and the electrode 11 is used to provide the reference capacitor. However the system described by Ehref and Hoyer requires storing values of the two measurement capacitors in their dry stage and thus does not allow the use of simple and cheap electronics. The present invention aims to provide an alternative capacitive position sensor that can be used to sense the level of a liquid or that can be used to sense the presence of a dielectric or metallic object in the vicinity of the sensor. Preferred embodiments of the sensor overcome or at least reduce the problems with the prior art sensor discussed above.

According to one aspect, the present invention provides a capacitive position sensor comprising: first and second sets of electrodes; excitation circuitry operable to generate and to apply a first excitation signal to an excitation electrode of said first set of electrodes and a second excitation signal to an excitation electrode of said second set of electrodes; and detection circuitry operable: i) to receive a first detection signal from a reception electrode of said first set of electrodes which first detection signal varies with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the first set of electrodes; ii) to receive a second detection signal from a reception electrode of said second set of electrodes, which second detection signal varies with the position along said measurement path of said inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the second set of electrodes; iii) to subtract the first detection signal from the second detection signal to generate a difference signal which varies with the position along said measurement path of said inhomogeneity; and iv) to process said difference signal to determine the position of said inhomogeneity along said measurement path. In one embodiment each set of electrodes are symmetrically disposed about an axis that is parallel to said measurement path, with the electrodes of the first set being disposed on one side of said axis and the electrodes of the second set being disposed on an opposite side of said axis. In another embodiment, the electrodes of the first set are symmetrically disposed about the measurement path relative to the electrodes of the second set, such that there is mirror like symmetry between the electrodes of the first and second sets.

In a preferred embodiment the excitation electrode of the first set has mirror symmetry with the excitation electrode of the second set and the reception electrode of the first set has mirror like symmetry with the reception electrode of the second set. Such an arrangement reduces errors caused with low levels of liquid that is inclined relative to the measurement path.

In another embodiment, the excitation or reception electrode of each set extends into a dead zone of the sensor beyond a measurement range of the sensor, wherein each set of electrodes includes a second reception or excitation electrode positioned adjacent said excitation or reception electrode in said dead zone. The electrodes of each set which extend into said dead zone are preferably elongate in a direction transverse to said axis, in order to minimise the size of the dead zone.

In one embodiment each set of electrodes comprises two pairs of electrodes, one pair of electrodes being provided for position measurement and one pair of electrodes being provided to correct for changes in permittivity surrounding the sensor. The electrodes of each pair that are provided for position measurement can be substantially identical and the electrodes of each pair that are provided to correct for changes in permittivity may not be identical. A reference capacitor is preferably associated with each pair that is provided to correct for changes in permittivity, to compensate for the electrodes of that pair not being identical.

The electrodes may be formed from conductive tracks on or in a non-conductive substrate and a groove may be provided in the substrate between Hie excitation electrode and the reception electrode of said first and second sets. In one embodiment the groove is a through groove. In another embodiment the groove is a blind groove.

In a preferred embodiment the detection circuitry determines an amplitude measure obtained using the difference signal and updates the position only if said amplitude measure is above a predetermined threshold value. Similarly, the detection circuitry preferably determines a phase measure using the determined difference signal and updates the determine position only if said phase measure is above a predetermined threshold value. In this way, errors caused when the amplitude or phase measurements are small can be avoided.

In one embodiment the surface area of the excitation electrodes and/or the reception electrodes are designed so that there is a linear relationship between the position of the inhomogeneity and a measurement parameter obtained the difference signal. According to another aspect, the invention provides a capacitive position sensor comprising: a first electrode that extends along a measurement path over a measurement range; first and second pairs of electrodes arranged along the measurement path adjacent said first electrode such that electrodes of each pair are interleaved along said measurement path with the electrodes of the other pair; excitation circuitry operable to generate and to apply first and second excitation signals of opposite polarity to one of said first electrode and said first and second pairs of electrodes; and detection circuitry operable: i) to receive signals from the other one of said first electrode and said first and second pairs of electrodes; ii) to process said signals to obtain a first detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said first pair and a second detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said second pair, which first and second detection signals vary with the position of an inhomogeneity along said measurement path; and iii) to process said first and second detection signals to determine the position of said inhomogeneity along said measurement path. In one embodiment the first pair of electrodes is provided for position measurement and the second pair of electrodes is provided to correct for changes in permittivity surrounding the sensor. In another embodiment, the electrodes of the first pair are substantially identical and the electrodes of the second pair are not identical.

According to another aspect, the invention provides a capacitive position sensor comprising: first and second subsets of electrodes, the electrodes of which extend along a measurement path of the sensor; excitation circuitry operable to generate and to apply an excitation signal to said first subset of said electrodes; and detection circuitry operable: i) to receive first and second signals from said second subset of said electrodes, which first and second signals vary with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the electrodes of the first and second subsets; and ii) to process said first and second signals to determine the position of said inhomogeneity along said measurement path; wherein said electrodes are formed on or in a non-conducting substrate and wherein a blind groove is provided between the electrodes of said first and second sub-sets.

The blind groove may be provided in the substrate on which the electrodes are mounted or by providing a through groove on the substrate and by providing a backing layer on the base of the substrate.

These and various other features and aspects of the invention will become apparent from the following detailed description of embodiments which are described with reference to the accompanying figures in which:

Figure 1 is a schematic diagram illustrating a set of electrodes forming part of a prior art liquid level sensor;

Figure 2 is a schematic diagram illustrating a set of electrodes forming part of a liquid level sensor according to one embodiment of the present invention;

Figure 3 schematically illustrates an electrical equivalent circuit illustrating one way to connect the electrodes shown in Figure 2 to excitation and detection circuitry used to detect the liquid level;

Figure 4A is a plot illustrating the way in which one signal obtained from the electrodes shown in Figure 2 varies with the liquid level;

Figure 4B is a plot illustrating the way in which another signal obtained from the electrodes shown in Figure 2 varies with the liquid level; Figure 4C is a plot illustrating the locus of points obtained by plotting the two signals obtained from the electrodes against each other for different liquid levels;

Figure 5 is a block diagram illustrating the main components of exemplary excitation and detection circuitry that can be used with the sensor shown in Figure 2;

Figure 6 is a block diagram illustrating the main components of a ratiometric calculator forming part of the detection circuitry shown in Figure 5;

Figure 7A is a block diagram illustrating the main components of a phase detector forming part of the ratiometric calculator shown in Figure 6;

Figure 7B illustrates the form of a logic circuit for zero crossing detection forming part of the circuitry shown in Figure 7A; Figure 7C is a plot illustrating the way in which the pulse width of the PWM signal output by the circuitry shown in Figure 7B varies with the time delay between the detected zero crossings;

Figure 8 schematically illustrates an electrical equivalent circuit illustrating an alternative way to connect the electrodes shown in Figure 2 to excitation and detection circuitry used to detect the liquid level;

Figure 9 is an electrical equivalent circuit illustrating an alternative way of connecting the electrodes shown in Figure 2 to the excitation and detection circuitry in which the electrodes previously connected to the excitation circuitry are connected to the detection circuitry and the electrodes previously connected to the detection circuitry are connected to the excitation circuitry;

Figure 10 is a block diagram illustrating excitation and detection circuitry that can be used to drive the electrodes in the inverted manner illustrated in Figure 9;

Figure 11 is a block diagram illustrating an alternative arrangement of the excitation and detection circuitry that can be used to drive the electrodes shown in Figure 2 in the inverted manner illustrated in Figure 9;

Figure 12 is a block diagram illustrating the excitation and detection circuitry that can be used in a further alternative for driving the electrodes shown in Figure 2 in the inverted manner illustrated in Figure 9; Figure 13 is a signal and circuit diagram illustrating in more detail the way in which the circuitry shown in Figure 12 operates to generate a signal whose phase varies with the position being measured;

Figure 14 is a block diagram illustrating the excitation and detection circuitry that can be used in a further alternative for driving the electrodes shown in Figure 2 in the inverted manner illustrated in Figure 9;

Figure 15, which comprises Figures 15A to 15D₅ illustrates the way in which the excitation pulse trains are generated by the excitation circuitry shown in Figure 14;

Figure 16, which comprises Figures 16A to 16H, illustrates the effects of offsets in the measurements on calculated phase angles and hence on detected positions;

Figure 17 illustrates an alternative set of electrodes that can be used in place of the electrodes illustrated in Figure 2;

Figure 18 is an electrical equivalent circuit illustrating one way to connect the electrodes shown in Figure 17 to excitation and detection circuitry used to detect the liquid level;

Figure 19 is an electrical equivalent circuit illustrating an alternative way of connecting the electrodes shown in Figure 17 to the excitation and detection circuitry, in which the electrodes previously connected to the excitation circuitry are connected to the detection circuitry and the electrodes previously connected to the detection circuitry are connected to the excitation circuitry;

Figure 2OA is a plot illustrating the way in which one of the signals obtained from the electrodes shown in Figure 17 varies with the liquid level;

Figure 2OB is a plot illustrating the way in which a second one of the signals obtained from the electrodes shown in Figure 17 varies with the liquid level;

Figure 2OC is a plot illustrating the locus of points obtained by plotting the two signals obtained from the electrodes against each other for different liquid levels;

Figure 21, which comprises Figures 21 A to 2 IH, illustrates the effects of offsets in the measurements obtained using the electrodes illustrated inn Figure 17 on calculated phase angles and hence on detected positions;

Figure 22A schematically illustrates the form of an alternative electrode assembly that can be used to sense position instead of the electrodes illustrated in Figure 17;

Figure 22B is a cross section along the line A-A of the electrode assembly illustrated in Figure 22A;

Figure 23 A is a plot illustrating the way in which one of the signals obtained from the electrodes shown in Figure 22 varies with the liquid level; Figure 23B is a plot illustrating the way in which a second one of the signals obtained from the electrodes shown in Figure 22 varies with the liquid level; Figure 23 C is a plot illustrating the locus of points obtained by plotting the two signals obtained from the electrodes shown in Figure 22 against each other for different liquid levels;

Figure 23D is an error plot between the measured phase and the corresponding liquid level, illustrating a non-linearity of the sensor;

Figure 24A is a plot illustrating the locus of points obtained by plotting the two signals obtained from the electrodes shown in Figure 22 against each other for different liquid levels and for two liquids having different dielectric constants;

Figure 24B illustrates a phase difference plot obtained for the electrodes illustrated in Figure 22, when two different liquids having different dielectric constants surround the electrodes;

Figure 24C illustrates a phase difference plot obtained when a first offset is introduced into the sensor illustrated in Figure 22;

Figure 24D illustrates a phase difference plot obtained when a second offset is introduced into the sensor illustrated in Figure 22;

Figure 25A schematically illustrates an alternative sensor in which the electrodes shown in Figure 22 are mounted on a layer of plastic;

Figure 25B is a cross section along the line A-A of the electrode assembly illustrated in Figure 25A;

Figure 26A is a plot illustrating the locus of points obtained by plotting the two signals obtained from the electrodes shown in Figure 25 against each other for different liquid levels and for two liquids having different dielectric constants;

Figure 26B is a plot illustrating the difference in the measured phase obtained over the measurement range for the two different liquids plotted in Figure 26A;

Figure 27A illustrates a further alternative sensor in which the electrodes illustrated in Figure 22 are embedded within a layer of plastic;

Figure 27B is a cross section along the line A-A of the electrode assembly illustrated in Figure 27A;

Figure 28 schematically illustrates the form of an alternative set of electrodes that can be used to sense position instead of the electrodes illustrated in Figure 2;

Figure 29A is a plot illustrating the way in which one of the signals obtained from the electrodes shown in Figure 28 varies with the liquid level;

Figure 29B is a plot illustrating the way in which a second one of the signals obtained from the electrodes shown in Figure 28 varies with the liquid level;

Figure 29C is a plot illustrating how the measured phase obtained using the signals illustrated in Figures 27A and 27B varies with the liquid level;

Figure 3OA schematically illustrates the form of an alternative set of electrodes that can be used to sense position instead of the electrodes illustrated in Figure 2; and

Figure 3OB is a cross section along the line A-A of the electrode assembly illustrated in Figure 3OA;

Figure 31 schematically illustrates the form of an alternative set of electrodes that can be used to sense position instead of the electrodes illustrated in Figure 2; and

Figure 32 schematically illustrates one way in which the electrodes illustrated in Figure 31 may be connected to excitation circuitry and detection circuitry of the sensor.

Overview

The embodiments of the invention enable the detection of the spatial position of an inhomogeneity in the dielectric constant (permittivity) in the space around a sensor. The sensor can be used to detect the position of the interface between a liquid (or other fiowable material such as grain or powder) and the air, the position and displacement of a dielectric or a metallic object adjacent to the sensor, the movement of an air bubble inside a liquid (e.g. in a level gauge), etc.

A detailed description will now be given of a number of different electrode structures that can be used as part of a capacitive position sensor. A number of alternative ways of connecting to the electrodes and processing the signals obtained from the electrodes will also be given. These embodiments will be described in relation to a liquid level sensor for illustration only.

Sensor Head A

Figure 2 schematically illustrates a capacitive liquid level sensor head 101 that is based on the sensor shown in Figure 1, except it includes two symmetrically arranged (back to back) sets of electrodes that are used for sensing the level of the liquid 103 within the container (not shown). As shown, the electrodes of the first set 21 are positioned to the left of a vertical axis of symmetry that lies parallel to the measurement path of the sensor and the second set 23 are positioned to the right of this axis of symmetry. The first set of electrodes 21 includes three electrodes 9, 11 and 12 which are connected to the excitation and processing electronics 22 via terminals 1 * to 4* and A* on the connection interface 20. The second set of electrodes 23 also includes three electrodes 10, 15 and 16 which are connected to the excitation and processing electronics 22 via terminals 5* to 8* and B* on the connection interface 20. Electrodes 9 and 10 extend over the entire depth of the container (defined between levels O and Q); electrodes 12 and 16 extend over an operating range of the sensor (defined between levels P and Q); and electrodes 11 and 15 extend over an inoperative range or dead zone (defined between levels O and P) of the sensor. Both electrodes 11 and 12 are positioned adjacent and form capacitors with electrode 9 and similarly electrodes 15 and 16 are positioned adjacent and form capacitors with electrode 10. As shown, the electrodes of the first set 21 are positioned to the left of a vertical axis of symmetry that lies parallel to a measurement path of the sensor and the electrodes of the second set 23 are positioned to the right of this axis of symmetry.

Figure 2 also shows that the connection interface 20 includes a pair of reference capacitors C₁₁ and C₁₂ which are connected between terminals A* and 2* and 4* respectively. As will be described in more detail below, reference capacitor C₁₁ is provided to (try to) nullify the dry capacitance (i.e. in the absence of any liquid in the container) of the capacitor formed by electrodes 9 and 11. Similarly, reference capacitor C₁₂ is provided to (try to) nullify the dry capacitance of the capacitor formed by the electrodes 9 and 12. A similar pair of reference capacitors C₁₅ and C₁₆ is also provided on the connection interface 20 between terminals B* and 6* and 8* respectively. Reference capacitor Ci₅ is provided to nullify the dry capacitance of the capacitor formed by electrodes 10 and 15 and reference capacitor C₁₆ is provided to nullify the dry capacitance of the capacitor formed by electrodes 10 and 16.

Figure 3 schematically illustrates the way in which the two sets of electrodes 21 and 23 are connected, in this embodiment, to excitation circuitry 24 and detection circuitry 26 within the excitation and processing electronics 22 via the connection interface 20. As shown, the excitation circuitry 24 is operable to apply a positive excitation voltage (+V₁) to electrode 9 and to the two reference capacitors C₁₁ and C₁₂; and to apply a negative excitation voltage (-V₂) to electrode 10 and to the two reference capacitors C₁₅ and C₁₆. As shown, electrode 11 and reference capacitor C₁₁ are connected to a first differential amplifier circuit 68^"1 of the detection circuitry 26. The capacitance of capacitor C₁₁ is chosen so that in the absence of any liquid in the container, the two signals input to the amplifier circuit 68^"1 will be the same and will therefore cancel each other out. However, in the presence of the liquid, the signal obtained from electrode 11 will be different to that obtained from reference capacitor C₁₁ and as a result a signal will be output from amplifier 68^~l.

In a similar manner, electrode 12 and reference capacitor C₁₂ are connected to differential amplifier circuit 68^"2; electrode 15 and reference capacitor C₁₅ are connected to differential amplifier circuit 68^~3; and electrode 16 and reference capacitor C₁₆ are connected to differential amplifier circuit 68^{^4}. As shown in Figure 3, the output from amplifier circuit 68^'3 is subtracted from the output from amplifier circuit 68^'1 by differential amplifier circuit 68^"5 to generate a first output signal V¹. The first output signal (V¹) will therefore vary with the capacitive coupling between electrodes 11 and 9 and 15 and 10. As the excitation signal applied to electrode 9 is of opposite polarity to the excitation signal applied to electrode 10, this will mean that the first output signal

(V¹) will represent the sum of the capacitive couplings between electrodes 11 and 9 and electrodes 15 and 10. In a similar fashion, the output from amplifier circuit 68^A is subtracted from the output from amplifier circuit 68^'2 by differential amplifier circuit 68^~6 to generate a second output signal V^π. The second output signal (V¹¹) will therefore represent the sum of the capacitive couplings between electrodes 12 and 9 and electrodes 16 and 10. As electrodes 11 and 15 and 12 and 16 are symmetrically arranged (with mirror symmetry) along the sensor, the signals obtained from these electrodes will vary in the same way as the level of the liquid 103 varies within the container. Therefore, twice the signal levels will be obtained compared to the signal levels obtained from the sensor design of Figure 1. Further, because of the differential combination of the signals obtained from the electrodes, any common mode signals will be removed such as might occur in a noisy environment.

Figures 4A and 4B illustrate the way in which these two output signals V¹ and V^π vary with the level (x) of the liquid 103 in the container. As shown, when there is no liquid in the container, the two output signals are at zero. As the level of liquid in the container increases between levels O and P, the capacitive coupling between electrodes 9 and 11 and electrodes 10 and 15 will increase approximately linearly whilst the capacitive coupling between electrodes 9 and 12 and electrodes 10 and 16 will remain constant. Therefore, as the level of liquid in the container increases between levels O and P the first output signal (V¹) will increase linearly and the second output signal (V¹¹) will remain at zero. Once the level of the liquid in the container is above level P, the capacitive coupling between electrodes 9 and 11 and electrodes 10 and 15 will remain constant whilst the capacitive coupling between electrodes 9 and 12 and electrodes 10 and 16 will vary linearly. Therefore, between levels P and Q, the first output signal (V¹) will remain constant and the second output signal will increase linearly.

The detection circuitry 26 can then determine the level of the liquid in the container using a ratiometric calculation of the first and second output signals. There are various different ratiometric calculations that can be performed, but in the preferred embodiments the value of a phase angle is determined between V¹ and V^π that monotonically varies with the level of the liquid in the container between levels P and Q. This phase angle is illustrated in Figure 4C, which is a plot of V¹ against V^π. As illustrated in Figure 4C, the phase angle (Φ) varies from zero degrees when the level of the liquid is at level P to approximately sixty degrees when the level of the liquid is at level Q. This phase angle can be determined from:

Φ = arctan

V¹ O)

The determined phase angle is then mapped, using an appropriate lookup table, to the actual liquid level which is then displayed on the level signal display 44. In this embodiment, and as will be described in more detail below, in order to reduce spurious phase errors when the liquid level is between levels O and P, amplitude and phase thresholds are used so that no phase measure is output until the signal amplitude is above the amplitude threshold and the determined phase angle is above the phase threshold. The amplitude threshold is illustrated in Figure 4C by the dotted circle 29 and the phase threshold is defined by the arc 28 labelled Φ*.

Excitation And Detection Circuitry - 1

As will be described in more detail below, the sensor design described above allows simple and cheap excitation and processing electronics 22 to be used to excite the electrodes and to process the two output signals to determine the liquid level in the container.

Figure 5 is a block diagram illustrating one form of the excitation circuitry 24 and the detection circuitiy 26 that can be used with the sensor head 101 shown in Figure 2. The capacitive sensor can operate over a range of excitation frequencies. However, for the electrode design shown in Figure 2, mutual capacitances are rather small and so the impedance of the sets of electrodes connected to the connection interface 20 will be relatively large. It is therefore preferable to operate at frequencies above IMHz or even 1 OMHz, in order to decrease the value of the impedance associated with these mutual capacitances. It is possible for the excitation circuitry 24 to generate and apply AC excitation signals to electrodes 9 and 10. However, this is not the most preferable option as this requires the use of expensive electronics and especially expensive differential amplifier circuits 68. Therefore, instead of using AC excitation signals, a pulse generator 60 is used that generates square form pulses of excitation signal (having a pulse width of the order of several hundred nanoseconds) with sharp rising and falling fronts (having widths of the order of several nanoseconds). Fast fronts of the excitation pulses correspond to the equivalent frequency of tens of MHz. By measuring signals induced at the fronts of the square form pulses, it is possible to further reduce the impedance associated with the mutual capacitances discussed above. The pulses of the drive voltages can have amplitudes in the range 0.1 V to 10V. The pulse repetition frequency that is used is less important as it only affects the overall signal levels that will be processed by the detection circuitry 26. Therefore, in order to minimise power consumption of the sensor, the pulse repetition frequency should be set to the minimum necessary to obtain the required level of signal to noise ratio for the measurements to be made.

As shown in Figure 5, the pulsed excitation signal 31 generated by the pulse generator 60 is applied to terminal A* of the connection interface 20 (and thus corresponds to +V₁) and is applied to an invertor 32 for generating an inverted version 33 of the pulsed excitation signal which is applied to terminal B* of the connection interface 20 (and thus corresponds to -V₂). In this embodiment, it is not essential for V₁ and V₂ to have the same magnitude, and therefore an inexpensive invertor circuit 32 can be used. As a result of the connections in the connection interface 20, the non-inverted pulsed excitation signal 31 (+Vi) is applied to electrode 9 and the inverted pulsed excitation signal 33 (-V₂) is applied to electrode 10. As a result of the application of these excitation signals to the electrodes 9 and 10, electric fields are generated around the sensor head 101 which capacitively couple with the electrodes 11, 12, 15 and 16. The amount of capacitive coupling between the electrodes 9 and 10 and the individual electrodes 11, 12, 15 and 16 depends on the dielectric constant (permittivity) in the space between those electrodes and hence on the level of the liquid in the container.

Figure 5 shows the differential amplifier circuits 68^'1 to 68^"6 forming part of the detection circuitry 26. As shown, the signals obtained from the terminals of the connection interface 20 are connected to the corresponding differential amplifier 68 through respective switches 62^"1 to 62^"8 which are all opened and closed in synchronism with the pulses of the excitation signal 31. As shown in Figure 7, in this embodiment, each of the differential amplifier circuits 68 includes a storage capacitor 64 (on which voltages from the corresponding terminals are accumulated which will vary slowly over time reflecting the movement of the liquid level along the length of the sensor head 101) and a differential amplifier 66. As a result of the use of the pulsed excitation signals 31 and 33 and the use of the switches 62, the detection circuitry 26 can use relatively cheap low frequency electronic components to amplify and process the signals obtained from the sensor head 101.

As shown in Figure 5, the signals (V¹ and V^π) output from the differential amplifiers 68^"5 to 68^"6 are applied to a ratiometric calculator 40 which determines the position (x) of the liquid level using the above arctangent function. This calculation may be performed using a microprocessor. However, microprocessors are expensive and therefore, in this embodiment, the ratiometric calculator 40 uses analogue processing techniques to generate a value that continuously varies with the liquid level (x). Figure 6 is a block diagram illustrating the way in which the ratiometric calculator 40 performs the calculation in this embodiment. As shown, the ratiometric calculator 40 includes a low frequency generator 48 that generates a low frequency (of the order of a few kHz) periodic signal. This low frequency periodic signal is applied directly to a first amplitude modulator 50^"1 and a 90 degree phase shifted version of the low frequency periodic signal is applied to a second amplitude modulator 50^"2. As shown in Figure 6, the 90 degree phase shift is obtained by passing the low frequency periodic signal though a 90 degree phase shifter 52. The amplitude modulator 50^"1 amplitude modulates the received low frequency periodic signal with the first output signal (V¹) obtained from the differential amplifier circuit 68^'5; and the amplitude modulator 50^'2 amplitude modulates the phase shifted low frequency periodic signal with the second output signal (V¹¹) obtained from the differential amplifier circuit 68^'6. The two amplitude modulated signals output from the amplitude modulators 50 are then added together by an adder 54. The output from the adder 54 can therefore be represented (approximated) by:

V¹ cos{2πβ)- V" sin(2πft) (2)

where f is the frequency of the low frequency periodic signal generated by the generator 48, which can be rewritten as:

Where A is a normalised amplitude term defined by:

As can be seen from equation (3) above, the output from the adder 54 includes a sinusoidal signal at the low frequency (f) whose phase is proportional to the ratiometric arctangent function defined in equation (1) given above. Therefore, in this embodiment, the ratiometric calculator 40 calculates equation (1) by using a phase detector 56 to detect the phase of this signal output from the adder 54 relative to the low frequency signal generated by the low frequency generator 48. The output of the phase detector 56 will therefore be a value that continuously changes with the level of the liquid within the container. However, as mentioned above, in order to reduce spurious phase measurements obtained when there is little or no liquid in the container, the peak amplitude (A) of the signal output from the adder 54 is compared with a threshold value by the amplitude threshold detector 55 and the signal obtained from the adder 54 is only passed to the phase detector (through switch 57) when the detector 55 determines that the peak amplitude (A) is above the threshold value. Similarly, the phase detector is arranged so that it does not output a phase measurement until the phase is greater than the phase threshold.

Figure 7A is a block diagram illustrating the main components of the phase detector 56 used in this embodiment. As shown, the AC signal 41 from the adder 54 is amplified by an amplifier 42 and then input to a first comparator 43^"1 where it is compared with a threshold voltage. Similarly, the AC signal 45 from the low frequency generator 48 is input to a phase shifter 58 which phase shifts the reference signal 45 by the above described phase threshold, to generate a phase shifted signal 59 which is input to a second comparator 43^"2 where it is also compared with the same threshold voltage. As will be seen below, by shifting the phase of the reference signal 45 in this way the phase detector 56 will automatically apply the desired phase thresholding discussed above. As shown, the square wave signals 46^"1 and 46^"2 output by the comparators 43 are shifted in time relative to each other by a value Δt corresponding to the phase shift between AC signal 41 and AC signal 59. As AC signal 59 has been shifted in phase by TΦ*/2π, the time delay Δt corresponds to a time delay between AC signal 41 and AC signal 45 of τ = Δt + TΦ*/2π. As shown, the two square wave signals 46 are input to a logic circuit for zero crossing detection 47 which generates a pulse width modulated sequence of pulses 51 whose pulse width depends upon the time delay τ. As shown in Figure 7A, this pulse width modulated signal 51 is input to a PWM/DC converter 53 which converts this pulse train into a DC value which monotonically varies with the value of τ and hence with the phase shift between the AC signal 41 and the AC signal 45.

Figure 7B illustrates the logic circuit 47 used in this embodiment for the zero crossing detection. As shown, the logic circuit 47 is formed from two AND gates 63^'1 and 63^"2 connected so that the output of each AND gate feeds into the input of the other. Figure

7C illustrates how the pulse duration of the PWM signal 51 output from the detection circuit varies with the time delay τ. As shown, no PWM signal will be output until the time delay (τ) reaches TΦ*/2π, which corresponds to the phase threshold (Φ*). From this point, the pulse width of the PWM signal varies linearly with the time delay (τ).

Thus, by applying the appropriate phase shift to the reference signal 45, the logic circuit 47 automatically performs the desired phase thresholding. This will also be the case if the phase shift is applied to the measured signal instead of the reference signal 45. What is important is to introduce an appropriate relative phase shift between the phase of the reference signal 45 and the phase of the measured signal 39. This may be achieved through the use of the phase shifter 58 or by the provision of a suitable signal generator that can generate the suitably shifted reference signal 59.

As mentioned above, the phase measure determined by the phase detector 56 will vary monotonically with the level of the liquid in the container. In order to equate this phase measure obtained from the phase detector 56 with the liquid level, the phase measure is, as shown in Figure 5, applied to a lookup table 42 which is determined in advance during a calibration routine. The thus determined liquid level is then input to the level signal display 44, which displays the determined level of the liquid 103 within the container.

Excitation and Detection Circuitry - 2

In the circuitry illustrated in Figure 5, the signals obtained from the electrodes and reference capacitors were subtracted from each other in differential amplifier circuits 68^"1 to 68^"4. As the signal levels input to these amplifier circuits will be quite large, relatively high quality differential amplifier circuits 68 are required in order to provide good common mode rejection. However, as V2 is of opposite polarity to V₁ it is possible to substantially reduce the common mode signals before they are applied to the differential amplifier circuits. The way in which this can be achieved is illustrated in Figure 8. As shown, in this arrangement, only two differential amplifier circuits 68^"1 and 68^"2 are required. The other subtractions are achieved by combining the signals from some of the terminals before they are applied to the respective amplifier circuits. For example, the signal input to the positive terminal of amplifier circuit 68^'1 is the signal (VE_{1 1}) obtained from electrode 11 plus the signal (Vc₁₅) obtained from the reference capacitor C₁₅; and the signal input to the negative terminal of amplifier circuit 68^"1 is the signal (Vcπ) obtained from the reference capacitor C₁₁ plus the signal (VBIS) obtained from electrode 15. Therefore, the output from amplifier 68^"1 will be (VEII+V_CI_SMVCII+VEI_S). Rearranging this gives (VEI₁-V_C1O-(VE₁₅-VC₁₅), which is the same output obtained from amplifier circuit 68^"5 shown in Figure 3. However, as V₁ and V₂ have opposite polarity, VEΠ will have opposite polarity to Va₅ and Vc₁₁ will have opposite polarity to V_E15. Thus the addition of these signals results in a significant reduction of common mode signals before they are applied to the amplifier circuits 68. The most reduction occurs when the magnitudes of V₁ and V₂ are the same. Therefore, in this embodiment as common mode signals are significantly reduced before being applied to the amplifier circuits 68, less expensive amplifier circuits 68 can be used.

Excitation And Detection Circuitry - 3

In the embodiments described above, the excitation signals (+V₁ and -V₂) were applied to electrodes 9 and 10 respectively and the signals obtained from the electrodes 11, 12, 15 and 16 were connected to the detection circuitry 26. In an alternative embodiment the roles of these electrodes can be exchanged so that the electrodes 11, 12, 15 and 16 are connected to the excitation circuitry 24 and the electrodes 9 and 10 are connected to the detection circuitry 26. One way of doing this is illustrated schematically in the electrical equivalent circuit shown in Figure 9. In this arrangement, the excitation signal +V₁ is applied in a first period (I) to electrodes 11 and 15 and subsequently in a second period (II) to electrodes 12 and 16; and the excitation signal -V₂ is applied in the first period (I) to reference capacitors C₁₁ and C₁₅ and subsequently in a second period (II) to reference capacitors C₁₂ and C₁₆. In this illustration, the switching between the first and second periods is controlled by the switches 72^"1 to 72^"4. The two signals obtained from terminals A and B are then subtracted in the differential amplifier circuit 68 to obtain V¹ in the first period and V^π in the second period. The remaining processing can be the same as in the first embodiment and a further description will be omitted.

Excitation And Detection Circuitry - 4

Figure 10 illustrates an alternative arrangement of the excitation and processing electronics 22 that can be used with the sensor shown in Figure 2 and when operated in the inverted manner discussed above. In this embodiment, the pulse generator 60 generates a train of voltage pulses 31^"1, each pulse having a similar pulse shape and duration as the pulses generated by the pulse generator 60 described with reference to Figure 5. As shown in Figure 10, the pulse train 31^"1 is also applied to an invertor 32^"1 to generate an inverted pulse train 33^'1. The pulse train 31^"Ms also applied to a time shifter 74 to generate a pulse train 31^"2 that is time shifted relative to the pulse train 31^"1. This time shifted pulse train 31^"2 is also applied to an invertor 32^"2 to generate the inverted pulse train 33^~2. As shown in Figure 10, pulse train 31^"1 is applied to terminals 1* and 6* of the connection interface 20; pulse train 33^"1 is applied to terminals 2* and 5* of the connection interface 20; pulse train 31^"2 is applied to terminals 3* and 8* of the connection interface 20; and pulse train 33^'2 is applied to terminals 4* and 7* of the connection interface 20. As a result of the time shift applied by the time shifter 74, the pulses will be applied to the appropriate electrodes in the desired time multiplexed manner. As shown in Figure 10, terminal A* of the connection interface 20 is connected to switch 62^"1 and to switch 62^"3 and terminal B* of the connection interface 20 is connected to switch 62^'2 and switch β2^Λ. Switches 62^"1 and 62^"2 are opened and closed in synchronism with the pulses of the pulse train 31^"1 and the switches 62^"3 and 62^"4 are opened and closed in synchronism with the pulses of the time shifted pulse train 31^"2. As a result, the voltage accumulated on storage capacitor 64^"1 will vary slowly over time reflecting the level of the liquid relative to electrodes 11 and 15, whilst the voltage accumulated on storage capacitor 64^"2 will vary slowly over time reflecting the level of the liquid relative to the sets of electrodes 12 and 16. As shown in Figure 10, the voltages accumulated on the storage capacitors 64 are amplified by differential amplifiers 66^'1 and 66^~2 to generate the above described output voltages V¹ and V^π, which are applied to the ratiometric calculator 40 as before. The remaining processing of the detection circuitry 26 is the same as the processing performed in the embodiment described with reference to Figure 5 and will not, therefore, be described again.

Excitation And Detection Circuitry - 5

Figure 11 is a block diagram illustrating alternative excitation and the detection circuitry 22 that can be used when the electrodes shown in Figure 2 are connected in the inverted manner illustrated in Figure 9. As shown, in this embodiment, the excitation circuitry 24 includes a signal generator 30, which generates an AC excitation signal 31^"1. A first invertor 32^"1 is provided for providing an inverted excitation signal 33^'1. The excitation circuitry 24 also includes a 90 degree phase shifter 52, which applies a 90 degree phase shift to the excitation signal 31^"Mo generate a phase shifted excitation signal 31^"2. This phase shifted excitation signal 31^"2 is also applied to an invertor 32^"2 to generate an inverted and phase shifted excitation signal 33^" . As shown in Figure 11, the excitation signal 31^"1 is applied to terminals 1* and 6* of the connection interface 20; the inverted excitation signal 33^"1 is applied to terminals 2* and 5* of the connection interface 20; the phase shifted excitation signal 31^"2 is applied to terminals 3* and 8* of the connection interface 20; and the inverted and phase shifted excitation signal 33^"2 is applied to terminals 4* and 7* of the connection interface 20. The excitation circuitry 24 used in this embodiment is therefore equivalent to the excitation circuitry shown in Figure 10, except the pulsed excitation signals are replaced with suitably phase shifted AC excitation signals. As shown in Figure 11, a differential amplifier 68 of the detection circuitry 26 is connected to terminals A* and B* of the connection interface 20. In view of the sinusoidal excitation signals and in view of the general principles of superposition, the output from the differential amplifier 34 can be represented by equation (2) or (3) given above, where f is the frequency of the excitation signals. In other words the output from the differential amplifier 68 includes a sinusoidal signal whose phase varies with the ratiometric arctangent function given in equation (1) above. As shown in Figure 11, the amplitude of the signal output from the differential amplifier 68 is compared against the amplitude threshold by the detector 55 and, provided its amplitude is above the threshold, it is applied to the phase detector 56, which measures the phase of this signal relative to the phase of the AC excitation signal 31^"1. The remaining processing carried out by the detection circuitry 26 is the same as in the other embodiments described above and will not, therefore, be described again.

Excitation And Detection Circuitry - 6

Figure 12 is a block diagram illustrating a further example of the excitation and detection circuitry that can be used when the electrodes shown in Figure 2 are connected in the inverted manner illustrated in Figure 9. In this embodiment pulse width modulated pulse trains are used as excitation signals. As shown, in this embodiment, the excitation circuitry 24 includes a pulse generator 60 that is arranged to generate a high frequency periodic pulse train 61, the pulses of which have relatively fast rising and falling fronts (of the order of several nanoseconds) and a relatively long pulse period (of the order of several hundred nanoseconds). The excitation circuitry 24 also includes a low frequency generator 48 that is arranged to generate a low frequency modulation signal 49 having a frequency of the order of several kHz. As shown in Figure 1, the pulse train 61 generated by the pulse generator 60 and the low frequency modulation signal 49 are applied to a modulator 82^"1 which uses the low frequency modulation signal 49 to pulse width modulate the pulse train 61, to generate a pulse width modulated pulse train 84. As shown, the pulse width modulated pulse train 84 is applied to terminals 1* and 6* of the connection interface 20 and is also applied to an invertor 32^'1. The inverted pulse train generated by the invertor 32^'1 is then applied to terminals 2* and 5* of the connection interface 20. As shown in Figure 12, the low frequency modulation signal 49 generated by the low frequency generator 48 is also applied to a 90 degree phase shifter 52 that applies a 90 degree phase shift to the modulation signal 49. The 90 degree phase shifted version of the modulation signal 49 is then applied to a pulse width modulator 82^'2 where it is used to pulse width modulate pulse train 61 generated by the pulse generator 60. Figure 12 illustrates the pulse width modulated signal 86 output from the modulator 82^"2. As shown, the pulse width modulated signal 86 is applied to terminals 3* and 8* of the connection interface 20 and is also applied to a second invertor 32^"2. The inverted version of the pulse width modulated signal 86 is then applied to terminals 4* and 7* of the connection interface 20.

As shown in Figure 12, both terminals A* and B* of the connection interface 20 are connected to the two switches 62^"1 and 62^"2. The position of the switches 62 are switched in synchronism with the rising and falling edges of the pulses in the pulse train 61^'1 obtained by phase shifting the pulse train 61 by 90 degrees using the phase shifter 52. At any instant in time, the signal obtained from terminal A* is passed through one of the switches 62^"1 and 62^"2 and the signal obtained from terminal B* is passed through the other one of the switches 62^"1 and 62^~2. The signals from switches 62^'1 and 62^"2 are then filtered and differentially amplified by the amplifier and filter circuitry 71. As shown further in Figure 13, both terminals A* and B* are coupled to ground through capacitors 64^"1, 64^"2, 64^"3 and 64^"4. Thus high frequency components of the currents I₁ and I₂ are shorted to ground through a small impedance. Whereas low frequency variations of the currents I₁ and I₂ (such as caused by the low frequency modulation of the excitation signals by the low frequency modulation signal 49) marked as I and I on Figure 13, are further amplified and combined by the circuitry 71. In the particular case where the two resistances R₂ and R₃ of the circuitry 71 are equal, the output current (Z₁) 73 from the circuitry 71 will become proportional to the difference in the currents I^~ and I^~ with simultaneous rejection of any common mode signals.

In addition to rejecting common mode signals, the circuit 71 will also remove signals due to resistive coupling between the electrodes caused by the liquid being conductive. This is because the capacitively coupled signals will be 90 degrees out of phase with the excitation signal (and hence are in phase with the switching signal 61^"1) whereas the resistively coupled signals will be in phase with the excitation signal (and hence 90 out of phase with the switching signal 61^"1). Consequently, the resistive coupling signals obtained at terminals A* and B* will also be removed as the output signal is proportional to the difference in the currents I^~ and I .

As the output signal is proportional to the difference in the currents I and I , the output from the filter and amplification circuitry 71 can therefore also be approximated by equation (3) above, with the frequency f corresponding to the frequency of the low frequency modulation signal 49. As shown in Figure 12, if the amplitude of this signal is above the amplitude threshold (as determined by the detector 55) then it is input to the phase detector 56 which measures the phase of this signal relative to the phase of the low frequency modulation signal 49 generated by the low frequency generator 48. The rest of the detection circuitry 26 used in this embodiment is the same as used in the previous embodiments and therefore a further description will not be given.

Excitation And Detection Circuitry - 7

Figure 14 is a block diagram illustrating a further example of the excitation and detection circuitry that can be used when the electrodes shown in Figure 2 are connected in the inverted manner illustrated in Figure 9. In this embodiment, the excitation circuitry 24 comprises a signal generator 65 (which may be a memory) which generates the desired excitation signals in response to being clocked by the clock 63. In this embodiment, the desired excitation signals have the same form as the low frequency modulated PWM signals 84 and 86 used in the embodiment described above with reference to Figure 12. In particular, the signal generator 65 generates: the PWM signal 84 which it applies to resistor R₁; the inverse of PWM signal 84 which it applies to resistor R₃; the PWM signal 86 which it applies to resistor R₄; and the inverse of PWM signal 86 which it applies to resistor R₆. Figures 15A and 15B illustrate in more detail part of the signals applied across resistors R₁ and R₃ and the resulting voltages V₁ and V₂ that are applied to terminals 1* and 6* and 2* and 5* of the connection interface 20 respectively. As shown in Figure 15B, the signal applied to terminals 1* and 6* varies between voltages Vj ¹ and Vi² and the signal applied to terminals 2* and 5* varies between voltages V₂ ¹ and V₂ ². The voltages applied to terminals 1*, 2*, 5* and 6* therefore always have a different magnitude from each other. As can be seen from Figures 15C and 15D, this is similarly true of the voltages V₃ ¹, V₃ ², V₄ ¹ and V₄ ² that are applied to terminals 3* and 8* and 4* and 7* respectively. Therefore electrode 11 and capacitor C₁₅ will be energised by voltage V₁; electrode 15 and capacitor Cn will be energised by voltage V₂; electrodel2 and capacitor C₁₆ will be energised by voltage V₃; and electrode 16 and capacitor C₁₂ will be energised by voltage V₄.

As those skilled in the art will appreciate, although pulses of excitation signal are applied to the electrodes and corresponding reference capacitors, the effective excitation occurs when the pulses transition between from a low value to a high value (corresponding to a positive excitation) and when the pulses transition from a high value to a low value (corresponding to a negative excitation), with the amplitude of the excitation being defined by the size of the step change between the low and high values. Thus the signal that will be obtained at terminal A* by applying V₁ to terminal 1 * can be represented by:

± AV_xEⁿ (5)

where AV₁ is the step change between levels V₁ ¹ and V₁ ² and E¹¹ represents the capacitive coupling between electrodes 11 and 9. A positive signal will be obtained when the transition is from V₁ ² to V₁ ¹ and a negative signal will be obtained when V₁ transitions from V₁ ¹ to V₁ ². Similarly, the signal that will be obtained at terminal B* by applying V₁ to terminal 6* can be represented by:

± ΔV_λC^ιs (6)

where C¹⁵ represents the capacitive coupling through reference capacitor C₁₅.

Similarly, the signal that will be obtained at terminal A* by applying V₂ to terminal 2* can be represented by:

± AV₂C^U (7)

where ΔV₂ is the step change between levels V₂ ¹ and V₂ ² and C¹¹ represents the capacitive coupling through reference capacitor C₁₁. A positive signal will be obtained when the transition is from V₂ ¹ to V₂ ² and a negative signal will be obtained when V₂ transitions from V₂ ² to V₂ ¹.

Similarly, the signal that will be obtained at terminal B* by applying V₂ to terminal 5* can be represented by: ± AV₂E¹⁵ (8)

where E¹⁵ represents the capacitive coupling between electrodes 10 and 15. As shown in Figure 15B, when V₁ transitions from a high value to a low value V₂ transitions from a low value to a high value and vice versa. Thus, the signal output from the differential amplifier circuit 71 (due to V₁ and V₂) at the transitions of signals V₁ and V₂ can be represented by:

± ((AF₁E¹¹ - AV₁C¹⁵ )+ (ΔF₂E¹⁵ - ΔV₂C^U )) (9)

However, as electrodes 11 and 15 are identical and as electrodes 9 and 10 are identical, reference capacitors C₁₁ and Cis will nominally have the same capacitance. Therefore, each of the terms in the brackets performs the desired dry value compensation, to leave a signal of the form:

+ ((AF₁ + AV₂ XE^11/15 - C^n/15 ))= IAV₁₂V¹ (10)

As mentioned above, in this embodiment, the pulses of V₁ and V₂ are PWM modulated by a low frequency signal. Therefore, the signal obtained from circuit 71 as a result of applying voltages V₁ and V₂ can be represented by:

± AV_nV' cos 2πft (11)

In a similar way, the signal obtained from circuit 71 as a result of the application of voltages V₃ and V₄ to terminals 3* and 8* and 4* and 7* respectively can be represented by:

± AV₃₄V" sin 2πft (12)

The total signal output from circuit 71 is therefore the sum of equations (11) and (12).

Provided ΔV₁₂ = ΔV₃₄, the result is a signal in the form of equation (3) above. ΔV₁₂ can be made to equal ΔV₃₄ by appropriate selection of the values of resistors R₁ to R₆. In particular, ΔV₁₂ is given by:

AK₁₂ (13)

And ΔV₃₄ is given by:

Therefore, ΔVi₂ can be made equal to ΔV₃₄ by making resistors R₂ and R₅ equal and by making (R^R₃) equal (R₄H-R₆). As shown in Figure 14, the amplitude of the signal output from the circuit 71 is compared with the amplitude threshold by the amplitude detector 55 and if it is above the threshold, the signal is passed through switch 57 to two phase detectors 56^"1 and 56^~2. In this embodiment, the phase of the PWM modulated signals Vi and V₂ is changed by 180 degrees during the measurement process. In particular, during a first time interval "cos" modulated PWM signals Vi and V₂ are applied and the signal obtained from the amplifier circuit 71 is passed to the first phase detector 56^"1 where the phase of the signal is measured relative to a "cos" reference signal supplied by the signal generator 65. During this first interval the second phase detector 56^"2 is effectively switched off by applying no reference signal to it. Then in a second time interval, "-cos" modulated PWM signals Vi and V₂ are applied and the signal obtained from the amplifier circuit 71 is passed to the second phase detector 56^'2, where the phase of the signal is measured relative to a "-cos" reference signal supplied by the signal generator 65. During this second interval the first phase detector 56^" * is effectively switched off by applying no reference signal to it. The phase measurements obtained from the two phase detectors 56 are then passed through low pass filters 69^"1 and 69^~2 (which effectively act as memories) and subtracted from each other by the differential amplifier 66^"3. As a result of the different phase of these excitation signals, this subtraction results in the addition of the desired phase measures whilst removing common phase offsets introduced by the electronics. The phase measure is then applied to the lookup table 42 as before to determine the corresponding liquid level which is displayed to the user on the display 44.

Phase Offset Due To Dry Value Reference Capacitors hi the above embodiments, each measurement electrode (11, 12, 15 and 16) requires its own separate dry value reference capacitor. These reference capacitors are mounted on the connection interface 20 and are formed using different manufacturing processes to the capacitors formed by the measurement electrodes. As a result of the different manufacturing processes, the reference capacitors will have different temperature characteristics than the capacitors formed by the measurement electrodes (11, 12, 15 and 16) and therefore, they will not make the desired dry capacitance cancellation in all operating conditions. This leads to phase offsets and hence errors in the measurement results. As will be described below, in some situations these phase offsets can lead to significant errors in the measurement results. A more detailed description of this offset problem will now be described with reference to Figure 16. In particular, Figure 16A shows, the desired or expected locus plot 76 and the actual locus plot 77 obtained if the dry capacitance of reference capacitors Cu and/or Ci₅ do not cancel out with the dry capacitance of the capacitors formed by the electrodes 9 and 11 and 10 and 15 respectively, and as a result the first output signal V¹ is reduced by ΔV¹. Figure 16B illustrates the resulting phase error plot as the liquid level is varied from levels O to Q. As can be seen, as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until some unknown level Wi when the measured phase angle exceeds the phase threshold (Φ*). At this point a positive phase error (because the measured phase leads the expected phase) will start to be introduced and will increase linearly, as shown in Figure 16B, until it reaches a maximum phase error when the liquid level reaches level Q. Figures 16C and 16D illustrate the situation when the dry capacitance of reference capacitors C₁₁ and/or C₁₅ do not cancel out with the dry capacitance of the capacitors formed by the electrodes 9 and 11 and 10 and 15 respectively, and as a result the first output signal V¹ is increased by ΔV¹. As shown as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*). At this point a negative phase error (because the measured phase lags the expected phase) will start to be introduced and will increase linearly in magnitude, as shown in Figure 16B, until it reaches a maximum phase error when the liquid level reaches level Q. Figures 16E and 16F illustrate the situation when the dry capacitance of reference capacitors C₁₂ and/or C₁₆ do not cancel out with the dry capacitance of the capacitors formed by the electrodes 9 and 12 and 10 and 16 respectively, and as a result the second output signal V^π is reduced by ΔV^π. As shown, in this situation, as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*). At this level a negative phase error (because the measured phase lags the expected phase) will start to be introduced which will increase in magnitude linearly as the liquid level rises, until the measured phase angle exceeds the phase threshold (Φ*). At this level the phase error will slowly reduce until the liquid level reaches level Q. Figures 16G and 16H illustrate the situation when the dry capacitance of reference capacitors C₁₂ and/or C₁₆ do not cancel out with the dry capacitance of the capacitors formed by the electrodes 9 and 12 and 10 and 16 respectively, and as a result the second output signal V^π is increased by ΔV^π. As shown, in this situation, as the liquid level rises up the container beyond level O, no phase error (ΔΦ₀) is introduced until the amplitude of the measured signals exceed the amplitude threshold (V A° - as represented by the circle 29) at level Y. As shown in Figures 16G, at this level the measured phase jumps to a relatively large phase (Φ_eπ-_or), corresponding to a phase error of (Φ_error- Φ*). As shown in Figure 16H, as the liquid level increases further towards level P, this phase error will reduce. As the liquid level increases beyond level P, however, the phase error will rise again until the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*). As the liquid level increases beyond level W₂, the phase error slowly reduces again until the liquid reaches level Q. Therefore, as can be seen from Figure 16G, in this situation, when the container is almost empty (at level Y), a phase measure is obtained that corresponds to the expected phase when the liquid is well beyond level P indicating that the container is almost half full. As those skilled in the art will appreciate, this corresponds to a significant error, especially when accuracy is required at low levels of liquid.

A number of alternative sensor designs will now be described which at least alleviate the above offset problem caused by mismatches between the capacitors formed by the measurement electrodes and the corresponding dry value reference capacitors.

Sensor Head B

Figure 17 schematically illustrates an alternative capacitive liquid level sensor head 101 that is based on the sensor shown in Figure 2, except that the dry value reference capacitors for the main measurement electrodes (12 and 16) have been incorporated into the electrode structure. As shown, the sensor includes two symmetrically arranged (back to back) sets of electrodes 21 and 23. The first set of electrodes 21 includes five electrodes 9, 11, 12, 13 and 14 which are connected to the excitation and processing electronics 22 via terminals 1* to 4* and A* on the connection interface 20. The second set of electrodes 23 also includes five electrodes 10, 15, 16, 17 and 18 which are connected to the excitation and processing electronics 22 via terminals 5* to 8* and B* on the connection interface 20. As shown, in this embodiment electrodes 9 and 10 are 'L' shaped with the long side extending over the entire depth of the container (defined between levels O and S). Electrodes 12, 13 and 14 and electrodes 16, 17 and 18 extend over the operating range of the sensor (defined between levels P and S); and electrodes 11 and 15 extend transverse to the measurement direction adjacent the shorter side of electrodes 9 and 10 respectively. As a result of the use of the 'L' shaped electrodes 9 and 10, the inoperative range or dead zone (defined between levels O and P) of the sensor is much smaller than that of the sensor illustrated in Figure 2. Electrodes 11, 12, 13 and 14 are positioned adjacent and form capacitors with electrode 9 and similarly electrodes 15, 16, 17 and 18 are positioned adjacent and form capacitors with electrode 10.

As shown in Figure 17, electrode 9 is connected to terminal A*; electrode 10 is connected to terminal B*; electrode 11 is connected to terminal 1*; electrode 12 is connected to terminal 3*; electrode 13 is connected to terminal 2*; electrode 14 is connected to terminal 4*; electrode 15 is connected to terminal 5*; electrode 16 is connected to terminal 7*; electrode 17 is connected to terminal 6*; and electrode 18 is connected to terminal 8*.

In this embodiment, electrodes 12 and 14 are identical and are connected to the processing electronics 22 in such a manner that they form a balanced pair of electrodes. In other words in the absence of any liquid in the container or when the container is full of liquid the signals obtained from electrodes 12 and 14 cancel each other out. Similarly electrodes 16 and 18 are identical and are connected to the processing electronics 22 in such a manner that they also form a balanced pair of electrodes. As will be described in more detail below, electrodes 11 and 13 are also paired, but they are not identical and so they are not balanced. An additional dry value reference capacitor C₁₁ is therefore provided to balance this pair of electrodes. Similarly, electrodes 15 and 17 are also paired, but they are not identical and so they are not balanced. An additional dry value reference capacitor Ci₅ is therefore provided to balance this pair of electrodes.

Figure 18 illustrates how the electrodes shown in Figure 17 may be connected to the excitation and processing electronics 22, such that electrodes 9 and 10 are connected to the excitation circuitry 24 and the other electrodes are connected to the detection circuitry 26. Similarly, Figure 19 illustrates the inverted connection where electrodes 9 and 10 are connected to the detection circuitry 26 and the other electrodes are connected to the excitation circuitry. In either case, the above described voltages V¹ and V^π will be generated. However, as a result of the different electrode arrangement, these voltages do not vary in the same manner as V¹ and V^π obtained with the sensor shown in Figure 2.

Figure 2OA illustrates the way in which the first output signal V¹ varies as the liquid level varies between level O and level S. As shown, when there is no liquid in the container, the first output voltage is at zero. This is because at this liquid level the capacitive coupling between electrodes 9 and 13 is cancelled out by the capacitive coupling between electrodes 9 and 11 in combination with the capacitive coupling through reference capacitor C₁₁; and because at this liquid level the capacitive coupling between electrodes 10 and 17 is cancelled out by the capacitive coupling between electrodes 10 and 15 in combination with the capacitive coupling through reference capacitor C₁₅. However, as the liquid level rises past level O, this balance is disturbed and the first output signal V¹ starts to rise. When the liquid level reaches level P, electrodes 11 and 15 will be fully submerged and V¹ will have reached its maximum value. As the liquid level rises further between levels P and Q, electrodes 11 and 15 are still fully submerged and as the liquid has not yet reached electrodes 13 and 17, the first output voltage will remain constant. However, once the liquid level rises past level Q, the first output signal V¹ starts to reduce and continues reducing as the liquid level rises, until the electrodes 13 and 17 are fully submerged. In this embodiment, electrodes 13 and 17 have approximately twice the surface area of electrodes 11 and 15 and therefore, by the time the liquid level has reached level R, the first output voltage will be at a peak negative value which is approximately equal in magnitude to the peak positive value reached at level P. The first output voltage then remains at this level as the liquid level increases toward the full level S.

Figure 2OB illustrates the way in which the second output signal V¹¹ varies as the liquid level varies between level O and level S. As shown, when there is no liquid in the container, the second output signal is also zero. This is because, as discussed above, electrodes 12 and 14 and 16 and 18 form balanced pairs of electrodes. Thus in the absence of any liquid to upset this balance, no signal will be obtained from these electrodes. However, as the liquid level rises past level P, this balance is disturbed and the second output signal V^π starts to rise. When the liquid level reaches level Q, electrodes 12 and 16 will be fully submerged and V^π will have reached its maximum value. As the liquid level rises further between levels Q and R, electrodes 12 and 16 are still fully submerged and as the liquid has not yet reached electrodes 14 and 18, the second output signal will remain constant. However, once the liquid level rises past level R, the second output signal V^π starts to reduce and continues reducing as the liquid level rises, until electrodes 14 and 18 are fully submerged. As electrodes 14 and 18 are identical to electrodes 12 and 16 respectively, by the time the liquid level has reached level S, the second output voltage will have reduced back to zero.

Figure 2OC is a locus plot obtained by plotting output signal V¹ against output signal V^π as the liquid level rises between level O and level S. As shown, the locus plot almost defines a closed square. Therefore, the measured phase angle, Φ, will vary from 0 degrees to 180 degrees as the liquid level rises from level O to level S. As a result of this greater range of phase angle variation, the measurements that are obtained will be more accurate as the rate of change of the measured phase angle with the level of the liquid is greater. Further, as will now be explained with reference to Figure 21, this sensor design is also less sensitive to the phase offset problem discussed above with reference to Figure 16. As shown in Figure 17, reference capacitors C₁₁ and C₁₅ are still provided in this embodiment. Therefore, V¹ may still vary by ±ΔV¹. Figure 21 A illustrates the desired or expected locus plot 76 and the actual locus plot 77 obtained if the reference capacitors Cn and/or C₁₅ do not provide the desired balance between electrodes 11 and 13 and electrodes 15 and 17 respectively, and as a result the first output signal V¹ is reduced by ΔV¹. Figure 21B illustrates the resulting phase error plot as the liquid level is varied from levels O to S. As can be seen, as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until some unknown level Wj when the measured phase angle exceeds the phase threshold (Φ*). At this point a positive phase error (because the measured phase leads the expected phase) will start to be introduced and will increase to a maximum value when the liquid level is midway between levels Q and R and then decrease back to zero by the time the liquid level has reached level S. Figures 21 C and 2 ID illustrate the situation when the reference capacitors C₁₁ and/or C₁₅ do not provide the desired balance between electrodes 11 and 13 and electrodes 15 and 17 respectively, and as a result the first output signal V¹ is increased by ΔV¹. As shown as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*). At this point a negative phase error (because the measured phase lags the expected phase) will start to be introduced and will increase in magnitude to a maximum value when the liquid level is midway between levels Q and R and then decrease back to zero by the time the liquid level has reached level S.

As shown in Figure 17, in this embodiment, there are no reference capacitors formed from a different manufacturing process associated with the second output signal V^π. Instead the electrodes associated with output signal V¹¹ are formed from balanced pairs of electrodes and so there should be a much smaller possible offset associated with output signal V^π. Figures 21E and 21 F illustrate the situation when a small offset ΔV^π reduces the second output signal. As shown, in this situation, as the liquid level rises up the container, no phase error (ΔΦ₀) is introduced until the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*). At this level a negative phase error (because the measured phase lags the expected phase) will start to be introduced which will increase in magnitude linearly as the liquid level rises, until the measured phase angle exceeds the phase threshold (Φ*). At this level the phase error will have reached its maximum negative value. As the liquid level increases further, the phase error reduces again to zero when the liquid is halfway between levels Q and R. A positive phase error will then be introduced which will increase linearly to a maximum value when the liquid level reaches level S.

Figures 21 G and 2 IH illustrate the situation when a small offset ΔV^π increases the second output signal V^π. As shown, in this situation, as the liquid level rises up the container beyond level O, no phase error (ΔΦ₀) is introduced until the level W₁ corresponding to the level when the measured phase angle exceeds the phase threshold (Φ*). At this point a positive phase error (because the measured phase leads the expected phase) will start to be introduced and will increase to a maximum value when the level W₂ corresponding to the level when the phase angle is expected to exceed the phase threshold (Φ*) is reached. As shown in Figures 2 IG, as a result of reducing the magnitude of the possible offset ΔV^π, when the signal levels first exceed the amplitude threshold (VA⁰ - represented by the circle 29), the measured phase angle (Φ_error) will be less than the phase threshold (Φ*). Therefore the same spike of phase error shown in Figure 16H does not occur with the sensor design illustrated in Figure 17. Sensor Head C

Figures 2 and 17 schematically illustrate two different sensor head designs. The electrodes may be formed from suitably shaped conductor plates or the electrodes (and their connections to the connection interface 20) may be formed using the conductive layers of one or more printed circuit boards. Such an arrangement would facilitate the positioning of the electrodes in the symmetric (back to back) arrangement due to the inherent parallel layer arrangement of multilayer printed circuit boards. Alternatively still, the conductors may be formed from a conductive ink printed onto suitable substrates which are then assembled to position the electrodes in the manner illustrated in Figures 2 and 17. Alternatively still, the electrode designs described above may be formed from conductors that are mounted in the same plane. This is illustrated in Figure 22A for a design of electrodes that is equivalent to those shown in Figure 17. The electrodes shown in Figure 22A have been labelled with the same reference numbers as their equivalent in the design shown in Figure 17.

As shown in the cross-section shown in Figure 22B, the electrodes 9 to 18 are mounted on a planar substrate 19 and through holes 89^'1 and 89^"2 are provided between electrode 9 and electrodes 11, 12, 13 and 14 and between electrode 10 and electrodes 15, 16, 17 and 18, respectively. This allows space for the liquid whose level is to be sensed to be positioned in the gap between the measurement electrodes and thus affect the mutual capacitance between these electrodes as the liquid level rises up the container. Further, as shown in Figure 22 A, electrodes 11 and 12 and electrodes 15 and 16 overlap each other along the measurement range of the sensor. This is because there is insufficient space at the base of the sensor to form electrodes of the required size (surface area) and therefore electrodes 11 and 15 have been extended up the sensor to overlap with electrodes 12 and 16.

Figure 23 A illustrates the way in which the first output signal V¹ varies as the liquid level increases from level O to level S and Figure 23 B illustrates the way in which the second output signal V¹¹ varies as the liquid level increases from level O to level S. As shown, the plots are similar but not identical to those shown in Figures 2OA and 2OB. As a result the locus plot obtained by plotting the first output signal V¹ against the second output signal V^π as the level of the liquid changes (and which is shown in Figure 23C) is not identical to the locus plot shown in Figure 2OC. Figure 23D illustrates, for information, the "non-linearity" of the sensor which is corrected using the lookup table 42. In particular, Figure 23D shows that with the design of electrodes illustrated in Figure 22, the relationship between the measured phase and the level of the liquid in the container is non-linear. The plot illustrated in Figure 23D is obtained during a calibration routine and used to generate the above described lookup table 42 that maps the measured phase angle to the corresponding liquid level.

Another advantage of this embodiment is that resistive coupling between the excitation electrodes and each detection electrode (due to conductivity of the liquid) will be attenuated by the differential amplifier circuits 68. This is because resistive coupling is less dependent on the separation between the excitation electrodes and each detection electrode and will therefore appear as a common mode signal in the inputs to the differential amplifier circuits 68. Phase Offset Due To Dielectric Variations

One design goal of capacitive liquid level sensors is to produce an inexpensive sensor that will work with different liquids having different dielectric constants. For example, when used as a fuel gauge, it is desirable if the sensor can work without needing any reconfiguration with, for example diesel (which has a low dielectric constant) and ethanol (which has a high dielectric constant). Figure 24 A illustrates the locus plot of V¹ against V¹¹ obtained for the sensor shown in Figure 22 A for diesel (plot 91) and a scaled down plot obtained for ethanol (plot 92). The ethanol plot is scaled down as the signal levels obtained when ethanol is the liquid are far greater than those obtained from diesel. As can be seen from the two plots 91 and 92, they do not follow the same path. This means that there will be an offset (ΔΦ_d) in the measured phase between the two liquids for the same level, as illustrated in Figure 24A at level R. A plot 94 showing how this offset (ΔΦ_d) varies over the full measurement range of the sensor is shown in Figure 24B. As shown, the offset (ΔΦ_d) rises initially to a peak positive value (corresponding to about 1% error) and then slowly reduces to zero and then increases to a peak negative value (corresponding to about 3% error) before reducing again towards the end of the measurement range (corresponding to point S).

The inventor has realised that by careful design of the electrodes, he can deliberately introduce offsets in the dry value capacitance calibration (ie ΔΦ₀ discussed above) to change the profile of the plot 94. In particular, by introducing a negative offset (ΔV^π) into the design of the sensor, the offset (ΔΦ₀) shown in Figure 21F will be added to the offset (ΔΦ_d) shown in Figure 24B to give a combined or total phase offset (ΔΦ_T1) that varies in the manner shown in plot 95 of Figure 24C. As can be seen, by deliberately introducing an offset in the balance of the electrodes affecting output signal V^π, the total phase offset of the sensor for the different liquids is reduced to a maximum of 2%. As those skilled in the art will appreciate, the offset ΔV^π can be introduced by adding an appropriate capacitor to the electrodes affecting V (ie electrodes 12, 14, 16 and 18) or more simply by altering the length of electrodes 14 and 18 so that they are not exactly identical to electrodes 12 and 16 and thus do not exactly balance with these electrodes. In some cases having accuracy at the beginning of the measurement range is all that is important. For example, for a fuel gauge sensor, the critical measurement is when the fuel tank is near or at empty. It does not matter if the measurement is less accurate when the tank is half full or full. hi addition to deliberately adding an offset to the electrodes that affect output signal V^π, a similar offset can be introduced to the electrodes that will affect output signal V¹. For example, by introducing a negative voltage offset (ΔV¹) into the design of the sensor, the offset (ΔΦ₀) shown in Figure 21 B will be added to the offset (ΔΦ_T1) shown in Figure 24C to give a total offset (ΔΦχ₂) that varies in the manner shown in plot 96 of Figure 24D. As can be seen from plot 96, by introducing this second offset, the accuracy of the sensor for different liquids is maintained over a longer portion of the measurement range of the sensor (approximately 60% of the measurement range). This offset can be added either by changing the capacitor values of capacitors C₁₁ and/or C₁₅, but is more preferably achieved by altering the length of electrodes 13 and 17 or electrodes 11 and 15. Sensor Head D

Referring back to Figure 24A, it can be seen that there is quite a considerable phase offset between levels O and Pl. The inventor has realised that this offset is caused by the effects of fringing electric fields. In particular, in high dielectric liquids such as ethanol, most of the electric field passes between the electrodes in the plane of the electrodes and there are negligible fringing electric fields (ie those which extend out of the gap between the electrodes into the liquid). However, in low dielectric liquids (such as diesel) the electric field extends further into the liquid. As a result, at the start of the measurement range (between levels O and Pl) when the liquid is diesel, electrode 9 will couple with electrode 12 and electrode 10 will couple with electrode 16. The inventor has realised that by attaching a non-conductive layer (made for example of plastic) to the bottom of the substrate, it is possible to reduce the extent of these fringing fields in low dielectric liquids (such as diesel). This non-conductive layer will have no effect when the liquid has a high dielectric constant and therefore the locus plots for the different liquids will become more similar (at least at the start of the measurement range).

Figures 25A and 25B illustrate the sensor head of Figure 22A having such a nonconducting layer 98 attached to the bottom of the substrate 19. Figure 26A illustrates the locus plot 92 obtained for the high dielectric liquid (which is unchanged from that shown in Figure 24A) and the locus plot 93 obtained for the low dielectric liquid. As can be seen in Figure 26A₅ the two locus plots 92 and 93 are much more similar to each other than plots 91 and 92, especially at the start of the measurement range (between levels O and P2). As the plots 92 and 93 are more similar, the phase offsets introduced with the sensor shown in Figure 25 due to different dielectric constants are much smaller. Figure 26B illustrates how this phase offset (ΔΦd) varies over the measurement range. As shown, the phase offset error is now more symmetric than the plots illustrated in Figure 24 and also has a smaller peak offset of about 1.2%.

Sensor Head E In the sensor designs illustrated in Figures 22 and 25, the electrodes were mounted on a planar substrate 19. In some applications, the liquid may include contaminants that affect the operation of the sensor if the contaminants contact or become stuck to one or more of the electrodes. Also, if the liquids are conductive, then significant levels of current may be drawn from the sensor. In these circumstances, it is possible to embed the electrodes within a plastic substrate, as illustrated in Figures 27A and 27B. In this case, it is preferable that the electrodes are placed in the middle of the plastic substrate 19, to keep the symmetry between the electrodes and the liquid surrounding the plastic. Additionally, in such an embodiment, it is possible to pattern the upper and/or lower surface of the plastic substrate in order to affect the operation of the sensor. This may be used, for example, to control effects caused by fringing electric fields and/or to control the locus plots so that they have a similar trajectory for liquids having different dielectric constants and/or to control the variation of any resistive coupling that is not filtered out by the detection circuitry, so that it varies in the same way as the capacitive coupling (and thus will not interfere with the measurements that are being taken).

Sensor Head F

In the above embodiments, there was a non-linear relationship between the measured phase and the corresponding liquid level. A lookup table 42 was therefore provided to map the determined phase angle to the corresponding liquid level. Figure 28 illustrates a set of electrodes that are designed so that there is a linear relationship between the measured phase and the corresponding liquid level, so that there is no need for the lookup table 42. As shown, electrodes 12, 14, 16 and 18 are "D" shaped and electrodes 13 and 17 are "hour glass" shaped. Figures 29A and 29B illustrate the way in which the two output signals V¹ and V vary as the liquid level rises up the container from level O to level S. Finally, Figure 29C illustrates how the measured phase angle (minus the phase threshold) varies as the liquid level rises from level O to level S. As shown, the phase angle remains at zero until the second dashed line corresponding to the level when the actual measured phase exceeds the phase threshold. At this point, the phase angle increases linearly with the level of the liquid in the container. Therefore, in this embodiment, the liquid level can be determined simply by scaling the measured phase angle and there is no need for the use of a lookup table 42.

Sensor Head G

Figures 3OA and 30B illustrate another sensor head which aims to emulate the sensor head shown in Figure 28 using planar electrodes. As shown, the surface areas of the electrodes are varied along the length of the sensor head. Where there is insufficient room on the substrate to achieve the desired rate of change of capacitance per unit length of the sensor, inter-digitised fingers of electrodes may be provided, such as in region 99.

Sensor Head H

In the above embodiments, sensor heads were provided that comprised two sets of electrodes arranged back to back in a symmetric manner along the measurement axis of the sensor. Figure 31 illustrates an alternative embodiment, where only one set of electrodes 21 is provided and Figure 32 illustrates how these electrodes may be connected to the excitation circuitry 24 and the detection circuitry 26. In this embodiment, the length of electrode 14 has been made slightly different to the length of electrode 12, so as to introduce a deliberate offset into the second output signal V^π. Similarly, the length of electrode 13 has been made slightly different to twice that of electrode 11, in order to deliberately introduce an offset into the first output signal V¹. These deliberate offsets have been introduced to allow the sensor to work, without requiring significant reconfiguration, with liquids having different dielectric constants. As shown in Figure 32, in this embodiment, the two excitation signals applied to the electrodes nominally have the same magnitude but different signs. This is to provide the desired balance (apart from the above offsets) between the paired electrodes - 11 and 13; and 12 and 14. Alternatively, in this embodiment, the desired offsets may be introduced by applying different magnitude excitation signals to the electrodes.

Modifications And Other Alternatives

A number of embodiments have been described above with reference to the accompanying Figures. As those skilled in the art will appreciate various modifications can be made to these embodiments. A number of these modifications will now be described for illustration only.

A number of different excitation and detection techniques have been described above. These techniques can be used with any of the sensor heads described above (with appropriate modification where necessary). As those skilled in the art will appreciate, various different excitation and detection techniques can be used with the sensors described above. A number of other suitable techniques are described in the applicant's earlier International patent application PCT/GB2006/001815, the contents of which are hereby incorporated by reference. For example, in the embodiment described with reference to Figure 12, PWM modulated pulses were applied to the excitation electrodes. Instead, amplitude modulated pulses may be applied. In this case, the signals obtained in the detection circuitry may be processed in the same way as shown in Figure 12 or they may be demodulated and then processed in the manner described above with reference to Figure 5. Similarly, instead of applying phase (or time) shifted excitation signals to the excitation electrodes, excitation signals having different frequencies may be used.

In the above embodiments, discussion was given to two output signals V¹ and V¹¹. As those skilled in the art will appreciate, many of the detection circuits described above do not actually calculate the value of these output signals, but instead calculate the phase angle between them, as defined in equation (1) above, by measuring the zero crossing point of a detected and processed signal relative to a reference signal. hi the above embodiments, the excitation and detection circuits included various electronic components. In an alternative embodiment, a programmable circuit (processor) controlled by software stored in a memory may implement these circuits. The software may be provided in any appropriate form and in any computer language. It may be supplied as a signal or stored on a computer readable medium such as a CD ROM.

In the above embodiments, one or more differential amplifier circuits subtracted the signals from complementary sets of electrodes. In an alternative embodiment, the signals may be subtracted by inverting one of the signals and then by adding it to the other signal. Other subtraction circuits may also be used.

In the above embodiments, the electrodes are connected to the excitation and detection circuitry via a connection interface. This connection interface is illustrated as being separate from the sensor head. As those skilled in the art will appreciate, the connection interface may be integrally formed with the sensor head and may, for example, be defined by conductor tracks on the sensor head. Further, the excitation and detection circuitry may also be mounted on the sensor head, so that the sensor is carried by a single substrate.

In the above embodiments, the detection circuitry determined the value of an amplitude measure and a phase measure of the received signals. If the amplitude measure was below the threshold, then the detection circuitry did not further process the signals to determine position. Similarly, if the phase measure was not above the phase threshold, then the detection circuitry would not use the signals to calculate a new position. As those skilled in the art will appreciate, this can be achieved either by preventing the detection circuitry from performing further calculations or simply by ignoring or not outputting any measurements obtained when the thresholds are not met. Additionally, in some embodiments, the detection circuitry may only compare the amplitude or the phase measurement with the corresponding threshold rather than both as in the above embodiments. In sensor head D a non-conductive backing layer was provided under the substrate 19. As those skilled in the art will appreciate, instead of providing a backing layer, grooves may be formed in the substrate to the required depth. The appended claims refer to the use of "blind grooves" for this feature. This language has been used to cover both the above possibilities.

Claims

1. A capacitive position sensor comprising: first and second sets of electrodes, each set comprising a plurality of electrodes that are spaced along a measurement path; excitation circuitry operable to generate and to apply a first excitation signal to an excitation electrode of said first set of electrodes and a second excitation signal of opposite polarity to said first excitation signal to an excitation electrode of said second set of electrodes; and detection circuitry operable: i) to receive a first detection signal from a reception electrode of said first set of electrodes which first detection signal varies with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the first set of electrodes; ii) to receive a second detection signal from a reception electrode of said second set of electrodes, which second detection signal varies with the position along said measurement path of said inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the second set of electrodes; iii) to subtract the first detection signal from the second detection signal to generate a difference signal which varies with the position along said measurement path of said inhomogeneity; and iv) to process said difference signal to determine the position of said inhomogeneity along said measurement path; wherein each set of electrodes are symmetrically disposed about an axis that is parallel to said measurement path, with the electrodes of the first set being disposed on one side of said axis and the electrodes of the second set being disposed on an opposite side of said axis.

2. A sensor according to claim 1, wherein said excitation electrodes of said first and second sets of electrodes extend over a measurement range of the sensor.

3. A sensor according to claim 1 or 2, wherein each set of electrodes is disposed about said axis such that, in use, the position along said measurement path of said inhomogeneity affects, in substantially the same way, the capacitive coupling between the excitation and reception electrodes of said first and second sets of electrodes.

4. A sensor according to any preceding claim, wherein the first and second sets of electrodes are symmetrically disposed about said axis such that there is mirror like symmetry between the first and second sets of electrodes.

5. A sensor according to claim 4, wherein the excitation electrode of said first set has mirror symmetry with the excitation electrode of said second set and wherein the reception electrode of said first set has mirror like symmetry with the reception electrode of said second set.

6. A sensor according to claim 4 or 5, wherein the excitation and reception electrodes of each set extend over a measurement range of the sensor.

7. A sensor according to claim 6, wherein the excitation electrode of each set extends into a dead zone of the sensor beyond a measurement range of the sensor, wherein each set of electrodes includes a second reception electrode positioned adjacent said excitation electrode in said dead zone and wherein said detection circuitry is operable to use signals obtained from said second reception electrodes to compensate for changes of permittivity surrounding the sensor.

8. A sensor according to claim 6, wherein the reception electrode of each set extends into a dead zone of the sensor beyond a measurement range of the sensor, wherein each set of electrodes includes a second excitation electrode positioned adjacent said reception electrode in said dead zone, wherein said excitation circuitry is operable to apply a third excitation signal of opposite polarity to said first excitation signal to the second excitation electrode of said first set such that said first detection signal represents the difference in capacitive coupling between the reception electrode and the first and second excitation electrodes of said first set and wherein said excitation circuitry is operable to apply a fourth excitation signal of opposite polarity to said second excitation signal to the second excitation electrode of said second set such that said second detection signal represents the difference in capacitive coupling between the reception electrode and the first and second excitation electrodes of said second set.

9. A sensor according to claims 7 or 8, wherein the electrodes of each set which extend into said dead zone are elongate in a direction transverse to said axis.

10. A sensor according to any preceding claim, comprising at least one reference capacitor associated with said excitation electrode or said reception electrode and connected between said excitation circuit and said detection circuit and operable to provide a dry value capacitance compensation for a capacitor formed by said excitation and reception electrodes.

11. A sensor according to any preceding claim, wherein said first set of electrodes comprises at least one pair of reception electrodes, wherein said detection circuitry is operable to subtract signals obtained from said pair of reception electrodes of said first set to produce said first detection signal, wherein said second set of electrodes comprises at least one pair of reception electrodes and wherein said detection circuitry is operable to subtract signals obtained from said pair of reception electrodes of said second set to produce said second detection signal.

12. A sensor according to any of claims 1 to 10, wherein said first and second sets of electrodes each comprises at least one pair of excitation electrodes and wherein said excitation circuitry is operable to apply excitation signals of opposite polarity to the excitation electrodes of each pair.

13. A sensor according to claim 11 or 12, wherein said at least one pair of electrodes of each set are substantially electrically balanced such that in the absence of said inhomogeneity in the vicinity of the sensor substantially no detection signals are produced by said detection circuitry.

14. A sensor according to claim 11 or 12, wherein said at least one pair of electrodes of each set are not electrically balanced and further comprising a reference capacitor connected between the excitation and detection circuitry operable to substantially balance the pair of electrodes such that in the absence of said inhomogeneity in the vicinity of the sensor substantially no detection signals are produced by said detection circuitry.

15. A sensor according to any preceding claim, wherein said first detection signal includes a component that varies with the position of said inhomogeneity along the measurement path and wherein said second detection signal includes a component that varies with the position of said inhomogeneity along the measurement path and which has opposite polarity to said component of said first detection signal, whereby said components which vary with the position of the inhomogeneity are added together by said detection circuitry upon the subtraction of said first and second detection signals whilst common mode signals are removed by said subtraction.

16. A sensor according to any preceding claim, wherein said excitation circuit is operable to generate and to apply a third excitation signal to a second excitation electrode of said first set of electrodes and is operable to apply a fourth excitation signal to a second excitation electrode of said second set of electrodes, wherein said first detection signal varies with the capacitive coupling between said first and second excitation electrodes and said reception electrode of said first set of electrodes and wherein said second detection signal varies with the capacitive coupling between said first and second excitation electrodes and said reception electrode of said second set of electrodes.

17. A sensor according to claim 16, wherein said difference signal includes first and second components which vary with the position of said inhomogeneity and wherein said detection circuitry is operable to process said difference signal to extract said first and second components and to determine the position of said inhomogeneity using a ratio of said first and second components.

18. A sensor according to claim 17, wherein said detection circuitry is operable to determine said position by calculating a ratiometric arctangent function of said first and second components.

19. A sensor according to claim 18, wherein said detection circuit is operable to combine said first and second components to generate a combined signal whose phase varies with the value of said ratiometric arctangent function and wherein said detection circuitry is operable to determine the value of said ratiometric arctangent function by determining the phase of said combined signal.

20. A sensor according to claim 16, wherein said difference signal includes a component whose phase varies with the value of said ratiometric function and wherein said detection circuitry is operable to determine the value of said ratiometric function by determining the phase of said component.

21. A sensor according to any of claims 1 to 15, wherein said difference signal is a first difference signal and wherein said detection circuit is operable: v) to receive a third detection signal from a second reception electrode of said first set of electrodes, which third signal varies with the position along said measurement path of said inhomogeneity; vi) to receive a fourth detection signal from a second reception electrode of said second set of electrodes, which fourth signal varies with the position along said measurement path of said inhomogeneity; vii) to subtract the third detection signal from the fourth detection signal to generate a second difference signal which varies with the position along said measurement path of said inhomogeneity; and vi) to process said first and second difference signals to determine the position of said inhomogeneity along said measurement path.

22. A sensor according to claim 21, wherein said detection circuitry is operable to process said difference signals to extract said first and second components and to determine the position of said inhomogeneity using a ratio of said first and second components.

23. A sensor according to claim 22, wherein said detection circuitry is operable to determine said position by calculating a ratiometric arctangent function of said first and second components.

24. A sensor according to claim 23, wherein said detection circuitry is operable to combine said first and second components to generate a combined signal whose phase varies with the value of said ratiometric arctangent function and wherein said detection circuitry is operable to determine the value of said ratiometric arctangent function by determining the phase of said combined signal.

25. A sensor according to any preceding claim, wherein each set of electrodes comprises two pairs of electrodes, one pair of electrodes being provided for position measurement and one pair of electrodes being provided to correct for changes in permittivity surrounding the sensor.

26. A sensor according to claim 25, wherein the electrodes of each pair that are provided for position measurement are substantially identical and wherein the electrodes of each pair that are provided to correct for changes in permittivity are not identical.

27. A sensor according to claim 26, wherein one electrode of each pair that is provided for position measurement is designed to be slightly shorter in length along the measurement path than the other electrode of the pair, to introduce an offset in said detection signals to compensate for offsets caused by different permittivities around the sensor.

28. A sensor according to claim 26 or 27, comprising a reference capacitor associated with each pair that is provided to correct for changes in permittivity, to compensate for the electrodes of that pair not being identical.

29. A sensor according to any preceding claim, wherein said detection circuitry is operable to process said difference signal to reduce components that are caused by resistive coupling between the electrodes of each set.

30. A sensor according to claim 29, wherein said detection circuitry is operable to mix said difference signal with a 90 degree phase shifted version of the excitation signal to reduce said resistive coupling components.

31. A sensor according to any preceding claim, wherein said excitation circuit is operable to generate said second excitation signal by inverting said first excitation signal.

32. A sensor according to any preceding claim, wherein said excitation circuit is operable to generate excitation signals that cyclically vary with time.

33. A sensor according to claim 32, wherein said excitation circuit is operable to generate AC excitation signals.

34. A sensor according to claim 32, wherein said excitation circuit is operable to generate excitation signals that comprise sequences of voltage pulses.

35. A sensor according to claim 34, wherein said excitation circuit is operable to generate excitation signals that comprise sequences of modulated voltage pulses.

36. A sensor according to claim 35, wherein said excitation circuit is operable to amplitude modulate or pulse width modulate said sequence of voltage pulses using a modulation signal having a lower frequency than the pulse repetition frequency of said pulses and wherein said detection circuitry is operable to detect modulations of said modulation signal caused the position of said inhomogeneity.

37. A sensor according to any preceding claim, wherein said detection circuit is operable to maintain said reception electrodes at a substantially constant potential and is operable to detect current obtained from the detection electrodes.

38. A sensor according to any preceding claim, wherein said electrodes are formed from conductive tracks on or in a non-conductive substrate.

39. A sensor according to claim 38, wherein a groove is provided in said substrate between the excitation electrode and the reception electrode of said first and second sets.

40. A sensor according to claim 38, wherein said groove is a through groove.

41. A sensor according to any of claims 38 to 40, wherein said electrodes are formed by printing conductive material onto said substrate.

42. A sensor according to any of claims 38 to 40, wherein said electrodes are formed from conductive tracks of a printed circuit board.

43. A sensor according to any of claims 38 to 41, wherein connections to said electrodes are formed by conductive material on or in said substrate.

44. A sensor according to any of claims 38 to 43, wherein said electrodes are formed from conductor tracks arranged in a single layer on or in said substrate.

45. A sensor according to any preceding claim, wherein said detection circuitry is operable to compare an amplitude measure obtained using said difference signal with a threshold value and is operable to process said difference signal to determine said position only if said amplitude measure is above a predetermined threshold value.

46. A sensor according to any preceding claim, wherein said detection circuitry is operable to determine a phase measure using the determined difference signal and is operable to determine said position only if said phase measure is above a predetermined threshold value.

47. A sensor according to any preceding claim, wherein the surface area of said excitation electrodes and/or said reception electrodes are designed so that there is a linear relationship between the position of said inhomogeneity and a measurement parameter obtained using said difference signal.

48. A transducer for use in a capacitive position sensor, the transducer comprising: a non-metallic substrate carrying a plurality of electrodes spaced along a measurement path and connection tracks to said electrodes for connecting the electrodes to excitation and detection circuitry of said sensor; wherein said electrodes are arranged in first and second sets, each set having an excitation electrode and a reception electrode which extend along a measurement path; and wherein said sets of electrodes are symmetrically disposed about an axis that is parallel to said measurement path, with the electrodes of the first set being disposed on one side of said axis and the electrodes of the second set being disposed on an opposite side of said axis.

49. A transducer according to claim 45, wherein said electrodes are formed by printing conductive material onto said substrate.

50. A capacitive position sensor comprising: first and second sets of electrodes, each set comprising a plurality of electrodes that are spaced along a measurement path; excitation circuitry operable to generate and to apply a first excitation signal to an excitation electrode of said first set of electrodes and a second excitation signal of opposite polarity to said first excitation signal to an excitation electrode of said second set of electrodes; and detection circuitry operable: i) to receive a first signal from a reception electrode of said first set of electrodes which first signal varies with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the first set of electrodes; ii) to receive a second signal from a reception electrode of said second set of electrodes, which second signal varies with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the excitation and reception electrodes of the second set of electrodes; iii) to subtract the first signal from the second signal to generate a difference signal which varies with the position along said measurement path of said inhomogeneity; and iv) to process said difference signal to determine the position of said inhomogeneity along said measurement path; wherein the electrodes of said first set are symmetrically disposed about said measurement path relative to the electrodes of said second set, such that there is mirror like symmetry between the electrodes of the first and second sets.

51. A capacitive position sensor comprising: a first electrode that extends along a measurement path over a measurement range; first and second pairs of electrodes arranged along the measurement path adjacent said first electrode such that electrodes of each pair are interleaved along said measurement path with the electrodes of the other pair; excitation circuitry operable to generate and to apply first and second excitation signals of opposite polarity to one of said first electrode and said first and second pairs of electrodes; and detection circuitry operable: i) to receive signals from the other one of said first electrode and said first and second pairs of electrodes; ii) to process said signals to obtain a first detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said first pair and a second detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said second pair, which first and second detection signals vary with the position of an inhomogeneity along said measurement path; and iii) to process said first and second detection signals to determine the position of said inhomogeneity along said measurement path; wherein said first pair of electrodes is provided for position measurement and the second pair of electrodes is provided to correct for changes in permittivity surrounding the sensor.

52. A capacitive position sensor comprising: a first electrode that extends along a measurement path over a measurement range; first and second pairs of electrodes arranged along the measurement path adjacent said first electrode such that electrodes of each pair are interleaved along said measurement path with the electrodes of the other pair; excitation circuitry operable to generate and to apply first and second excitation signals of opposite polarity to one of said first electrode and said first and second pairs of electrodes; and detection circuitry operable: i) to receive signals from the other one of said first electrode and said first and second pairs of electrodes; ii) to process said signals to obtain a first detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said first pair and a second detection signal that varies with the difference in capacitive coupling between said first electrode and the respective electrodes of said second pair, which first and second detection signals vary with the position of an inhomogeneity along said measurement path; and iii) to process said first and second detection signals to determine the position of said inhomogeneity along said measurement path; wherein the electrodes of said first pair are substantially identical and wherein the electrodes of the second pair are not identical.

53. A sensor according to claim 52, wherein one electrode of said first pair is designed to be slightly shorter in length along the measurement path than the other electrode of the pair, to introduce an offset in said detection signals to compensate for offsets caused by different permittivities around the sensor.

54. A sensor according to claim 52 or 53, comprising a reference capacitor associated with the second pair and connected between the excitation and detection circuitry, which reference capacitor is provided to compensate for the electrodes of the second pair not being identical.

55. A capacitive position sensor comprising: first and second subsets of electrodes, the electrodes of which extend along a measurement path of the sensor; excitation circuitry operable to generate and to apply an excitation signal to said first subset of said electrodes; and detection circuitry operable: i) to receive first and second signals from said second subset of said electrodes, which first and second signals vary with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the electrodes of the first and second subsets; ii) to process said first and second signals to determine the position of said inhomogeneity along said measurement path; and iii) to output the determined position; wherein said detection circuitry is operable to determine an amplitude measure for said first and second signals, to determine if said amplitude measure is less than a threshold value and to prevent processing of said signals to determine said position, or to prevent outputting of said determined position, when said amplitude measure is less than said threshold value.

56. A capacitive position sensor comprising: first and second subsets of electrodes, the electrodes of which extend along a measurement path of the sensor; excitation circuitry operable to generate and to apply an excitation signal to said first subset of said electrodes; and detection circuitry operable: i) to receive first and second signals from said second subset of said electrodes, which first and second signals vary with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the electrodes of the first and second subsets; ii) to process said first and second signals to determine the position of said inhomogeneity along said measurement path; and iii) to output the determined position; wherein said detection circuitry is operable to determine a phase measure from said first and second signals, to determine if the phase measure is less than a threshold value and to prevent processing of said signals to determine said position, or to prevent outputting of said determined position, when said phase measure is less than said threshold value.

57. A capacitive position sensor comprising: first and second subsets of electrodes, the electrodes of which extend along a measurement path of the sensor; excitation circuitry operable to generate and to apply an excitation signal to said first subset of said electrodes; and detection circuitry operable: i) to receive first and second signals from said second subset of said electrodes, which first and second signals vary with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the electrodes of the first and second subsets; and ii) to process said first and second signals to determine the position of said inhomogeneity along said measurement path; wherein said electrodes are formed on or in a non-conducting substrate and wherein a blind groove is provided between the electrodes of said first and second sub- sets.

58. A capacitive position sensor comprising: first and second subsets of electrodes, the electrodes of which extend along a measurement path of the sensor; excitation circuitry operable to generate and to apply an excitation signal to said first subset of said electrodes; and detection circuitry operable: i) to receive first and second signals from said second subset of said electrodes, which first and second signals vary with the position along said measurement path of an inhomogeneity which affects the capacitive coupling between the electrodes of the first and second subsets; and ii) to process said first and second signals to determine the position of said inhomogeneity along said measurement path; wherein each of said first and second subsets of electrodes includes an electrode or an elongate portion that is elongate and whose axis extends in a direction transverse to said measurement path.

59. A sensor according to any of claims 50 to 58, comprising the sensor features of any of claims 1 to 47.

60. A method of sensing the position of an inhomogeneity characterised by using a sensor according to any of claims 1 to 47 or 50 to 58 or a transducer according to claim

48 or 49.