WO2006061213A1 - Method and device for producing an output signal having reduced out-of-band radiation - Google Patents

Abstract

The invention relates to a method and to a device for producing an output signal (160) based on a complex, modulated input signal (102) which is divided into an amplitude signal (104) and a phase signal (106), which provide an amplitude signal path (120) and a phase signal path (130). A first group running time distortion is brought about in the signal path which is counteracted by introducing a second group running time distortion by means of a second filter in the amplitude signal path (120) or phase signal path (30). Also, the output signal (160) based on the signals is produced on an outlet of the amplitude signal path (120) and on an outlet of the phase signal path (130).

Description

 Method and device for generating an output signal with reduced out-of-band radiation

description

The present invention relates to a method and apparatus with reduced out-of-band radiation, e.g. a transmitter of complex modulated signals with reduced out-of-band radiation.

Next generation communication standards such as EDGE or UMTS, unlike GSM and DECT, use modulation schemes without a constant envelope, e.g. linear envelope modulation methods, which usually have both a modulation of the phase and the amplitude. Therefore, high linearity, low efficiency power amplifiers must be used to meet the requirements of out-of-band radiation. Class AB linear power amplifiers achieve a reasonable level of efficiency, over 50%, when operating close to the saturation point. In most cases, however, the transmission power of the handsets will be widely controlled, especially if they are near a base station. In this case, the efficiency falls far below 10%.

Polar transmitter architectures, as used in current GSM terminals or GSM handsets, offer advantages over the linear or quadrature modulation architectures. In this case, a complex modulated signal is separated into an amplitude signal and a phase signal, that is converted into a complex modulated signal in polar coordinate representation. This can be done for example in a digital part of the transmitter by a CORDIC algorithm (CORDIC = COordinate Rotation Digital Computer) or in an analog part by a limiter and an envelope detector. For complex signals or complex numbers in general, instead of the term Amplitude also uses the terms magnitude or magnitude, instead of the term phase also the term argument and in communications engineering in particular, the terms envelope and envelope signal are used instead of the term amplitude signal.

With the phase information or the phase signal, for example, a phase locked loop is modulated. This phase signal has a constant envelope after separation and therefore can be amplified distortion-free with an efficient, nonlinear power amplifier. The envelope or the amplitude signal can be impressed on the phase signal again by modulating a supply voltage of the power amplifier with the envelope or the amplitude signal. A common polar transmitter architecture 300 is shown schematically in FIG. 3 shows a conversion device 310, an amplitude signal path 320 comprising a first digital-to-analog converter 322 and a voltage follower 324, a phase signal path 330, a second digital-analog converter 332, a voltage-controlled oscillator 334, which also is called a "Voltage Controlled Oscillator" (VCO), and a phase-locked loop 336, also referred to as "Phase Locked Loop (PLL)", and a high-frequency power amplifier 340.

The amplitude signal path is also referred to as amplitude path, amplitude path or amplitude branch and the phase signal path also as phase path, phase path or phase branch.

Conversion device 310 converts a complex modulated Cartesian coordinate representation input signal, defined by its in-phase signal 3501 and its quadrature signal 350Q, into a corresponding complex modulated signal in polar coordinate representation, represented by an amplitude signal 350A and Phase signal 350P is defined. The amplitude signal 350A and the phase signal 350P are further processed separately, the amplitude signal 350A in the amplitude signal path 320 and the phase signal 350P in the phase signal path 330.

The amplitude signal 350A at an input of the amplitude signal path is converted by the first digital-to-analog converter 322 into an analog amplitude signal, wherein the analog amplitude signal forms an input signal of the voltage follower 324. The voltage follower 324 controls the supply voltage of the high frequency power amplifier 340, thus effecting an amplitude modulation corresponding to the original digital amplitude signal 350A.

The phase signal 350P at one input of the phase signal path 330 is converted into an analog phase signal by the second digital-to-analog converter 332 and generates a high frequency phase modulated signal by means of the voltage controlled oscillator 334 and the phase locked loop 336, which together form a phase modulator. As described above, this high-frequency, phase-modulated signal is modulated by means of the high-frequency power amplifier to amplify the amplitude signal and thus forms as output an amplified version of the original signal 350A, 350P.

This technology is also referred to as Envelope Elimination and Restoration (EER), or "Envelope Removal and Restoration." EER technology was first introduced in the single-sideband transmission by envelope elimination and restoration by L.R. Barge in Proc. of the IRE, Vol. 40, July 1952, pp. 803-806.

Corresponding amplifiers or transmitters are also referred to as EER amplifiers or EER transmitters. In order for the original signal in the RF power amplifier 340 to be regenerated in a polar transmitter or EER transmitter, an amplitude signal and a phase signal of the output signal 360 must be present outside the modulation bandwidth. extinguish each other. This only works if the propagation times in the two paths or signal paths, the amplitude signal path and the phase signal path, are exactly compensated.

Splitting the complex modulated input signal into the amplitude signal and the phase signal is a nonlinear function. 4 shows simulated spectra of an original EDGE signal 410, as well as the corresponding amplitude signal (AM signal) 420 and the corresponding phase signal (PM signal) 430 after conversion from a Cartesian coordinate representation to a polar coordinate representation. Along the X-axis the frequency is plotted in kHz, on the Y-axis the amplitude in dB. It can clearly be seen that both the amplitude signal 420 and the phase signal 430 have a significantly wider spectrum than the original EDGE signal 410.

5 shows simulations of the frequency spectra using the example of an EDGE signal in the case of faulty delay compensation between the amplitude signal path and the phase signal path. Along the X-axis the frequency is plotted in kHz, on the Y-axis the amplitude in dB. The frequency spectra of the output signal 360 at an erroneous transit time equalization of 1/128, s. Frequency spectrum 510, a faulty delay compensation of 1/64 symbol duration, s. Frequency spectrum 520, and with a faulty runtime compensation of 1/32, s. Frequency spectrum 530, have a significant spectral expansion compared to the frequency spectrum 540 of the original EDGE input signal. Even with a runtime error of 1/32 symbol duration, the spectral mask 550 prescribed in the ETSI specification is violated.

Therefore, the propagation delays in the amplitude signal path and the phase signal path must be set very accurately. Various methods are known for the compensation of the different transit times in the amplitude signal path and the phase signal path. A common approach is to introduce a constant or frequency-independent duration. This is synonymous with the introduction of a constant or frequency-independent group delay or a linear phase response and can be done for example in the digital part of the transmitter by delay elements or delay elements.

In the following, selected publications are listed and their solutions described to exemplify the state of the art.

In the 1998 publication "High-Efficiency L-Band Kahn-Technique Transmitter" by FH Raab, BE Sigmon, RG Myers and RM Jackson in the IEEE MTT-S International Microwave Symposium Digest of 1998, there is a frequency-independent delay in the phase signal path "In the classical EER transmitter, a time delay corresponding to the group delay of the output filter in the class-S modulator is introduced in the phase signal path to tune the amplitude and phase in the output signal."

Also, the publication "High-efficiency Multimode RF / VHF Transmitters for Communication and Jamming", by Frederick H. Raab, Daniel J. Rupp, IEEE Military Communications Conference, 1994, MILCOM '94, Conference Record, 1994, describes a frequency-independent one Delay in the phase path.

US patent 6084468 describes a transmitter according to the EER principle, wherein in the amplitude signal path the low-frequency amplitude signal components or envelope components are generated by a class-S modulator, the high-frequency amplitude signal components by a class B amplifier. The delay in the amplitude path is in the phase path through a delay line, ie by frequency-independent delay balanced, balanced. Alternatively, one delay line each can be used in the amplitude signal path and the phase signal path. As a result, the absolute delay of the line is not so important, but much more the difference between the two lines, as is advantageous for example in temperature drift.

In US Patent 2004 / 0247041A1 it is assumed that the amplitude signal path and the phase signal path of the polar modulator and the polar transmitter for the amplitude signal and for the phase signal have a transit time or delay which are different for the amplitude signal branch and the phase signal branch can, but are independent of the signal frequency constant or frequency-independent. Thus, an adjustable delay must be incorporated in one of the signal paths in order to ensure that related amplitude signals and phase signals in the amplitude signal path and in the phase signal path arrive simultaneously at the power amplifier.

The patent further describes an idea of how to make a temporal resolution of this delay finer than the length of the clock period of digital signal processing. The patent assumes that the group delay, ie the cycle time of the signals, as stated above, is frequency-independent for all components, and thus the delay to be inserted also has a group delay that is as constant as possible over the complete, for the respective signal is interesting frequency range. In the patent, it is explicitly pointed out in paragraph [0005] that the group delay delay can only be approximately constant and that this is always a bit annoying.

The publication "An Application of Envelope Elimination and Restoration Technique for Class-E RF Power Amplifiers" by Gokhan Remzi Aricioglu, Osman Palamutcuogullari, International Conference on Electrical and Electronics Engineering. gineering ELECO '2003, December 2003 describes on page 4 that a delay equalizer should be inserted in the amplitude path, but does not disclose any technical teaching as to how this can be achieved.

Dietmar Rudolph also describes in his script for the lecture "Digital Radio Systems" in Part 10 ^λ ΕER Technique in Digital Transmission ", Chapter 8 on page 22, that an equalization of the propagation time due to the low-pass filter in the amplitude path is required. However, there is no teaching as to how this can be achieved: "The phase response of the TP [low-pass] filter in the A [amplitude] branch is assumed to be linear ... IA, it will therefore be necessary to level the amplitude in the amplitude branch by means of an equalizer. Instead, D. Rudolph suggests a feedback as a solution in Chapter 10, "Improvements by Neglecting" on pages 23 and 24 of the same part. By means of negative feedback the disturbance in the output signal is reduced and the transmission behavior is linearized. In this case, a fundamental structure for a linearization by means of feedback has an additional receiver and reference demodulator, which converts the transmission signal into a corresponding in-phase signal and a corresponding quadrature signal and leads back to an adapted modulation and signal shaping.

Kang-Chun Peng, Je-Kuan Jau and Tzyy-Seng Horng describe in their publication "A Novel EER Transmitter Using Two-Point Delta-Sigma Modulation Scheine for WLAN and 3G Applicatons", IEEE MTT-S International Microwave Symposium Digest. 2002, feedback in the amplitude signal path to keep the delay small, and a so-called predistortion filter is inserted in the phase signal path to smooth the transfer function of the phase path with respect to its amplitude gain and its phase response (delay).

The publication "An IC for Linearizing RF Power Amplifiers Using Envelope Elimination and Restoration" by David Su, William McFarland, IEEE Journal of Solid State Circuits, Vol. 12, December 1998 teaches that the delay could be compensated by a low-pass filter in the amplitude path, by a constant delay or a constant, frequency-independent transit time in the phase signal path, but preferably a feedback in Amplituden- signal path, on page 5: "The dominant source of the delay in the amplitude signal path in Fig. 3 (b) is the switching power supply low-pass filter. Instead of using an explicit delay in the phase signal path to match that of the amplitude signal path, this concept uses feedback to reduce the delay of the amplitude signal. "

US patent 4972440 describes a transmitter according to the principle of polar modulation, wherein phase distortions in the amplitude modulation are compensated by an additional phase offset, which is realized by means of a PROM (PROM = Programmable Read OnIy Memory).

US Pat. No. 5,861,777 describes a transmitter according to the EER principle, wherein phase distortion in the amplitude modulation is compensated by predistortion in a high frequency driver amplifier. Furthermore, the transmitter has a static or frequency-independent delay compensation for the low-pass in the amplitude path.

Both patents, US Pat. No. 4,972,440 and US Pat. No. 5,861,777, propose solutions to the problem that a non-linear power amplifier, referred to as Power Amplifier (PA), when modulating the amplitude, will normally also slightly shift the phase, ie The amplitude modulation (AM) of the power amplifier is also the output of a small unwanted phase modulation (PM) or (AM / PM). In US Pat. No. 4,972,440, not only is the unwanted phase modulation (AM / PM) produced by the amplitude modulation, but also also compensates for an undesired additional amplitude modulation (AM / AM) generated in the actual amplitude modulation, ie the problem that the amplitude of the envelope at the power amplifier output does not change perfectly proportionally to the voltage at the control input or the ratio control voltage to high-frequency amplitude behaves a bit nonlinear. The two US patents describe solutions for how power amplifier distortions can be reduced due to the varying instantaneous powers, but do not take into account frequency-dependent distortions or do not contain any solutions for frequency-dependent distortions.

Filtering is usually necessary both in the amplitude signal path and in the phase signal path.

For example, the BIAS control or voltage follower 324, see FIG. 3, has a low-pass behavior on the power amplifier 340 for linear recovery of the envelope information or of the amplitude signal for stability reasons. If, for the purpose of improving the overall efficiency of the arrangement, a class S amplifier is used as amplitude modulator or high-frequency power amplifier, then the switching losses increase with increasing transmission bandwidth. Therefore, a reduced bandwidth is advantageous here.

In addition, for example, the first digital-to-analog converter 322, see FIG. 3, generates quantization noise and mirror products in the amplitude signal path 320. The amount of quantization noise is dependent on the word width of the first digital-to-analog converter 322, the position of the mirror products of the sampling frequency. These spurious products are impressed on the power amplifier or RF power amplifier 340 along with the amplitude signal, the RF signal 360, and are directly visible in the output spectrum. Therefore, a low pass filter is present at an output of the first digital Analog converter 322 necessary to suppress these interfering products sufficiently. The lower the cutoff frequency and the higher the order of the filter, the lower the sample rate and word width in the first digital-to-analog converter can be selected.

These filters and the filter characteristics, e.g. the BIAS control or the voltage follower 324, but not only affect the amplitude frequency response, but also generate group delay distortions, so have no constant group delay, but a frequency-dependent group delay and thus a nonlinear phase frequency response.

Fig. 6 shows a simulated spectrum of an EDGE signal at a 400 kHz first order low pass filter. The frequency in kHz is plotted on the X-axis and the amplitude in dB is plotted on the Y-axis. In this case, the spectrum 610 represents the spectrum of the original EDGE signal or input signal, and the spectrum 620 the spectrum of the output signal 360 with a 400 kHz first-order low-pass filter in the amplitude signal path, the amplitudes of the spectrum 620 being in the ranges around +/- 500 kHz above the spectrum mask specified in the ETSI specification and thus violate the spectral mask. The group delay of the 400 kHz filter is compensated by a static or frequency-independent delay already as far as possible. The simulation of the frequency spectrum 620 with a 400 kHz first-order filter in the amplitude signal path shows a comparable spectral broadening as a static transit time error, as shown in FIG. 5 in particular for a delay of 1/32 symbol duration as the frequency spectrum 530. Thus, due to the 40OkHz low pass filter, as with the runtime error of 1/32 symbol duration, the spectral mask prescribed in the ETSI specification is violated. A disadvantage of the above-described solutions with constant propagation times is that they can not compensate for any frequency-dependent differences between the amplitude signal path and the phase signal path, and thus the required exact tuning of the transit times between the amplitude signal path and the phase signal path can not be fulfilled ,

Further known methods are based on the feedback of the amplitude signal or the feedback of the amplitude signal and the phase signal. A disadvantage of the method by means of feedback, the high cost, for example, due to the separate receiver and the separate reference demodulator, which are necessary to reduce the noise in the output signal or to linearize the transmission behavior of the transmitter.

In summary, therefore, it can be stated that the known methods or transmitters can either neglect or not compensate for the group delay distortions and the frequency-dependent transit time differences between the amplitude signal path and the phase signal path and therefore produce output signals with high outer band radiation, or the group delay distortions, in particular of the amplitude signal path only reduce with great effort.

The object of the present invention is to provide an efficient concept for exactly matching the propagation times and in particular the frequency-dependent propagation times of the amplitude signal path and the phase signal path so as to reduce the out-of-band radiation.

This object is achieved by a method for generating an output signal with reduced outer band emission according to claim 1, by a computer program according to claim 13 and by a device according to claim 14. According to the invention there is therefore provided a method of generating an output signal based on a complex modulated input signal decomposed into an amplitude signal and a phase signal provided to an amplitude signal path and a phase signal path, causing a first group delay distortion in one of the signal paths the output signal is generated based on the signals at an output of the amplitude signal path and an output of the phase signal path and wherein the method comprises introducing a second group delay distortion into the phase signal path or into the amplitude signal path to counteract the first group delay distortion.

The invention therefore further provides an apparatus for generating an output signal, the apparatus being based on a complex modulated input signal decomposed into an amplitude signal and a phase signal having an amplitude signal path having an input for receiving the amplitude signal, a phase signal path having an input to Receiving the phase signal, wherein one of the signal paths causes a first group delay distortion, a combiner that generates the output signal based on a signal at an output of the amplitude signal path and a signal at an output of the phase signal path, and a device configured to perform a second group delay distortion to counteract the first group delay distortion.

The present invention is based on the finding that frequency-dependent transit time differences between the amplitude signal path and the phase signal path can be reduced by a filter with frequency-dependent group delays or group delay distortions, and can be exactly compensated for with appropriate dimensioning.

Group delay distortions can occur both in the amplitude signal path and in the phase signal path, whereby natural according to some circuit elements produce larger group delay distortions than others. A first filter or an amplitude signal processing unit or a voltage regulator of the combiner are, for example, circuit elements which can generate significant group propagation time distortions in the amplitude signal path. In addition, of course, a phase signal processing unit, for example, designed as a phase-locked loop, or a digital-to-analog converter generate group delay distortions, but which are generally less than the group delay distortions described above in the Amplitudensignalweg.

For the reduction of the frequency-dependent transit time differences between the amplitude signal path and the phase signal path, therefore, the counteracting with regard to the significant group delay distortions is essential, however, an exact compensation is only necessary when all group delay distortions, i. by an adjustment of a first total group delay distortion of the amplitude signal path, which comprises the group delay distortions of all circuit elements in the amplitude signal path, and a second total group delay distortion of the phase signal path, which comprises the group delay distortions of all circuit elements of the phase signal path. This is true regardless of whether the means for generating a second group delay delay, e.g. a second filter in which the amplitude signal path or the phase signal path is arranged and also irrespective of whether the amplitude signal path or the phase signal path have the individual group delay distortions, in particular the significant group delay distortions.

In one embodiment of the present invention, the amplitude signal path includes a first digital-to-analog converter and a first filter, wherein the first filter effects the first group delay distortion, and the phase signal path comprises a second digital-to-analog converter, the second filter being located in the amplitude signal path and since- is dimensioned such that it compensates the group delay distortion of at least the first filter or an amplitude signal processing unit, but preferably, as described above, the first total group delay distortion.

In embodiments according to the invention in which one of the signal paths has a significant group delay distortion with respect to the other signal path, the second filter is preferably arranged in the other signal path.

For example, in preferred embodiments of the present invention in which the significant group delay distortion occurs in the amplitude signal path, e.g. through the first filter, therefore, the second filter is preferably arranged in the phase signal path. If one introduces the second filter with the same group delay distortion into the phase signal path or generally into the other signal path, the frequency-dependent propagation times caused by the filtering in the two paths are identical. This leads to a significantly improved cancellation of the amplitude signal and the phase signal outside the modulation bandwidth.

For example, in preferred embodiments where the second filter is disposed in the phase signal path, the second filter is dimensioned such that the second group delay distortion substantially corresponds to the first group delay distortion or the first total group delay distortion, or the second total group delay distortion corresponds to the first total group delay distortion , The same applies vice versa for the second filter in the amplitude signal path.

In a particularly preferred embodiment of the present invention, the second filter is designed as a digital filter, since this is the easiest to implement. Preferred embodiments of the present invention will be explained below in more detail with reference to the accompanying drawings. Show it:

Fig. 1 is a schematic block diagram of a device according to the invention;

Fig. 2a is a schematic block diagram of a preferred embodiment of a device according to the invention;

2b shows a basic block diagram of a further embodiment of a device according to the invention;

3 is a schematic block diagram of a possible transmitter;

4 shows a simulated frequency spectrum of an EDGE signal and the corresponding amplitude signal and the corresponding phase signal after the conversion of the original EDGE signal into a polar coordinate representation;

5 shows a simulated frequency spectrum of an EDGE signal with a delay between the amplitude signal and the phase signal of 1/128, 1/64 and 1/32 symbol lengths relative to the frequency spectrum of the original EDGE signal; and

6 shows a simulated spectrum of an EDGE signal with a 400 kHz first order low-pass filter a) in the amplitude signal path, b) in the amplitude signal path and phase signal path with respect to the original EDGE signal. 1 shows a basic block diagram of a device 100 according to the invention, embodied here for example as a transmitter, with a complex modulated input signal 102, which is decomposed into an amplitude signal 104 and a phase signal 106, which in analog or digital form an amplitude signal path 120 and a Phase signal path 130 can be provided, wherein a signal path causes a first group delay distortion, by means of a second filter 121 in the phase signal path 120 or a second filter 131 in the Amplitudensignalweg 130, a second group delay distortion is introduced to counteract the first group delay distortion or the run times and in particular the frequency-dependent maturities of the amplitude signal path and the phase signal path to match exactly. In this case, the output signal 160 is generated based on the signals at an output of the amplitude signal path 120 and at an output of the phase signal path 130 by means of a combiner 140.

2 shows a schematic illustration or a basic block diagram of a preferred device according to the invention, here formed for example in a polar transmitter architecture, with a conversion device 210, an amplitude signal path 220, a first digital-to-analog converter 222, a first filter 223 and an amplitude signal processing unit 224, a phase signal path 230, which has a second filter 231a, 231b, a second digital / analog converter 232 and a phase signal processing unit 233, and a combiner 240. In this case, the amplitude signal processing unit 224 or the composite forms Amplitude signal processing unit 224 and combiner 240 an amplitude modulator. Furthermore, the first filter 223 is designed as a low-pass filter.

As previously explained, the complex modulated input signal is in the form of an in-phase signal 2501 and its quadrature signal 250Q and is converted by means of the conversion signal. device from the Cartesian coordinate representation or IQ representation into polar coordinates or an amplitude signal A (t) 250A and a phase signal Φ (t) 250P decomposed.

The amplitude signal 250A is supplied to the amplitude signal path 220 and converted by the first digital-to-analog converter 222 from a digital signal into a corresponding analog amplitude signal and fed to a first filter 223. The characteristic of the low-pass filter or first filter 223 in the amplitude path is generally predetermined since, as described above, it serves to suppress interfering products or is tuned to the control behavior of the amplitude signal processing unit or of the amplitude modulator. The filtered, analog amplitude signal is then supplied to the amplitude signal processing unit 224, which can be designed as a voltage follower or impedance converter, voltage control, BIAS control, driver amplifier or amplifier or also referred to as such.

The combiner 240 is formed in FIG. 2 as a high-frequency power amplifier. Alternatively, however, it may also be embodied as a pure combiner, for example if the amplitude signal has already been amplified by an amplifier in the amplitude signal path, or it may be designed for other carrier frequency ranges. In this case, for example, the method with the look-up table predistortion of US Pat. No. 4,974,440 can additionally be used in order to reduce the power-dependent distortions of the power amplifier discussed above. However, this is not shown separately in FIG. 2, since it is assumed for the exemplary embodiment that the power amplifier 240 or also 340 in FIG. 1 either has only such negligible negligence, which is unlikely in a real power amplifier , or this is compensated by a suitable measure. The phase signal processing unit 233 is implemented in this embodiment as a phase modulator comprising a voltage controlled oscillator 234 and a phase locked loop 236. Alternatively, however, other embodiments for, for example, a phase modulation are possible.

In another embodiment of the invention, the constant envelope phase signal 250P in the phase signal path is converted to a corresponding Cartesian or IQ signal, and this in turn is converted into a high frequency, phase modulated signal by means of an IQ mixer.

The phase signal 250P is supplied to the phase signal path 230. In a particularly preferred embodiment of the device according to the invention, the second filter 231a is realized in a digital part or arranged between the input and the second digital-to-analog converter 232 of the phase signal path 230, since it can be realized there most easily. By optimizing the coefficients, the simulation of the group delay or the group delay distortions of the amplitude signal path 220 and, in particular, the filter or low-pass filter 223 and the amplitude signal processing unit 224 and the first digital-to-analog converter 222 can generally be easily achieved by optimizing the coefficients. However, the second filter 231b can also be arranged at a different location in the phase signal path between the second digital-to-analog converter 232 and the high-frequency mixer 240, for example in the analog baseband.

The influences of the group delay distortions, for example of the amplitude signal path 220, can be largely compensated by a second filter 231a, 231b having the same group delay in the phase signal path 230, for example by an identical lowpass or an allpass. If there is already a filter in the phase signal path 230, for example a loop filter of the phase-locked loop, then this becomes necessary when dimensioning of the additional second filter, so that the second total group delay distortion substantially corresponds to the first total group delay distortion.

In the following, an illustrative example of device dimensioning in which the bandwidth of the phase modulator is much higher than the bandwidth of the amplitude path, i. whose group delay behavior could be neglected. The dimensioning of the second filter for equalizing the first group delay is therefore a simple simulation of the low-pass behavior of the amplitude signal path, which is modulated, for example, as a first-order pole in the frequency domain. This pole is approximated digitally by a second-order HR filter, i. the group delay of the digital filter matches the analogue only in the lower frequency range. The approach is good enough to see the improvement. With an HR filter of higher order, which is better able to simulate the group delay behavior of the analog low-pass filter, the approximation can be further improved.

FIG. 6 shows a simulated spectrum of an EDGE signal when using a 400 kHz first-order low-pass filter. The frequency in kHz is plotted on the X-axis and the amplitude in dB is plotted on the Y-axis. In this case, the spectrum 610 represents the spectrum of the original EDGE signal or input signal, the spectrum 620 the spectrum of the output signal with a 400 kHz first order low-pass filter alone in the amplitude signal path, the group delay of the low-pass filter being compensated by a constant delay and the Spectrum 640 the spectrum of the output signal at a 400 kHz first order lowpass filter in the amplitude signal path and additionally a same in the phase signal path. It can be clearly seen that the spectrum 640 has a much lower out-of-band emission than the spectrum 620 or, in contrast to the spectrum 620, the amplitudes far below that of FIG spectrum spectrum prescribed in the ETSI specification.

The second filter 231a, 231b in the phase signal path 230 can be dimensioned such that the second group delay distortion has a constant group delay or frequency-independent delay component, which is dimensioned so that the group delay of the amplitude signal path and the phase signal path in their frequency-dependent and their frequency-independent portions substantially coincide, so that an exact matching of the propagation times of the amplitude signal path and the phase signal path is achieved.

Genevieve Baudoin, Corinne Berland, Martine Villegas, Antoine Diet, IEEE MTT-S International Microwave, in the publication "Influence of time and processing between phase and envelope signal in linearization systems using Envelope Elimination and Restoration, application to hiperlan2" Symposium Digest, 2003, section C on page 4 simulates a filter in the phase signal path, but not together with an amplitude filter and not intentionally added to improve the spectrum, but as a model for the unavoidable low-pass effect of the phase modulator or high-frequency mixer therefore, no solution according to the invention for the problem indicated.

Alternatively, the second filter can also be arranged in the same signal path, in this case the amplitude signal path. However, this has the disadvantage over the preferred embodiment with a second filter in the other signal path that higher frequencies are thereby increased, that is, the second filter has a high-pass characteristic, which in turn places higher demands on a digital-to-analog conversion. In addition, increasing the slew rate will be detrimental if a class S amplifier is to be used. The second filter It can either be arranged in the digital part between the input and the first digital / analog converter 222 of the amplitude signal path 220 or be arranged in the analog part between the first digital / analog filter 223 and an output of the amplitude signal path 220. These possibilities of arranging the second filter in the amplitude signal path are not shown in FIG. In the solution with a second filter in the amplitude signal path 220, a device, for example a group delay element, can additionally be introduced into the phase signal path 230, which compensates for a possible constant group delay time difference between the amplitude signal path and the phase signal path.

Another possibility is to compensate the group delay distortions in the amplitude signal path by an all-pass filter as a second filter. Dietmar Rudolph, for example, points to an equalization of the group delay in the amplitude path in his script for the lecture "Digital Radio Systems", Chapter 10, but does not describe how it is done higher complexity of the allpass filter in the amplitude signal path compared to a lowpass or allpass filter in the phase signal path.

Although above exemplary embodiments have been described above, in which the amplitude signal path has the more significant group delay distortion, and the second filter is preferably arranged in the other, the phase signal path, it should be noted that the present invention is not limited to these embodiments, but in particular Embodiments applies, in which the phase signal path has the more significant group delay distortion, and the second filter in the other, in this case the amplitude signal path, is introduced. FIG. 2b essentially corresponds to FIG. 2, wherein the conversion of the IQ coordinates into polar coordinates takes place here in analog form, for example by means of a limiter and an envelope detector, and accordingly the digital / analogue converters 222, 232 in front of the conversion unit 210 are arranged.

Embodiments of the invention may alternatively also have a second filter in both signal paths, the amplitude signal path and the phase signal path, in order to be able to tune the individual filters more flexibly.

As previously discussed, a transmit architecture that uses constant envelope signals has v. A. the advantage of higher efficiency, since nonlinear power amplifiers can be used, which have a higher efficiency than linear power amplifiers, v.a. in operation with a lot of sub-control or backoff.

High-frequency signals that do not have a constant envelope need linear amplifiers with less efficiency, viz. at backoff. But they have other advantages, such as higher spectral efficiency, i. a higher data transmission rate per bandwidth, or the ability to implement CDMA (Code Division Multiplex Access) using e.g. send several phones simultaneously on one frequency and the base station on the basis of the spreading codes can separate the individual signals again.

The polar system allows the amplification of linear signals, i. with variable envelope, with non-linear amplifiers that are more energy efficient than linear PAs.

In contrast to the known solutions of polar systems, which are based on a compensation of a frequency-independent delay difference, it is the case of the method according to the invention or the device according to the invention that an analog component is a frequency-dependent group. has a running time. In most cases, as previously mentioned, the amplitude signal processing unit 224 and the voltage regulator 224, respectively, are for amplitude control of the power amplifier or combiner 240, which exhibits the strongest lowpass response.

In the literature, it has always been proposed to compensate for this frequency-dependent group delay by an all-pass in the amplitude signal branch, which has exactly opposite runtime behavior, so that a combined group delay of both elements and thus almost the entire amplitude path becomes frequency-independent again.

The core of the device according to the invention or of the method according to the invention consists, as explained above, in that one can also achieve an improvement in the spectrum of the output signal or transmission signal 260, if the frequency-dependent group delay, for example in phase signal path 130, 230 is bent so that it corresponds to the frequency-dependent group delay in the Amplitudensignalweg 120, 220, for example by introducing a low pass or allpass 131, 231a, 231b. This does not give a perfect signal at the end, because now both signal paths, ie the amplitude signal path 120, 220 and the phase signal path 130, 230 have a frequency-dependent transit time, but a better one than before, since both signal paths behave at least the same.

The advantage over the solution with an allpass in the amplitude signal path is that the low-pass characteristic in the phase signal path 130, 230 can be realized with significantly less effort, for example multipliers, than the all-pass in the amplitude signal path 120, 220, in particular in digital signal processing.

If this has ensured that both signal paths have the same characteristics with respect to the frequency-dependent group delay, then in general still ensure that the absolute delay or frequency-independent delay is the same in both signal paths, this can be achieved, for example, by known methods, eg by delay elements or integrated into the second filter 131, 231a, 231b as described above.

In summary, it can be stated that with a device according to the invention or with a method according to the invention, time differences and, in particular, frequency-dependent transit time differences in the amplitude signal path 120, 220 and the phase signal path 130, 230 are exactly compensated by the second filter 121, 131, 231a, 231b can. In preferred exemplary embodiments of the device or method according to the invention, the second filter 131, 231a, 231b is arranged in the phase signal path 130, 230, and in a particularly preferred embodiment also equals the frequency-independent transit time differences between the amplitude signal path 120, 220 and the phase signal path 130, 230 exactly. Thus, the spectra outside the modulation bandwidth cancel out significantly better, so that a lower outer band radiation or improved outer band properties can be achieved.

Claims

claims A method of generating an output signal (160, 60) based on a complex modulated input signal (102) which is decomposed into an amplitude signal (104, 250A) and a phase signal (106, 250P) corresponding to an amplitude signal path (120, 220) and a phase signal path (130, 230), wherein a first group delay distortion is effected in one of the signal paths, and wherein the output signal (160, 260) is based on the signals at an output of the amplitude signal path (120, 220). and an output of the phase signal path (130, 230), the method comprising the following step: Introducing a second group delay distortion into the phase signal path (130, 230) or into the amplitude signal path (120, 220) to counteract the first group delay distortion. 2. The method of claim 1, wherein an analog-to-digital conversion is effected before or after a conversion of the complex modulated signal into an amplitude signal and a phase signal. 3. The method of claim 1 or 2, wherein the amplitude signal (104, 250A) along the amplitude signal path (120, 230) of a digital-to-analog conversion with subsequent first filtering is subjected, the first filtering causes the first group delay distortion, further the phase signal (106, 250P) undergoes digital-to-analog conversion along the phase signal path (130, 230). A method according to any one of claims 1 to 3, wherein the second group delay distortion is introduced into the phase signal path (130, 230) and substantially corresponds to the first group delay distortion. 5. The method of claim 1, wherein the amplitude signal path causes a first overall group delay distortion comprising the first group delay distortion and wherein the second group delay distortion is introduced into the phase signal path and substantially matches the first overall group delay distortion. 6. The method of claim 1, wherein the amplitude signal path causes a first total group delay distortion comprising the first group delay distortion and the second group delay distortion is introduced into the phase signal path and is selected such that a second total group delay of the phase signal path (130, 230), which comprises the second group delay distortion, substantially corresponds to the first total group delay. 7. Method according to one of claims 1 to 6, in which the group delay distortion in the phase signal path (130, 230) is effected by means of all-pass filtering or low-pass filtering. 8. The method according to any one of claims 1 to 7, wherein the second group delay distortion in the phase signal path (130, 230) is effected by a digital filtering. 9. The method according to any one of claims 1 to 7, wherein the group delay distortion in the phase signal path (130, 230) is effected by an analog filtering ■. 10. The method of claim 1, wherein the group delay distortion is introduced into the amplitude signal path. 11. The method according to any one of claims 1 to 10, further comprising the following step: Converting a complex modulated signal in Cartesian representation into a corresponding complex modulated signal in polar representation and breaking it into an amplitude signal (104, 250A) and a phase signal (106, 250P). 12. A method according to any of claims 1 to 12, wherein the amplitude signal path (120, 220) effects amplitude signal processing and the phase signal path (130, 230) effects phase signal processing contributing to first and second total group delay, respectively , A computer program according to a program code for carrying out a method according to claims 1 to 12, when the computer program runs on a computer. 14. Apparatus for generating an output signal (160, 260) based on a complex modulated input signal (102) decomposed into an amplitude signal (104, 250A) and a phase signal (106, 250P), comprising: an amplitude signal path (120, 220) having an input for receiving the amplitude signal (104, 250A); a phase signal path (130, 230) having an input for receiving the phase signal (106, 250P); wherein one of the signal paths causes a first group delay distortion; a combiner (140, 240) based on a signal at an output of the amplitude signal path (120, 120); 220) and a signal at an output of the phase signal path (130, 230) generates the output signal (160, 260); and means (121, 131, 231a, 231b) adapted to effect a second group delay distortion counteracting the first group delay distortion. 15. The apparatus of claim 14, wherein the means for effecting a second group delay distortion (131, 231a, 231b) is arranged in the phase signal path (130, 230) and the second group delay distortion substantially corresponds to the first group delay distortion. 16. The apparatus of claim 14, wherein the amplitude signal path (120, 220) effects a first total group delay distortion comprising the first group delay distortion, and wherein the means for effecting a second group delay distortion (131, 231a, 231b) in the phase signal path (130, 230) and the second group delay distortion substantially corresponds to the first total group delay. 17. The apparatus of claim 14, wherein the amplitude signal path (120, 220) effects a first total group delay distortion comprising the first group delay distortion, and the means for effecting a second group delay distortion (131, 231a, 231b) in the phase signal path (130 , 230), and the second group delay distortion is selected such that a second total group delay distortion of the phase signal path (130, 230) comprising the second group delay distortion corresponds to the first total group delay distortion. An apparatus according to any one of claims 14 to 17, wherein said means for effecting a second group delay distortion (131, 231a, 231b) is a lowpass or allpass. 19. An apparatus according to any one of claims 14 to 18, wherein the means for effecting a second group delay distortion (131, 231a) is a digital filter connected between the input and a second digital-to-analog converter (232) of the phase signal path (130 , 230) is arranged. 20. Device according to one of claims 14 to 18, wherein the means for effecting a second group delay distortion (131, 231b) is an analog filter which is between a second digital-to-analog converter (232) and the output of the phase signal path (130 , 230) is arranged. The apparatus of claim 14, wherein the means for effecting a second group delay distortion (121) is located in the amplitude signal path (120, 220). Apparatus according to any one of claims 14 to 21, comprising conversion means (210) adapted to convert a complex modulated signal (2501, 250Q) in Cartesian representation into a corresponding complex modulated signal in polar representation and into an amplitude signal (104, 250A) and a phase signal (106, 250P). 23. Device according to one of claims 14 to 22, wherein the amplitude signal path (120, 220) an amplitude signal processing unit (224) and the phase signal path (130, 230) a phase signal processing unit (233), which to a first or second total group delay distortion. 24. Device according to one of claims 14 to 23, wherein the amplitude signal path (120, 220) has a first Digital-to-analog converter (222) and a first filter (223), wherein the first filter (223), the first Group delay distortion causes the phase signal path (130, 230) comprises a second digital-to-analog converter (232) and the means for effecting the second Group delay distortion is a second filter (231a, 231b).