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WO2003067707A1 - Antenne cadre a large bande - Google Patents

Antenne cadre a large bande Download PDF

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Publication number
WO2003067707A1
WO2003067707A1 PCT/US2002/000154 US0200154W WO03067707A1 WO 2003067707 A1 WO2003067707 A1 WO 2003067707A1 US 0200154 W US0200154 W US 0200154W WO 03067707 A1 WO03067707 A1 WO 03067707A1
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WIPO (PCT)
Prior art keywords
edge
antenna
transceiver
ground
host device
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PCT/US2002/000154
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English (en)
Inventor
Hans G. Schantz
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Time Domain Corp
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Time Domain Corp
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Priority to AU2002255474A priority Critical patent/AU2002255474A1/en
Priority to PCT/US2002/000154 priority patent/WO2003067707A1/fr
Publication of WO2003067707A1 publication Critical patent/WO2003067707A1/fr
Anticipated expiration legal-status Critical
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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/30—Resonant antennas with feed to end of elongated active element, e.g. unipole
    • H01Q9/42—Resonant antennas with feed to end of elongated active element, e.g. unipole with folded element, the folded parts being spaced apart a small fraction of the operating wavelength
    • H01Q9/43—Scimitar antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/30—Resonant antennas with feed to end of elongated active element, e.g. unipole
    • H01Q9/40—Element having extended radiating surface
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B2001/6908—Spread spectrum techniques using time hopping

Definitions

  • the present invention relates generally to electromagnetic energy radiation and reception using a single element antenna, and especially relates to electromagnetic energy radiation and reception with a single element antenna effected using impulse radio energy. Still more particularly the present invention provides a single element antenna suited for broadband energy radiation and reception, and particularly well suited for broadband energy radiation and reception employing impulse radio energy.
  • impulse radio impulse radio communications systems
  • Impulse radio was first fully described in a series of patents, including U.S. Patent Nos. 4,641,317 (issued February 3, 1987), 4,813,057 (issued March 14, 1989), 4,979,186 (issued December 18, 1990) and 5,363,108 (issued November 8, 1994) to Larry W. Fullerton.
  • a second generation of impulse radio patents include U.S. Patent Nos. 5,677,927 (issued October 14, 1997) to Fullerton etal; and 5,687,169 (issued November 11, 1997) and 5,832,035 (issued November 3, 1998) to Fullerton. These patent documents are incorporated herein by reference.
  • the pulse-to-pulse interval is varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component.
  • the pseudo-random code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code of an impulse radio system is used for channelization, energy smoothing in the frequency domain and for interference suppression.
  • an impulse radio receiver is a direct conversion receiver with a cross correlator front end.
  • the front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage.
  • the data rate of the impulse radio transmission is typically a fraction of the periodic timing signal used as a time base. Because each data bit modulates the time position of many pulses of the periodic timing signal, this yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit.
  • the impulse radio receiver integrates multiple pulses to recover the transmitted information.
  • impulse radio depends, in part, on processing gain to achieve rejection of unwanted signals. Because of the extremely high processing gain achievable with impulse radio, much higher dynamic ranges are possible than are commonly achieved with other spread spectrum methods, some of which must use power control in order to have a viable system Further, if power is kept to a minimum in an impulse radio system, this will allow closer operation in co-site or nearly co-site situations where two impulse radios must operate concurrently, or where an impulse radio and a narrow band radio must operate close by one another and share the same band.
  • impulse radio technology Many applications for impulse radio technology, including communication applications, position determination applications, locating (e.g., radar) applications and other applications require lightweight, compact broadband antennas with broad beam transmit/receive characteristics. As with any antenna, impedance matching to feed elements is necessary for efficient operation. Moreover, in the case of impulse radio technology applications, the antenna must not be subject to ringing in response to application of pulses - either in a transmit mode or in a receive mode.
  • UWB ultra wideband
  • ultra wideband (UWB) antennas teaches using element antennas such as monopoles, dipoles, conical antennas and bow-tie antennas for ultra wideband systems. However, they are generally characterized by low directivity and relatively limited bandwidth unless either end loading or distributed loading techniques are employed, in which case bandwidth is increased at the expense of radiation efficiency.
  • Conventional antennas are designed to radiate only over the relatively narrow range of frequencies used in conventional narrow band systems. Such narrow band systems may, for example, employ fractional bandwidths no more than about 25%. If an impulse signal, such as a signal of the sort employed for impulse radio purposes, is fed to such a narrow band antenna, the antenna tends to ring. Ringing severely distorts signal pulses and spreads them out in time. Impulse radio signals are preferably modulated by pulse timing, so such distortion of pulses is not desirable.
  • Broadband antennas are advantageous for many purposes, including then- use with impulse radio systems.
  • Conventional design in broadband antennas follows a commonly accepted principle that the impedance and pattern properties of an antenna will be frequency independent if the antenna shape is specified only in terms of angles. That is to say, a self-similar or self-complimentary antenna will be a broadband antenna.
  • This principle explains known broadband antennas like biconical and bow tie antennas, but also applies to other broadband antennas like log periodic, log spiral, and conical spiral antennas.
  • a biconical antenna is a classic example of a prior art broadband antenna with an omni-directional pattern.
  • a typical biconical antenna with a 60° half angle will have a 100 ⁇ input with a voltage standing wave ratio (VSWR) of ⁇ 2: 1 over a 6: 1 bandwidth.
  • VSWR voltage standing wave ratio
  • a significant drawback with such a biconical antenna is that such an antenna is typically implemented with a diameter equal to the wavelength at the lower frequency limit ( ⁇ ; ), thus requiring that the antenna be
  • atypical monocone antenna will not provide a good match if it is much less than 0.2 ⁇ in diameter.
  • a monocone antenna does not have very stable performance over a broad band.
  • large antennas are difficult to fit into a small portable or hand held devices.
  • TEM horn antennas often suffer from frequency dispersion as well.
  • a horn antenna is inherently a large structure, often several wavelengths in dimension.
  • a hom antenna may be made smaller by dielectric loading, but such loading adds weight which is often undesirable.
  • a horn antenna is a highly directive antenna and cannot provide the less directive coverage required for many portable or mobile applications.
  • a TEM feed may be combined with a parabolic dish to create a ribbed horn "impulse radiating antenna" (IRA).
  • IRA impulse radiating antenna
  • Such antennas can have bandwidths on the order of a couple of decades, and very high gain, but their large size and high directivity make them inappropriate for portable or mobile use. Because spherical antennas must be fed by a radial waveguide, they often exhibit poor matching characteristics unless an elaborate and difficult-to- manufacture impedance matching structure is used.
  • a typical impedance matching structure also tends to further impair antenna performance by making the antenna more likely to ring. It is very difficult to construct a feed that maintains a constant matched impedance over a broad bandwidth, something essential to an ultra wideband (UWB) antenna. It is a commonly accepted design criteria in electromagnetic applications, and especially in radio communication applications, that an antenna should match a 50 ⁇ impedance feed providing signals to (or receiving signals from) the antenna. Some video applications require matching a 75 ⁇ impedance feed.
  • Resistive loading is an alternate technique commonly employed to achieve impedance matching in broadband antennas. Resistive loading succeeds in reducing reflection, but at the cost of throwing away typically around half the power that may be transmitted by an antenna Such a design trade-off has become accepted in design approaches in prior art antennas. It has been generally believed that resistive loading must be employed for a small broadband antenna in order to achieve good impedance matching. Non-resistively loaded small ultra wideband antennas are known, but they tend to have poor impedance matching and high voltage standing wave ratios (VSWR's). A lower value for VSWR is a better value; the optimum value of VSWR is 1:1. The prior art teaches that resistive loading must be used in an element antenna in order to achieve wide bandwidth. It is commonly believed that high radiation efficiency and high bandwidth are mutually exclusive.
  • Henning Harmuth Henning F. Harmuth, "Nonsinusoidal Waves for Radar and Radio Communication," New York: Academic Press, 1981, pp. 108-110.
  • the underlying goal of Harmuth's antenna was to isolate the radiating currents on the loop from currents on the return behind the ground plane.
  • Pochanin suggested employing a ferrite plate as a ground plane and providing a current carrying radiating element with a radius of curvature normal to the ground plane.
  • Farr designed a structure incorporating a radius of curvature normal to the backplane, a broad planar aspect perpendicular with the radius of curvature and a load impedance intermediate the radiating element and the backplane.
  • Both the Pochanin and Farr designs use lossy materials - Pochanin' s ferrite backplane and Farr's load impedance - to achieve their design goals.
  • Neither of the Pochanin or Farr antennas uses a simple conducting loop terminated against a ground plane as contemplated by the present invention.
  • a loop structure has been employed in exciting modes in a waveguide
  • UWB ultra wide-band
  • UWB radar device may be more efficiently, more quickly and more conveniently located for field operations if it involves a compact antenna that exhibits efficient matching and operating characteristics.
  • An array of single element antennas for use in radar imaging or motion detection should employ single element antennas with patterns that are substantially similar to the desired field of view of the complete radar system
  • single element antennas have a wide field of view in the horizontal direction and a more narrow field of view in the vertical direction. It is also preferable if such radar- employed antenna elements have a horizontal polarity.
  • An antenna for transferring electromagnetic energy intermediate a host device and a medium substantially adjacent to the antenna includes (a) a ground element preferably generally coplanar with a ground plane; and (b) a substantially planar transceiver element generally coplanar with a transceiver plane.
  • the transceiver element intersects the ground element at a first end in a joint that has a first terminus and a second terminus.
  • a first edge of the transceiver departs from the first terminus in a first arcuate path in a first direction from the ground plane.
  • a second edge of the transceiver departs from the second terminus in a second arcuate path generally in the first direction.
  • the first edge and the second edge each include at least one arc-set.
  • Each respective arc-set includes a first arc having a first radius describing a respective first edge sector of the first edge and a second arc having a second radius describing a respective second edge sector of the second edge.
  • the first radius and the second radius define a transverse separation between the first edge sector and the second edge sector.
  • the first edge and the second edge terminate in a terminal structure at a second end distal from the first end.
  • the terminal structure is in spaced relation with respect to the ground element to establish a gap intermediate the transceiver and the ground element.
  • the antenna also includes (c) a feed structure.
  • the feed structure conveys the electromagnetic energy intermediate the transceiver and the host device.
  • the antenna of the present invention is configured as a layer of copper arranged upon a dielectric substrate to form a generally planar transceiving element that is affixed to a generally planar ground plane.
  • the plane of the transceiver element is preferably substantially perpendicular with the plane of the ground element.
  • the thickness of the dielectric substrate may be advantageously altered to adjust the speed of signal propagation in elements supported by the dielectric material.
  • An energy guiding means is preferably embodied in a structure that conveys electromagnetic energy. Examples of an energy guiding means include, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or a connector or coupler that enables connection to a transmission line.
  • An energy channeling structure is preferably embodied in a structure that couples electromagnetic energy between an apparatus and an adjacent free space or medium
  • Examples of a channeling structure include, by way of illustration and not by way of limitation, radiating elements, receiving elements, reflectors, directors and horns.
  • a transition means is preferably embodied in a structure that receives radio frequency (RF) energy, transmits RF energy or receives and transmits RF energy.
  • RF radio frequency
  • a host radio is a RF device that receives RF energy, transmits RF energy or receives and transmits RF energy.
  • An antenna may be integrally included with or within a host radio or that antenna may be situated remotely from the host radio at an arbitrary distance yet coupled with the host radio, such as by using an energy guiding means.
  • the term "host radio” does not per se indicate any particular relation between a radio and an associated antenna. In particular, the term “host radio” does not preclude an antenna remotely located from a radio or standing alone with respect to a radio.
  • the term "host device" intentionally indicates an element that may be embodied in other than a radio.
  • Examples of host devices other than radios include, for example, radar devices, location transducer devices, and other devices employing electromagnetic energy transmitted, received or transmitted and received using an antenna
  • the present invention is embodied in antennas having a structure characterized by the inventor as "monoloop" antennas.
  • Monoloop antennas are planar single element antennas that are preferably well matched to the standard 50
  • Monoloop antennas are efficient, physically small and radiate in a broad beam pattern. Such antennas exhibit some spatial dispersion, but they emit a waveform that is relatively short and non-temporally dispersive.
  • Monoloop antennas generally include a planar radiating loop, a ground plane reflector and a feed structure for providing signals between the antenna and a host device.
  • a planar radiating loop is preferably a generally planar, approximately semi-circular arc of a suitable conducting material.
  • the plane in which the planar radiating loop is oriented is preferably normal to the plane of the ground plane.
  • the preferred typical shape of the radiating loop is close to circular, but various elliptical, ovoidaL Archimedian and log spiral shapes may also be employed to advantage. It is important to note that the present invention is configured in contrast to teaching of the prior art relating to antenna construction. Rather than being configured to block or minimize reflection from the ground plane, the present invention is oriented to take advantage of the reflections from the ground plane.
  • the ground plane of the present invention is preferably a suitably conductive sheet that reflects energy from the planar radiating loop.
  • the ground plane is a flat conducting plane.
  • a variety of alternate configurations are also useful including a cylindrical reflector, a parabolic reflector, a hyperbolic reflector or a corner reflector.
  • the feed structure of the present invention includes a transmission line or other energy guiding means, a gap of an appropriate size and a preferably blunt gap intersection or interface intermediate the planar radiating loop and the underlying ground plane. Gap interfaces having smaller diametral dimensions (i.e., less blunt, more pointed configurations) may be employed, but such less blunt gap interface structures present higher input impedance that can be on the order of 100 ⁇ - 150 ⁇ .
  • FIG. 1 A illustrates a representative Gaussian Monocycle waveform in the time domain.
  • FIG. IB illustrates the frequency domain amplitude of the Gaussian Monocycle of FIG. 1A
  • FIG. 2A illustrates a pulse train comprising pulses as in FIG. 1A.
  • FIG. 2B illustrates the frequency domain amplitude of the waveform of FIG. 2A.
  • FIG. 3 illustrates the frequency domain amplitude of a sequence of time coded pulses.
  • FIG. 4 illustrates a typical received signal and interference signal.
  • FIG. 5A illustrates a typical geometrical configuration giving rise to multipath received signals.
  • FIG. 5B illustrates exemplary multipath signals in the time domain.
  • FIGS 5C - 5E illustrate a signal plot of various multipath environments.
  • FIGS. 5F illustrates the Rayleigh fading curve associated with non-impulse radio transmissions in a multipath environment.
  • FIG. 5G illustrates a plurality of multipaths with a plurality of reflectors from a transmitter to a receiver.
  • FIG. 5H graphically represents signal strength as volts vs. time in a direct path and multipath environment.
  • FIG. 6 illustrates a representative impulse radio transmitter functional diagram
  • FIG. 7 illustrates a representative impulse radio receiver functional diagram.
  • FIG. 8A illustrates a representative received pulse signal at the input to the correlator.
  • FIG. 8B illustrates a sequence of representative impulse signals in the correlation process.
  • FIG. 8C illustrates the output of the correlator for each of the time offsets of FIG. 8B.
  • FIG. 9 illustrates a vertically oriented current-carrying conductor with its associated electric radiation field and magnetic radiation field.
  • FIG. 10 illustrates a horizontally oriented current-carrying conductor with its associated electric radiation field and magnetic radiation field.
  • FIG. 11 illustrates a representative prior art single element antenna structure with its associated electric radiation field.
  • FIG. 12 is a side elevation view of a single element antenna according to the preferred embodiment of the present invention.
  • FIG. 13 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a first representative set of signal transmission paths.
  • FIG. 14 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a second representative set of signal transmission paths.
  • FIG. 15 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a third representative set of signal transmission paths.
  • FIG. 16 is a perspective view of the single element antenna of the present invention with its associated electric radiation field.
  • FIG. 17 is a perspective view of a first alternate embodiment of a single element antenna according to the present invention.
  • FIG. 18 is a side view of a second alternate embodiment of a single element antenna according to the present invention.
  • FIG. 19 is a side view of a third alternate embodiment of a single element antenna according to the present invention.
  • FIG. 20 is a side view of a fourth alternate embodiment of a single element antenna according to the present invention.
  • FIG. 21 is a side view of a fifth alternate embodiment of a single element antenna according to the present invention.
  • Impulse Radio Basics This section is directed to technology basics and provides the reader with an introduction to impulse radio concepts, as well as other relevant aspects of communications theory. This section includes subsections relating to waveforms, pulse trains, coding for energy smoothing and channelization, modulation, reception and demodulation, interference resistance, processing gain, capacity, multipath and propagation, distance measurement, and qualitative and quantitative characteristics of these concepts. It should be understood that this section is provided to assist the reader with understanding the present invention, and should not be used to limit the scope of the present invention.
  • Impulse radio refers to a radio system based on short, low duty cycle pulses.
  • An ideal impulse radio waveform is a short Gaussian monocycle. As the name suggests, this waveform attempts to approach one cycle of radio frequency (RF) energy at a desired center frequency. Due to implementation and other spectral limitations, this waveform may be altered significantly in practice for a given application. Most waveforms with enough bandwidth approximate a Gaussian shape to a useful degree.
  • RF radio frequency
  • Impulse radio can use many types of modulation, including AM, time shift (also referred to as pulse position) and M-ary versions.
  • the time shift method has simplicity and power output advantages that make it desirable.
  • the time shift method is used as an illustrative example.
  • the pulse-to-pulse interval can be varied on a pulse-by-pulse basis by two components: an information component and a pseudo-random code component.
  • an information component and a pseudo-random code component.
  • conventional spread spectrum systems make use of pseudo-random codes to spread the normally narrow band control the bandwidth to meet desired properties such as out of band emissions or in- band spectral flatness, or time domain peak power or burst off time attenuation.
  • FIG. 1 A This waveform is representative of the transmitted pulse produced by a step function into an ultra wideband antenna.
  • the basic equation normalized to a peak value of 1 is as follows:
  • ⁇ is a time scaling parameter
  • t is time
  • f m o n o(t) is the waveform voltage
  • e is the natural logarithm base.
  • FIG. IB The corresponding equation is:
  • the center frequency (f c ), or frequency of peak spectral density is:
  • pulses may be produced by methods described in the patents referenced above or by other methods that are known to one of ordinary skill in the art. Any practical implementation will deviate from the ideal mathematical model by some amount. In fact, this deviation from ideal may be substantial and yet yield a system with acceptable performance. This is especially true for microwave implementations, where precise waveform shaping is difficult to achieve.
  • These mathematical models are provided as an aid to describing ideal
  • the pseudorandom code for impulse radio communications is not necessary for energy spreading because the monocycle pulses themselves have an inherently wide bandwidth. Instead, the pseudo-random code is used for channelization, energy smoothing in the frequency domain, resistance to interference, and reducing the interference potential to nearby receivers.
  • the impulse radio receiver is typically a direct conversion receiver with a cross correlator front end in which the front end coherently converts an electromagnetic pulse train of monocycle pulses to a baseband signal in a single stage.
  • the baseband signal is the basic information signal for the impulse radio communications system It is often found desirable to include a subcarrier with the baseband signal to help reduce the effects of amplifier drift and low frequency noise.
  • the subcarrier that is typically implemented alternately reverses modulation according to a known pattern at a rate faster than the data rate. This same pattern is used to reverse the process and restore the original data pattern just before detection.
  • This method permits alternating current (AC) coupling of stages, or equivalent signal processing to eliminate direct current (DC) drift and errors from the detection process. This method is described in detail in U.S. Patent No. 5,677,927 to Fullerton et al.
  • each data bit typically time position modulates many pulses of the periodic timing signal. This yields a modulated, coded timing signal that comprises a train of identically shaped pulses for each single data bit.
  • the impulse radio receiver integrates multiple pulses to recover the transmitted information.
  • Impulse radio refers to a radio system based on short, low duty cycle pulses.
  • the resulting waveform approaches one cycle per pulse at the center frequency.
  • each pulse consists of a burst of cycles usually with some spectral shaping to
  • Impulse radio systems can deliver one or more data bits per pulse; however, impulse radio systems more typically use pulse trains, not single pulses, for each data bit. As described in detail in the following example system, the impulse radio transmitter produces and outputs a train of pulses for each bit of information.
  • FIG. 2A shows a time domain representation of this sequence of pulses 102.
  • Fig 2B which shows 60 MHz at the center of the spectrum for the waveform of FIG. 2 A, illustrates that the result of the pulse train in the frequency domain is to produce a spectrum comprising a set of lines 204 spaced at the frequency of the 1 Mpps pulse repetition rate.
  • the envelope of the line spectrum follows the curve of the single pulse spectrum 104 of FIG. IB.
  • the power of the pulse train is spread among roughly two hundred comb lines. Each comb line thus has a small fraction of the total power and presents much less of an interference problem to receiver sharing the band.
  • FIG. 2A impulse radio systems typically have very low average duty cycles resulting in average power significantly lower than peak power.
  • the duty cycle of the signal in the present example is 0.5%, based on a 0.5 ns pulse in a 100 ns interval.
  • FIG. 3 is a plot illustrating the impact of a pseudo-noise (PN) code dither on energy distribution in the frequency domain (A pseudo-noise, or PN code is a set of time positions defining the pseudo-random positioning for each pulse in a sequence of pulses).
  • PN code when compared to FIG 2B, shows that the impact of using a PN code is to destroy the comb line structure and spread the energy more uniformly. This structure typically has slight variations which are characteristic of the specific code used.
  • the PN code also provides a method of establishing independent communication channels using impulse radio.
  • PN codes can be designed to have low cross correlation such that a pulse train using one code will seldom collide on more than one or two pulse positions with a pulses train using another code during any one data bit time. Since a data bit may comprise hundreds of pulses, this represents a substantial attenuation of the unwanted channel.
  • any aspect of the waveform can be modulated to convey information.
  • Amplitude modulation, phase modulation, frequency modulation, time shift modulation and M-ary versions of these have been proposed. Both analog and digital forms have been implemented. Of these, digital time shift modulation has been demonstrated to have various advantages and can be easily implemented using a correlation receiver architecture.
  • Digital time shift modulation can be implemented by shifting the coded time position by an additional amount (that is, in addition to PN code dither) in response to the information signal. This amount is typically very small relative to the PN code shift. In a 10 Mpps system with a center frequency of 2 GHz., for example, the PN code may command pulse position variations over a range of 100 ns; whereas, the information modulation may only deviate the pulse position by 150 ps.
  • each pulse is delayed a different amount from its respective time base clock position by an individual code delay amount plus a modulation amount, where n is the number of pulses associated with a given data symbol digital bit. Modulation further smooths the spectrum, minimizing structure in the resulting spectrum.
  • impulse radios are able to perform in these environments, in part, because they do not depend on receiving every pulse.
  • the impulse radio receiver performs a correlating, synchronous receiving function (at the RF level) that uses a statistical sampling and combining of many pulses to recover the transmitted information.
  • Impulse radio receivers typically integrate from 1 to 1000 or more pulses to yield the demodulated output.
  • the optimal number of pulses over which the receiver integrates is dependent on a number of variables, including pulse rate, bit rate, interference levels, and range.
  • the PN coding also makes impulse radios highly resistant to interference from all radio communications systems, including other impulse radio transmitters. This is critical as any other signals within the band occupied by an impulse signal potentially interfere with the impulse radio. Since there are currently no unallocated bands available for impulse systems, they must share spectrum with other conventional radio systems without being adversely affected.
  • the PN code helps impulse systems discriminate between the intended impulse transmission and interfering transmissions from others.
  • FIG. 4 illustrates the result of a narrow band sinusoidal interference signal 402 overlaying an impulse radio signal 404.
  • the input to the cross correlation would include the narrow band signal 402, as well as the received ultra wideband impulse radio signal 404.
  • the input is sampled by the cross correlator with a PN dithered template signal 406. Without PN coding, the cross correlation would sample the interfering signal 402 with such regularity that the interfering signals could cause significant interference to the impulse radio receiver.
  • the transmitted impulse signal is encoded with the PN code dither (and the impulse radio receiver template signal 406 is synchronized with that identical PN code dither)
  • the correlation samples the interfering signals pseudo-rando ly.
  • the samples from the interfering signal add incoherently, increasing roughly according to square root of the number of samples integrated; whereas, the impulse radio samples add coherently, increasing directly according to the number of samples integrated.
  • integrating over many pulses overcomes the impact of interference.
  • Impulse radio is resistant to interference because of its large processing gain.
  • processing gain which quantifies the decrease in channel interference when wide-band communications are used, is the ratio of the bandwidth of the channel to the bit rate of the information signal.
  • a direct sequence spread spectrum system with a 10 kHz information bandwidth and a 10 MHz channel bandwidth yields a processing gain of 1000 or 30 dB.
  • far greater processing gains are achieved with impulse radio systems, where for the same 10 kHz information bandwidth is spread across a much greater 2 GHz. channel bandwidth, the theoretical processing gain is 200,000 or 53 dB.
  • V 2 tot is the total interference signal to noise ratio variance, at the receiver
  • N is the number of interfering users
  • cr 2 is the signal to noise ratio variance resulting from one of the interfering signals with a single pulse cross correlation
  • Z is the number of pulses over which the receiver integrates to recover the modulation.
  • impulse radio is its resistance to multipath fading effects.
  • Conventional narrow band systems are subject to multipath through the Rayleigh fading process, where the signals from many delayed reflections combine at the receiver antenna according to their seemingly random relative phases. This results in possible summation or possible cancellation, depending on the specific propagation to a given location.
  • This situation occurs where the direct path signal is weak relative to the multipath signals, which represents a major portion of the potential coverage of a radio system In mobile systems, this results in wild signal strength fluctuations as a function of distance traveled, where the changing mix of multipath signals results in signal strength fluctuations for every few feet of travel.
  • Impulse radios can be substantially resistant to these effects. Impulses arriving from delayed multipath reflections typically arrive outside of the correlation time and thus can be ignored. This process is described in detail with reference to Figs. 5A and 5B.
  • the direct path representing the straight line distance between the transmitter and receiver is the shortest.
  • Path 1 represents a grazing multipath reflection, which is very close to the direct path.
  • Path 2 represents a distant multipath reflection.
  • elliptical (or, in space, ellipsoidal) traces that represent other possible locations for reflections with the same time delay.
  • FIG. 5B represents a time domain plot of the received waveform from this multipath propagation configuration.
  • This figure comprises three doublet pulses as shown in FIG. 1
  • the direct path signal is the reference signal and represents the shortest propagation time.
  • the path 1 signal is delayed slightly and actually overlaps and enhances the signal strength at this delay value. Note that the reflected waves are reversed in polarity.
  • the path 2 signal is delayed sufficiently that the waveform is completely separated from the direct path signal. If the correlator template signal is positioned at the direct path signal, the path 2 signal will produce no response. It can be seen that only the multipath signals resulting from very close reflectors have any effect on the reception of the direct path signal.
  • the multipath signals delayed less than one quarter wave are the only multipath signals that can attenuate the direct path signal. This region is equivalent to the first Fresnel zone familiar to narrow band systems designers.
  • Impulse radio however, has no further nulls in the higher Fresnel zones. The ability to avoid the highly variable attenuation from multipath gives impulse radio significant performance advantages.
  • Fig 5 A illustrates a typical multipath situation, such as in a building, where there are many reflectors 5A04, 5A05 and multiple propagation paths 5A02, 5 A01.
  • a transmitter TX 5A06 transmits a signal which propagates along the multiple propagation paths 5A02, 5A04 to receiver RX 5A08, where the multiple reflected signals are combined at the antenna.
  • FIG. 5B illustrates a resulting typical received composite pulse waveform resulting from the multiple reflections and multiple propagation paths 5A01, 5A02.
  • the direct path signal 5A01 is shown as the first pulse signal received.
  • the multiple reflected signals (“multipath signals”, or “multipath") comprise the remaining response as illustrated.
  • FIG. 5C illustrates the received signal in a very low multipath environment. This may occur in a building where the receiver antenna is in the middle of a room and is one meter from the transmitter. This may also represent signals received from some distance, such as 100 meters, in an open field where there are no objects to produce reflections. In this situation, the predominant pulse is the first received pulse and the multipath reflections are too weak to be significant.
  • FIG. 5D illustrates an intermediate multipath environment. This approximates the response from one room to the next in a building.
  • FIG. 5E approximates the response in a severe multipath environment such as: propagation through many rooms; from comer to corner in a building; within a metal cargo hold of a ship; within a metal truck trailer; or within an intermodal shipping container.
  • the main path signal is weaker than in FIG. 5D.
  • the direct path signal power is small relative to the total signal power from the reflections.
  • An impulse radio receiver in accordance with the present invention can receive the signal and demodulate the information using either the direct path signal or any multipath signal peak having sufficient signal to noise ratio.
  • the impulse radio receiver can select the strongest response from among the many arriving signals.
  • dozens of reflections would have to be canceled simultaneously and precisely while blocking the direct path - a highly unlikely scenario.
  • This time separation of multipath signals together with time resolution and selection by the receiver permit a type of time diversity that virtually eliminates cancellation of the signal.
  • performance is further improved by collecting the signal power from multiple signal peaks for additional signal to noise performance.
  • the received signal is a sum of a large number of sine waves of random amplitude and phase.
  • the resulting envelope amplitude has been shown to follow a Rayleigh probability distribution as follows.”
  • r is the envelope amplitude of the combined multipath signals
  • p ⁇ ⁇ is the RMS amplitude of the combined multipath signals
  • FIG. 5G and 5H This is illustrated in FIG. 5G and 5H in a transmit and receive system in a high multipath environment 5G00, wherein the transmitter 5G06 transmits to receiver 5G08 with the signals reflecting off reflectors 5G03 which form multipaths 5G02.
  • the direct path is illustrated as 5G01 with the signal graphically illustrated at 5H02 with the vertical axis being the signal strength in volts and horizontal axis representing time in nanoseconds.
  • Multipath signals are graphically illustrated at 5H04.
  • the transmitter 602 comprises a time base 604 that generates a periodic timing signal 606.
  • the time base 604 typically comprises a voltage controlled oscillator (VCO), or the like, having a high timing accuracy and low jitter, on the order of picoseconds (ps).
  • VCO voltage controlled oscillator
  • the voltage control to adjust the VCO center frequency is set at calibration to the desired center frequency used to define the transmitter's nominal pulse repetition rate.
  • the periodic timing signal 606 is supplied to a precision timing generator 608.
  • the precision timing generator 608 supplies synchronizing signals 610 to the code source 612 and utilizes the code source output 614 together with an internally generated subcarrier signal (which is optional) and an information signal 616 to generate a modulated, coded timing signal 618.
  • the code source 612 comprises a storage device such as a random access memory (RAM), read only memory (ROM), or the like, for storing suitable PN codes and for outputting the PN codes as a code signal 614.
  • RAM random access memory
  • ROM read only memory
  • maximum length shift registers or other computational means can be used to generate the PN codes.
  • An information source 620 supplies the information signal 616 to the precision timing generator 608.
  • the information signal 616 can be any type of intelligence, including digital bits representing voice, data, imagery, or the like, analog signals, or complex signals.
  • a pulse generator 622 uses the modulated, coded timing signal 618 as a trigger to generate output pulses.
  • the output pulses are sent to a transmit antenna 624 via a transmission line 626 coupled thereto.
  • the output pulses are converted into propagating electromagnetic pulses by the transmit antenna 624.
  • the electromagnetic pulses are called the emitted signal, and propagate to an impulse radio receiver 702, such as shown in FIG. 7, through a propagation medium, such as air, in a radio frequency embodiment.
  • the emitted signal is wide-band or ultra wideband, approaching a monocycle pulse as in FIG. 1A.
  • the emitted signal can be spectrally modified by filtering of the pulses. This filtering will usually cause each monocycle pulse to have more zero crossings (more cycles) in the time domain.
  • the impulse radio receiver can use a similar waveform as the template signal in the cross correlator for efficient conversion.
  • FIG. 7 Receiver An exemplary embodiment of an impulse radio receiver 702 (hereinafter called the receiver) for the impulse radio communication system is now described with reference to FIG. 7. More specifically, the system illustrated in FIG. 7 is for reception of digital data wherein one or more pulses are transmitted for each data bit.
  • the receiver for the impulse radio communication system is now described with reference to FIG. 7. More specifically, the system illustrated in FIG. 7 is for reception of digital data wherein one or more pulses are transmitted for each data bit.
  • the receiver 702 comprises a receive antenna 704 for receiving a propagated impulse radio signal 706.
  • a received signal 708 from the receive antenna 704 is coupled to a cross correlator or sampler 710 to produce a baseband output 712.
  • the cross correlator or sampler 710 includes multiply and integrate functions together with any necessary filters to optimize signal to noise ratio.
  • the receiver 702 also includes a precision timing generator 714, which receives a periodic timing signal 716 from a receiver time base 718. This time base 718 is adjustable and controllable in time, frequency, or phase, as required by the lock loop in order to lock on the received signal 708.
  • the precision timing generator 714 provides synchronizing signals 720 to the code source 722 and receives a code control signal 724 from the code source 722.
  • the precision timing generator 714 utilizes the periodic timing signal 716 and code control signal 724 to produce a coded timing signal 726.
  • the template generator 728 is triggered by this coded timing signal 726 and produces a train of template signal pulses 730 ideally having waveforms substantially equivalent to each pulse of the received signal 708.
  • the code for receiving a given signal is the same code utilized by the originating transmitter 602 to generate the propagated signal 706.
  • the timing of the template pulse train 730 matches the timing of the received signal pulse train 708, allowing the received signal 708 to be synchronously sampled in the correlator 710.
  • the correlator 710 ideally comprises a multiplier followed by a short term integrator to sum the multiplier product over the pulse interval.
  • the output of the subcarrier demodulator 732 is then filtered or integrated in a pulse summation stage 734.
  • the pulse summation stage produces an output representative of the sum of a number of pulse signals comprising a single data bit.
  • the output of the pulse summation stage 734 is then compared with a nominal zero (or reference) signal output in a detector stage 738 to determine an output signal 739 representing an estimate of the original information signal 616.
  • the baseband signal 712 is also input to a lowpass filter 742 (also referred to as lock loop filter 742).
  • a control loop comprising the lowpass filter 742, time base 718, precision timing generator 714, template generator 728, and correlator 710 is used to generate a filtered error signal 744.
  • the filtered error signal 744 provides adjustments to the adjustable time base 718 to time position the periodic timing signal 726 in relation to the position of the received signal 708.
  • transceiver embodiment substantial economy can be achieved by sharing part or all of several of the functions of the transmitter 602 and receiver 702. Some of these include the time base 718, precision timing generator 714, code source 722, antenna 704, and the like.
  • FIGS. 8A-8C illustrate the cross correlation process and the correlation function
  • FIG. 8A shows the waveform of a template signal.
  • FIG. 8B shows the waveform of a received impulse radio signal at a set of several possible time offsets.
  • FIG. 8C represents the output of the correlator (multiplier and short time integrator) for each of the time offsets of FIG. 8B.
  • this graph, FIG. 8C does not show a waveform that is a function of time, but rather a function of time- offset, i.e., for any given pulse received, there is only one corresponding point which is applicable on this graph. This is the point corresponding to the time offset of the template signal used to receive that pulse.
  • the characteristics of impulse radio significantly improve the state of the art.
  • the present invention is particularly valuable when used in a radio network employing impulse radio; the present invention is compact and exhibits efficient broad beam non-dispersive radio transmission and receive characteristics with reduced ringing in the presence of impulse signals.
  • a vital component for any radio communication system is the antenna system or systems employed for transmitting and receiving radio frequency (RF) signals.
  • RF radio frequency
  • characteristics that relate to good transmitting quality for a particular antenna apply with equal relevance to receiving characteristics of the antenna.
  • Characteristics that are preferably optimized for antennas employed with an impulse radio communication system are that the antennas should be a broadband antenna that is small and compact, well-matched (preferably impedance-matched with a 50 ohm load), efficient without a propensity for ringing when subjected to pulsed signals, non-dispersive in its transceiving operations, and having a field of view appropriate for the desired application. From a practical standpoint, an antenna system should be easy to make with reliable quality in production volumes (as contrasted with volumes appropriate for prototype manufacture).
  • the present invention is embodied in antennas having a structure characterized by the inventor as "monoloop" antennas.
  • Monoloop antennas are planar single element antennas that are preferably well matched to the standard 50 ⁇ impedance design parameter employed in communication apparatuses.
  • Monoloop antennas are efficient, physically small and radiate in a broad beam pattern. Such antennas exhibit some spatial dispersion, but they emit a waveform that is relatively short and non-temporally dispersive.
  • Monoloop antennas generally include a planar radiating loop, a ground plane reflector and a feed structure for providing signals to the antenna from a host device.
  • a planar radiating loop is preferably a generally planar, approximately semi-circular arc of a suitable conducting material.
  • the plane in which the planar radiating loop is oriented is preferably normal to the plane of the ground plane.
  • the preferred typical shape of the radiating loop is close to circular, but various elliptical, ovoidal, Archimedian and log spiral shapes may also be employed to advantage. It is important to note that the present invention is configured in contrast to teaching of the prior art relating to antenna construction. Rather than being configured to block or minimize reflection from the ground plane, the present invention is oriented to take advantage of the reflections from the ground plane.
  • FIG. 9 illustrates a vertically oriented current-carrying conductor with its associated electric radiation field and magnetic radiation field.
  • an electrical conductor 10 is oriented vertically and carries an electrical current that changes with respect to time at a rate dl/dt while flowing in a direction indicated by an arrow 12 to create an electric radiation field 14.
  • Electric radiation field 14 is created in a vertical orientation; a magnetic radiation field 16 is also created in a horizontal orientation.
  • Electric radiation field 14 has a radiation field strength E and is vectorally oriented as indicated by arrows 15 in FIG. 9.
  • Magnetic radiation field 16 has a radiation field strength H and is vectorally oriented about electrical conductor 10 as indicated by an arrow 17 in FIG. 9.
  • FIG. 10 illustrates a horizontally oriented current-carrying conductor with its associated electric radiation field and magnetic radiation field.
  • an electrical conductor 20 is oriented horizontally and carries an electrical current that changes with respect to time at a rate dl/dt while flowing in a direction indicated by an arrow 22 to create an electric radiation field 24.
  • Electric radiation field 24 is created in a horizontal orientation; a magnetic radiation field 26 is also created in a vertical orientation.
  • Electric radiation field 24 has a radiation field strength E and is vectorally oriented as indicated by an arrow 25 in FIG. 10.
  • Magnetic radiation field 26 has a radiation field strength H and is vectorally oriented as indicated by an arrow 27 in FIG. 10.
  • FIG. 11 illustrates a representative prior art single element antenna structure with its associated electric radiation field.
  • an antenna apparatus 30 includes a ground element 32 and a transceiving element 34.
  • Transceiving element 34 has a thickness t that is significantly smaller than its width W.
  • Transceiving element 34 is thus a planar element oriented in an arcuate arrangement having a radius R from a center 36 that is located substantially at ground element 32.
  • Radius R is substantially perpendicular with the plane of transceiving element 34 throughout the length of transceiving element 34.
  • Transceiving element 34 is electrically connected with ground element 32 at a first end 33 of transceiving element 34.
  • the attachment is effected at an attachment locus 38 via a load impedance 40.
  • a second end 35 of transceiving element 34 includes a feed structure 42 by which transceiving element 32 conveys signals with a host device (not shown in FIG. 11) during operation.
  • a host device not shown in FIG. 11
  • current may flow in transceiving element 32 alternately in the directions indicated by arrows 44. Changing current flowing in the directions indicated by arrows 44 will support an electric radiation field 46 having a radiation field strength E in a vertical orientation vectorally directed alternately as indicated by arrows 48.
  • FIG. 12 is a side elevation view of a single element antenna according to the preferred embodiment of the present invention.
  • an antenna apparatus 50 includes a ground element 52 and a transceiver element 54.
  • Transceiver element 54 is affixed to ground element 52 at a first end 56 of transceiver element 54.
  • Transceiver element 54 is spaced from ground element 52 by a gap 59 having a gap distance G at a second end 58 of transceiver element 54.
  • Transceiver element 54 is preferably substantially planar in a transceiver plane (not shown in FIG. 12) substantially containing transceiver element 54.
  • Ground element 52 is preferably substantially planar in a ground plane 61 substantially containing ground element 52.
  • transceiver element 54 is configured as a layer of copper arranged upon a dielectric substrate to form generally planar transceiving element 54 that is affixed to a generally planar ground plane.
  • Transceiver element 54 intersects ground element 52 in a joint 51 bounded by a first terminus 53 and a second terminus 55.
  • transceiver element 54 and ground element 52 are perpendicular.
  • Transceiver element 54 is bounded by a first edge 60 and a second edge 62 in the transceiver plane.
  • First edge 60 departs in a departure direction from first terminus 53 at ground element 52 and is generally defined by a radius R] .
  • Second edge 62 departs in the same departure direction from second terminus 55 at ground element 52 and is generally defined by a radius R 2 .
  • first edge 60 and second edge 62 having the dimension (R 2 - Rj)-
  • First edge 60 and second edge 62 terminate in a termination structure 66 at second end 58.
  • termination structure 66 is an arc-section that joins first edge 60 with second edge 62.
  • termination structure 66 has a diameter at least equal to (R 2 - K ).
  • a larger separation distance (R 2 - Ri) permits a greater broadband operating capability for transceiver element 54.
  • Termination structure 66 is coupled with a feed structure 68.
  • feed structure 68 is a coaxial feed arrangement for conveying signals to and from transceiver element 54.
  • Feed structure 68 may be embodied in other configurations such as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • Feed structure 68 preferably includes a signal carrying conductor 70 surrounded by a shield 72 with an insulator 74 intermediate conductor 70 and shield 72.
  • a ground connection 76 is established intermediate shield 72 and ground element 52 when feed structure 68 is installed.
  • antenna apparatus 50 defines a separation distance D between ground element 52 and the maximum excursion of first edge 60 from ground element 52.
  • separation distance D is equal to radius Ri.
  • circumferential path C is the generally arcuate path a signal must pass along when traversing transceiver element 54 during operation as a transmitter or a receiver in operation. It is important that circumferential path C be an appropriate length vis-avis the wavelength of signals accommodated by transceiver element 54 during operation. Preferably, circumferential path C should be approximately equal with one-quarter the wavelength of frequencies handled by transceiver element 54.
  • FIG. 13 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a first representative set of signal transmission paths.
  • an antenna apparatus 50a includes a ground element 52 and a transceiver element 54.
  • Transceiver element 54 is affixed to ground element 52 at a first end 56 of transceiver element 54.
  • Transceiver element 54 is spaced from ground element 52 at a second end 58 of transceiver element 54 to establish a gap 59 intermediate ground element 52 and transceiver element 54.
  • Transceiver element 54 is preferably substantially planar in a transceiver plane (not shown in FIG. 13) substantially containing transceiver element 54.
  • transceiver element 54 is configured as a layer of copper arranged upon a dielectric substrate to form generally planar transceiver element 54.
  • Ground element 52 is preferably substantially planar in a ground plane 61 substantially containing ground element 52.
  • Transceiver element 54 is bounded by a first edge 60 and a second edge 62 in the transceiver plane.
  • a first signal path 80 emanates from transceiver element 54 (for illustration purposes) from a locus 82 generally in the middle of transceiver element 54.
  • a second signal path 90 emanates from a locus 84, also generally in the middle of transceiver element 54 (for illustration purposes).
  • Second signal path 90 includes a first path segment 92 from locus 84 to a reflection locus 86 situated on ground element 52, and a second path segment 94 from reflection locus 86 to outside transceiver element 54.
  • First signal path 80 and second path segment 94 are substantially parallel and preferably colocated (they are separated in FIG. 13 to facilitate description).
  • first signal path 80 and second signal path 90 there may be interference between signals following first signal path 80 and second signal path 90 if first signal path 80 and second signal path 90 are of appropriately different lengths.
  • First locus 82 and second locus 84 are situated on a circle 85 having a radius r centered on reflection locus 86.
  • First locus 82 and second locus 84 are separated on circle 85 by a circumferential sector ⁇ C.
  • second signal path 90 is longer than first signal path 80 by an amount ( ⁇ C + 2r).
  • the distance ( ⁇ C + 2r) is an appropriate multiple of the wavelength of the signal emanating from loci 82, 84 on signal paths 80, 90, there may be interference between signals on signal paths 80, 90.
  • a dielectric support substrate for transceiver element 54 may be employed to adjust phase relationships between direct signals (e.g., signals traveling via signal path 80) and reflected signals (e.g., signals traveling via signal path 90).
  • direct signals e.g., signals traveling via signal path 80
  • reflected signals e.g., signals traveling via signal path 90
  • FIG. 14 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a second representative set of signal transmission paths.
  • an antenna apparatus 50b includes a ground element 52 and a transceiver element 54.
  • Transceiver element 54 is affixed to ground element 52 at a first end 56 of transceiver element 54.
  • Transceiver element 54 is spaced from ground element 52 at a second end 58 of transceiver element 54 to establish a gap 59 intermediate ground element 52 and transceiver element 54.
  • Transceiver element 54 is preferably substantially planar in a transceiver plane (not shown in FIG. 14) substantially containing transceiver element 54.
  • transceiver element 54 is configured as a layer of copper arranged upon a dielectric substrate to form generally planar transceiver element 54.
  • Ground element 52 is preferably substantially planar in a ground plane 61 substantially containing ground element 52.
  • Transceiver element 54 is bounded by a first edge 60 and a second edge 62 in the transceiver plane.
  • a first signal path 100 emanates from transceiver element 54 (for illustration purposes) from a locus 102 generally in the middle of transceiver element 54.
  • a second signal path 110 emanates from a locus 104, also generally in the middle of transceiver element 54 (for illustration purposes).
  • Second signal path 110 includes a first path segment 112 from locus 104 to a reflection locus 106 situated on ground element 52, and a second path segment 114 from reflection locus 106 to outside transceiver element 54.
  • First signal path 100 and second path segment 114 are substantially parallel and preferably colocated (they are separated in FIG. 14 to facilitate description).
  • first signal path 100 and second signal path 110 there may be interference between signals following first signal path 100 and second signal path 110, if first signal path 100 and second signal path 110 are of appropriately different lengths.
  • Locus 102 and locus 104 are situated on a circle 105 having a radius r centered on reflection locus 106. Locus 102 and locus 104 are separated on circle 105 by a circumferential sector ⁇ C.
  • second signal path 110 is differs in length from first signal path 100 by an amount (2r- ⁇ C).
  • the distance (2r- ⁇ C) is an appropriate multiple of the wavelength of the signal emanating from loci 102, 104 on signal paths 100, 110, there may be interference between signals on signal paths 100, 110.
  • a dielectric support substrate for transceiver element 54 may be employed to adjust phase relationships between direct signals (e.g., signals traveling via signal path 100) and reflected signals (e.g., signals traveling via signal path 110).
  • direct signals e.g., signals traveling via signal path 100
  • reflected signals e.g., signals traveling via signal path 110.
  • FIG. 14 is representative of selected signal paths only. Other relationships among signal paths are also possible, as represented in FIGs. 13 and 15.
  • FIG. 15 is a plan view of the single element antenna illustrated in FIG. 12, illustrating a third representative set of signal transmission paths.
  • an antenna apparatus 50c includes a ground element 52 and a transceiver element 54.
  • Transceiver element 54 is affixed to ground element 52 at a first end 56 of transceiver element 54.
  • Transceiver element 54 is spaced from ground element 52 at a second end 58 of transceiver element 54 to establish a gap 59 intermediate ground element 52 and transceiver element 54.
  • Transceiver element 54 is preferably substantially planar in a transceiver plane (not shown in FIG. 15) substantially containing transceiver element 54.
  • transceiver element 54 is configured as a layer of copper arranged upon a dielectric substrate to form generally planar transceiver element 54.
  • Ground element 52 is preferably substantially planar in a ground plane 61 substantially containing ground element 52.
  • Transceiver element 54 is bounded by a first edge 60 and a second edge 62 in the transceiver plane.
  • a first signal path 120 emanates from transceiver element 54 (for illustration purposes) from a locus 122 generally in the middle of transceiver element 54.
  • a second signal path 130 also emanates from locus 124.
  • Second signal path 130 includes a first path segment 132 from locus 122 to a reflection locus 126 situated on ground element 52, and a second path segment from reflection locus 126 to outside transceiver element 54.
  • First signal path 120 and second path segment 134 are substantially colinear. There may be interference between signals following first signal path 120 and second signal path 130, if first signal path 120 and second signal path 130 are of appropriately different lengths.
  • Locus 122 is situated on a circle 125 having a radius r centered on reflection locus 126.
  • second signal path 130 is longer than first signal path 120 by an amount (2r).
  • distance (2r) is an appropriate multiple of the wavelength of the signal emanating from locus 122 on signal paths 120, 130
  • Presence of a dielectric substrate supporting a copper transceiving element establishes a longer effective path for signals traversing transceiver element 54. That is, a dielectric substrate support structure establishes an effectively longer time delay that is manifested in an increase of the parameter ⁇ C.
  • a dielectric support substrate for transceiver element 54 may be employed to adjust phase relationships between direct signals (e.g., signals traveling via signal path 120) and reflected signals (e.g., signals traveling via signal path 130).
  • the situation illustrated in FIG. 15 is representative of selected signal paths only. Other relationships among signal paths are also possible, as represented in FIGs. 13 and 14.
  • FIG. 16 is a perspective view of the single element antenna of the present invention with its associated electric radiation field.
  • an antenna apparatus 50 includes a ground element 52 and a transceiver element 54.
  • Transceiver element 54 is preferably configured as a layer of copper 63 arranged upon a dielectric substrate 65 to form generally planar transceiver element 54.
  • Transceiver element 54 is affixed to ground element 52 at a first end 56 of transceiver element 54.
  • Transceiver element 54 is spaced from ground element 52 by a gap 59 at a second end 58 of transceiver element 54.
  • Transceiver element 54 intersects ground element in a joint 51 bounded by a first terminus 53 and a second terminus 55.
  • transceiver 54 and ground element 52 are perpendicular.
  • Transceiver element 54 is bounded by a first edge 60 and a second edge 62 in the transceiver plane.
  • First edge 60 departs in a departure direction from first terminus 53 at ground element 52
  • second edge 62 departs in the same departure direction from second terminus 55 at ground element 52. There is thus defined a separation distance between first edge 60 and second edge 62.
  • First edge 60 and second edge 62 terminate in a termination structure 66 at second end 58.
  • termination structure 66 is an arc-section that joins first edge 60 with second edge 62.
  • termination structure 66 has a diameter at least equal to the separation distance between first edge 60 and second edge 62 at second end 58.
  • Termination structure 66 is coupled with a signal carrying conductor 70 that is part of a feed structure (not shown in FIG. 16).
  • antenna apparatus 50 may provide signals via signal carrying conductor 70 to transceiver element 54 for transmission to a medium adjacent to transceiver element 54.
  • current may alternately flow in transceiver element 54 in the directions indicated by arrows 64.
  • Alternate current flow in the directions indicated by arrows 64 will support an electric radiation field 67 having a radiation field strength E in a vertical orientation vectorally directed as indicated by arrows 69.
  • Such an orientation for electric radiation field 67 is advantageous when antenna 50 may be oriented to have a wide field of vision (i.e., a wide field orientation of electric radiation field 67) in a horizontal plane while exhibiting a relatively narrow field of vision in a vertical plane.
  • Such an arrrangement is particularly useful in certain applications, such as a position-determining ranging apparatus like a radar apparatus.
  • FIG. 17 is a perspective view of a first alternate embodiment of a single element antenna according to the present invention.
  • an antenna apparatus 1750 includes aground element 1752 and a transceiver element 1754.
  • Transceiver element 1754 is affixed to ground element 1752 at a first end 1756 of transceiver element 1754.
  • Transceiver element 1754 is spaced from ground element 1752 by a gap 1759 at a second end 1758 of transceiver element 1754.
  • Transceiver element 1754 intersects ground element in a joint 1751 bounded by a first terminus 1753 and a second terminus 1755.
  • transceiver 1754 and ground element 1752 are perpendicular.
  • Transceiver element 1754 is bounded by a first edge 1760 and a second edge 1762 in the transceiver plane.
  • First edge 1760 departs in a departure direction from first terminus 1753 at ground plane 1752.
  • First edge 1760 has a radius r with respect to a center point 1761.
  • Second edge 1762 departs in the same departure direction from second terminus 1755 at ground plane 1752. There is thus defined a separation distance between first edge 1760 and second edge 1762.
  • First edge 1760 and second edge 1762 terminate in a termination structure 1766 at second end 1758.
  • termination structure 1766 is an arc-section that joins first edge 1760 with second edge 1762.
  • termination structure 1766 has a diameter at least equal to the separation distance between first edge 1760 and second edge 1762 at second end 1758.
  • Termination structure 1766 is coupled with a signal carrying conductor 1770 that is part of a feed structure (not shown in FIG. 17).
  • Ground element 1752 includes a substantially planar land 1757 and a substantially semi-cylindrical channel structure 1779 that departs from planar land 1757 in a direction opposite from the departure direction of first edge 1760 and second edge 1762.
  • Channel structure 1779 has a longitudinal axis 1763; longitudinal axis 1763 passes through center point 1761.
  • Channel structure 1779 has a radius that is substantially equal to radius r of first edge 1760.
  • channel structure 1779 and first edge 1760 are substantially symmetrically arranged about center point 1761 in a plane containing transceiver element 1754. In such a symmetric arrangement, certain transmit and receive characteristics of antenna apparatus 1750 may be enhanced for certain frequencies.
  • FIG. 18 is a side view of a second alternate embodiment of a single element antenna according to the present invention.
  • an antenna apparatus 1850 includes aground element 1852, a first transceiver element 1854a and a second transceiver element 1854b.
  • First transceiver element 1854a is affixed to ground element 1852 at a first end 1856a of first transceiver element 1854a.
  • Second transceiver element 1854b is affixed to ground element 1852 at a first end 1856b of second transceiver element 1852b.
  • a gapl878 having a gap dimension G is established by a spaced relation between a second end 1858a of first transceiver element 1854a and a second end 1858b of second transceiver element 1854b.
  • Transceiver elements 1854a, 1854b are preferably substantially coplanar in a transceiver plane (not shown in FIG. 18) substantially containing transceiver elements 1854a, 1854b.
  • Ground element 1852 is preferably substantially planar in a ground plane 1861 substantially containing ground element 1852.
  • First transceiver element 1854a intersects ground element 1852 in a joint 1851a bounded by a first terminus 1853a and a second terminus 1855a.
  • first transceiver element 1854a and ground element 1852 are perpendicular.
  • First transceiver element 1854a is bounded by a first edge 1860a and a second edge 1862a in the transceiver plane.
  • First edge 1860a departs in a departure direction from first terminus 1853a at ground element 1852; second edge 1862a departs in the same departure direction from second terminus 1855a at ground element 1852.
  • termination structure 1866a is an arc-section that joins first edge 1860a with second edge 1862a.
  • termination structure 1866a has a diameter at least equal to the separation distance at second end 1858a
  • Second transceiver element 1854b intersects ground element 1852 in a joint 1851b bounded by a first terminus 1853b and a second terminus 1855b.
  • second transceiver element 1854b and ground element 1852 are perpendicular.
  • Second transceiver element 1854b is bounded by a first edge 1860b and a second edge 1862b in the transceiver plane.
  • First edge 1860b departs in a departure direction from first terminus 1853b at ground element 1852; second edge 1862b departs in the same departure direction from second terminus 1855b at ground element 1852.
  • First edge 1860b and second edge 1862b terminate in a termination structure 1866b at second end 1858b.
  • termination structure 1866b is an arc-section that joins first edge 1860b with second edge 1862b.
  • termination structure 1866b has a diameter at least equal to the separation distance at second end 1858b.
  • First termination structure 1866a is coupled with a feed lead 1871.
  • Feed lead 1871 connects with a feed structure 1868.
  • Feed lead 1871 may be embodied in such configurations as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • a feed bridge 1879 connects termination structure 1866b with feed structure 1868 via feed lead 1871.
  • feed structure 1868 is a coaxial feed arrangement for conveying signals to and from first transceiver element 1854a
  • Feed structure 1868 may be embodied in other configurations such as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • Feed structure 1868 preferably includes a signal carrying conductor 1870 surrounded by a shield 1872 with an insulator 1874 intermediate conductor 1870 and shield 1872.
  • Aground connection 1876 is established intermediate shield 1872 and ground element 1852 when feed structure 1868 is installed.
  • FIG. 19 is a side view of a third alternate embodiment of a single element antenna according to the present invention.
  • an antenna apparatus 1950 includes a ground element 1952 and a transceiver element 1954.
  • Ground element 1952 includes a first ground element segment 1952a and a second ground element segment 1952b.
  • Preferably ground element segments 1952a, 1952b are perpendicular (they may meet at another angle than 90 degrees) and form what is commonly known as a comer reflector.
  • Transceiver element 1954 is affixed to second ground element segment 1952b at a first end 1956 of transceiver element 1954.
  • Transceiver element 1954 is spaced from first ground element segment 1952a by a gap 1959 at a second end 1958 of transceiver element 1954.
  • Transceiver element 1954 is preferably substantially planar in a transceiver plane (not shown in FIG. 19) substantially containing transceiver element 1954. Each respective ground element segment 1952a, 1952b is preferably substantially planar. Transceiver element 1954 intersects second ground element segment 1952b in a joint 1951 bounded by a first terminus 1953 and a second terminus 1955. Preferably, transceiver element 1954 and second ground element segment 1952b are perpendicular. Transceiver element 1954 is bounded by a first edge 1960 and a second edge 1962 in the transceiver plane. First edge 1960 departs in a departure direction from first terminus 1953 at second ground element segment 1952b. Second edge 1962 departs in the same departure direction from second terminus 1955 at second ground element segment 1952b.
  • first edge 1960 and second edge 1962 There is a separation distance between first edge 1960 and second edge 1962.
  • First edge 1960 and second edge 1962 terminate in a termination structure 1966 at second end 1958.
  • termination structure 1966 is an arc-section that joins first edge 1960 with second edge 1962.
  • termination structure 1966 has a diameter at least equal the separation distance at second end 1958.
  • Termination structure 1966 is coupled with a feed structure 1968.
  • feed structure 1968 is a coaxial feed arrangement for conveying signals to and from transceiver element 1954.
  • Feed structure 1968 may be embodied in other configurations such as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • Feed structure 1968 preferably includes a signal carrying conductor 1970 surrounded by a shield 1972 with an insulator 1974 intermediate conductor 1970 and shield 1972.
  • a ground connection 1976 is established intermediate shield 1972 and ground element segment 1952a when feed structure 1968 is installed.
  • FIG. 20 is a side view of a fourth alternate embodiment of a single element antenna according to the present invention.
  • an antenna apparatus 2050 includes a ground element 2052, a first transceiver element 2054a and a second transceiver element 2054b.
  • Ground element 2052 includes a first ground element segment 2052a and a second ground element segment 2052b.
  • Preferably ground element segments 2052a, 2052b are perpendicular (they may meet at another angle than 90 degrees) and form what is commonly known as a comer reflector.
  • First transceiver element 2054a is affixed to first ground element segment 2052a at a first end 2056a of first transceiver element 2052a
  • Second transceiver element 2054b is affixed to second ground element segment 2052b at a first end 2056b of second transceiver element 2054b.
  • a gap 2078 having a gap dimension G is established by a spaced relation between a second end 2058a of first transceiver element 2054a and a second end 2058b of second transceiver element 2054b.
  • Transceiver elements 2054a, 2054b are preferably substantially coplanar in a transceiver plane (not shown in FIG. 20) substantially containing transceiver elements 2054a, 2054b.
  • Each respective ground element segment 2052a, 2052b is preferably substantially planar.
  • First transceiver element 2054a intersects first ground element segment 2052a in a joint 2051a bounded by a first terminus 2053a and a second terminus 2055a.
  • first transceiver element 2054a and first ground element segment 2052a are perpendicular.
  • First transceiver element 2054a is bounded by a first edge 2060a and a second edge 2062a in the transceiver plane.
  • First edge 2060a departs in a departure direction from first terminus 2053a at first ground element segment 2052a; second edge 2062a departs in the same departure direction from second terminus 2055a at first ground element segment 2052a.
  • termination structure 2066a is an arc- section that joins first edge 2060a with second edge 2062a
  • termination structure 2066a has a diameter at least equal to the separation distance at second end 2058a.
  • Second transceiver element 2054b intersects second ground element segment 2052b in a joint 2051b bounded by a first terminus 2053b and a second terminus 2055b.
  • second transceiver element 2054b and second ground element segment 2052b are perpendicular.
  • Second transceiver element 2054b is bounded by a first edge 2060b and a second edge 2062b in the transceiver plane.
  • First edge 2060b departs in a departure direction from first terminus 2053b at second ground element segment 2052b; second edge 2062b departs in the same departure direction from second terminus 2055b at second ground element segment 2052b.
  • First edge 2060b and second edge 2062b terminate in a termination structure 2066b at second end 2058b.
  • termination structure 2066b is an arc-section distance that joins first edge 2060b with second edge 2062b.
  • termination structure 2066b has a diameter at least equal to the separation distance at second end 2058b.
  • Termination structure 2066a is coupled with a feed lead 2071.
  • a feed bridge 2079 connects termination structure 2066b with feed structure 2068 via feed lead 2071.
  • Feed lead 2071 connects with a feed structure 2068.
  • feed structure 2068 is a coaxial feed arrangement for conveying signals to and from transceiver element 2054a
  • Feed structure 2068 may be embodied in other configurations such as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • Feed structure 2068 preferably includes a signal carrying conductor 2070 surrounded by a shield 2072 with an insulator 2074 intermediate conductor 2070 and shield 2072.
  • a ground connection 2076 is established intermediate shield 2072 and ground element 2052 when feed structure 2068 is installed.
  • FIG. 21 is a side view of a fifth alternate embodiment of a single element antenna according to the present invention.
  • an antenna apparatus 2150 includes a ground element 2152, a first transceiver element 2154a and a second transceiver element 2154b.
  • Ground element 2152 is a curved shape, preferably spherical or parabolic in at least two dimensions, but other curved configurations may be employed as well to form what is commonly known as a directional reflector.
  • First transceiver element 2154a is affixed to ground element 2152 at a first end 2156a of first transceiver element 2152a
  • Second transceiver element 2154b is affixed to ground element 2152 at a first end 2156b of second transceiver element 2154b.
  • a gap 2178 having a gap dimension G is established by a spaced relation between a second end 2158a of first transceiver element 2154a and a second end 2158b of second transceiver element 2154b.
  • Transceiver elements 2154a, 2154b are preferably substantially coplanar in a transceiver plane (not shown in FIG. 21) substantially containing transceiver elements 2154a, 2154b.
  • First transceiver element 2154a intersects ground element 2152 in a joint 2151a bounded by a first terminus 2153a and a second terminus 2155a
  • First transceiver element 2154a is bounded by a first edge 2160a and a second edge 2162a in the transceiver plane.
  • First edge 2160a departs in a departure direction from first terminus 2153a at ground element 2152; second edge 2162a departs in the same departure direction from second terminus 2155a at ground element 2152.
  • termination structure 2166a is an arc-section that joins first edge 2160a with second edge 2162a
  • termination structure 2166a has a diameter at least equal to the separation distance at second end 2158a.
  • Second transceiver element 2154b intersects ground element 2152 in a joint 2151b bounded by a first terminus 2153b and a second terminus 2155b.
  • Second transceiver element 2154b is bounded by a first edge 2160b and a second edge 2162b in the transceiver plane.
  • First edge 2160b departs in a departure direction from first terminus 2153b at ground element 2152;
  • second edge 2162b departs in the same departure direction from second terminus 2155b at ground element 2152.
  • First edge 2160b and second edge 2162b terminate in a termination structure 2166b at second end 2158b.
  • termination structure 2166b is an arc-section distance that joins first edge 2160b with second edge 2162b.
  • termination structure 2166b has a diameter at least equal to the separation distance at second end 2158b.
  • Termination structure 2166a is coupled with a feed lead 2171.
  • a feed bridge 2179 connects termination structure 2166b with feed structure 2168 via feed lead 2171.
  • Feed lead 2171 connects with a feed structure 2168.
  • feed structure 2168 is a coaxial feed arrangement for conveying signals to and from transceiver element 2154a
  • Feed structure 2168 may be embodied in other configurations such as, by way of illustration and not by way of limitation, coaxial cable, stripline, microstrip, twin lead, twisted pair fiber optic cable, wave guide or other transmission line, or any connector or coupler that enables connection to a transmission line.
  • Feed structure 2168 preferably includes a signal carrying conductor 2170 surrounded by a shield 2172 with an insulator 2174 intermediate conductor 2170 and shield 2172.
  • a ground connection 2176 is established intermediate shield 2172 and ground element 2152 when feed structure 2168 is installed.
  • the antenna should be a broadband antenna that is small and compact, well -matched - preferably impedance-matched with a 50 ohm load, efficient without a propensity for ringing when subjected to pulsed signals, non-dispersive in its transceiving operations, and radiates in a broad beam) one must consider the ease of manufacture in reliable quantities provided by a planar antenna
  • the antennas disclosed in the present invention are capable of radiating very short, non-time-dispersive pulses, they are ideal for use in an array.
  • Conventional elements in arrays exhibit undesirable grating lobes as later lobes of a pulse waveform interfere with earlier lobes.
  • the antennas that are the subject of the present disclosure can emit short non- time-dispersive pulses that significantly mitigate the grating lobe problem.
  • Such short pulse waveforms allow the antennas of the present invention to be advantageously used in conjunction with comer, planar, convex cylindrical or concave cylindrical reflectors.
  • defocusing leads to undesired grating lobes.
  • the short, non-time- dispersive pulses of the antennas of the present invention allow a reflected waveform to be defocused without leading to the grating lobes experienced when using conventional antennas. Defocusing a waveform without creating grating lobes permits higher gain and directivity than are achievable using prior art antenna elements.

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  • Variable-Direction Aerials And Aerial Arrays (AREA)

Abstract

L'invention concerne une antenne permettant de transférer de l'énergie électromagnétique, qui comprend: a) un élément de terre (52) situé dans un plan de terre (61), b) un élément émetteur-récepteur (54) situé dans un plan d'émetteur-récepteur, qui croise l'élément de terre au niveau d'une première extrémité dans un joint (51) possédant une première extrémité (53) et une seconde extrémité (55), et c) une structure d'alimentation (68) qui transporte l'énergie électromagnétique entre l'émetteur-récepteur et le dispositif hôte. Un premier bord d'émetteur-récepteur (60) part du joint selon un premier chemin dans une première direction. Un second bord d'émetteur-récepteur (62) part du joint selon un second chemin dans la première direction. Chaque bord comprend au moins un premier secteur de bord possédant un premier rayon et un second secteur de bord possédant un second rayon. Les rayons définissent un séparation entre les secteurs de bord. Les premier et second bords aboutissent à une structure terminale (66) au niveau d'une seconde extrémité espacée de l'élément de terre pour former un espace.
PCT/US2002/000154 2002-01-03 2002-01-03 Antenne cadre a large bande Ceased WO2003067707A1 (fr)

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Application Number Priority Date Filing Date Title
AU2002255474A AU2002255474A1 (en) 2002-01-03 2002-01-03 Broadband loop antenna
PCT/US2002/000154 WO2003067707A1 (fr) 2002-01-03 2002-01-03 Antenne cadre a large bande

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Application Number Priority Date Filing Date Title
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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7327318B2 (en) 2006-02-28 2008-02-05 Mti Wireless Edge, Ltd. Ultra wide band flat antenna

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US2615134A (en) * 1946-01-09 1952-10-21 Rca Corp Antenna
US3015101A (en) * 1958-10-31 1961-12-26 Edwin M Turner Scimitar antenna
DE3529914A1 (de) * 1985-08-21 1987-03-05 Siemens Ag Mikrowellenstrahler
DE19857191A1 (de) * 1998-12-11 2000-07-06 Bosch Gmbh Robert Halfloop-Antenne

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US2615134A (en) * 1946-01-09 1952-10-21 Rca Corp Antenna
US3015101A (en) * 1958-10-31 1961-12-26 Edwin M Turner Scimitar antenna
DE3529914A1 (de) * 1985-08-21 1987-03-05 Siemens Ag Mikrowellenstrahler
DE19857191A1 (de) * 1998-12-11 2000-07-06 Bosch Gmbh Robert Halfloop-Antenne

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7327318B2 (en) 2006-02-28 2008-02-05 Mti Wireless Edge, Ltd. Ultra wide band flat antenna

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