TW202339406A - High voltage gain DC converter which uses two coupling inductors for voltage doubling without requiring operation at a maximum conduction ratio - Google Patents

Abstract

A high voltage gain DC converter includes a first coupling inductor, a second coupling inductor, two switches, a first clamping diode, a second clamping diode, a first voltage doubling capacitor, a second voltage doubling diode, a second voltage doubling capacitor, a first output stage, a second output stage, and a third output stage. The input parallel architecture can share the input current, so it is suitable for high input current applications. The first switch and the second switch adopt an interleaved operation, which can cause the ripple current on the primary side of the coupling inductor to have a cancelling effect with each other and can reduce the input current ripple. The first voltage doubling capacitor, the second voltage doubling diode, and the second voltage doubling capacitor provide the voltage doubling function. Three coupling inductors are used for voltage doubling and are superimposed for output with a first output stage, a second output stage, and a third output stage to further increase the voltage gain.

Description

High Voltage Gain DC Converter

The present invention relates to a voltage conversion technology, in particular to a high voltage gain DC converter.

Referring to Figure 1, a conventional boost converter is shown. The conventional boost converter operates at an extremely high conduction ratio to achieve a higher voltage gain. , the parameters V _O , V _in , and D are the output voltage, input voltage, and the duty conduction ratio of the switch respectively. However, in practice, they are affected by parasitic components. When the conduction ratio exceeds 0.9, the voltage gain does not increase but decreases, which is not high. Therefore, high boost converters that do not require extremely high conduction ratios and at the same time meet the requirements of high voltage gain are future research directions.

Therefore, an object of the present invention is to provide a high voltage gain DC converter that can overcome the shortcomings of the prior art.

Therefore, the high voltage gain DC converter includes a first coupling inductor, a second coupling inductor, a first switch, a second switch, a first clamping diode, a second clamping diode, a third A regenerative diode, a first regenerative capacitor, a first voltage doubling diode, a first voltage doubling capacitor, a second regenerative diode, a second regenerative capacitor, and a second voltage doubling diode body, a second voltage doubling capacitor, a first output stage, a second output stage and a third output stage.

Each coupled inductor has a first winding, a second winding and a third winding, and each winding has a first end and a second end, wherein the first end of the first winding of the first coupled inductor and The first end of the first winding of the second coupled inductor is electrically connected together to receive a DC input voltage, and the second end of the second winding of the first coupled inductor is electrically connected to the second end of the second winding of the second coupled inductor. The second end of the third winding of the first coupling inductor is electrically connected to the second end of the third winding of the second coupling inductor.

The first switch has a first end electrically connected to the second end of the first winding of the first coupling inductor, and a second end grounded, and the first switch is controlled to switch between a conductive state and a non-conductive state. , when the first switch is turned on, the first winding of the first coupling inductor receives a current for charging. The second switch has a first end electrically connected to the second end of the first winding of the second coupling inductor, and a second end grounded, and the second switch is controlled to switch between a conductive state and a non-conductive state. .

The first clamping diode has an anode electrically connected to the second end of the first winding of the first coupling inductor, and a cathode. The second clamping diode has an anode electrically connected to the second end of the first winding of the second coupling inductor, and a cathode. The cathode of the first clamping diode is connected to the second clamping diode. The cathode is electrically connected to a common contact.

The first regenerative diode has an anode electrically connected to the first end of the second winding of the second coupling inductor, and a cathode. The first regenerative capacitor is electrically connected between the first end of the second winding of the first coupling inductor and the cathode of the first regenerative diode. The first voltage multiplying diode has an anode electrically connected to the cathode of the first regenerative diode, and a cathode. A first voltage doubling capacitor is electrically connected between the first end of the second winding of the second coupling inductor and the cathode of the first voltage doubling diode.

The second regenerative diode has an anode electrically connected to the first end of the third winding of the first coupling inductor, and a cathode. The second regenerative capacitor is electrically connected between the first end of the third winding of the second coupling inductor and the cathode of the second regenerative diode. The second voltage doubler diode has an anode electrically connected to the cathode of the second regenerative diode, and a cathode. The second voltage doubling capacitor is electrically connected between the first end of the third winding of the first coupling inductor and the cathode of the second voltage doubling diode.

A first output stage is electrically connected to the common contact for generating a first DC voltage based on the discharge from the first winding; a second output stage is electrically connected to the cathode of the first voltage multiplier diode and The common contact is used to generate a second DC voltage based on the discharge from the second winding and the first voltage multiplier capacitor. The third output stage is electrically connected to the cathode of the second voltage multiplier diode, and is used to generate a third voltage in the form of a direct current based on the discharge from the third winding and the second voltage multiplier capacitor. The first output stage is also stacked with the second output stage and the third output stage to generate an output voltage that is proportional to the sum of the first voltage, the second voltage, and the third voltage.

The effect of the present invention is to use two coupled inductors for voltage multiplication, so that the voltage gain has two degrees of design freedom: the coupled inductor turns ratio and the switch conduction ratio. Therefore, it is not necessary to operate at a huge conduction ratio to achieve high voltage gain.

Before the present invention is described in detail, it should be noted that in the following description, similar elements are designated with the same numbering.

Referring to FIG. 2, an embodiment of the high voltage gain DC converter of the present invention includes a first coupling inductor 1, a second coupling inductor 2, a first switch S ₁ , a second switch S ₂ , and a first clamp. Bit diode D ₁ , a second clamping diode D ₂ , a first regeneration diode D 11 , a first regeneration capacitor C ₁₁ , a first voltage multiplication diode D ₁₂ , a first multiplication diode D ₁ Piezocapacitor C ₁₂ , a second regeneration diode D ₂₁ , a second regeneration capacitor C ₂₁ , a second voltage doubling diode D ₂₂ , a second voltage doubling capacitor C ₂₂ , and a first output stage O1, a second output stage O2, a third output stage O3, and a control unit 4.

The first coupled inductor 1 has a first winding N ₁ , a second winding N ₂ and a third winding N ₃ , and the second coupled inductor 2 has a first winding P ₁ , a second winding P ₂ and a third winding N 3 . Winding P ₃ , each winding N ₁ ~N ₃ has a first end and a second end, wherein the first end of the first winding N ₁ of the first coupled inductor 1 and the third end of the second coupled inductor 2 The first end of a winding P ₁ is electrically connected together to receive a DC input voltage, and the second end of the second winding N ₂ of the first coupled inductor 1 is electrically connected to the second winding P 2 of the second coupled inductor ₂ The second end of the third winding N ₃ of the first coupled inductor 1 is electrically connected to the second end of the third winding P ₃ of the second coupled inductor 2 . The first end of each first winding N ₁ , P ₁ is a dot end, and the second end of each first winding N ₁ , P ₁ is a non-dot end. The first end of each second winding N ₂ , P ₂ is a dot end, and the second end of each second winding N ₂ , P ₂ is a non-dot end. The first end of each third winding N ₃ , P ₃ is a dot end, and the second end of each third winding N ₃ , P ₃ is a non-dot end.

The first switch S ₁ has a first end electrically connected to the second end of the first winding N ₁ of the first coupling inductor 1 and a second end grounded, and the first switch S ₁ is controlled to switch between Between the conduction state and the non-conduction state, when the first switch S ₁ is turned on, the first winding N ₁ of the first coupling inductor 1 receives a current for charging. The first switch S ₁ is an N-type power semiconductor transistor, and the first terminal of the first switch S ₁ is the drain, and the second terminal of the first switch S ₁ is the source.

The second switch S ₂ has a first end electrically connected to the second end of the first winding P ₁ of the second coupling inductor 2 and a second end grounded, and the second switch S ₂ is controlled to switch between Between the conductive state and the non-conductive state; the second switch S ₂ is an N-type power semiconductor transistor, and the first terminal of the second switch S ₂ is the drain, and the second terminal of the second switch S ₂ is the source. Extremely.

The first clamping diode D ₁ has an anode electrically connected to the second end of the first winding N ₁ of the first coupling inductor 1 and a cathode. The second clamping diode D ₂ has an anode electrically connected to the second end of the first winding P ₁ of the second coupling inductor 2 and a cathode. The cathode of the first clamping diode D 1 is connected to the second end of the first winding P ₁ of the second coupling inductor 2 . The cathode of the second clamping diode _D2 is electrically connected to a common contact.

The first regenerative diode D ₁₁ has an anode electrically connected to the first end of the second winding P ₂ of the second coupling inductor 2 and a cathode. The first regeneration capacitor C ₁₁ is electrically connected between the first end of the second winding N ₂ of the first coupling inductor 1 and the cathode of the first regeneration diode D ₁₁ . The first voltage multiplying diode D ₁₂ has an anode electrically connected to the cathode of the first regenerative diode D ₁₁ and a cathode. The first voltage-doubling capacitor C ₁₂ is electrically connected between the first end of the second winding P ₂ of the second coupling inductor 2 and the cathode of the first voltage-doubling diode D ₁₂ .

The second regenerative diode D ₂₁ has an anode electrically connected to the first end of the third winding N ₃ of the first coupling inductor 1 and a cathode. The second regeneration capacitor C ₂₁ is electrically connected between the first end of the third winding P ₃ of the second coupling inductor 2 and the cathode of the second regeneration diode D ₂₁ . The second voltage-doubling diode D ₂₂ has an anode electrically connected to the cathode of the second regenerative diode D ₂₁ and a cathode. The second voltage doubling capacitor C ₂₂ is electrically connected between the first end of the third winding N ₃ of the first coupling inductor 1 and the cathode of the second voltage doubling diode D ₂₂ .

The first output stage O1 is electrically connected to the common contact for generating a first DC voltage based on the discharge from the first windings N ₁ and P ₁ . The first output stage O1 includes a first output capacitor C _o1 . The first output capacitor C _o1 is electrically connected between the common node and ground to provide the first voltage.

The second output stage O2 is electrically connected between the cathode of the first voltage-doubling diode D ₁₂ and the common contact, and is used to generate signals from the second windings N ₂ and P ₂ and the first voltage-doubling capacitor C ₁₂ The discharge generates a second voltage that is DC. The second output stage O2 includes a first output diode D _o1 and a second output capacitor C _o2 .

The first output diode D _o1 has an anode electrically connected to one end of the first voltage doubling capacitor C ₁₂ and a cathode. The second output capacitor C _o2 is electrically connected between the cathode of the first output diode D _o1 and the first output capacitor C _o1 for providing the second voltage.

The third output stage O3 is electrically connected to the cathode of the second voltage doubling diode D ₂₂ to generate a direct current based on the discharge from the third winding N ₃ , P ₃ and the second voltage doubling capacitor C ₂₂ the third voltage. The third output stage O3 includes a second output diode D _o2 and a third output capacitor C _o3 .

The second output diode D _o2 has an anode electrically connected to one end of the second voltage doubling capacitor C ₂₂ and a cathode. The third output capacitor C _o3 is electrically connected between the cathode of the second output diode D _o2 and the second output capacitor C _o2 to provide the third voltage, wherein the first output capacitor C _o1 and the second output capacitor C o2 The second output capacitor C _o2 and the third output capacitor C _o3 are stacked together.

The first output stage O1 is also superimposed with the second output stage O2 and the third output stage O3 to generate an output voltage _Vo . The output voltage _Vo is proportional to the first voltage, the second voltage and the The sum of the third voltage.

The control unit 4 generates a first pulse wave modulation signal for switching the first switch S ₁ and a second pulse wave modulation signal for switching the second switch S ₂ . The first pulse wave modulation signal and the third pulse wave modulation signal are The two pulse modulated signals have the same cycle time. A portion of the cycle time of the first and second pulse wave modulation signals overlap. The switching timing diagram of the first switch S ₁ and the second switch S ₂ will be further described in ten stages below.

Refer to Figure 3, which is an equivalent circuit diagram of this embodiment, illustrating the magnetizing inductors L _m1 and L _m2 and their leakage inductances L _k1 and L _k2 in the non-ideal equivalent circuits of the two coupled inductors. Among them, the parameter V _in represents the input voltage, and the parameter V _o represents the output voltage.

Referring to Figure 4, which is an operation timing diagram of this embodiment, in which parameters v _gs1 and v _gs2 respectively represent the voltages of the first and second pulse signals that control whether the first switch S ₁ and the second switch S ₂ are turned on. The parameters v _ds1 and v _ds2 respectively represent the two-terminal cross-voltage of the first switch S ₁ and the second switch S ₂ , and the parameter _TS is the period time of the first pulse signal. The parameter i _in represents the input current. The parameters i _Lk1 and i _Lk2 respectively represent the leakage inductance current flowing through the first winding N ₁ of the first coupling inductor 1 and the first winding P ₁ of the second coupling inductor 2 . The parameter i _D1 represents the current flowing through the first clamping diode D ₁ , the parameter i _D2 represents the current flowing through the second clamping diode D ₂ , and the parameter i _D11 represents the current flowing through the first regenerative diode D ₁₁ The current, parameter i _D21 represents the current flowing through the second regenerative diode D ₂₁ , the parameter i _D12 represents the current flowing through the first voltage doubling diode D ₁₂ , and the parameter i _D22 represents the current flowing through the second voltage doubling diode The current of body D ₂₂ , the parameter i _Do1 represents the current flowing through the first output diode D _o1 , and the parameter i _Do2 represents the current flowing through the second output diode D _o2 .

The following are circuit diagrams of each of the ten stages of operation in this embodiment, and each stage is described separately.

The first stage (time: t ₀ ~t ₁ ):

Referring to Figure 4 and Figure 5, the first phase begins at t=t _0. The first switch S ₁ is switched on, and the second switch S ₂ remains on. Due to the existence of leakage inductance L _k1 , and at time t ₀ The leakage inductance current i _Lk1 is 0, so the first switch S ₁ has zero current switching (ZCS) soft-cutting performance of being turned on. The leakage inductor current i _Lk1 rises rapidly. When i _Lk1 <i _Lm1 , the energy stored in the magnetizing inductor L _m1 is coupled and transmitted to the second winding N ₂ and the third winding N ₃ . The first regenerative diode D ₁₁ , the first output diode D _o1 and the second voltage doubler diode D ₂₂ are turned on, the currents i _D11 , i _Do1 and i _D22 decrease, and the leakage inductances L _k1 and L _k2 are controlled The decreasing rate of the current of the first regenerative diode D ₁₁ , the first output diode D _o1 and the second voltage doubling diode D ₂₂ thus alleviates the reverse recovery problem of these diodes. The remaining diodes, such as the first clamping diode D ₁ , the second clamping diode D ₂ , the first voltage doubling diode D ₁₂ , the second regenerative diode D ₂₁ and the second output diode D _o2 is reverse biased and does not conduct.

When t=t ₁ , the current i _Lk1 rises to satisfy i _Lk1 = i _Lm1 so that the currents i _D11 , i _Do2 and i _D22 drop to 0, the first regenerative diode D ₁₁ , the first output diode D _o1 and the This stage ends when the voltage-double diode D ₂₂ naturally transitions to non-conducting state through zero current switching (ZCS).

The second stage ( ):

Referring to Figure 4 and Figure 6, the second stage begins at t=t ₁ , the first switch S ₁ and the second switch S ₂ are both in a conductive state, and all diodes are reverse biased and not conductive. The input voltage V _in spans the magnetizing inductors L _m1 and L _m2 and the leakage inductors L _k1 and L _k2 . The leakage inductor currents i _Lk1 and i _Lk2 rise linearly. From an energy point of view, the input voltage source has a positive impact on the two magnetizing inductors L _m1 , L _m2 stores energy. The first output capacitor C _o1 , the second output capacitor C _o2 , and the third output capacitor C _o3 connected in series provide energy to the load.

The third stage ( ):

Referring to Figure 4 and Figure 7, the first switch S ₁ is turned on, while the second switch S ₂ is switched to non-conducting, and the leakage inductor current The parasitic output capacitance C _s2 of the second switch S ₂ is charged. Because the parasitic capacitance value is very small, the parasitic capacitance voltage v _ds2 rises rapidly from 0. When t=t ₃ , when the parasitic capacitance voltage rises to the first voltage V _Co1 , the second clamping diode D ₂ transitions to conduction, and the third stage ends.

The fourth stage ( ):

Referring to Figure 4 and _{Figure 8, the fourth stage begins at t=t3, the second clamping diode D2 transitions to conduction, and the cross-voltage vds2 of the} second switch S2 is clamped at v _Co1 . The energy of the leakage inductor is transferred to the first output capacitor C _o1 , the leakage inductor current i _Lk2 in _the fourth stage decreases, and the stored energy of the magnetizing inductor L _m2 is transferred to the second winding P ₂ and the third winding P through the first winding P 1 ₃ . The second winding P ₂ causes the diode D ₁₂ to turn on, and the current i _D12 charges the capacitor C ₁₂ and simultaneously discharges the first regenerative capacitor C ₁₁ . In the third winding P ₃ , the second regenerative diode D ₂₁ and the second output diode D _o2 are turned on, the current i _D21 charges the second regenerative capacitor C ₂₁ , and the current i _Do2 charges the second piezoelectric Capacitor C ₂₂ is discharged. When the leakage inductance current i _Lk2 drops to 0 and the second clamping diode D ₂ transitions to non-conducting state, the fourth stage ends.

The fifth stage ( ):

Referring to Figure 4 and Figure 9, the fifth stage begins at t=t4. At this time, the energy stored in the leakage inductor L _k2 is released, and the second clamping diode D ₂ transitions to non-conducting state. The magnetizing inductor current i _Lm2 is completely reflected from the first winding P ₁ to the second winding P ₂ and the third winding P ₃ . The circuit operation of the voltage multiplication module composed of the second winding P ₂ and the third winding P ₃ is the same as that in the fourth stage. The current flowing through the first switch S ₁ at this stage is equal to the sum of the currents of the magnetizing inductors L _m1 and L _m2 , that is, .

The sixth stage ( ):

Referring to Figure 4 and Figure 10, the sixth stage begins at t=t ₅ , the second switch S ₂ is switched on, and the first switch S ₁ remains on. Due to the existence of the leakage inductance L _k2 , the second switch S ₂ It has zero current switching (ZCS) soft cutting performance for conduction. The leakage inductor current i _Lk2 rises rapidly. When i _Lk2 < i _Lm2 , the energy stored in the magnetizing inductor L _m2 is still transmitted to the second winding P ₂ and the third winding P ₃ through the first winding P ₁ . The first voltage multiplier diode D ₁₂ , the second regenerative diode D ₂₁ and the second output diode D _o2 are turned on, the currents i _D12 , i _D21 and i _DO2 decrease, and the leakage inductances L _k1 and L _k2 are controlled. The decreasing rate of the current of the first voltage multiplier diode D ₁₂ , the second regenerative diode D ₂₁ and the second output diode D _o2 is thus alleviated, thereby alleviating the reverse recovery problem of these diodes. The remaining diodes are reverse biased and do not conduct. When the currents i _D12 , i _D21 and i _Do3 drop to 0, the first voltage doubling diode D ₁₂ , the second regenerative diode D ₂₁ and the second output diode D _o2 transition to zero current switching (ZCS). When it becomes non-conductive, the sixth stage ends.

The seventh stage ( ):

Referring to Figure 4 and Figure 11, the seventh stage begins at t=t ₆ , switches S ₁ and S ₂ are both turned on, and all diodes are reverse biased and not turned on. The input voltage V _in spans the magnetizing inductors L _m1 and L _m2 and the leakage inductors L _k1 and L _k2 . The leakage inductor currents i _Lk1 and i _Lk2 rise linearly. From an energy point of view, the input voltage source has a negative impact on the first coupling inductor 1. Magnetizing inductors L _m1 and L _m2 store energy. The first output capacitor C _o1 , the second output capacitor C _o2 , and the third output capacitor C _o3 are connected in series to provide energy to the load. When the first switch _S1 is switched non-conductive, the seventh phase ends.

The eighth stage ( ):

Referring to Figure 4 and Figure 12, the eighth stage begins at t=t ₇ , the first switch S ₁ is switched to non-conducting, and the leakage inductor current i _Lk1 charges the parasitic output capacitance C _s1 of the first switch S ₁ because the parasitic capacitance value It is very small, so the parasitic capacitance voltage v _ds1 rises rapidly from 0. When the parasitic capacitance voltage v _ds1 rises to the first voltage V _Co1 , the first clamping diode D ₁ transitions to conduction.

The ninth stage ( ):

Referring to Figure 4 and Figure 13, the ninth stage begins at t=t ₈ , the first clamping diode D ₁ transitions to conduction, and the cross-voltage v _ds1 of the first switch S ₁ is clamped at v _Co1 . The energy of the leakage inductor is transferred to the output side, the leakage inductor current i _Lk1 decreases in the ninth stage, and the stored energy of _the magnetizing inductor L _m1 is transferred to the second winding N ₂ and the third winding N ₃ through the first winding N 1 . In the second coupling inductor 2, the first regenerative diode D ₁₁ and the first output diode D _o1 are turned on, the current i _D11 charges the first regenerative capacitor C ₁₁ , and the current i _Do1 discharges the capacitor C ₁₂ . In the third coupling inductor 3, the diode D ₂₂ is turned on, and the current i _D22 charges the second voltage doubling capacitor C ₂₂ and simultaneously discharges the second regenerative capacitor C ₂₁ . When the leakage inductance current i _Lk1 drops to 0 and the first clamping diode D ₁ transitions to non-conducting state, the ninth stage ends.

The tenth stage ( ):

Referring to Figure 4 and Figure 13, the ninth stage begins at t= _t9 . At this time, the energy of the leakage inductance is completely released, and the first clamping diode _D1 transitions to non-conducting state. The magnetizing inductor current i _Lm1 is completely reflected from the first winding N ₁ to the second winding N ₂ and the third winding N ₃ . The circuit operation of the voltage multiplication module composed of the second winding N ₂ and the third winding N ₃ is the same as that in the ninth stage. At this time, the current of the second switch S ₂ is equal to the sum of the currents of the magnetizing inductors L _m1 and L _m2 , that is, . When the first switch _S1 is switched on, this phase ends and the next switching cycle is entered.

＜Voltage Gain Analysis＞

The voltage of the output capacitor C _o1 can be regarded as the output voltage of a traditional boost converter. According to the magnetizing inductor L _m1 , it satisfies the principle of volt-second balance, so the output voltage V _Co1 can be deduced as

In the fourth stage, the magnetizing inductor voltages of the coupled inductor are respectively ; ,

Therefore, the voltage of the second regenerative capacitor C ₂₁ can be obtained .

In addition, in the fourth stage, Kirchhoff’s voltage law (KVL) can be used to obtain ; .

In the ninth stage, the magnetizing inductor voltages of the coupled inductor are respectively

;

Therefore, the voltage of the first regenerative capacitor C ₁₁ can be obtained

In addition, in the ninth stage, the voltage doubling module composed of the second winding N ₂ and the third winding N ₃ can be obtained by using KVL

;

The voltage gain that can be estimated is

As shown in Figure 15, the relationship curve between voltage gain and conduction ratio under different coupling coefficients. When the coupling inductor turns ratio , this converter has three different coupling coefficients , , and In this case, the relationship curve between voltage gain and conduction ratio shows that the coupling coefficient has little effect on voltage gain. If the leakage inductance of the coupling inductor is ignored, that is , the ideal voltage gain M of the converter is as follows:

(When the coupling coefficient )

From the formula of ideal voltage gain M, we can know that the voltage gain of this converter has two degrees of design freedom: coupled inductor turns ratio and conduction ratio . By adjusting the coupled inductor turns ratio of the converter , can achieve high voltage boost without the converter needing to operate at a maximum conduction ratio.

As shown in Figure 16, when the coupling coefficient k=1, the voltage gain and conduction ratio and coupled inductor turns ratio From the graph, it can be seen that: when and When , the voltage gain is 17.5 times, when and When , the voltage gain is 32.5 times.

＜Switching voltage stress analysis＞

If the ripple voltage of the capacitor and the leakage inductance of the coupling inductor (coupling coefficient ), the switching element is considered ideal. It can be seen from the fourth and ninth stages that the first switch and second switch The voltage stresses are as follows:

From the fourth stage, it can be found that the voltage stress of the first clamping diode D ₁ , the first output diode D _o1 , the first regenerative diode D ₁₁ , and the second voltage doubling diode D ₂₂ can be expressed respectively. as follows: , ; ; .

From the ninth stage, it can be found that the voltage stress of the second clamping diode D ₂ , the first voltage doubling diode D ₁₂ , the second regenerative diode D ₂₁ , and the second output diode D _o2 can be expressed respectively. as follows: ; ; ; .

Since the voltage stress of the power switch and diode of the traditional interleaved boost converter is the output voltage V _o , from the above formula we can know: the voltage stress of the first switch S ₁ and the second switch S ₂ in this embodiment It is only 1/(6n+1) times of the output voltage _Vo , so MOSFETs with low rated voltage and low on-resistance can be used to reduce switching conduction losses. Moreover, the voltage stress of the diodes in this embodiment is much lower than the output voltage V _o , and they are diodes with low voltage stress. Power diodes with lower forward conduction voltage drop can be used to reduce conduction losses.

To sum up, the above embodiments have the following advantages:

1. A voltage doubling module composed of two coupling inductors, a doubling capacitor, and a doubling diode (among them, the second winding N ₂ and P ₂ and the first voltage doubling diode D ₁₂ and the first doubling capacitor C _12. The first regenerative diode D ₁₁ forms a voltage doubling module. The second windings N ₃ and P ₃ are connected with the second voltage doubling diode D ₂₂ , the second voltage doubling capacitor C ₂₂ and the second regenerative diode The body D ₂₁ forms another voltage multiplication module), so that the voltage gain has two degrees of design freedom: the coupled inductor turns ratio and the switch conduction ratio. Therefore, to achieve high voltage gain, it is not necessary to operate at a huge conduction ratio. 2. Due to the low voltage stress of the first switch S ₁ and the second switch S ₂ (much lower than the output voltage), low-rated voltage MOSFETs with smaller on-resistance can be used to reduce the conduction loss. The switch can conduct zero current switching (ZCS) to reduce switching losses. 3. Due to the low voltage stress of the diode (much lower than the output voltage), a diode with a smaller forward conduction voltage drop can be used to reduce the conduction loss. Leakage inductance alleviates the reverse recovery problem of the diode and improves the reverse recovery loss of the diode. 4. The leakage inductance energy of the coupling inductor can be recovered and transmitted to the output side, which not only avoids voltage surges on the switch, but also improves efficiency. 5. The parallel input structure of the converter reduces component current stress, and the two power switches (the first switch S ₁ and the second switch S ₂ ) adopt interleaved operation to reduce the input current ripple.

However, the above are only examples of the present invention and should not be used to limit the scope of the present invention. All simple equivalent changes and modifications made based on the patent scope of the present invention and the content of the patent specification are still within the scope of the present invention. within the scope covered by the patent of this invention.

1: First coupling inductor 2: Second coupling inductor N ₁ , P ₁ : First winding N ₂ , P ₂ : Second winding N ₃ , P ₃ : Third winding S ₁ : First switch S ₂ : Second Switch D ₁ : _first clamping diode D ₂ : second clamping diode D ₁₁ : first regeneration diode C 11: first regeneration capacitor D ₁₂ : first voltage doubler diode C ₁₂ : First voltage doubling capacitor D ₂₁ : Second regenerative diode C ₂₁ : Second regenerative capacitor D ₂₂ : Second voltage doubling diode C ₂₂ : Second voltage doubling capacitor O1: First output stage C _o1 : First output capacitor O2: Second output stage D _o1 : First output diode C _o2 : Second output capacitor O3: Third output stage D _o2 : Second output diode C _o3 : Third output capacitor 4: Control unit V _in : input voltage V _o : output voltage L _m1 : magnetizing inductance L _m2 : magnetizing inductance v _gs1 : voltage of the first pulse signal v _gs2 : voltage of the second pulse signal v _ds1 : voltage of the first switch Terminal voltage v _ds2 : voltage across the second switch i _Lk1 : leakage inductance current i _Lk2 : leakage inductance current i _D1 : current flowing through the first clamping diode i _D2 : flowing through the second clamping diode Current i of the pole body _Do1 : Current i flowing through the first output diode _D11 : Current i flowing through the first regenerative diode _D21 : Current i flowing through the second regenerative diode _D12 : Current i flowing through the first regenerative diode Current i of the voltage doubling diode _D22 : Current i flowing through the second voltage doubling diode _Do2 : Current flowing through the second output diode

Other features and effects of the present invention will be clearly presented in the embodiments with reference to the drawings, in which: Figure 1 is a circuit diagram of a conventional boost converter; Figure 2 is a circuit diagram of an embodiment of the high voltage gain DC converter of the present invention; Figure 3 is an equivalent circuit diagram of this embodiment; Figure 4 is an operation timing diagram of this embodiment; Figure 5 is a circuit diagram of the first stage of operation of this embodiment; Figure 6 is a circuit diagram of this embodiment operating in the second stage; Figure 7 is a circuit diagram of this embodiment operating in the third stage; Figure 8 is a circuit diagram of this embodiment operating in the fourth stage; Figure 9 is a circuit diagram of this embodiment operating in the fifth stage; Figure 10 is a circuit diagram of this embodiment operating in the sixth stage; Figure 11 is a circuit diagram of this embodiment operating in the seventh stage; Figure 12 is a circuit diagram of this embodiment operating in the eighth stage; Figure 13 is a circuit diagram of this embodiment operating in the ninth stage; Figure 14 is a circuit diagram of the embodiment operating in the tenth stage; Figure 15 is a graph showing the relationship between different coupling coefficients and voltage gains in this embodiment; and FIG. 16 is a voltage gain curve diagram of the coupled inductor turns ratio and conduction ratio in this embodiment.

1: First coupling inductor

2: Second coupling inductor

N ₁ , P ₁ : first winding

N ₂ , P ₂ : Second winding

N ₃ , P ₃ : third winding

S ₁ : first switch

S ₂ : Second switch

D ₁ : First clamping diode

D ₂ : Second clamping diode

D ₁₁ : First regenerative diode

C ₁₁ : First regenerative capacitor

D ₁₂ : first voltage doubling diode

C ₁₂ : First voltage multiplier capacitor

D ₂₁ : Second regenerative diode

C ₂₁ : Second regenerative capacitor

D ₂₂ : Second voltage doubling diode

C ₂₂ : Second voltage doubling capacitor

O1: first output stage

C _o1 : first output capacitor

O2: second output stage

D _o1 : first output diode

C _o2 : second output capacitor

O3: The third output stage

D _o2 : Second output diode

C _o3 : The third output capacitor

4:Control unit

V _in : input voltage

V _o :Output voltage

Claims (10)

A high voltage gain DC converter containing: A first coupled inductor and a second coupled inductor, each coupled inductor has a first winding, a second winding and a third winding, each winding has a first end and a second end, wherein the third winding The first end of the first winding of a coupled inductor is electrically connected to the first end of the first winding of the second coupled inductor to receive a DC input voltage, and the second end of the second winding of the first coupled inductor is electrically connected together. The second end of the second winding of the second coupled inductor is electrically connected, and the second end of the third winding of the first coupled inductor is electrically connected to the second end of the third winding of the second coupled inductor, A first switch having a first end electrically connected to the second end of the first winding of the first coupled inductor and a second end grounded, and the first switch is controlled to switch between a conductive state and a non-conductive state state, when the first switch is turned on, the first winding of the first coupling inductor receives a current for charging; a second switch having a first end electrically connected to the second end of the first winding of the second coupling inductor, and a second end grounded, and the second switch is controlled to switch between a conductive state and a non-conductive state between states; a first clamping diode having an anode electrically connected to the second end of the first winding of the first coupling inductor, and a cathode; A second clamping diode has an anode electrically connected to the second end of the first winding of the second coupling inductor, and a cathode, and the cathode of the first clamping diode is connected to the second clamping diode. The cathode of the pole body is electrically connected to a common contact; The first regenerative diode has an anode electrically connected to the first end of the second winding of the second coupling inductor, and a cathode; a first regenerative capacitor electrically connected between the first end of the second winding of the first coupling inductor and the cathode of the first regenerative diode; A first voltage multiplier diode having an anode electrically connected to the cathode of the first regenerative diode, and a cathode; a first voltage doubling capacitor, electrically connected between the first end of the second winding of the second coupling inductor and the cathode of the first voltage doubling diode; a second regenerative diode having an anode electrically connected to the first end of the third winding of the first coupling inductor, and a cathode; a second regenerative capacitor, electrically connected between the first end of the third winding of the second coupling inductor and the cathode of the second regenerative diode; a second voltage doubling diode having an anode electrically connected to the cathode of the second regenerative diode, and a cathode; a second voltage doubling capacitor, electrically connected between the first end of the third winding of the first coupling inductor and the cathode of the second voltage doubling diode; a first output stage electrically connected to the common contact for generating a first DC voltage based on the discharge from the first winding; A second output stage, electrically connected between the cathode of the first voltage doubling diode and the common contact, for generating a DC voltage based on the discharge from the second winding and the first voltage doubling capacitor. second voltage; a third output stage, electrically connected to the cathode of the second voltage multiplier diode, for generating a third DC voltage based on the discharge from the third winding and the second voltage multiplier capacitor; The first output stage is also stacked with the second output stage and the third output stage to generate an output voltage that is proportional to the sum of the first voltage, the second voltage, and the third voltage. The high voltage gain DC converter as claimed in claim 1, wherein the first output stage includes: A first output capacitor is electrically connected between the common contact and ground to provide the first voltage. The high voltage gain DC converter as claimed in claim 2, wherein the second output stage includes: a first output diode having an anode electrically connected to one end of the first voltage doubling capacitor, and a cathode; and A second output capacitor is electrically connected between the cathode of the first output diode and the first output capacitor for providing the second voltage. The high voltage gain DC converter as claimed in claim 3, wherein the third output stage includes: a second output diode having an anode electrically connected to one end of the second voltage doubling capacitor, and a cathode; and A third output capacitor is electrically connected between the cathode of the second output diode and the second output capacitor to provide the third voltage, wherein the first output capacitor and the second output capacitor are connected to the second output capacitor. The third output capacitors are stacked together. The high voltage gain DC converter as claimed in claim 1, wherein the first end of each first winding is a dotted end, and the second end of each first winding is a non-dotted end. The high voltage gain DC converter as claimed in claim 1, wherein the first end of each second winding is a dotted end, and the second end of each second winding is a non-dotted end. The high voltage gain DC converter as claimed in claim 1, wherein the first switch is an N-type power semiconductor transistor, and the first terminal of the first switch is a drain, and the second terminal of the first switch is a drain. It is the source. The high voltage gain DC converter of claim 1, wherein the second switch is an N-type power semiconductor transistor, and the first terminal of the second switch is a drain, and the second terminal of the second switch is a drain. It is the source. The high voltage gain DC converter as claimed in claim 1, further comprising a control unit that generates a first pulse modulation signal for switching the first switch and a second pulse wave for switching the second switch. Modulation signal, the first pulse wave modulation signal and the second pulse wave modulation signal have the same cycle time. The high voltage gain DC converter as claimed in claim 9, wherein part of the cycle time of the first and second pulse modulation signals overlap.

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