RU2851698C1 - Tunable cascode self-contained generator of harmonics - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: invention relates to radio engineering and can be used for synthesis of frequencies in electronic warfare systems of microwave range, in particular, this device relates to tunable transistor self-maintained generators of harmonics, which simultaneously generate approximately equal power oscillations of fundamental frequency and frequencies of selected harmonics. In the tunable cascode self-contained generator of harmonics with an active element on two transistors described by the system of wave [S]-parameters connected according to the “common emitter-common base” scheme, inductance (115) is introduced, which on one side is connected to common bus, and on other side is to resistor (106) and capacitor (122) free terminals, as well as to varicap anode (125), which cathode is connected to capacitor (110) and inductance (118) common point, which in turn are respectively connected between transistor (100) base and varicap (124) cathode. Cathode of varicap (123) is connected to common point of capacitor (108) and section of microstrip line (119), the other end of which is connected to capacitor (109). Decoupling elements for supplying control voltages to varicaps with capacitances C_V are made in the form of three L-shaped low-pass filters L_CC_C on inductances L_C (112), (113) and (114), as well as on capacitance C_C (116), parallel to power supply (129), and their values satisfy inequality

where and ƒ_C is the characteristic frequency of the L_CC_C chain.

EFFECT: increase of operating frequencies of harmonics at expansion of their tuning bands.

1 cl, 16 dwg

Description

The proposed device pertains to radio engineering and can be used, for example, to synthesize frequencies in microwave electronic warfare systems. Specifically, this device is a tunable transistor harmonic oscillator (THO), which simultaneously generates approximately equal-power oscillations of the fundamental frequency and the frequencies of the extracted harmonics.

A tunable microwave oscillator is known (see Fig. 3 in the article by Kyung-Whan Yeom An investigation of the high-frequency limit of a miniaturized commercial voltage-controlled oscillator used in a 900-MHz-band mobile-communication handset / KW Yeom // IEEE Transactions on Microwave Theory and Techniques. - 1998. - Vol. 46. - No. 8 (August). - P. 1165 - 1168). The device (see Fig. 1) consists of two transistors 1 and 2, which are connected according to the "common collector - common emitter" (CC-CE) circuit for a microwave signal, and according to the DC signal, using resistors 3 - 6, they are set to the same operating modes as in the "common emitter - common base" (CE-CB) cascode circuit. As a result, this analogue has a good frequency tuning range with satisfactory decoupling between its output load resistance and that of resonator 7 in the frequency-setting circuit on elements 7-14. To decouple the microwave and DC circuits, this device uses microstrip lines (MSL) 14 and 15 together with blocking capacitors 16 and 17. Elements 18 and 19 form a matching circuit between the output resistance of the oscillator and the resistance of the standard path. The input terminals of the supply and control voltages are designated by the numbers 20 and 21, respectively. This device can be considered a harmonic oscillator only conditionally, since the harmonic levels here are determined only by the nonlinear operating modes of the transistors and are usually distributed among themselves inversely proportional to the value of k ² , where k is the harmonic number [1]. Based on an analysis of the equivalent transistor circuit for this analog, it was established that, in addition to its elements, resistive element 6, which sets the overall oscillator current, significantly influences the practical maximum oscillation frequency. The value of resistor 6 has a particularly strong influence on frequency when its values are small. To minimize this effect, microwave transistors with higher cutoff frequencies should be selected for the oscillator. To increase the maximum oscillation frequency, the value of element 6 should be increased [2]. Thus, in this device, an increase in operating frequencies is achieved either by selecting transistors with higher cutoff frequencies or by increasing the resistor that sets the current operating mode, which is not always feasible.

Since in this analog the increase in the operating frequencies of generation is limited by the choice of the cutoff frequencies of the transistors and not always feasible operating modes, the disadvantage of the autogenerator is the relatively low operating frequencies with inefficient generation of harmonics and expansion of their tuning bands.

A tunable cascode microwave oscillator is known (see Baranov, A.V. Voltage-controlled cascode oscillator // Russian Federation Patent for Invention No. 2644067 with priority from 04.05.2017, IPC - 2006.01 H03B 5/12, Published 07.02.2018, Bulletin No. 4). The device (see Fig. 2) consists of two transistors 22 and 23, which are connected according to a cascode circuit of the OE-OB type. This harmonic oscillator also has good values of generation frequency tuning bands and decoupling between its output load resistances and resonant circuits. The second analog includes five resistors 24-28, which ensure the transistors operate on direct current. Capacitor 29 serves as a blocking capacitor, capacitors 30-33 serve as decoupling capacitors, and elements 34-36 serve as power supply decoupling elements for the high-frequency circuits. The resistances of elements 34-36 are either inductive or purely active, and their values should be significantly greater than the output load resistance. Furthermore, the device in Fig. 2 contains inductors 37 and 38, capacitors 39 and 40, and four varicaps 41-44, all of which are the primary frequency-setting elements. The harmonic oscillator output is designated by 45, and the supply and control voltage input terminals are designated by 46 and 47, respectively.

This device is a system of two mutually synchronized oscillators operating on a single load. Even in the case of using identical three-point circuits in this analogue, the cascode implementation of direct coupling between the transistors allows the characteristic points of the two three-point circuits to be separated from each other, eliminating the mutual influence of the oscillators created on their basis. In the first and second oscillators (on the generalized active element 48 or 49), identical (capacitive type) three-point circuits are used: triangular a) and star-shaped b), respectively (see Fig. 3, where the characteristic points are designated by the letters a, b and c) [1]. The three-point circuit of the first self-oscillator corresponds to the equivalent circuit in Fig. 3 a), if the capacitance C _OC is designated as the capacitance of element 40, and the value of capacitance C _AE of element 50 is understood to be the equivalent capacitance of the series circuit containing the inductance of element 38 and the capacitances of varicaps 42 and 44, in addition, the inductance L _KK of element 51 is considered to be the equivalent inductance of the series circuit formed by the capacitances of varicaps 41, 43 and the inductance of element 37. At the same time, the three-point circuit of the second self-oscillator corresponds to the equivalent circuit in Fig. 3 b), if the capacitance of capacitor 39 is designated as C _B , and the capacitance C _K of element 52 is understood to be the equivalent capacitance of the circuit containing the inductance of element 38 and the capacitances of varicaps 41 and 42, at the same time, the inductance L _E of element 53 is considered to be the equivalent inductance of the circuit formed by the inductance of element 37 and capacitances of capacitor 40 and varicaps 43, 44.

When selecting the nominal values of frequency-setting elements in accordance with the formula:

Two interconnected oscillators operate in the analog under consideration at the same frequency ƒ ₀ and on the same common load - resistive element 24. In this case, the same active element, formed by two transistors 22 and 23, is used in both oscillators. It should be noted that, with the exception of one element - capacitor 39, the remaining frequency-setting elements are used simultaneously in both oscillators. If the oscillators operate at high frequencies, then in order to comply with relation (1), it is necessary to fulfill additional conditions under which the connection points of varicaps 41-44 to the housing must be located as close to each other as possible. Otherwise, the design inductances that occur between the specified points must be taken into account when calculating the values of inductive elements 37, 38. The most accurate compliance with (1) is achieved using a second power source through additional adjustment of the voltages on the varicaps. The resulting frequency adjustment range, where condition (1) is met, as a rule, turns out to be smaller compared to the overall frequency tuning range of the oscillator. By satisfying relation (1) in this analogue, a technical effect is achieved - a reduction in the spectral power density of the phase fluctuations of the cascode-type OE-OB generator devices. However, the fulfillment of condition (1) does not guarantee high power levels of the emitted harmonics, which are determined, on the one hand, by the nonlinear operating mode of the generalized active element, and on the other hand, by the use of a low-pass filter on elements 38, 41 and 42. In the analogue under consideration, an increase in the operating frequencies of generation can be achieved either by selecting transistors with higher cutoff frequencies, as in the first analogue, or by selecting varicaps that provide frequency tuning in a higher-frequency region or in a wider band, as in work [3].

The disadvantage of the second analogue is the impossibility of achieving (especially using a standard element base) increased operating frequencies of harmonic generators with their efficient generation and extended frequency tuning ranges.

The closest technical solution to the proposed one is a tunable cascode microwave harmonic oscillator (see Baranov, A.V. Tunable cascode harmonic oscillator // Russian Federation Patent for Invention No. 2774408 with priority dated September 14, 2021, IPC - 2006.01 H03B 5/12, Published June 21, 2022, Bulletin No. 18). The device (see Fig. 4) consists of two transistors 54 and 55 connected in a cascode OE-OB circuit, five resistors 56 - 60, which ensure their operation mode by direct current. Capacitor 61 is a blocking capacitor, capacitors 62 - 65 act as isolating elements, and elements 66 - 68 are decoupling circuits for the RF power supply. The resistances of elements 66-68 are either inductive or purely active, and their values must be significantly greater than the output load resistance. Furthermore, the prototype contains frequency-setting elements, the main ones being inductors 69 and 70, MPL sections 71 and 72, capacitors 73 and 74, and four varicaps 75-78. The device's 50-ohm output is designated by the number 79, and the corresponding supply and control voltage input terminals are designated by the numbers 80 and 81.

Unlike the second analog, where the mutual synchronization of two oscillators results in the creation of an oscillation of only the fundamental frequency, the prototype is a system of two mutually synchronized harmonic oscillators 82 and 83 operating on a single load 84 (see Fig. 5a). One harmonic oscillator is configured to generate oscillations of the fundamental frequency and its even harmonics, while the second harmonic oscillator is configured to generate oscillations of the fundamental frequency and its odd harmonics. Each of the harmonic oscillators can be represented as a model in Fig. 5b). For the first AGG 82, we assume that the amplifying element 85 with the gain G(jω) operates simultaneously on two circuits 86 and 87 with gains H ₁ (j(jω) and H _2k (jω). In this case, one circuit is configured to generate an oscillation of the fundamental frequency w ₀ =2πƒ ₀ , and the other - to generate its even (2k)-th harmonic. And for the second AGG 83, we assume that the amplifying element 85 with the gain G(jω) operates simultaneously on two circuits 86 and 87 with gains H ₁ (jω) and H _2k+1 (jω). Moreover, here one circuit is configured to generate an oscillation of the fundamental frequency ω ₀ , and the other - to generate its odd (2k+1)-th harmonic. For the prototype, in which the oscillations of the autogenerators 82 and 83 at the fundamental frequency are mutually synchronized, expressions are obtained linking input V _ex (ω) and output V ₀ (ω) voltages and confirming the possibility of simultaneous generation of oscillations with frequencies ω ₀ , 2kω ₀ and (2k+1)ω ₀ . Using these expressions, it is possible to obtain multi-frequency oscillations with approximately equal levels of their output powers. Such equality of levels can be achieved if the transfer coefficients of the active and passive circuits in Fig. 5 b), as well as the dynamic range of the output power of the selected transistor, differ little from each other at the fundamental frequency and at the (2k)-th and (2k+1)-th harmonics. For identical star-shaped capacitive circuits of two autogenerators (see Fig. 5 a)) in the prototype, the generation frequencies at the selected frequency and its harmonics were found from the condition [1]:

where ImZ _B1 , ImZ _K , ImZ _E1 and ImZ _B2 , ImZ _K and ImZ _E2 are the imaginary parts of the minimum total input resistances measured at the ports of the circuit in Fig. 6 or at the terminals of the base and emitter of the first transistor (Z _B1 ,Z _E1 ) and the base, emitter and collector of the second transistor (Z _B2 , Z _E2 , Z _K ). The technique for designing autogenerators [1] used for three-port models is, in fact, a modification of the techniques proposed for two-port (two-point) models: the Kurokawa oscillator model [4] or his improved model [5]. In three-port models, the active elements are usually described by a system of wave [S]-parameters, and their equivalent circuits can be considered particular rather than generalized, since they are valid not for any, but only for a selected microwave transistor with known [S]-parameters [1]. In such models, the elements of the transistor circuit, called "negatrons", are used, which have negative real parts of the conductivity Y _neg or resistance Z _neg (see Fig. 3 a) and b)), which are approximately equal in magnitude to similar characteristics of the corresponding load circuit. The total values of these quantities form each resistance included in the system (2). Moreover, the conductivities Y _neg are usually used in generator models based on triangular three-point circuits, and the resistances Z _neg - in the models of autogenerators, where star-shaped triangular circuits are realized [1]. For the three-point models of the AGG in Fig. 6, as the negatron resistances, we simultaneously use on ports _P1 - _P5 all possible input resistances of the transistor terminals that satisfy the relations: Z _B1 → 0, Z _E1 → 0, Z _B2 → 0, Z _E2 → 0, Z _K → 0. To implement these relationships in the circuit in Fig. 6, transistors 88 and 89 described by the [S]-parameters are used, as well as models of components 90 - 99, which correspond to the elements of the circuit in Fig. 4. The implementation of these relationships in practice leads to the same result as the fulfillment of the conditions obtained on the basis of the analysis of the model in Fig. 5 b). Taking into account the comments noted for system (2), in the prototype, the relationship between the moduli of the quantities ImZ _B1 , ImZ _K and ImZ _E1 with the moduli of the quantities ImZ _B2 , ImZ _K and ImZ _E2 at the fundamental frequency ω ₀ is established, as well as their relationship with each other at the frequencies of its even 2kω ₀ and odd (2k+1)ω ₀ harmonics at k=1, 2, … using the following relationships:

The prototype shows that in the AGG on two cascode-connected microwave transistors, which are described by wave [S]-parameters, the fulfillment of the system of relations (3) is a necessary condition for the implementation in each of the interconnected harmonic oscillators of the system in Fig. 5 a) of the same type (capacitive) star-shaped three-point circuits, such as in Fig. 3 b). Moreover, in comparison with known harmonic oscillators, the effective generation of all harmonics takes place without the obligatory adjustment of various (unloaded, loaded, over-limit and others) circuits at their outputs. In addition, close power levels of oscillations of the fundamental frequency and its harmonics are ensured without the use of dividers and multipliers of the oscillation frequency with subsequent amplification of their power [1]. In comparison with known harmonic generators, the prototype, as well as the second analogue, additionally has a useful property - a 3 dB lower level of spectral power density of phase noise. Unfortunately, with effective harmonic generation in the prototype, an increase in the operating frequencies of its harmonics is achieved either by selecting transistors with higher cutoff frequency values, as in the first analog, or by selecting varicaps that perform frequency tuning in a higher frequency region or in a wider band, as in [3].

The technical effect that the proposed solution is aimed at achieving is an increase in the operating frequencies of harmonics while expanding their tuning bands.

This effect is achieved by the fact that in a tunable cascode harmonic oscillator containing an active element in which transistors 100 and 101 are connected according to the “common emitter - common base” scheme, where the collector of transistor 100 is connected to the emitter of transistor 101 through a section of microstrip transmission line 120, resistors 102, 103, 104 and 105 are connected in series between the positive terminal of power source 128 and its negative terminal, which is a common bus, in which the common point of resistor 102 and the positive terminal is connected to the common bus through capacitor 107, the common point of resistors 102 and 103 is connected to the collector of transistor 101 and simultaneously to capacitor 108, which in turn is connected with the other end to a section of microstrip line 119, the common point of resistors 103 and 104 is connected to the base transistor 101 and through capacitor 121 to the common bus, the common point of resistors 104 and 105 is connected to the base of transistor 100, the emitter of transistor 100 is connected on one side through capacitor 111 to the cathode of varicap 126, on the other side - to the common connection point of resistor 106 and capacitor 122, in addition, the device contains a series-connected anode of varicap 123 and inductance 117, the other end of which is connected to the common bus, a series-connected capacitor 110 and inductance 118, a separate varicap 125, as well as three decoupling elements, the common terminal of which is connected to the positive terminal of control voltage source 129, the negative terminal of which is the common bus, the free terminals of two decoupling elements are connected to the cathodes of varicaps 124, 126, their anodes - to the common bus, the free terminal of the third decoupling element connected through a capacitor 109 to the output of device 127, among other things, the minimum total resistances Z _B1 →0, Z _E1 →0, Z _B2 →0, Z _E2 →0, Z _K →0, calculated at the corresponding points of the collector Zk of transistor 101, as well as the bases Z _B1 , Z _B2 and emitters Z _E1 , Z _E2 of transistors 100 and 101, which are described by a system of wave [S]-parameters, are related by the relations:

where k is an integer, ImZ _B1 (ω), ImZ _B2 (ω), ImZ _E1 (ω), ImZ _E2 (ω) and ImZ _K (ω) are the imaginary parts of the resistances Z _B1 , Z _B2 , Z _E1 , Z _E2 and Z _K of the elements of star-shaped capacitive three-point equivalent circuits, which are formed at a cyclic frequency co equal to either the fundamental frequency ω ₀ or even 2kω ₀ or odd (2k+1)ω ₀ frequencies, according to the invention, an inductance 115 is introduced, which is connected on one side to the common bus, and on the other - to the free terminals of the resistor 106 and the capacitor 122, as well as to the anode of the varicap 125, the cathode of which is connected to the common point of the capacitor 110 and the inductance 118, which in turn are connected respectively between the base of the transistor 100 and the cathode of the varicap 124, in addition, the cathode of the varicap 123 is connected to the common point of the capacitor 108 and the section of the microstrip line 119, the other end of which is connected to the capacitor 109, among other things, the decoupling elements for supplying control voltages to the varicaps with capacitances C _B are made in the form of three L-shaped L _C C _C low-pass filters on inductances L _C 112, 113 and 114, as well as on capacitance C _C 116, parallel to the power source 129, and their values satisfy the inequality:

Where a ƒ _Ц - characteristic frequency of the L _Ц С _Ц -circuit.

The schematic diagram of the proposed device is shown in Fig. 7. The tunable harmonic oscillator consists of two transistors 100 and 101 connected in a cascode configuration of the OE-OB type, and five resistors 102-106, which ensure their operation in direct current mode. Capacitor 107 serves as a blocking capacitor, capacitors 108-111 serve as separating capacitors, and elements 112-116 serve to decouple the microwave paths and control circuits. When the relationships established below are met, elements 112-114 and 116 also contribute to an increase in the fundamental frequency and the frequencies of its harmonics of the output oscillation. In addition, the device in Fig. 7 contains frequency-setting elements: inductors 117 and 118, MPL sections 119 and 120, capacitors 121 and 122, and also four varicaps 123 - 126. The 50-Ohm output of the device is designated by the number 127, and the corresponding terminals for inputting the supply and control voltages are designated by the numbers 128 and 129.

The proposed device operates as follows. As a harmonic oscillator, this device differs little from the prototype. The equivalent circuits considered for the prototype and the second analog (Fig. 3b), Fig. 5a), and Fig. 5b) and the similar model (Fig. 6) in Fig. 8 are also suitable for describing the operation of this device. Below, we analyze the frequency properties of the proposed harmonic oscillator.

It is known [1] that for the equivalent three-point circuit of the autogenerator in Fig. 3 b), its fundamental oscillation generation frequency ƒ ₀ can be estimated as follows:

where L _E , C _K , C _B are the inductance and capacitance of the equivalent circuits in the corresponding emitter, collector, and base circuits in the AGG circuit in Fig. 7 or its model in Fig. 8. For multiples of the fundamental frequency, a similar expression can be written, which includes the same elements whose values were measured at the harmonics. From equation (4), it follows that in order to increase the generation frequencies in an AGG, it is necessary to decrease the values of the elements of the equivalent circuits.

In the proposed device, we will evaluate the influence of the varicap control circuits on the operating frequencies of the fundamental oscillation and its harmonics, and also evaluate the possibilities of expanding the tuning bands of the generated frequencies with their help. For this purpose, in Fig. 9 a) we will consider a circuit model, where a varicap is represented in the form of a variable capacitor C _B , to which an L-shaped low-pass filter on L _C C _C elements is connected to supply the blocking voltage. The total resistance We write the varicap as follows:

Where

In equations (6) and (7) β=С _Ц /С _В , and ƒ _Ц is the characteristic frequency of the L _Ц С _Ц -circuit.

The sign of the reactivity in expression (5) does not change only in those cases when the expression in brackets in the denominator of equation (6) is greater than zero, or when the entire denominator of equation (6) is greater than zero. The first case corresponds to the operation of the autogenerator in a relatively low-frequency region, when the second is its operation in the region of higher frequencies, at which the inequality is satisfied:

When the autogenerator operates in the region of higher frequencies, which are of greater interest (or when condition (8) is satisfied), the values of α are large, the total resistances - also, and this is equivalent to smaller values of the total capacitance of the varicap. That is, for large values of β, which correspond to the operation of varicaps at high frequencies, when their capacitances are minimal, it is possible to decrease the values of С _К and С _Б in expression (4). Obviously, in tunable oscillators, β cannot be increased simply by increasing the values of С _Ц . This will lead to a decrease in the characteristic frequency and, consequently, to a narrowing of the oscillator's frequency range in accordance with inequality (8). Moreover, in a tunable oscillator, an increase in the values of С _Ц will worsen its frequency switching time.

We will establish relationships similar to equations (5) - (8) for the model of the L-shaped L _C C _C -circuit in Fig. 9 b), where the inductance L _B is used as the variable value using the varicap. The total input impedance of this inductance

has the form:

Where

Here γ=L _B /L _C , and the characteristic frequency ƒ _C of this circuit is calculated using formula (7).

The sign of the reactivity in expression (9) does not change only in those cases when the expression in brackets in the denominator of equation (10) is greater than zero, or when the entire denominator of equation (10) is greater than zero. The first case corresponds to the operation of the autogenerator in the region when the second is its operation in the frequency range where the inequality is satisfied Usually, this inequality is satisfied automatically if we select an L _C C _C -circuit with parameters for which relation (8) is valid. Thus, for the circuit considered in Fig. 9 b), the characteristic frequency is not a limiter for the operation of the oscillator in the high-frequency region. At high frequencies, the value of δ becomes slightly less than unity, and the total resistance also decreases slightly in relation to the resistance of the original inductance L _B . The corresponding decrease in the total inductance leads to some decrease in the value of L _E in formula (4) and, consequently, to an insignificant (compared to the first case) effect of increasing the frequency ƒ ₀ .

Thus, in contrast to known passive circuits [6], the implementation of decoupling elements proposed in the harmonic oscillator can be considered as "forcing" L _C S _C - links or as L _C S _{C -} circuits that prolong the operation of varicaps with minimal capacitances at the "tails" of their control characteristics. Then it becomes possible (albeit to varying degrees) to reduce the values of C _K , C _B and L _E in expression (4), and, consequently, to increase the fundamental frequency and its harmonics.

An example of a specific device implementation. Let us consider a tunable harmonic oscillator, which simultaneously generates oscillations at the fundamental frequency, and at the second, third, fourth, and fifth harmonics in the vicinity of frequencies from 1 to 6 GHz. In accordance with the AGG circuit in Fig. 7 and its model in Fig. 8, this oscillator is implemented on an FR-4 glass-textolite substrate in a VK377 package, which is standard for some foreign-made voltage-controlled oscillators (VCOs) (see Fig. 10). The device prototype is implemented using two silicon bipolar transistors 100 and 101 of the 2T682A-2 type, which have oscillation gain factors guaranteed by technical specifications, both at the fundamental frequency and at its selected harmonics. Thus, when used in a common-emitter circuit, its cutoff frequency is ~6 GHz, and in a common-base circuit, the transistor operates at up to 9 GHz. Using the recommendations of the book [7], the methodology for designing a harmonic oscillator proposed in work [1], and the wave [S]-parameters of the transistor given in the technical specifications, we will determine the values of the circuit elements in Fig. 8 that correspond to the minimum values of the total resistances Z _B1 → 0, Z _E1 → 0, Z _B2 → 0, Z _E2 → 0 and Z _K → 0. We will perform such a calculation in each of the five ports simultaneously at all selected frequencies. Parameters of the circuit elements close to the optimal ones are given in the harmonic oscillator model in Fig. 8. Varicaps 2V174A9 and 2V174G9 were selected as elements 123 and 124, and varicaps BB555 were selected as elements 125 and 126. Their operation in the selected modeling program [7] was simulated as follows. If the capacitance variation range of the 2V174A9 varicap is chosen as a conventional unit, then the coefficients 2 and 4 can be assigned to the ranges of the 2V174G9 and BB555 varicaps. Taking this into account, when changing one capacitance of the 2V174A9 varicap, the other varicap capacitances are automatically adjusted. The parameters of the circuit elements shown in Fig. 8 were obtained with a 2V174A9 varicap capacitance (element 123) of 0.7 pF. As a harmonic oscillator, the prototype device demonstrates satisfactory operation, despite the fact that, compared to the prototype, the process of generating oscillations of the fundamental frequency and its harmonics is slightly less efficient, since relations (3) are satisfied with somewhat larger deviations from the ideal values. This is confirmed by the output oscillation spectra obtained using the FSUP-26 spectrum analyzer (ROHDE&SCHWARZ), when the control voltages on the varicaps are equal to 0.5, 5, and 10 V (see Figs. 11, 12, and 13, respectively). In the presented spectra, starting from frequencies of ~3.5 GHz, a certain decrease in the levels of higher harmonics is observed, characteristic of the selected microwave transistors. At the same time, when varying the control voltage over the entire range from 0 to 28 V, the developed prototype generates oscillations of the fundamental frequency from 740 to 1510 MHz and its four harmonics with a total power of 7 to 12 mW at a supply voltage of +12 V and a consumption current of ~17 mA. That is, the proposed device is tunable in an octave frequency band, while the prototype ensures fundamental frequency tuning with an overlap coefficient of no more than 1.6. When using the same set of varicaps in the prototype, the minimum frequency is 820 MHz, and its maximum frequency does not exceed 1.2 GHz. Among other things, the use of varicap 125 expands the tuning band down from 820 MHz to 740 MHz. In the model in Fig. 8, the control voltage supply circuits for the operation of varicaps 123 and 124 are selected in accordance with the recommendations that follow from formulas (5) - (8). Thus, the total capacitance of element 116 is 32 pF, and the inductance values of elements 112 and 113 are equal to 15 nH. Then, at a characteristic frequency of the L _C S _C circuit of ~ 230 MHz, its maximum value reaches ~ 1.57 GHz, if

This means that with varicapacitances of 123 and 124 ~0.7 pF and ~1.4 pF, respectively, generation at the fundamental frequency of ~1500 MHz is possible, the sign in expression (5) does not change, the total value of capacitance С _В /α decreases, which leads to a decrease in the values of С _К and С _Б in expression (4) and, consequently, to the desired increase in the value of ƒ ₀ . The reduction of the capacitances C _K and C _B of the equivalent circuits in the collector and base circuits can be demonstrated by comparing the frequency dependences for the imaginary parts of the input impedance, which are measured at the collector and base (see ports 1 and 3 in Fig. 8) for different values of inductive elements 112 and 113. Fig. 14 and Fig. 15, respectively, for the collector (port 1) and base (port 3) circuits show graphs of these dependences at 15 nH (curves 1) and 150 nH (curves 2). Obviously, with element inductances of 150 nH, the harmonic oscillator does not reach maximum frequencies, and the L _C C _C circuits operate as conventional decoupling elements. Therefore, at frequencies from 1.2 to 2.5 GHz, smaller values of the capacitances C _K and C _B correspond to L _C C _C circuits with inductances of 15 nH. A comparative analysis of the frequency graphs of the imaginary parts of the input impedance of the L _C C _C filter, which in the emitter circuit (port 4) is made in accordance with the model in Fig. 9 b), confirms the possibility of a slight increase in the frequency ƒ ₀ if the value of L _C is changed from 150 to 15 nH (see Fig. 16, curves 2 and 1, respectively).

Thus, the presented example of a specific implementation of a tunable harmonic oscillator using two transistors connected in a cascode configuration confirms the possibility of increasing the fundamental frequency and its harmonic frequencies with wider tuning bands. The use of "boosting" circuits as decoupling elements, which extend the life of varicaps with minimal capacitance at the "tails" of their control characteristics, is theoretically justified. Based on simulation and experimental confirmation, it has been shown that using such circuits, the prototype's frequency overlap ratio increases from 1.6 to 2, and the maximum frequency of its first harmonic increases from 1.2 to 1.5 GHz while maintaining the original set of higher harmonics.

Sources of information

1. Baranov, A.V. Transistor autogenerators of harmonic microwave oscillations / A.V. Baranov, M.A. Krevsky. - M .: Goryachaya Liniya - Telecom, 2021. - 276 p.

2. Grebennikov, A. RF and microwave transistor oscillator design / A. Grebennikov. -Chichester, England: John Wiley & Sons, Ltd, 2007. - 441 p.

3. Gorevoy, A.V. Generator of the 1 - 2 GHz range with an increased steepness of the control characteristic / A.V. Gorevoy // TUSUR Reports. - 2011. - No. 1 (23). - P. 44-49.

4. Kurokawa, K. Some basic characteristics of broadband negative resistance oscillator circuits / K. Kurokawa // The Bell System Technical Journal. - 1969, July-August. - P. 1937 - 1955.

5. Chang, C.-R. Computer-aided analysis of free-running microwave oscillators / C. -R. Chang, M.B. Steer, S. Martin, E. Reese // IEEE Trans. on Microwave Theory and Techniques. - 1991. -Vol.MTT-39, No. 10.-P. 1735-1745.

6. Baranov, A.V. Passive circuits with distributed and lumped parameters / A.V. Baranov, E.L. Priver. - M .: Goryachaya Liniya - Telecom, 2023. - 264 p.

7. Razevig V.D., Potapov Yu.V., Kurushin A.A. Design of microwave devices using Microwave Office. Edited by V.D. Razevig. - Moscow: Solon-Press, 2003. - 496 p.

Claims (6)

A tunable cascode harmonic oscillator comprising an active element in which transistors (100) and (101) are connected in a "common emitter - common base" configuration, where the collector of transistor (100) is connected to the emitter of transistor (101) through a section of microstrip transmission line (120), resistors (102), (103), (104) and (105) connected in series between the positive terminal of power supply (128) and its negative terminal which is a common bus, in which the common point of resistor (102) and the positive terminal is connected to the common bus through a capacitor (107), the common point of resistors (102) and (103) is connected to the collector of transistor (101) and simultaneously to capacitor (108), which in turn is connected at the other end to a section of microstrip line (119), the common point of resistors (103) and (104) is connected to the base of the transistor (101) and through the capacitor (121) to the common bus, the common point of the resistors (104) and (105) is connected to the base of the transistor (100), the emitter of the transistor (100) is connected on one side through the capacitor (111) to the cathode of the varicap (126), on the other side - to the common connection point of the resistor (106) and capacitor (122), in addition, the device contains a series-connected anode of the varicap (123) and an inductance (117), the other end of which is connected to the common bus, a series-connected capacitor (110) and inductance (118), a separate varicap (125), as well as three decoupling elements, the common terminal of which is connected to the positive terminal of the control voltage source (129), the negative terminal of which is the common bus, the free terminals of the two decoupling elements are connected to the cathodes of the varicaps (124), (126), their anodes - to the common bus, the free terminal of the third decoupling element is connected through a capacitor (109) to the output of the device (127), among other things, the minimum total resistances Z _B1 →0, Z _E1 →0, Z _B2 →0, Z _E2 →0, Z _K →0, calculated at the corresponding points of the collector Z _K of the transistor (101), as well as the bases Z _B1 , Z _B2 and emitters Z _E1 , Z _E2 of the transistors (100) and (101), which are described by the system of wave [S]-parameters, are related by the relations: where k is an integer, ImZ _B1 (ω), ImZ _B2 (ω), ImZ _E1 (ω), ImZ _E2 (ω) and ImZ _K (ω) are the imaginary parts of the resistances Z _B1 , Z _B2 , Z _E1 , Z _E2 and Z _K of the elements of star-shaped capacitive three-point equivalent circuits, which are formed at a cyclic frequency ω equal to either the fundamental frequency ω ₀ , or even 2kω ₀ or odd (2k+1)ω ₀ frequencies, characterized in that an inductance (115) is introduced, which is connected on one side to the common bus, and on the other to the free terminals of the resistor (106) and capacitor (122), as well as to the anode of the varicap (125), the cathode of which is connected to the common point of the capacitor (110) and the inductance (118), which in turn are connected respectively between the base of the transistor (100) and the cathode of the varicap (124), in addition, the cathode of the varicap (123) is connected to the common point of the capacitor (108) and a section of the microstrip line (119), the other end of which is connected to the capacitor (109), among other things, the decoupling elements for supplying control voltages to the varicaps with capacitances C _B are made in the form of three L-shaped L _C C _C low-pass filters on inductances L _C (112), (113) and (114), as well as on capacitance C _C (116), parallel to the power source (129), and their values satisfy the inequality Where and ƒ _Ц is the characteristic frequency of the L _Ц С _Ц -circuit.