RU2297013C1 - Multi-channel correlation-filter receiving arrangement with selection of moving targets - Google Patents

Abstract

FIELD: the invention refers to radio receiving technique of processing of quasi-continuous pulse-Doppler signals and may be used in radar systems using sounding signals with comb-shaped spectrum.

SUBSTANCE: the multi-channel correlation-filter receiving arrangement with selection of moving targets has in-series connected a compensator of spectral lines of interference and multi-channel arrangement of correlation-filter processing of received signals of pulse-Doppler radar. The compensator of spectral lines of interference has in the direct channel in-series switched a multiplier-mixer, an algebraic adder, a wide-band filter, a modulator and an amplifier and in the compensating channel - in-series switched an analog-digital converter, a digital filter of low frequencies and digital-analog converter. The first input of the analog-digital converter is connected with the input of the multiplier-mixer which is the input of the arrangement, the output of the digital-analog converter is connected with the second input of the algebraic adder, controlling inputs of the digital-analog converter and digital filter of low frequencies are combined and switched to the source of digitization pulses. The second input of the multiplier-mixer is connected through a phase inverter with a heterodyne of the radar and to the second input of the modulator is switched the output of the heterodyne of the weighting function.

EFFECT: simplification of the arrangement, increasing stability of its work and functional possibilities with more simple and reliable means.

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Description

The invention relates to a radio reception technique for processing quasi-continuous pulsed-Doppler signals and can be used in radar systems using sounding signals with a comb spectrum.

A multi-channel correlation-filter receiving device is known that contains a sequentially included compensator of spectral interference lines and a multi-channel correlation-filter processing device for received signals of a pulse-Doppler radar station, as well as a multi-channel synthesizer of a tunable equidistant heterodyne frequency grid and a weight function generator. The spectral interference line compensator contains a direct-connected algebraic adder, a broadband filter, a modulator and a first amplifier in the forward channel, and a heterodyne-tunable narrow-band comb filter, a second amplifier and a phase shifter, and each channel of a multi-channel heterodyne-tunable narrow-band combiner is sequentially connected in a direct channel the filter contains a series-connected first analog multiplier, a narrow-band filter and a second analog ne a multiplier, the first inputs of the first analog multipliers of all channels being combined and connected to the first input of the algebraic adder, which is the input of the device and connected to the output of the signal source of the pulse-Doppler radar, the outputs of the second analog multipliers of all channels of the heterodyne tunable narrow-band comb filter are combined and connected with the input of the second amplifier, and the output of the phase shifter with the second input of the algebraic adder, the second inputs of the first and second analog each channel's multipliers are combined and connected to the corresponding output of the multichannel synthesizer tunable equidistant frequency grid; the output of the generator of the weight function is connected to the second input of the modulator; A multichannel device for correlation-filter processing of received signals of a pulse-Doppler radar contains in each channel a series-connected gating cascade in range and a set of band-pass filters with adjacent bands in the interval of Doppler frequencies [1].

In the described device (prototype of the invention), the passive interference spectrum is suppressed, the input signal is time-gated and the Doppler frequency is filtered in each gate. In it, the output multichannel device for correlation-filter processing of received signals (SLS) can be implemented in both analog and digital form with preliminary digitization of the received signal using analog-to-digital converters. In the analog implementation of a set of correlation filter channels (CFC) of the SLR, the main suppression of the spectral lines of passive interference is performed using an analog compensator for the spectral interference lines and, additionally, using a single-band analog filter that is turned on at the output of each gating stage before a set of band-pass Doppler filters. In the digital implementation of the KFK UOS kit, the main suppression of the spectral lines of passive interference is also performed using an analog compensator of the spectral lines of interference, and gating, additional suppression of the spectral lines of passive interference and Doppler filtering of the received signals are digitally performed.

The reason that impedes the obtaining of the technical result indicated below when using the well-known multi-channel correlation filter receiver is the cumbersomeness of the hardware implementation of the analog compensator of the spectral interference lines, since it requires multichannel structure and the corresponding multichannel synthesizer tunable equidistant heterodyne frequency grid, as well as insufficient stability of its characteristics due to the influence of parametric and climatic factors in analog hardware. In addition, the prototype does not have universality, since the number of suppressed spectral lines of interference is always finite, and the notch band of each channel of the compensator for spectral lines of interference cannot be rebuilt during operation, which limits the functionality of the device when it is used.

The invention consists in the following.

The objective of the invention is to simplify the compensator for spectral interference lines, increase the stability of its operation and functionality to increase the noise immunity of the device by simpler and more reliable means.

The specified technical result is achieved by the fact that in the well-known multi-channel correlation-filter receiving device containing sequentially included a compensator for spectral interference lines and a multi-channel device for correlation-filter processing of received signals of a pulse-Doppler radar, as well as a phase shifter and a generator of a weight function, moreover, a compensator of spectral interference lines contains in direct channel sequentially connected algebraic adder, broadband filter, modulator and amplifier and the output of the generator of the weight function is connected to the second input of the modulator, according to the invention, a multiplier-mixer is introduced into the direct channel of the spectral interference compensator, the first input of which is the input of the device and connected to the output of the signal source of the pulse-Doppler radar, and the output is connected to the first input of the algebraic adder, the compensating channel of the compensator for the spectral interference lines is made in the form of a series-connected analog-to-digital converter (ADC), a digital filter lower their frequencies (low-pass filter) and digital-to-analog converter (DAC), while the first ADC input is connected to the input of the multiplier-mixer, the digital-to-analog converter output is connected to the second input of the algebraic adder, the control input of the ADC and the control input of the digital low-pass filter are combined and connected to the sampling pulse source, the input the phase shifter is connected to the local oscillator signal source, and its output is connected to the second input of the mixer multiplier.

The canonical recursive digital low-pass filter of the first or second order is used as a digital low-pass filter.

Causal relationships of the features of the invention with the technical result are expressed in the following. Instead of a part of the circuit of an analog compensator of spectral interference lines in the claimed device, a circuit is included, which is a digital moving target selector (SDC) of a compensation type, containing a multiplier-mixer and an algebraic adder in a direct channel, and ADC and a digital low-pass filter in a compensating channel and the DAC, as well as the phase shifter, through which the multiplier-mixer is connected to the local oscillator of the pulse-Doppler radar. Such a compensation digital type SDC possesses an infinite notch comb response due to a digital low-pass filter and therefore suppresses all spectral lines of passive interference regardless of the bandwidth of the broadband filter. In this case, in the steady state, the amplitude of the interference pulses is suppressed, which are emitted by the digital low-pass filter, and the spectral lines and the amplitude of the pulses of the Doppler signals remain unchanged, since they are not allocated by the digital low-pass filter in the compensating channel of the SDC. In addition, the use of a digital compensating channel eliminates parametric and climatic factors for the operation of the SDC, stabilizes all its characteristics and makes it easy to change its parameters, since they are determined by the weight coefficients of the digital low-pass filter. An important advantage of the invention is the fact that such a digital SDS allows you to work directly with radio signals at a higher intermediate frequency than the prototype, since due to the frequency conversion to ADC during sampling and in the mixer multiplier during heterodyning, the output of the digital SDS significantly decreases intermediate frequency without additional equipment, which simplifies further signal processing, especially with the digital implementation of KFK UOS. In this case, ADCs are used, the bit depth of which is less than that required in the device without the use of moving targets selection.

The invention is illustrated by drawings, in which: FIG. 1 is a functional diagram of a multi-channel correlation-filter receiving device with selection of moving targets; figure 2 and 3, respectively, the temporal and spectral characteristics that explain the operation of the device.

A multichannel correlation filter receiver with moving target selection (Fig. 1) contains sequentially connected digital SDC 1 of compensation type, a broadband filter 2, modulator 3, amplifier 4 and a multichannel device for correlation filter processing of received radar signals 5, as well as a weight function generator 6, the output of which is connected to the second input of the modulator 3. Digital SDS 1 contains in a direct channel a series-connected multiplier-mixer 7 and an algebraic adder 8, as well as phase shifters spruce 9; in the compensating channel, it contains ADC 10, digital low-pass filter 11 and DAC 12 connected in series. The first input of the multiplier-mixer 7 is the input of the device, it is connected to the output of the received signal from a pulse-Doppler radar (not shown) and is also connected to the input of the ADC 10. The second input of the multiplier-mixer 7 is connected to the output of the phase shifter 9, the input of which is connected to the source signal of the local oscillator with a frequency f _HET (not shown). The output of the DAC 12 is connected to the second input of the algebraic adder 8, the output of which is the output of the digital SDC 1 and connected to the input of the broadband filter 2. The control input of the ADC 10 and the control input of the digital low-pass filter 11 are combined and connected to a source of sampling pulses (not shown), following repetition rate f _DK .

A multichannel device for correlation and filter processing of received radar 5 signals (analog or digital with preliminary digitization of the signal) contains a set of N identical CPKs, each of which consists of a series-connected range gating cascade (analog or digital) and a set of M narrow-band filters with adjacent bands (analog or digital) [2, 3]. As a digital low-pass filter 11, a well-known canonical recursive digital low-pass filter of the first or second order can be used [4]. In particular, the second-order low-pass filter can be made in the form of series-connected first, second adders and a multiplier by a normalizing coefficient; serially connected first and second delay devices; the first, second, third and fourth multipliers by the weight coefficient; the output of the first adder is also connected to the input of the first delay device, the output of which is also connected to the inputs of the first and third multipliers by a weight factor, the outputs of which are connected to the second inputs of the second and first adders, the output of the second delay device is connected to the inputs of the second and fourth weight factor multipliers, the outputs of which are connected to the third inputs of the second and first adders, respectively; the first input of the first adder is the first input of the low-pass filter, its control input is connected to the synchronizer, and the output of the multiplier by the normalizing coefficient is the output of the second-order low-pass filter. Delay devices in the low-pass filter can be made in the form of multi-link structures of digital shift registers, while the digital low-pass filter 11 can generally be performed, for example, in the form of a well-known second-order block diagram [5] with the number of shift register links m in each digital bit equal to the product of the sampling frequency f _DK by the pulse repetition period of a quasi-continuous signal T = 1 / F, where F is the pulse repetition rate, i.e. m = f _DK T. If the oscillations of the received and compensating signals are out of phase at the first and second inputs of the algebraic adder 8, then it performs the function of summing the signals, otherwise, the function of subtracting signals. When summing the input signals, the algebraic adder 8 can be performed on resistors or active elements with a total load [6], and when subtracting the input signals in the form of a differential amplifier [7]. As the multiplier-mixer 7 can be used an analog multiplier, implemented in the form of a standard microcircuit, for example, type K174PS1 [8].

A multi-channel correlation-filter receiving device with selection of moving targets works as follows (figure 1). An additive mixture (sum) of a packet of coherent radio pulses of passive interference and Doppler signals, generally coinciding in time, with a duration of τ and a repetition period T = 1 / is received at its input and, accordingly, in the multiplier-mixer 7 and ADC 10 of the digital SDC 1 F with the frequency of filling the radio pulses of the interference f _IF and the radio pulses of the Doppler signal f _IF + F _D , where f _IF is the intermediate frequency at the path input, F _D is the Doppler frequency shift of the signal from a moving object. Moreover, the amplitude of the interference pulses significantly exceeds the amplitude of the signal pulses and they are indistinguishable in time (figa, oscillogram 1). However, their comb spectra differ due to the Doppler shift of the comb spectrum of the radio pulses of the Doppler signal with the center frequency of the spectrum f _IF + F _D relative to the stationary comb spectrum of the radio pulses of interference with the center frequency of the spectrum f _IF . From the output of the ADC 10, the digitally converted signal enters the digital low-pass filter 11, which has a narrow-band comb frequency response with a crest period of F = 1 / T, which emits a comb spectrum of interference and does not pass the comb spectrum of the Doppler signal. Therefore, after conversion to an analog form in the DAC 12, at the output of the compensating channel of the digital SDC 1, only interference radio pulses are extracted, which in the algebraic adder 8 are subtracted from the mixture of the radio interference pulses and the Doppler signal, as a result of which the interference pulses are suppressed a specified number of times, and the Doppler pulses signal pass without suppression. It should be noted that in order to isolate interference pulses in the digital low-pass filter 11 with the filling frequency f _IF , this frequency must be a multiple of the repetition frequency of the crests of this filter equal to F, i.e. f _IF = n _f F, where n _f is an integer. In pulsed-Doppler systems, this is usually performed, since in them both intermediate frequencies and pulse repetition frequencies are formed multiple of the same reference frequency f ₀ .

When turning on and off the pulse train of a mixture of a quasicontinuous signal, transients occur at the leading and trailing edges of the packet [8], as shown in Fig. 2a (waveform 2), due to the inertia of the narrow-band digital low-pass filter 11. The presence of transient processes does not allow the effective use of the suppression result interference pulses, therefore, they should be excluded from further signal processing. To cut out transients, use a modulator 3 controlled by a weight function from the generator 6. The weight function can be rectangular or smooth, for example, in the form of a Hamming function, the beginning of which is delayed relative to the beginning of the packet during the transient process T _PP , and the end coincides with the end of the received packet . Figure 2a shows the signals at the output of modulator 3 for the case of a rectangular weight function (waveform 3) and for the case of the Hamming function (waveform 4), where T _PP = 50 T, burst duration T ₀ = 200 T and the duration of the weight function T _B = 150 T, and a significant suppression of interference pulses in the steady state (more than 30 dB) is also seen. After modulator 3, the level of residuals from the compensation of interference pulses increases by the amount of mixture suppression in the digital SDC 1 using amplifier 4, which thereby increases the dynamic range of the device by the same amount.

The frequency conversion of the input radio signal in digital SDC 1 is as follows. When the filling frequency of the pulses of the input radio signal f _IF and the sampling frequency of this signal in the ADC 10 f _DC at the output of the DAC 12, a set of direct and mirror spectra is formed:

- direct spectra at frequencies f _IF -f _DC , f _IF , f _IF + f _DC , etc .;

- mirror spectra at frequencies 2f _DK -f _IF , 3f _DK -f _IF , etc.

Since the entire frequency grid in pulse-Doppler radars is a multiple of the reference frequency f ₀ , then, assuming f _IF = nf ₀ and f _DC = (n-1) f ₀ , we obtain direct spectra at frequencies f ₀ , nf ₀ , (2n -1) f ₀ , etc. and mirror spectra at frequencies (n-2) f ₀ , (2n-3) f ₀ , etc. Figure 3a shows spectrograms of processes in digital SDC 1 with n = 6, while the spectrum of the input signal is located at a frequency of 6f ₀ (spectrogram 1), and the sampling spectra at the output of DAC 12 are at frequencies f ₀ (spectrogram 3), 6f ₀ (spectrogram 3b), 11f ₀ (spectrogram 3d), etc. (direct spectra) and at frequencies 4f ₀ (spectrogram 3a), 9f ₀ (spectrogram 3c), etc. (mirror spectra) at a sampling frequency of 5f ₀ (spectral line 7). As can be seen from the spectrograms, the most intense is the direct sampling spectrum at a frequency f ₀ , the intensity of the remaining spectra rapidly decreases with frequency. This is due to the meander shape of the sampling pulses commonly used in standard ADCs. To compensate for interference pulses, it is necessary to transfer the interference spectrum at the input of the algebraic adder 8 to the frequency f ₀ . For this, a direct multiplier-mixer 7 is provided in the direct channel of the digital SDC 1, to the second input of which a local oscillator signal with a frequency f _HET = f _DK = 5f ₀ is received (Fig. 3a, spectral lines, respectively, 6 and 7), while the output of the multiplier-mixer 7, two spectra are formed at frequencies f _IF ± f _HET , i.e. at frequencies f ₀ and 11f ₀ (figa, respectively, spectrograms 2a and 2b), the intensity of which is 2 times less than the intensity of the spectrum of the input signal at a frequency of 6f ₀ (figa, spectrogram 1). Therefore, to obtain a given suppression depth P of interference pulses, it is necessary that the transmission coefficient of the compensating channel on all ridges of the frequency response be K ₀ = (P-1) / P, which, for example, at P = 30 (30 dB) is K ₀ = 29 / 30 = 0.966. This is achieved by fixing the desired value of the normalizing coefficient of the transfer function of the digital low-pass filter 11 of the compensating channel. Broadband filter 2 has a center frequency f ₀ and a passband of no more than 2 / τ and therefore sifts out all other spectra acting in the direct and compensating channels of digital SDC 1. As a result of weight processing with gating in modulator 3, the intensity of the interference spectra decreases significantly, as can be seen from figa, where spectrum 4 is the result of weight processing by a rectangular function, spectrum 5 is the result of weight processing by a smooth Hamming function.

On figb shows the waveforms explaining the process of compensation of impulse noise in the digital SDC 1 in steady state in the scale of the pulse duration:

waveform 1 - interference radio pulse at the input of digital SDS 1 of duration τ = T / 10 with a filling frequency f _IF = 6f ₀ ;

waveform 2 - pulse at the output of the multiplier-mixer 7, representing oscillations in the form of the sum of the differential frequency f _IF -f _HET = f ₀ and the total frequency f _IF + f _HET = 11f ₀ ;

waveform 3 - pulse at the output of the DAC 12 in the form of a stepwise oscillation with a repetition frequency f ₀ representing the result of sampling the input radio pulse 1 with a sampling frequency f _DK = 5f ₀ and filtering it in the digital low-pass filter 11; the length of the steps is τ _DC = 1 / f _DC ; step oscillations 3 and total oscillations 2 are combined so that the amplitude and phase of the middle oscillation line 3 coincides with the average oscillation line 2, while the oscillation phase 2 is adjusted by shifting the oscillation phase of the heterodyne f _HET at the second input of the multiplier-mixer 7 using a phase shifter 9, which can be performed using a simple RC circuit;

Oscillogram 4 represents the pulse of the result of filtering the difference between the oscillations 2 and 3 in the algebraic adder 8 at the output of the broadband filter 2 with a fill frequency f ₀ , while all other conversion spectra of the input signal in digital SDC 1 are effectively suppressed.

On figb shows spectrograms illustrating the process of clearing the interference spectrum in the range of Doppler frequencies f ₀ ± F:

spectrogram 1 represents the spectrum of the burst of pulses of the received interference signal, converted in the multiplier-mixer 7 to a frequency f ₀ , in the form of main lobes following with a pulse repetition rate F = 1 / T, and side lobes between them with a level below the level of the main lobes, depending from the number of pulses in a packet;

spectrogram 2 represents the spectrum of the stepwise oscillation of pulses of an interference burst at the output of the DAC 12 of the compensating channel of the SDC at a frequency f ₀ , where it can be seen that the main lobes of this spectrum coincide with the main lobes of the spectrum of pulses at the output of the direct channel multiplier-mixer 7, and the side lobes fall faster behind account comb frequency response of the digital low-pass filter 11;

spectrogram 3 represents the spectrum of the pulse train at the output of broadband filter 2, where it is seen that the main lobes are substantially suppressed at all repetition frequencies as a result of subtraction of spectra 1 and 2, and the side lobes remain unchanged;

spectrogram 4 represents the spectrum of a packet of suppressed interference pulses at the output of modulator 3 during rectangular weight processing that cuts out transients of the SDC, while the main lobes of the interference spectrum remain unchanged, and the side lobes are significantly reduced;

Spectrogram 5 represents the spectrum of the same packet, but with smooth weighting by the Hamming function with gating, one can see that the spectrum of suppressed interference pulses from the side lobes is completely cleared in the dynamic range of 100 dB relative to the level of the main lobes of the spectrum of the interference pulse packet at the input of the receiving device.

Thus, the implementation of the invention allows to significantly increase the noise immunity of the receiving path of the pulse-Doppler radar with a significant simplification of the interference compensator, to increase its dynamic range by interference, the stability of its notch characteristics and to gain the ability to control their parameters, as well as reduce the requirements for the bit capacity and speed of the secondary ADC, necessary for the digital implementation of KFK. This is ensured by the fact that in the proposed digital SDC 1, a low-speed high-speed ADC 10 can be used that is sufficient for preliminary suppression of a packet of interference pulses to the level of the maximum Doppler signal, while otherwise it would be required for the digital implementation of the CPK without preliminary selection of moving targets multi-bit high-speed ADC. For example, in the absence of selection of moving targets, a 17-bit high-speed ADC is required to realize the dynamic range of a CPK of 100 dB. In the claimed device, it is sufficient to have a slower 10 ... 12-bit ADC at the input of the digital CFC, providing a dynamic range for the Doppler signal of 60-70 dB, and a 5 ... 7-bit ADC 10 in the digital SDC 1, providing preliminary suppression 30-40 dB interference pulses to the maximum Doppler signal level, which is much cheaper.

Information sources

1. RU No. 2205422, G 01 S 13/52, 13/626, H 04 B 1/10, 2003.

2. Reference radar. Ed. M. Skolnik, New York, 1970: Per. from English (in four volumes) / Under the general ed. K.N. Trofimova; Volume 3. Radar devices and systems / Ed. A.S. Vinitsky. - M .: Owls. Radio 1978, p. 369, Fig. 6.

3. P.A. Bakulev, V.M. Stepin. Methods and devices for moving targets selection. M .: Radio and communications, 1986, pp. 140-141, Fig. 5.20.

4.S.I. Baskakov. Radio circuits and signals. M.: Higher School, 1983, pp. 489-491.

5. A. Anthony. Digital filters: analysis and design. M .: Radio and communications, 1983, pp. 292-293.

6. Design of radar receiving devices. Ed. M.A.Sokolova. M.: Higher School, 1984, pp. 131-136.

7. Radio receivers. Ed. L.G. Barulina. M .: Radio and communications, 1984, pp. 101-102.

8. A.A. Trukhachev. Radar signals and their applications. M .: Military Publishing House, 2005, pp. 112-113.

Claims (2)

1. A multi-channel correlation-filter receiving device with selection of moving targets, containing sequentially included a broadband filter, a modulator, an amplifier and a multi-channel device for correlation-filter processing of received signals of a pulse-Doppler radar, as well as a weight function generator, the output of which is connected to the second input of the modulator, characterized in that the device includes a digital moving target selector (SDC) containing in the direct channel sequentially connected multiplier-mix There is an algebraic adder, as well as a phase shifter, and in the compensating channel there are series-connected analog-to-digital converters, a digital low-pass filter and a digital-to-analog converter, while the first input of the multiplier-mixer is the input of the device and is connected to the output of the signal source of the pulse-Doppler radar , the first input of the analog-to-digital converter is connected to the first input of the multiplier-mixer, the output of the digital-to-analog converter is connected to the second input of the algebraic about the adder, the control input of the analog-to-digital converter and the control input of the digital low-pass filter are combined and connected to the source of sampling pulses, the input of the phase shifter is connected to the source signal of the local oscillator, and its output is connected to the second input of the mixer multiplier, the output of the algebraic adder is the output of the digital SDC compensation type and connected to the input of a broadband filter. 2. The device according to claim 1, characterized in that the canonical recursive digital low-pass filter of the first or second order is used as a digital low-pass filter.

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