RU186407U1 - Relative phase modulation adaptive pseudo random signal demodulator - Google Patents

Abstract

The utility model relates to telecommunications technology and can be used to receive discrete signals with relative phase modulation in spread spectrum systems (pseudorandom signals) during carrier frequency shifts in communication channels with aircraft under conditions of intentional (organized) interference. The technical result of the utility model is to increase the noise immunity of signal reception by adapting the demodulator to changes in conditions when frequency offsets occur or are absent. The device contains a first multiplier (1), an SRP generator with a synchronization block (2), the first (3) and the second (4 ) delay elements, first (5) and second (6) phase shifters by π / 2, second (7,13), third (9,11) and quadruple (15,16) multipliers, integrators (8,10,12,14 ), addition blocks (17.18), threshold block (19), switching block (20), accumulation block (21), decision block (22).

Description

The invention relates to communication technology and can be used to receive discrete signals with relative phase modulation in spread spectrum systems (with pseudo-random signals) during carrier frequency shifts in communication channels with aircraft under conditions of intentional (organized) interference.

The known demodulator of phase-difference modulation signals of the second order (A.S. 240043, IPC H03D, 1969), a demodulator with phase-difference modulation (A.S. 1670799, IPC H04L 27/22, 1991), a digital demodulator of phase-difference modulation signals of the second order (A. P. 1716616, IPC H04L 27/22, 1992), a digital demodulator of phase-difference modulation signals of the first and second order (Patent 1838884, IPC H04L 27/22, 1993), an adaptive receiver of data signals (Patent 2019048, IPC H04L 25/40, 1994 ), a method for demodulating signals with relative phase shift keying and a device for its implementation (Patent 2168869, IPC H03D, 2001), a method for correlating receiving signals with relative phase modulation and a device for determining it (Patent 2237978, IPC H04L 27/22, 2004), a method for transmitting information using noise-like signals and a device for its implementation (Patent 2362273, IPC H04L 22 / 18, 2009).

Some of these technical solutions implement signals with relative second-order phase modulation (OFM-2) and can be used to process signals received via communication channels with arbitrary frequency shifts due to the movement of aircraft (Doppler effect). But in the context of deliberate organized interference, they will be inoperative.

One of the listed technical solutions, using noise-like (pseudo-random) signals, can provide their reception at signal-to-noise ratios less than unity, but in the presence of the Doppler effect it is also inoperative. In addition, the above mentioned adaptive signal receiver will not provide the required noise immunity under the influence of deliberate interference and frequency shifts.

Thus, under the conditions of the complex effect of interference — deliberate organized and frequency shifts due to movements of moving objects, the above technical solutions will be inoperative, not providing the given noise immunity of signal reception.

In [1], it was proved that for discrete phase-modulated signals the optimal interference is also a discrete phase-modulated oscillation, the frequency and duration of which coincide with the same signal parameters. With out-of-phase and more powerful interference, the demodulator will register it instead of the transmitted signal.

To ensure the necessary noise immunity of reception under the conditions of signal-like intentional interference, spread spectrum signals generated using pseudorandom sequences are currently used [2-7]. When processing at the reception in the demodulator a combination of these signals and deliberate organized interference, this interference is transformed into a kind of natural noise provided by the model of white Gaussian noise (BGS). Therefore, the subsequent processing of the signals does not cause difficulties, since the methods of their reception under conditions of natural noise interference are developed and very effective.

Relative phase modulation of the second order (OFM-2) can be used to generate noise-like (pseudorandom) signals under the action of deliberate interference in communication channels with moving objects (aircraft). This type of modulation is also called phase difference modulation (FRM-2) [8] or in foreign literature - Double differential phase shift keying (DDPSK) [9]. OFM-2 signals, as shown in [8, 9], are resistant (insensitive) to arbitrary changes in the initial phase of signals and arbitrary frequency shifts caused by the movement of objects (Doppler effect).

OFM-2 signals are formed in such a way that the information is determined by the difference of the second order of phases of the signal packets. The information parameter here is the difference between the phase differences, determined by three neighboring premises. With relative phase modulation of the OFM, in contrast to the OFM-2, information is counted only by two adjacent signal bursts.

Suppose that due to the movement of the named object, the oscillation frequency of the signal ω has changed by some arbitrary value Δω. In this case, if the phase of the (k-1) th sending is ϕ _k-1 _1 + ϕ _s , ϕ _s is the initial phase of the signal, ϕ _k-1 is the information phase, k = 1,2, ... then the phase k- of the first parcel will be equal to (ϕ _k + ϕ _s + ΔωT), T is the duration of the parcel, ΔωT is the phase shift by the duration of one parcel, and the phase of the next (k + 1) th parcel is (ϕ _k-1 + ϕ _c + 2ΔωT )

When OFM, when the signal is formed based on the comparison of two adjacent premises

Δ ¹ ϕ _{k + 1} = ϕ _{k + 1} -ϕ _k + ΔωT,

Δ ¹ ϕ _k = ϕ _k -ϕ _k-1 + ΔωT,

it is seen that the so-called first differences Δ ¹ ϕ _{k + 1} and Δ ¹ ϕ _k do not depend on the initial phase of the signal ϕ ₀ , but depend on the frequency shift Δω.

With OFM-2, the second phase difference (the difference between the first phase differences)

Δ ² ϕ _{k + 1} = Δ ¹ ϕ _{k + 1} -Δ ¹ _k = ϕ _{k + 1} -2ϕ _k + ϕ _k-1

it does not depend on the initial phase ϕ ₀ and on arbitrary frequency shifts [8]. This property of OFM-2 signals makes it possible to use them in communication channels with moving objects, in which arbitrary signal frequency shifts occur due to the movement of objects.

Thus, the use of spread spectrum signals (pseudo-random signals) generated on the basis of OFM-2 makes it possible to provide a certain level of noise immunity of the demodulator under conditions of frequency shifts and under the influence of deliberate interference. However, this noise immunity value is not always acceptable in accordance with modern requirements, especially in cases where a moving object (aircraft) freezes at a given point in the airspace and performs the task. In this case, it is more advisable to switch to the use of OFM signals, which:

a) easier to implement;

b) have better noise immunity [8].

This can be done by adapting to the applicable signal transmission conditions.

The closest in technical essence to the proposed device is an autocorrelation signal demodulator with a single phase-difference second-order modulation (see: Yu. B. Okunev. Theory of phase-difference modulation. - M.: Communication, 1979. - 216 s; Fig. 4.11 on p. 140 )

The said demodulator contains a delay code element, two receive branches, each of which consists of second connected multipliers, integrators, fourth multipliers connected to the inputs of the first addition unit, the output of the first delay element connected directly to the second input of the second multiplier in the first branch, and in the second branch through the first phase shifter on π / 2.

In the above source (Y. Okunev, Theory of phase difference modulation) on p. 148-150 it is shown that the prototype autocorrelation demodulator provides a certain noise immunity of receiving signals with relative second-order phase modulation (OFM-2) or, as called in the source of the FRM-2, under conditions of natural interference and frequency offsets. It was also shown that the noise immunity of receiving OFM-2 signals turns out to be worse than the noise immunity of signals with relative phase modulation (OFM), which, however, cannot provide any signal reception at frequency offsets.

At the same time, under the conditions of organized deliberate interference, when the ratio of signal powers P _c and interference P _{p is} less than unity, q = P _c / P≤1, the prototype demodulator, like the above-mentioned analogues, is inoperative.

To eliminate this drawback, you can apply pseudo-random (noise-like) signals. In addition, in static modes, when a moving object (aircraft) freezes at some point in the airspace, and at this moment there is no signal frequency shift, you can switch from OFM-2 signals to OFM signals, which will improve the noise immunity of reception under the conditions of intentional organized signal-like interference. Such a transition can be implemented automatically, thereby the demodulator will adapt to changing conditions when frequency offsets occur or are absent.

The purpose of the invention is to increase the noise immunity of receiving pseudo-random signals under the conditions of organized deliberate interference, having a structure similar to signals, and exceeding them in power in communication channels with frequency shifts at various stages of moving moving objects (aircraft).

To achieve this goal, in the well-known autocorrelation demodulator containing the first delay element, two receive branches, each of which consists of second series multipliers, integrals, fourth multipliers connected to the inputs of the first addition unit, the output of the first delay element is connected to the second input the second multiplier in the first branch directly, and in the second branch through the first phase shifter on π / 2, the first multiplier is introduced, the first input of which is the input of device and through a pseudo-random sequence generator with a synchronization unit is connected to its second input, and the output is connected to the combined inputs of the first delay element and second multipliers in the first and second receiving branches, in addition, the second delay element, the second phase shifter on π / 2, connected in series to the second the addition unit, the threshold unit, the switching unit, the accumulation unit and the deciding unit, the output of which is the output of the device, while the output of the first power unit is connected to the second inputs of the com unit mutations and the second block of embeddings, the first inputs of which are connected to the integrator output of the first branch of reception, as well as the third and fourth branches of reception, each of which contains series-connected third multipliers, integrators whose outputs are connected to the second inputs of the fourth multipliers, while the first inputs of the third and the fourth branch of the reception are connected to the output of the first delay element, the output of the second delay element is connected to the second inputs of the third multipliers in the third branch of the reception directly, and in h tvertoy receiving branch via the second phase shifter to π / 2.

In FIG. 1 presents a structural diagram of the proposed device. In FIG. 2 presents the results of evaluating the effectiveness of the proposed device.

An adaptive pseudo-random signal demodulator with relative phase modulation contains a first delay element 3, two receiving branches, each of which consists of second series multipliers 7 and 13, integrators 8 and 14, fourth multipliers 15 and 16 connected to the inputs of the first addition block 17, the output of the first delay element 3 is connected to the second input of the second multiplier 7 in the first branch directly, and in the second branch through the first phase shifter 5 by π / 2, in addition, the first multiplier 1, the first in the path of which is the input of the device and is connected to its second input through a pseudo-random sequence generator 2 with a synchronization unit, and the output is connected to the combined inputs of the first delay element 3 and second multipliers 7 and 13 in the first and second reception branches, in addition the second delay element 4, a second phase shifter 6 by 7 π / 2, serially connected to the second addition unit 18, the threshold unit 19, the switching unit 20, the accumulation unit 21 and the decision unit 22, while the output of the first addition unit 17 is connected to the second inputs of the unit switching 20 and the second addition unit 18, the first inputs of which are connected to the output of the integrator 8 of the first branch of reception, as well as 3 and 4 branches of the reception, each of which contains sequentially connected third multipliers 9 and 11, integrators 10 and 12, the outputs of which are connected to the second the inputs of the fourth multipliers 15 and 16, while the first inputs of the third multiplier 9 and 11 are connected to the output of the first delay element 3, and the output of the second delay element is connected directly to the second inputs of the multiplier 9, and the multiplier 11 through the second phase shifter 6 by π / 2, while the output of the deciding unit 22 is the output of the device.

The new set formed by introducing the blocks and elements mentioned above allows you to process pseudo-random signals based on relative phase modulation (OFM) or relative second-order phase modulation (OFM-2), which leads to a positive effect - increasing the noise immunity of receiving pseudorandom signals in conditions of deliberate interference in communication channels with frequency shifts at various stages of moving moving objects (aircraft) - OFM-2 mode when the object is moving, OFM mode case. When the aircraft freezes motionless at a given point in airspace.

The proposed device operates as follows. The received signal is multiplied by the pseudo-random sequence y generated by the pseudo-random sequence generator with synchronization unit 2. The output of the first multiplier 1, the signal from which the pseudo-random sequence is removed after the multiplication procedure, is fed simultaneously to the first delay element 3 and the second multipliers 7 and 13 in via the combined input the first and second branches of the reception, respectively. Simultaneously with the received signal, a deliberate (organized) interference acts on these elements 3, 7, 13, which, as a result of the conversion (the first multiplier 1 and the pseudo-random sequence generator with synchronization block 2), takes on a form similar to natural noise in the communication channel, i.e., it becomes random (stochastic) process.

Block 2 synchronization with a pseudo-random sequence generator can be implemented in the form of known synchronization devices (see, for example, patent for utility model 118495, IPC H04L 7/02, 2012).

The second inputs of the second multipliers 7 and 13, which perform the functions of phase detectors, are supplied with a signal delayed by the duration of one parcel T from the output of the first delay element, and directly to the second input of the multiplier 7, and through the first phase shifter 5 to the second input of the second multiplier 13 π / 2, so that in the first branch to implement autocorrelation processing of the cosine component, and in the second branch of the reception, the components of the sine.

In addition, the signal from the output of the first delay element 3 is fed to the input of the second delay element 4 for the duration of the sending T and at the same time to the first inputs of the third multipliers 9 and 11 in the third and fourth reception branches, respectively.

By analogy with the second multipliers 7 and 13, the third multipliers 9 and 11 also serve as phase detectors. A signal delayed by the duration of two 2T transmissions is sent to their second inputs, and to the multiplier 9 in the third branch of the receiver after passing the first 3 and second 4 delay elements directly, and the multiplier 11 in the fourth branch of the receiver through the second phase shifter 6 to tg / 2. Thus, by analogy with the signal processing in the first and second reception branches, the third and fourth reception branches also carry out autocorrelation processing of the cosine and sine signal components. But unlike the first and second branches of the reception, where the signal is processed - the current and delayed by the time T, the third and fourth branches of the reception process the signal with a delay of T and 2T, respectively.

The sending of the signal in a mixture with interference after the phase detectors of the multipliers 7, 13, 9, and 11 pass through the integrators 8, 10, 12, and 14, which are essentially low-pass filters. As a result, the inputs of the fourth multiplier 15 are fed the elements of the processed signal, having the form

The inputs of another fourth multiplier 16 receive elements of the processed signal of the form

where u _k (t), u _k-1 (t), u _k-2 (t) are additive mixtures of signal and interference bursts -

current (k), delayed by the duration of one (k-1) and two (k-2) packages, respectively; T is the duration of the signal; “*” - sign of complex conjugation (Hilbert transform); γ _k is the current element of the pseudo-random sequence generated by the SRP generator with synchronization unit 2.

After the corresponding multiplication in the multipliers 15 and 16 and addition in the addition unit 17, the total signal values are fed to the combined second input of the addition unit 18 and the first input of the switching unit 20. At the same time, the signal is sent in the mixture with interference from the output of the integrator 8 to the first input of the unit addition 18 and the third input of the switching unit 20.

As noted above, at the input of the proposed device there are sequentially successive signal transmissions

u _k-1 (t) = U _c cos + (ωt + ϕ _k-1 + ϕ _s ),

u _k (t) = U _c cos (ωt + ϕ _k + ϕ _s + ΔϕT),

u _k-1 (t) = U _with cos (ωt + ϕ _{k + 1} + 2ΔϕТ),

where all the parameters of the packages were determined earlier.

Let the aircraft, in particular, the unmanned aerial vehicle (UAV), be located at some point in the airspace, and there is no frequency shift, Δω = 0. Let in this case, the first phase differences between adjacent packages are equal to π, i.e.

Δ ¹ ϕ _{k + 1} = ϕ _{k + 1} -ϕ _k = π,

Δ ¹ ϕ _k = φ _k -ϕ _k-1 = π

Then, at the output of addition block 17 (point 1), the signal will be proportional to the cosine of the second phase difference [8] Δ ² ϕ _{k + 1} = Δ ¹ ϕ _{k + 1} -Δ ¹ ϕ _k = 0 and has a value

At the output of the integrator 8 in the first branch of the reception (point 2), the signal is proportional to the cosine of the first phase difference [8]

At the first input of addition unit 18, a signal u ₁ (t) with a negative sign is supplied to the second input of block 18 - a signal u ₂ (t) with a positive sign. Therefore, with the equality of their amplitudes at the input of the threshold block 19, the signal will be absent. In this case, the switching unit 20 will connect the output of the integrator 8 (point 2) to the accumulation unit 21 and there will be sent the pseudo-random signal with relative phase modulation (PS-OFM) received by the circuit elements of the proposed device (blocks 1, 2, 3, 7, 8) performing the function of the autocorrelation PS-OFM signal demodulator.

If now, due to the movements of the aircraft (Flight mode), a frequency shift Δω ≠ 0 appears in the communication channel, then the first phase differences Δ ¹ ϕ _{k + 1} and Δ ¹ ϕ _k between the signal transmissions, as shown above, will depend on of this frequency shift Δω. As a result, the output of the integrator 8 (point 2) changes the value of the signal. In contrast, the second phase difference between the signal bursts Δ ² ϕ _{to + 1} does not depend on an arbitrary frequency shift Δω, which was shown earlier. In other words, at the output of addition block 17 (point 1), the signal does not change. Then two signals will arrive at the inputs of addition block 18, different in both sign and amplitude. The resulting signal from its output will go to the input of the threshold block 19 and make the switching unit connect to the input of the addition block 21 the signal from the output of the addition block 17 (point 1), which corresponds to the implementation of that part of the proposed device that performs the functions of autocorrelation processing of a pseudorandom signal with a relative phase second-order modulation (PS-OFM-2), insensitive to frequency shifts in the communication channel with moving aircraft. The threshold value in the threshold block 19 is selected empirically. The presented procedure will be implemented with other values and ratios of the phase difference of the signal.

In both versions of the processing of the received signal — OFM-2 mode, the accumulation unit 21 is designed to add sequentially incoming n packets of a pseudo-random signal.

Recall that a pseudo-random signal is a set of successively following n elementary premises, the form of which is hidden with the help of a pseudo-random sequence superimposed on information symbols in the transmission.

In the proposed device, a pseudo-random sequence is removed from the received signal, as a result of which it is a cumulative information symbol (sending a signal) in the form of “0” or “1”, consisting of n elements, some of which are distorted by interference. Since the jammer does not know the law of changing the pseudo-random sequence superimposed on the signal, it remains for him to act at random in order to disrupt the signal transmission process. Therefore, increasing the number of elementary premises n, it is possible to reduce the damaging effect of interference. Note that hiding the law of pseudo-random changes is a fundamental condition in such devices.

Thus, for the correct registration of the signal in the proposed device, for example, corresponding to the information symbol “0”, “zero” premises must prevail in it in a set of n elementary premises. "Single" packages in this case will correspond to the result of the interference. For final registration in the deciding unit 22, the cumulative signal received from the accumulation unit 21 is compared with a threshold value. If the cumulative signal exceeds a threshold, then the signal corresponding to the “zero” information symbol is recorded, otherwise a decision is made to transmit the “single” information symbol.

The decisive unit 22 can be implemented on a comparator, one signal of which receives a signal, and the second voltage threshold. When the signal exceeds the threshold voltage, a high logic level appears at the output of the comparator, otherwise it is low. When implementing block 22, you can apply the integrated circuit K521CA2 (see: Goroshkov B.I. Radioelectronic devices. Handbook. - M .: Radio and communications, 1985. P. 314, Fig. 13.34b).

Accumulation block 21 can be implemented as an integrator on operational amplifiers with a reset, which accumulate the signal over the time interval nТ, where n is the base (number of elements) of the total (composite) pseudorandom signal, T is the duration of its one element (see: A. Sikarev A., Lebedev O. N. Microelectronic devices for the processing of complex signals. - M.: Radio and communication, 1983. P. 198-210, Fig. 7.10; Goroshkov B.I. Radioelectronic devices. Handbook. - M .: Radio and Communication, 1985. S. 349, Fig. 15, 42).

The switching unit 20 can be implemented on a multiplexer, i.e. on a device that has several signal inputs, one control input and one output. In some cases, the multiplexer is called a switch (see: Ugryumov E.P. Digital circuitry. - SPb .: BHV-Petersburg, 2010. P. 54-56, Fig. 2.9).

The threshold block 19 can be performed on a comparator by analogy with the decision block 22. To implement block 19, you can use the integrated circuit K133LAZ (see: B. Goroshkov Radio-electronic devices. Reference book. - M .: Radio and communication, 1985. P. 312, Fig. 13.31a, 13.31c).

To evaluate the achieved positive effect, i.e. increasing the noise immunity of receiving pseudo-random signals under the conditions of organized deliberate interference having a structure similar to signals and exceeding them in power in communication channels with frequency shifts at various stages of aircraft movements, we find the calculated relationships for error probabilities.

Let pseudo-random signals form on the transmitting side

kind of

where rect (t - (kl) T) is a unit impulse of duration T that specifies the time interval of the PS-OFM-2 signal, π _k (t) = (- l) ^γk , (k-1) T≤t≤kT,

γ _k ∈ {0,1} is an element of the pseudo-random sequence, α _k = α _k-1 ⊕Δ _k ,

Δ _k = Δ _k-1 ⊕γ _k , ⊕ is the modulus two addition sign, α _k ∈ {0,1} is the quantity,

defining the rule of signal formation, ϕ _с (t) is the phase shift of the signal, in the general case taking into account the Doppler effect, i.e. frequency offsets.

For processing these signals under conditions of frequency shifts of the carrier wave and the action of organized signal-like interference

N (t) = U _p π _p (t) sin [ωt + ϕ _p (t)],

where π _n (t) = (- 1) ^δk , δ _k ∈ {0,1} are equally probable and mutually independent cases

tea quantities that determine the discrete phase modulation of interference, the proposed device implements a signal processing procedure

- правило определения в решающемWhere

- decisive rule

block 22 of the sign (polarity) λ - the total value of the signal at the input of the accumulation block 21 by comparing with the threshold in the decision block 22.

- совокупности сигнала и помехи на выходе другого четвертого перемножителя 16. Знак «+» показывает сложение названных компонент сигнала в блоке сложения 17, символ «∑» с пределами от 1 до n указывает на накопление элементов сигнала в блоке накопления 21, «signλ,» означает принятие решения в решающем блоке 22, если λ>0, то регистрируется информационная посылка сигнала «0», в противном случае - «1».Here X _k X _k-1 - the signal and interference at the output of the fourth multiplier 15,

- the signal and interference aggregate at the output of the other fourth multiplier 16. The “+” sign shows the addition of the named signal components in the addition block 17, the “∑” symbol with limits from 1 to n indicates the accumulation of signal elements in the accumulation block 21, “signλ,” means a decision is made in decision block 22, if λ> 0, then the information sending of the signal “0” is recorded, otherwise - “1”.

Given the pseudo-random transformations at the input of the proposed device, one can determine the mathematical expectation and variance of the quantity X

M {λ} = (1 + q) qn, D {λ} = (l + 8q + I3q ² + 6q ³ ) n,

where q is the ratio of the power of the signal element to the power of the organized intentional interference; n is the number of elements of a pseudo-random signal (signal base).

Using a Gaussian approximation, which follows from the nature of these pseudo-random transformations during signal processing in the proposed device, we obtain the final formula for the probability of receiving errors of pseudo-random OFM-2 signals

- интеграл вероятностей.Where

is the probability integral.

Calculations by relation (1) show that under conditions of signal frequency shifts and organized signal-like interference, twice the signal power, i.e. q = 0,5 and base dither signal FMR-2 n - 100, the likelihood of erroneous reception _oui P ≈ 6,21⋅10 ^-3. It is clear that in modern conditions this quantity cannot fully satisfy the requirements of fidelity of signal reception. But at the stage of the aircraft’s movement, when there are signal frequency offsets, this can be reconciled to some extent.

Another thing is when the aircraft occupies a certain point in the airspace and solves the tasks without moving in space. In this case, the proposed device, in accordance with the above procedure, switches to another mode of receiving signals, namely, it starts to process PS-OFM signals that provide greater noise immunity of reception in the absence of signal frequency offsets.

In this situation, the proposed device will implement a pseudo-random signal processing procedure

u_k(t), u_h__x{t) - текущая и задержанная с помощьюWhere

u _k (t), u _h _ _x {t) - current and delayed with

the first delay element 3 for the duration T of one transmission, the combination of signal and interference, X _k are the signal and interference components at the output of the integrator 8 (point 2), γ _k is the current element of the pseudo-random sequence received at the second input of the first multiplier 1, the symbol " ∑ "with limits from 1 to n indicates the accumulation of signal elements in accumulation block 21," sign ε "means the decision is made in decision block 22.

For expression in square brackets, one can find the mathematical expectation and variance

M {ε} = nq, D {ε} = n (2qcos ² θ + l),

where q is the ratio of the power of the signal element to the interference power, θ is the phase difference of the signal and the interference.

By analogy with the previous expression (1), we obtain the relation for the probability of error for pseudo-random OFM signals when there is no frequency shift

where - F (x) - is defined in (1).

по (1) и

по (2) для случая наиболее опасного воздействия преднамеренной противофазной помехи θ=ϕ_с-ϕ_n=0 при использовании в предлагаемом устройстве ПС-ОФМ-2 сигналов при наличии сдвига частоты и ПС-ОФМ при отсутствии сдвига частоты соответственно представлены на фиг. 2. В качестве примера для обоих сигналов - ПС-ОФМ-2 и ПС-ОФМ выбрано одинаковое значение базы n=100 и n=200 и диапазон изменений соотношений сигнал/помеха 0<q<1.Error Probability Calculation Results

by (1) and

according to (2) for the case of the most dangerous effect of intentional out-of-phase interference θ = ϕ _with -ϕ _n = 0 when using in the proposed device PS-OFM-2 signals in the presence of a frequency shift and PS-OFM in the absence of a frequency shift, respectively, are shown in FIG. 2. As an example, for both signals - PS-OFM-2 and PS-OFM, the same base value n = 100 and n = 200 and a range of signal-to-noise ratios of 0 <q <1 are selected.

а для ПС-ОФМ

т.е. на два порядка меньше.It can be seen that with interference twice the signal power q = P _s / P _n = 0.5 of the probable error for PS-OFM-2, as noted above,

and for PS-OFM

those. two orders of magnitude less.

Under the aforementioned conditions, the action of deliberate organized interference, exceeding the interference power, as mentioned above, the prototype is inoperative. The proposed device as seen from the calculation results shown in FIG. 2 allows one to increase the noise immunity of receiving pseudorandom signals at various stages of the aircraft’s movements, i.e. in the presence and absence of frequency offsets and under the conditions of deliberate interference by automatically switching from one to another mode - use of PS-OFM-2 or PS-OFM signals.

Thus, the use of new elements indicated in the characterizing part of the claims favorably distinguishes the proposed technical solution from the prototype and allows to obtain a positive effect in the form of increased noise immunity of receiving pseudorandom signals under the conditions of organized deliberate interference having a structure similar to the signal and exceeding them by power in communication channels with frequency shift at various stages of aircraft movements.

Literature and Sources

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Block designation

1. The first multiplier.

2. PSP generator with synchronization block.

3. The first element of delay.

4. The second element of delay.

5. The first phase shifter on π / 2.

6. The second phase shifter on π / 2.

7. 13. The second multipliers.

9, 11. Third multipliers.

8. 10, 12, 14. Integrators.

15, 16. Fourth multipliers.

17, 18. Blocks of addition.

19. Threshold block.

20. The switching unit.

21. The accumulation unit.

22. The decisive unit.

Claims (1)

An adaptive pseudo-random signal demodulator with relative phase modulation, comprising a first delay element, two receive branches, each of which consists of second series multipliers, integrators, fourth multipliers connected to the inputs of the first addition unit, and the output of the first delay element is connected to the second input the second multiplier in the first branch of the reception directly, and in the second branch of the reception through the first phase shifter by π / 2, characterized in that the first variable an amplifier, the first input of which is the input of the device and is connected to its second input through a pseudo-random sequence generator with a synchronization unit, and its output is connected to the combined inputs of the first delay element and second multipliers in the first and second reception branches, in addition, the second delay element, the second π / 2 phase shifter, serially connected second addition unit, threshold unit, switching unit, accumulation unit and decision unit, while the output of the first addition unit is connected to the second inputs of the unit switching and the second addition unit, the first inputs of which are connected to the integrator output of the first branch of the reception, as well as the third and fourth branches of the reception, each of which contains series-connected third multipliers, integrators, the outputs of which are connected to the second inputs of the fourth multipliers, while the first inputs of the third and the fourth branches of the reception are connected to the output of the first delay element, and the output of the second delay element is connected to the second input of the third multipliers in the third branch of the reception directly, and in the fourth branch of the reception through the second phase shifter at π / 2, while the output of the decision block is the output of the device.

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