KR920011005B1 - Nector control method and system for an induction motor - Google Patents

Abstract

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Description

VECTOR CONTROL METHOD AND APPARATUS FOR INDUCTOR MOTORS

1 is an equivalent block diagram of a three-phase flow motor.

2 is an equivalent block diagram of a vector controlled flow motor.

Figure 3a is a block diagram of a conventional non-interfering vector control device in which three power amplifiers are used to drive a three-phase flow motor, and (b) shows a conventional non-interference in which two power amplifiers are used to drive an induction motor. (C) is a block diagram of a conventional non-interfering vector control device in which a pulse width modulation (PWM) inverter is used to drive a three-phase induction motor;

4 is a block diagram showing a main portion of a non-interference operator and a two-phase converter in the induction motor non-interference vector control apparatus according to an embodiment of the present invention.

FIG. 5 is a block diagram used in the apparatus of FIG. 4, and FIG. 5A is a block diagram showing an example of a proportional differentiator, and FIG. 5B is a block diagram showing an example of a first order delay branch.

FIG. 6A is a diagram illustrating the rotor angle frequency on the horizontal axis, the rotor torque on the vertical axis, and FIG. 6C is classified on the four quadrants to indicate the sign of the rotor angle frequency. Table 6c is a diagram showing an example of the four-quadrant operation, and when the reference rotor each frequency is switched from forward rotation to reverse rotation, the operating mode of the motor passes from the first upper limit operation to the second upper limit operation, and then to the third. Table showing an example of four upper limit operation converted to upper limit operation.

7 is a block diagram of a triangular wave generator and a positive and cosine wave generator used in the induction motor non-interfering vector control apparatus of the present invention.

FIG. 8A shows a digital triangular wave signal output from the up and down counters used in the triangular wave generator of FIG. 7, and FIG. 8B shows a digital positive and cosine wave signal output from a ROM used in the generator of FIG. Drawing.

FIG. 9 is a block diagram of a triangular wave generator and a sinusoidal wave generator in which a pulse generator is installed for initial secondary magnetic flux application used in the induction motor non-interfering vector control apparatus of the present invention.

The present invention relates to a method and a device for controlling an induction motor, and more particularly, a primary current corresponding to a secondary magnetic flux and a primary current corresponding to a secondary driving current are individually and two vectors are orthogonal. The present invention relates to a non-interfering vector control method and apparatus of an induction motor.

The non-interference described above means that the mutual interference between the secondary magnetic flux and the secondary driving current is eliminated from each other.

In recent years, in addition to the extraordinary advances in high-power electronics, the vector control method, in particular, has been proposed to drive induction motors at all speeds with a rapid response characteristic equivalent to direct current motors.

In this vector control method, the primary current of the motor is divided into the primary excitation current for generating the secondary magnetic flux and the primary drive current for generating the secondary driving current, so that their vectors are orthogonal to each other. Independently controlled.

Furthermore, in this vector control method, as in the case of a DC motor, the magnitude of the secondary magnetic flux is controlled at a constant level, and the secondary drive current is independently increased or decreased.

In the above-described vector control method of the induction motor, since there is mutual interference between the secondary magnetic flux and the secondary drive current, the magnitude of the secondary magnetic flux is not constant in practice. In order to solve this problem, a so-called non-interfering vector control method is used.

In this method, the mutual interference of the secondary magnetic flux and the secondary driving current, that is, the cross term of the vector, is eliminated. Theoretically, there are three necessary sufficient conditions to make the vector of the secondary magnetic flux and the secondary drive current non-interfering. These conditions are usually met by adding a non-interfering operator to a conventional vector control device.

However, in the above-described non-interfering vector control method or apparatus, since the non-interfering calculator 2-3 converters are provided separately, the configuration of the apparatus becomes complicated.

Furthermore, in order to drive an induction motor exactly the same as a direct current, four upper limit operation (freely driving the motor in the forward and reverse directions) is required. However, in this vector control method, a triangular inertia (sine wave and cosine) is used to convert a primary voltage of a two-phase synchronic rotation coordinate into a primary voltage of two-phase fixed coordinates. Since par) is used, it is especially important when the above four quadrants are changed in order to continuously change the positive sign of these triangular water in correspondence with the rotational direction of the electric motor.

Moreover, pulse width modulated (PWM) inverters are often used to drive high torque three-phase induction motors. In this PWM inverter, a triangular wave signal is used to generate a modulated signal for a reference three-phase primary voltage. However, in the conventional vector control method and apparatus, the triangular wave signal must be synchronized with the supply voltage, so that the number of triangular wave signals output during one cycle of the supply voltage decreases as the frequency of the supply voltage decreases. . This causes a problem in that the response characteristics are degraded due to an increase in high frequency current and an increase in control delay or unnecessary time due to a long period of the triangular wave signal.

In addition, in the above-described PWM inverter control method, when the motor is stopped, the supply voltage frequency becomes zero and no triangular wave signal is generated. Thus, the PWM induces secondary magnetic flux before starting the motor. No signal is generated.

Since the second magnetic flux is induced with time delay after supplying the supply voltage, its operation response characteristic is not satisfactory.

SUMMARY OF THE INVENTION The present invention has been made in view of these problems, and a first object of the present invention is to provide a vector control method and apparatus for an induction motor having a simple device configuration by combining a non-interference calculator and a 2-3 phase converter.

Another object of the present invention is to provide a vector control method for an induction motor which enables four-quadrant operation by continuously changing the fixed sign of the triangular water in accordance with the rotational direction of the motor. In providing a device.

It is still another object of the present invention to provide the number of trivalent (carrier) wave signals generated during one cycle of the supply voltage to improve the response characteristics of the paddle at which the motor is rotating at a relatively low speed. The present invention provides a vector control method and apparatus for an induction motor having a PWM inverter that increases with a decrease.

Furthermore, another object of the present invention is to generate an initial secondary magnetic flux before starting the motor, so that when the motor is rotating at a near zero speed, an initial supply voltage signal of a predetermined frequency is applied to the triangular signal generator. The present invention provides a vector control method and apparatus for an induction motor having an inverter.

In order to achieve the above-mentioned first object, the method and apparatus of the present invention are based on the first reference current of the rotating α-β coordinate and the one of the fixed dq coordinates according to the calculation formula based on the positive and cosine waves. Has a procedure or means for calculating the difference voltage.

In order to achieve the above-mentioned second object, the method and apparatus of the present invention detects the rotational direction of the motor and generates an up or down direction corresponding to the direction in order to generate a sinusoidal cosine wave including information of the rotational direction of the motor. (down) has a level or means for outputting a signal.

In order to achieve the above-mentioned third object, the method and apparatus of the present invention generate a signal having a voltage level proportional to the magnitude of the supply voltage frequency, voltage-frequency convert the generated signal, and sequentially convert the converted signal. It divides and generates a plurality of switching signals according to the magnitude of the supply voltage frequency. When the supply voltage frequency is low, a signal having a small frequency of division is selected, and a signal having a large frequency of division is selected when the supply voltage frequency is high. And a procedure and means for selectively outputting the above-described switched signal corresponding to the switched signal.

In order to achieve the above-mentioned fourth object, the method and apparatus of the present invention provide a method or means for outputting a pulse signal of a predetermined frequency to a triangular signal generator when the supply frequency is zero to apply an initial secondary magnetic flux to the motor. Have

In the vector control method and device of induction motors, the combination of non-interference operation and 2-3 phase conversion simplifies the device configuration, and also enables four-quadrant operation by switching the government code of the triangular irrigation signal corresponding to the rotation direction. Furthermore, in the case of a three-phase induction motor driven by a PWM inverter, the number of triangular wave (carrier) signals generated during one cycle of the supply voltage signal increases with the decrease of the supply voltage frequency, and the motor is rotating at low speed. Improve the response characteristics when

Furthermore, when the motor is stopped, an initial supply of a predetermined frequency is applied to a triangular wave signal generator to generate an initial secondary magnetic flux before starting the motor.

It is a simplified model based on the biaxial theory of induction motors and can be treated mathematically. In this theory, harmonics, iron loss, magnetic saturation, etc. are all ignored, and the three-phase electric value including the secondary axis is converted into the two-phase electric value on the primary side under the condition that the signal is a trigonometric function. According to this two-axis theory, the voltage-current equation and torque on the primary side of the basket-type three-phase induction motor are expressed as follows on the α-β coordinates rotating in synchronization with the primary voltage.

Where the subscript α is the component on the α axis, the subscript β is the component on the β axis, 1 is the value on the primary side, 2 is the secondary value converted to the primary value, e is the voltage, i is the current, and λ is Magnetic flux, r is resistance M is mutual inductance, L is inductance, L _σ = (L ₁ L ₂ -M ² ) / L ₂ , P is differential or Laplace operator, P = d / dt, W _O is primary The instantaneous angle of the supply voltage vector or the angular frequency of the voltage-controlled power supply, W _r is the rotational angular frequency.Equation (1) shows the relationship between the primary voltage and the primary current and the relationship between the secondary voltage and the secondary magnetic flux. Included and displayed. This α-β coordinate is most suitable when driven by an induction motor with a voltage controlled power supply.

1 is an equivalent block diagram of a cage induction motor deployed on the α-β coordinate. The detailed description of the block diagram creation method of FIG. 1 is described in the following document. In other words, Onishi, Sugiura and Miyaji, “About Non-Interference Control in Induction Motor Drives,” Japanese IEE Report, RM-81, February 1981. It can be seen that the torque is proportional to the _differential current i _2α or i _2β if the secondary magnetic flux λ _2β or λ _2α is constant according to the above formula (2).

In order to control an induction motor like a DC motor, it is necessary to use a vector control method. In this method, the two vectors of secondary flux λ ₂ and secondary drive current i ₂ are controlled separately so that they are orthogonal to each other. The α and β axes may be determined in any way, but the α axis is preferably determined in the direction of the secondary magnetic flux and the β axis is determined in the direction of the secondary drive current i ₂ . In this way, the conditions under which λ ₂ and i ₂ are orthogonal are as follows.

The foregoing indicates that the secondary magnetic flux λ _2α is constantly controlled only on the α axis, and the secondary drive current i _2β is controlled variable only on the β axis, _thus becoming the same as the DC motor.

The condition of the above formula (3) can be obtained by controlling the slip angular frequency W _s as follows.

Where W _s is the slip angular frequency.

As described above, when the α and β axes are determined, the primary current α axis component i _1α corresponds to the secondary magnetic flux lambda _2α , and the primary current β axis component i _1β corresponds to the secondary drive current i _2β .

Equations (3) and (4) above are essential conditions for non-interference between the secondary magnetic flux and the secondary current, but the electromotive force for the mutual interference (cross term) between the α-axis component and the β-axis component on the primary side is It is not a perfect condition because it still remains.

The conditions of equations (3) and (4) apply here to the block diagram of FIG.

In Figure 1, point a is r ₂ = 0 and e _2β = 0,

Therefore, it becomes zero. At point b, lambda _2α is constant and becomes zero because Pλ _2α = 0. Point c is zero because λ _2β = 0. The point d is zero because i _2α = 0. Furthermore, for the part surrounded by the dotted line, A is calculated as follows.

On the other hand, the part B surrounded by the dotted line is calculated as follows.

e _1α = (calculate arrow B ₁ ) + (calculate arrow B ₂ ) = i _{1βL σ} W _O + ○ (since point a is zero)

As described above, the block diagram of the induction motor of FIG. 1 can be simplified as shown in FIG. 2 under the condition of Equation (3) (4).

_Referring to FIG. 2, (1) the secondary α-axis and the magnetic flux λ _2α are not determined solely by the primary α-axis e _α but the component + L _σ W _s i _β by the primary β-axis current i _β . Affected by interference, and (2) the secondary β-axis current i _2β is not determined solely by the primary β-axis voltage e _1β and is a component of the primary α-axis current i _1α -L ₁ W _s i _1α It can be seen that it is affected by the interference.

In order to solve this problem, the compensation of e ₁ β of the primary voltage e _1α is compensated in advance for the interference effect, that is, the secondary magnetic flux λ _2α is non-interfered from the primary voltage e _1β and the secondary drive current i _2β is It is preferable to make non-interference from the primary voltage e1 ( _alpha ).

As can be readily understood from FIG. 2, the non-interference control described above can be achieved by canceling these two intersection terms. That is, the primary voltages e _1α , e _1β can be achieved by combining the reference primary currents i _1α , i _2β with the following _erase values.

^{_{Here, e * 1α = i * 1α}} · r 1

e ^* 1 _β = i ^* _{1βr 1} + L _σ P

The upper point difference ^* is the reference or setpoint and is applied from the outside when the motor is driven.

The above expression represents a non-interfering condition. Gritty detail, is subtracted from the reference primary current i ^* 1 _β (2 secondary current i _2β the control) and the supply voltage angular frequency W _O L and coefficient value _σ obtained by rising to determine the primary voltage e _1α.

Similarly, in order to determine the primary voltage e _1β , a value obtained by multiplying the reference primary current i ^* _1α by the supply voltage angular frequency W _O and the coefficient L ₁ , the primary current i ^* based on L ₁ W _O i ^* _{1α The} value obtained by multiplying _1β and the primary resistance r ₁ is added to r ₁ i ^* _1β .

Since λ _2α is constant (or i _1α constant) in equation (4), the slip frequency W _s is proportional to the primary current i _1β .

In other words, equations (4) and (5) are sufficient to make the secondary magnetic flux and the secondary current non-interfering and drive the secondary magnetic flux λ _2α (i.e. i _1α ) constant when driving the induction motor with a voltage controlled power supply. Condition.

In order to realize non-interference control, the value of the primary current i _1α corresponding to the secondary magnetic flux λ _2α and the primary current i _1β corresponding to the secondary current i _2β are required as control information in addition to the rotational angular frequency W _r . . In actual control, it is practical to use a reference value or a command value.

Referring to FIG. 2, the reference primary current i ^* _1α can be obtained from the following equation based on the reference secondary magnetic flux λ ^* _2α . Similarly, the reference primary current I ^* _1β can be obtained from the following equation based on the reference torque T ^* and the reference secondary magnetic flux λ ^* _2α .

Because in Figure 2

Since the secondary reference flux λ ^* _2α is constant, the reference torque T ^* can be determined by the primary reference current i ^* _1β obtained through the PI speed controller (proportional branch). In this proportional integrator, the difference between the reference rotational angular frequency W ^* r and the actual rotational angular frequency Wr is integrated.

3A is an example of a block diagram of a non-interfering vector control apparatus of a three-phase induction motor. Induction motor 1A is driven by a primary voltage supplied from power amplifier 2A. The three-phase AC voltage 2a is applied to the amplifier 2A through a suitable rectifier 2b. The non-interference calculator 3 receives the primary reference current i ^* _1β (variable secondary current i _2β ) and calculates the non-interference value in order to output the primary voltages e _1α and e _1β according to equation (5). do.

Reference numerals 3 ₁ and 3 ₄ denote coefficient (r ₁ ) multipliers, and 3 ₂ and 3 ₅ denote multipliers.

Reference numerals 3 ₃ and 3 ₆ are inductance (L _σ or L ₁ ) multipliers. Reference section 4 is a speed sensor for detecting the rotational angular frequency (Wr) of the induction motor (1A). Reference numeral 5 is a proportional integrator (PI) that is the difference between the reference (target) rotational angular frequency (W ^* r) and the actual rotational angular frequency (Wr) to determine the primary reference current (i ^* _1β (i _2β )). To integrate.

Reference numeral 6 is a supply voltage angular frequency calculator that calculates the supply voltage angular frequency (W _O ), which is a calculator (6 ₁ ) and a coefficient (1 / τ ₂ ) multiplier (1 / τ ₂ ) that calculates the primary reference current (i ^* _1α ). 6 ₂ ). The supply voltage angular frequency W _O of equation (4) is calculated by the calculator 6 as follows.

Since T ₁ = L ₂ r ₂

Reference numeral 7A is a two-phase converter that generates a three-phase reference voltage (e ^* _a , e ^* _b , e ^* _c ) based on an _{uninterrupted} primary voltage (e _{1αe 1 β} ) and a trigonometric function. .

The reference voltages e ^* _a , e ^* _b , e ^* _c are obtained from the following equation.

Equation (6) indicates that the primary voltages e _1α , e _1β on the two-phase synchronous rotation α-β coordinates can be converted into two-phase fixed dq coordinates.

Equation (7) shows that the primary voltages (e _1d , e _1q ) on the two-phase fixed dq coordinates can be converted to the usual fixed three-phase primary reference voltages (e ^* _a , e ^* _b , e ^* _c ). It is displaying.

Reference numeral 8 is a sinusoidal wave generator, which will be described later in detail.

The power amplifier 2A comprises three independent amplifiers 2 _1A , 2 _2A , 2 _3A and the primary reference voltage signals e ^* _a , e ^* _b , e ^* _c ) for driving the induction motor 1A. Amplify each.

Generally, three-phase induction motor 1A is used, but when a relatively small torque is required, two or more induction motors 2B can also be used as shown in FIG. 3b.

In this case, another coordinate converter 7B is used.

That is, the primary voltages e _1α and e _1β are converted into primary voltages e ^* _1d and e ^* _q without converting the two-phase primary voltage into a three-phase primary voltage. The calculated reference signals e ^* _a , e ^* _b , e ^* _c are all amplified directly by the power amplifiers 2 _1β and 2 _1β , respectively, and drive the two-phase electric motor 1B.

Moreover, the two-phase power supply 2a 'can be used in the apparatus of FIG. 3B.

3c is a block of a non-interfering vector device for a three-phase induction motor coupled with a PWM inverter 2C. In such a device, a triangular wave signal generator 9 is further included. The PWM inverter 2C has three e ^* _a , e ^* _b , e ^* _c PWM (pulse width modulation) based on the reference voltage (e ^* _a , e ^* _b , e ^* _c ) and the triangular wave signal (Tr ₁ ). Generates the signal. In order to generate the PWM signal, the voltage level of each of the reference signals is individually compared with the voltage level of the triangular wave signal. The pulse width modulated pulse is generated only while the reference voltage signal exceeds the triangular wave signal at the voltage level. More specifically, during the H (ON) level time of the PWM signal, it starts from when the instantaneous voltage level of the reference signal exceeds the level of the triangular wave signal until it is enhanced below the level of the triangular wave signal. Therefore, the higher the reference voltage signal voltage level is, the longer the pulse width of the PWM signal is. In response to these PWM signals, a switching element such as an SCR in the inverter 2C is fired to generate a primary voltage signal. The non-interfering vector control condition state between the secondary flux and the secondary drive current is generated by this signal. Under the control, the speed and torque of the induction motor 1A are controlled.

Based on the above description, the 1st characteristic of this invention is demonstrated below.

The first feature is that the configuration of the device possible in the non-interfering vector control device is simplified.

In the conventional device configuration shown in Figs. 3A and 3C, the non-interference calculator 3 and the two-phase converter 7A are separately provided to calculate equations (6) and (7). However, in the first embodiment, two calculations are performed at the same time. More specifically, the reference voltages (e ^* _a , e ^* _b , e ^* _c ) applied to the voltage controlled inverter 2C are directly calculated with respect to equation (6) as follows.

Equations (8) and (9) above describe the non-interference condition and the primary voltage on the fixed dq coordinates including the non-interference condition of equation (5) without separately calculating the conversion from the α-β coordinate to the dq coordinate. It can be directly obtained from the primary reference current (i _1α , i _1β ).

4 is a detailed block diagram of the main part of the apparatus of the present invention. The non-interfering arithmetic operator (3D) has four proportional differentiators (3 _1D , 2 _2D , 3 _3D , 3 _4D ), four multipliers (5 ₅ , 3 _6D , 3 _7D , 3 _8D ) and two integrators ( 3 _9D , 3 _10D ), and e _1d or e _1q are obtained according to equation (8) (9).

Proportional differentiation (3 _1D- 3 _4D ) multiplies sin W _O t or cos W _O t with (r ₁ + L _σ S) or (r ₁ + L ₁ S). An example of the differentiator using the operational amplifier OA is shown in FIG. 5A. The transfer function determined by the feedback resistance R _f , the input resistance R ₁ , and the input capacitance C is determined to match the coefficients r ₁ and L ₁ or L _σ , respectively.

A multiplier (3 _5D- 3 _8D ) calculates the product of the proportional differential output and the primary reference current (i ^* _1β ) or (i ^* _1α ). The integrators 3 _9D and 3 _10D are the first order delay elements. These elements are provided to remove high frequency noise components superimposed on the positive and cosine wave signals. An example of these elements using an operational amplifier is shown in FIG. 5B.

The time constant T (= CR) of the first delay integrator is determined to be sufficiently small compared with the constant and cosine wave signals. That is, since the time constant is small enough to remove high frequency components due to noise, the influence of the delay on the induction motor in the control frequency range may be negligible. These noise canceling elements are necessary because high frequency noise components are included in the positive and cosine wave signals when the static and cosine wave signal generators are digitally configured.

By the non-interference calculator 3D described above, voltages e _1d and e _1q can be obtained according to equations (8) and (9).

The two-phase converter 7d includes a first inverting amplifier 7 _1D having a gain of 1/2, a second amplifier 7 _2D having a gain of 2/2, and a third inverting amplifier 7 _3D having a gain of 1. And the triple reference control voltages e ^* _a , e ^* _b , e ^* _c corresponding to dq coordinate primary voltages e _1d , e _1q are calculated according to equation (7). More specifically, e ^* _a is obtained directly from the non-interfering operator 3D, and e ^* _b is the output of the first inverted heavy amplifier 7 _1D (-l / 2 e _1d ) and the second amplifier 7 _1D. Is obtained by adding the output (1 3 / 2e _1q ) of the third inverting amplifier 7 _3D connected to), and e ^* _c is the output (-l / 2e _1d ) of the first inverting amplifier 7 _1D It is obtained by adding the output 3 / 2e _1q of 2 amplifier 7 _2D .

)를 발생하도록 상기 3개의 기준전압(e^* _a, e^* _b, e^* _c)은 3각파 발생기 (9)로부터 출력되는 3각파 신호(Tr₁)의 전압레벨과, 인버어터 (2C)내에 구성된 3개의 비교기 (2₁, 2₂, 2₃)에 의해서 비교된다.Three PWM control signals (B _a , B _b , B _c ) which separately call the three main switching elements of (2C) or (

The three reference voltages e ^* _a , e ^* _b , e ^* _c ) are generated in the inverter 2C and the voltage level of the triangular wave signal Tr ₁ output from the triangular wave generator 9. Comparison is made by means of three comparators 2 ₁ , 2 ₂ , 2 ₃ configured.

Next, the second feature of the present invention will be described. Like the DC motor, four quadrant operation is required to drive the induction motor. This four-quadrant operation will be described below. As shown in FIG. 6A, when the rotation angle frequency Wr is taken as the horizontal axis, and the motor torque T is taken as the longitudinal axis, the motor is rotated in the forward rotation direction at the first upper limit. It is driven, braking while rotating in the forward rotation direction at the second upper limit, driven in the reverse rotation direction at the third upper limit, and braking while rotating in the reverse rotation direction at the fourth upper limit.

In other words, as shown in FIG. 6B, the torque T is positive at the first upper limit, the rotational angular frequency Wr is positive, and at the second upper limit, T is negative, and Wr is positive. In the third upper limit, T is negative, Wr is negative, and in the fourth upper limit, T is negative Wr.

FIG. 6C is an example when the motor during forward rotation is switched in the reverse direction at time t ₁ . When the vibrator is rotating at the first upper limit (T> ○, Wr> ○), if the reference frequency (speed) + W ^* r is switched to -W ^* r, the motor operates in the second upper limit operation (T <○, Wr>). ○) (the motor is braked or the torque is absorbed).

When the rotational speed becomes zero, the third upper limit operation (T> ○, Wr <○) (drives in the reverse direction) is performed.

In the non-interfering vector control device of the induction motor according to the present invention, in order to perform the four-quadrant operation, the actual or reference motor angular frequency (speed) (Wr or W ^* r) is determined from positive rotation to positive, and reverse rotation. Is determined to be negative.

Under these conditions, the primary reference current (i ^* _1β ) (corresponding to the secondary drive current) is always positive when the reference angular frequency (W ^* r) is higher than the actual angular frequency (Wr) regardless of the rotational direction of the motor. Becomes That is, + I ^* _1β represents positive reference torque, and slip angular frequency Ws = W _O -Wr (W _O is supply voltage angular frequency) is also positive. In contrast, -i ^* _1β is always negative when less than W ^* r. That is, -i ^* _1β represents negative reference torque, so Ws is also negative.

In short, all the angular frequency ^{(Wr, W * r, W} O, Ws) is treated in consideration of the information-part code.

Next, the third feature of the present invention will be described. The third feature is that the ratio of the positive or cosine wave signal frequency to the triangular signal frequency is adjusted corresponding to the magnitude of the supply voltage angular frequency W _O while synchronizing the positive or cosine wave signal to the triangular wave signal. The generator 8 and the triangular wave generator 9 are constituted. In other words, the number of triangular wave (carrier) signals generated in one cycle of the supply voltage signal is increased as the supply voltage frequency decreases, thereby improving the response characteristics when the motor is rotating at low speed.

In order to obtain a PWM signal with three reference voltages (e ^* _a , e ^* _b , e ^* _c ) and a triangular wave signal (Tri), these signals must be synchronized with each other. Otherwise, direct current components and even-order harmonics are easily generated by modulation, causing torque fluctuations.

In order to synchronize the three-phase reference voltage with the triangular signal, the sinusoidal sin W _O t and the cosine wave cos W _O t used in calculating the reference voltage must be synchronized with the triangular signal Tri. In other words, the triangular wave signal of the positive and cosine wave must be generated by a clock pulse generated in synchronization with each frequency W _O of the supply voltage.

However, in the conventional PWM controller information or one cycle 3 may have more that since the fixedly determined by the supply voltage angular frequency (W _O) (motor speed) lower triangular wave signal frequency of the triangular wave signal generated in a cosine-wave signal is also Lowers.

Since the longer the period of the triangular signal, the longer the unnecessary time (the longer the OFF time of the PWM signal), the lower frequency triangular signal has an adverse effect on the response speed in controlling the speed of the induction motor. That is the supply voltage frequency (W _O) lower the lower the response speed is lowered, and there arises a problem that the harmonic current increases further.

7 is an embodiment of the sin / cos wave signal generator and the triangular wave generator of the present invention. The triangular wave signal generator comprises an absolute value detector 11, a VF converter 12, a divider 13, 14, 15, a switching unit 16, a switching signal generator 17, and an up-down ( up-down counter 18, DA converter 20, inverting amplifier 21, switch 22, and another divider 23.

The absolute value detector 11 generates a voltage signal proportional to the magnitude of W _O regardless of the positive and negative values of the supply voltage angular frequency W _O.

Positive W _O values are obtained when the motor rotates forward, while negative W _O values are obtained when the motor rotates reversely.

The V-F converter 12 converts the output signal of the absolute value detector 11 into a signal having a frequency proportional to its voltage level. Therefore, the higher the supply voltage frequency is, the higher the frequency of the pulse signal output from the V-F converter t2 is.

The dividers 13, 14, and 15 divide the pulse signal frequency of the VF converter 21 in half. Switching unit 16 is switched in response to the VF converter 12 and the divider pulse signal from the 13, 14 and 15, the switching signal generator, the output 17 from the switching signal (AS _O).

In FIG. 7, the double lines indicate a plurality of conductive lines.

When the supply voltage frequency W _O is very low, the switching signal generator 17 outputs the first switching signal AS ₁ to close the contact 16 ₁ of the switching unit 16, and the supply voltage frequency W _O. ), the second switching signal (AS ₂₎ to output them to output the contact (16 ₂₎ the third switching signal (AS ₃₎ closed and, when the middle the supply voltage frequency (W _O) of a contact (16 ₃ when low ), And when the supply voltage frequency W _O is high, the fourth switching signal AS ₄ is output to close the contacts 16 ₄ , respectively.

I.e., outputs the switching signal generator 17 is supplied to the voltage-frequency (W _O), a switching signal (AS ₁ -AS ₄₎ apart to correspond to the size.

The up-down counter 18 repeatedly adds or subtracts the clock pulse signal of the switching unit 18 and simultaneously outputs a carry signal to the up-down switching terminal VD of the counter itself. For example, when the counter 18 is composed of 3 bits, when the count signal of 8 = 2 ³ is added and a carry signal is generated and the counter 18 is switched, the counter 18 subtracts the pulse signal.

As indicated by the dotted line in the figure of FIG. 8A, when the counter 18 subtracts the pulse signal of 8, the pulse signal is added to the next. The DA converter 20 converts the digital signal of the up-down counter 18 into an analog signal. The inverting amplifier 21 whose gain is? 1? Inverts the output of the DA converter 20. The switch 22 selects an output signal of any one of the inverting amplifiers 21, in response to the switching signal AS ₁₀ from the divider 23, and selects a triangular wave (carrier) signal (Tri). ) This is because the up-down counter 18 repeatedly outputs the half wave of the triangular wave signal, as shown by the solid line in FIG. 8a, each half wave signal is alternately rotated, i.e., for each half period of the triangular wave signal of the DA switch 20. This is because it needs to be reversed.

The sin / cos wave signal generator includes a rotation direction detector 24, another up-down counter 25, a divider 26, two ROMs 27 and 28, and two DA converters 29, 30, two inverting amplifiers 31 and 32, two switches 33 and 34, and two dividers 35 and 36.

The rotation direction detector 24 detects the rotation direction of the motor and generates an up signal when the motor rotates in the forward direction and a down signal when the motor rotates in the reverse direction.

The up-down counter 25 counts the pulse signal supplied from the divider 15 through another 1/6 divider 26 in response to the signal from the rotation direction detector 24. In other words, the counter 25 adds a pulse signal when the motor rotates forward and subtracts the pulse signal when the motor rotates backward.

The ROM 27 stores sample data for half a period of the cosine wave and continuously outputs sample data one by one in response to the count signal of the up-down counter 25. Furthermore, the ROM 27 outputs the signal indicated by the cos [pi] signal after all of the half cycle sample data is output. For example, when the ROM 27 stores eight sample values, the cos [pi] signal is output after seven sample values are output.

The DA converter 25 converts the digital value of the ROM 27 into an analog value. The inverting amplifier 31 of gain 1 inverts the output value of the DA converter 29. The switch 33 selects one of the output values of the DA converter 25 or the inverting amplifier 31 and outputs the cosine wave signal in response to the cos pi signal AS ₂₀ output from the frequency divider 35. . This is because since the ROM 27 repeatedly outputs the half wave of the cosine wave, it is necessary to invert each of the half wave signals alternately, that is, inverting the cosine wave signal of the DA converter 29 for each half period, as indicated by the solid line of FIG. Because.

Similarly, the RPM 28 stores sample values for half a period of the sine wave, and outputs sample values one by one in succession in response to the count signal of the up-down counter 25. Furthermore, the ROM 28 outputs a sinπ signal after all half-cycle sample values have been output.

For example, when the ROM 28 stores eight sample values, the sin? Signal is output after seven sample values are output.

The DA converter 30 analog-converts the digital value of the ROM 28. The inverting amplifier 32 of gain 1 inverts the output value of the DA converter 30. In response to the sin pi signal AS ₃₀ output from the divider 36, the switch 34 selects either of the output values of the sinusoidal signal sin w _O t by the DA converter 30 or the inverting amplifier. Outputs

This is because since the ROM 28 repeatedly outputs the sine wave half wave, it is necessary to invert each half wave signal alternately, that is, for each half cycle of the sine wave signal of the DA converter 30, as indicated by the solid line of FIG. . Furthermore, the sinπ signal is applied to the terminal of the up-down counter 25 to repeat the above-described operation. This is because the ROM 28 stores only the sample data for the half period of the positive and cosine waves.

The second characteristic (four quadrant operation) of the present invention is achieved as follows.

Rotational direction detector 24 detects the sign of the supply voltage frequency (W _O). W is _O is positive, because the motor is rotated in the forward rotation detector 24 is an up-and outputs an up (addition coefficients) signal to the down counter 5. In this case, the counter 25 adds a clock signal and outputs respective addressing signals to the ROMs 27 and 28 for generating respective sample data of the sine or cosine wave. The result is the + sin W _O t and + cos W _O t signals. Therefore, the reference primary voltages e _1d and e _1q for forward running the motor are calculated based on Equations (8) and (9). If W _O is negative, this means that the motor rotates in reverse rotation, so the detector 24 outputs a down (subtraction factor) signal to the up-down counter 25. In this case, the counter 25 subtracts the clock signal and outputs each addressing signal to the ROMs 27 and 28 for generating respective sample data of the sine wave or the cosine wave. As a result, sin (-W _O t) = sin W _O t and cos (W _O t) = cosW _O t signals are generated. Therefore, the reference primary voltages e _1d and e _1q for reverse rotation of the motor are calculated based on Equations (8) and (9). In short, the rotation of the motor can be switched from the forward direction to the reverse or vice versa only by switching the counter 25 from the addition coefficient to the subtraction coefficient or vice versa. Furthermore, it is notable that the initial values of the sine and cosine waves do not change when the addition or subtraction coefficients are switched, so that the rotation direction of the motor can be smoothly changed.

A third feature of the present invention (relationship between the triangular wave signal Tri and the trigonometric function signal) will be described below using numerical values. If the sample data for the half period of the sine wave is quantized to 112 and stored in the ROM 28, one cycle of the sine wave signal is output when the up-down counter 25 counts 224 clock waves. Furthermore, when the counter 18 counts the clock pulses associated with the supply voltage frequency W _O of 16 (= 2 ⁴ ), a quarter period of the triangular wave signal is output. That is, when the counter 18 counts 64 (16x4) clock pulses, one period of the triangular wave signal is output. Since the supply voltage frequency data W _O is simultaneously input to the two counters 18 and 25 as clock pulses, the pulse ratio (occurring for one cycle) of the sine wave of the triangular wave is 2 (64) to 7 ( 224).

Since the frequency data W _O input to the counter 18 is switched by the switching unit 16 corresponding to the magnitude of W _O , the above-described pulse ratio of the sine wave of the triangular wave changes corresponding to W _O. For example, when the contact 16 ₁ is closed, since the clock pulses input to the counter 25 are always divided into 1/48 (1/8 x 1/6), the number of input pulses of the counter 18 and the counter 25 The ratio of the number of input pulses is 1: 1/48 or 48/2: 1/7 or 168: 1. In other words, 168 [(48 x 7) / 2] triangular waves are output. In other words, the number of triangular wave signals generated during one period of the positive and cosine wave signals decreases with increasing supply voltage angular frequency W _O.

Thus in the composed function generator can supply the voltage spread is divided into a (W _O) a plurality of range (in particular micro, small, medium, and high), and the reduction of the number W _O of the triangular wave generated during one period of the W _O Since the frequency increases, the frequency of the triangular (carrier) wave signal Tri can be increased freely, thereby improving the response characteristic of the motor speed control and reducing the harmonic current.

Next, a fourth feature of the present invention will be described.

The fourth feature is to apply an initial supply voltage signal having a predetermined frequency to the triangular wave generator 9 when the supply voltage frequency W ₀ is zero, thereby initializing the generator 9 to impart an initial magnetic flux before starting the motor. will be.

In a conventional PWM control when the motor speed is zero or the torque reference 01 a supply voltage frequency (W _O) is do not occur spirit and triangular wave signal. Therefore, it is not possible to generate a PWM signal that maintains the secondary magnetic flux before starting the motor. In other words, since the secondary magnetic flux cannot be generated instantaneously due to the inductance during motor starting, the starting characteristics deteriorate due to the delay of the secondary magnetic flux generation.

In FIG. 9, the pulse generator 37 is further connected to the _fifth contact 1625 of the switching unit 16. Therefore, when the motor stops and the voltage frequency W ₀ is zero, the fixed frequency pulse signal W ₁ is applied to the up-down counter 18 and a triangular wave signal Tri having a fixed frequency is generated.

Next, the reason why the secondary magnetic flux λ ₂ can be generated even during the motor stop when the reference primary current i ^* _1α is applied when the triangular wave signal having the predetermined frequency is generated will be explained.

Given the conditions,

The reference voltages (e ^* _a , e ^* _b , e ^* _c ) are DC voltages. Therefore, the PWM signal is obtained between the DC voltage and the fixed-frequency triangular wave signal, so that the primary initial reference voltage is applied to the motor. That is, the initial secondary magnetic flux occurs. Since the secondary magnetic flux is always generated before starting the motor, there is no delay caused by the operation start time of the secondary magnetic flux and the motor can be started instantaneously. Furthermore, in this case, the start response characteristic is determined only by the operation start time of the primary current i ^* _1β corresponding to the secondary drive current i ₂ . Moreover is the DC signal containing the modulation component exists due to the triangular wave signal (Tri) does not signal (Tri) is synchronized with the voltage frequency (W _O) when the result in response to the pulse signal of the pulse generator (37).

In this case, however, triangular wave frequency is relatively set to a high value, and W _O is the number of the triangular wave signal contained in one period of the positive and cosine wave signal at zero since there adverse influences of the direct current component in practice is negligible have.

As described above, in the non-interfering vector control apparatus for driving an induction motor of the present invention, since the non-interference calculator 3 and the 2-3 phase converter 7 are combined, the device configuration can be simplified.

It is possible to switch the motor from forward rotation to reverse rotation and vice versa because the quadrant operation can be performed by continuously changing the positive sign of the trigonometric function signal corresponding to the rotation direction of the motor.

In addition, the frequency ratio of the triangular wave (carrier) signal and the trigonometric function signal can be adjusted to correspond to the magnitude of the supply voltage frequency, so that the response characteristics of the speed control can be improved and the harmonic current can be excluded, especially in the low speed region. have.

In addition, since the fixed frequency triangular wave signal is generated during the motor stop, three reference voltages e ^* _a , e ^* _b , e ^* _c corresponding to the second magnetic flux can be generated and the motor can be started at a high response speed.

Claims (10)

The speed sensor 4 which detects the rotational angular frequency (Wr) of the induction motor and the secondary reference drive current corresponding to the difference between the reference rotational angular frequency (W ^* r) and the detected rotational angular frequency (Wr) A proportional integrator 5 generating a primary reference current i ^* _1β to obtain a reference torque, a primary reference current i ^* _1β , a primary reference current i i corresponding to the secondary magnetic flux, i ^* _1α , Supply voltage angle for calculating the primary supply voltage angular frequency (W _O ) based on the detected rotational angular frequency (Wr) and the ratio (τ ₂ ) of the secondary inductance (L ₂ ) and the secondary resistance (r ₂ ) A frequency calculator 6, a sin / cos signal generator 8 for generating a sinusoidal wave and a cosine wave having the calculated angular frequency W _O , a primary reference current i ^* _1α , i ^* _1β and and a computing circuit _(D 3, _D 7) for calculating dp coordinate primary voltage e _1d, e _1q in accordance with the food is based on the cosine wave signal,

In the non-interference vector control apparatus of an induction motor including power amplifying means (2A, 2B) for driving an induction motor based on the calculated primary reference voltages e _1d , e _1q , the signal generator 8 is (a). A rotation direction detector 24 which detects the rotational direction of the motor to generate an up or down signal, and (b) a clock which is synchronized with the supply voltage signal of each frequency W _O in response to the up or down signal of the rotational direction detector. A first up-down counter 25 for adding or subtracting pulses, (c) a first ROM 27 for continuously outputting sample data of the cosine wave in response to the count signal of the first counter, and (d A second ROM 28 for continuously outputting sine wave sample data in response to the count signal of the first counter, and (e) a first D for converting digital sample data output from the first ROM into an analog value of the cosine wave signal. -A converter (29), and (f) the second ROM. Is achieved by including a first 2D-A converter 30 for converting the digital sample data outputted from the analog value of the sinusoidal signal, the arithmetic circuit

Where r ₁ is the primary resistance, L ₁ is the primary inductance, L _σ is the equivalent leakage inductance (L ₁ L ₂ -M ² ) / L ₂ , M is the mutual inductance, and S is the Laplace operator. first calculation means _3D for calculating the dq coordinate primary voltages e _1d and e _1q , and

Second operation means for calculating the three-phase primary reference voltages e ^* _a , e ^* _b , e ^* _c based on the primary reference voltages e _1d , e _1q calculated for driving the three-phase induction motor according to the equation (7 _D) to include, and triangular wave signal generating means (9) are sin / cos wave signal and being adapted to generate a triangular wave signal that is synchronous, the power amplifier is a calculated three-phase primary voltages e ^* _a, e PWM inverter means 2 for generating three pulse-modulated control signals in response to ^* _b , e ^* _c and triangular wave signals; The triangular wave generating means 9 includes (a) an absolute value detector 11 for generating a signal of a voltage level proportional to the magnitude of the supply voltage angular frequency W _O regardless of a positive or negative sign, and (b) the A VF converter 12 for converting an output of an absolute detector into a signal having a frequency proportional to its output voltage level, and (c) a plurality of serially connected dividers 13 and 14 for dividing a frequency of a signal of the VF converter. 15), (d) a switching signal generator 17 for outputting a plurality of switching signals according to the magnitude of the supply voltage angular frequency W _O , and (e) a conversion signal having a small frequency division frequency when the supply voltage frequency is low. When high, the converter 16 selectively connected to the serially connected divider for selectively outputting the output signal of the VF converter so that the converted signal having a large frequency division frequency is output, and the carrier signal generated in (f) itself. A second to add or subtract the converter signal by And an up-down counter (18) and (g) a third DA converter (20) for converting the counted digital value to an analog value. The method according to claim 1, wherein the first calculation means (3 _D ) comprises: (a) first proportional difference means (3 _1D ) for calculating a first product of a sine wave and (r ₁ + L _σ S), and (b) A second proportional differential means (3 _2D ) for calculating a second product of the cosine wave and (r ₁ + L ₁ S), and (c) a third proportional derivative for calculating a third product of the cosine wave and (r ₁ + L _σ S) Fourth proportional means 3 _{4D for} calculating a fourth product of the means 3 _3D , and (d) the sinusoidal wave and (r ₁ + L ₁ S), and (e) the first product (r ₁ + L _σ S ) a first multiplier (3 _5D ) that multiplies sin W _O t by the primary reference current i ^* _1β , and (f) the second reference (r ₁ + L ₁ S) cos W _O t is the primary reference current i ^* A second multiplier 3 _6D multiplying _1α , (g) a fourth multiplier 3 _7D multiplying the third (r ₁ + L _σ S) cos W _O t by the primary reference current i ^* _1β and (h) a fourth multiplier (3 _8D ) that multiplies the fourth quadrature (r ₁ + L ₁ S) sin W _O t by the primary reference current i ^* _1α , and (i) the first multiplication result (r ₁ + _{L σ S) sin W O t} · i * 1β to the second multiplied result (r ₁ + L ₁ S) fixed dq left by subtracting from _O W t · sin i ³ _1α And a subtracter for obtaining the voltage e _1q, (j) the third multiplied result _{(r 1 + L σ S)} cos W O t. Non-interfering vector control device for induction motors, comprising: an adder which adds i ^* _1β and the fourth multiplication result (r ₁ + L ₁ S) sin W _O t · i ^* _1α to obtain a fixed dq coordinate voltage e _1q . The method of claim 2 wherein said second computing means (7, _D) is, (a) in response to the voltage e _1d signal (-l / 2 e _1d) a first inverting amplifier (7 of the gain (gain) 1 for generating a _1D ) and (b) a signal in response to the voltage e _1q

the amplifier gain 2 for generating an e _1q) (7 _2D) with, (c) Voltage

e in response to _1q

a second inverting amplifier 7 _3D of gain 1 generating e _1q ), (d) signal (−1/2 e _1d ) and signal (−

and a second adder for generating a reference voltage e ^* _b by adding e _1q ). 3. The first operating means (3 _D ) according to claim 2, characterized in that (a) a first integrator (3 _9D ) having a first order delay in response to the signal sin W _O t and (b) the signal cos W _O t. A high frequency component formed of a second integrator 3 _10D having a first order delay in response to the second integrator, wherein the time constant of the first and second integrators is sufficiently smaller than the supply voltage angular frequency W _O and overlaps the Non-interference vector control device of induction motor, characterized in that to remove. 2. The pulse generator (37) according to claim 1, wherein the pulse generator (37) outputs a pulse signal of a predetermined frequency to the switch (16) to apply an initial secondary magnetic flux when the supply voltage angular frequency is substantially zero or the motor is stopped. Non-interference vector control device of an induction motor comprising a. Detects the rotational angular frequency Wr of the induction motor, and the primary reference current i ^* corresponding to the secondary reference driving current based on the difference between the reference tortor angular frequency W ^* _r and the detected rotational angular frequency Wr to generate the reference torque _1β is determined, and the primary reference current i ^* _1β , the constant primary reference current i ^* _1α corresponding to the secondary magnetic flux, the rotation angle frequency Wr, the ratio of the secondary inductance L ₂ and the secondary resistance r ₂ τ ₂ The primary supply voltage angular frequency W _O is calculated on the basis of, and the positive and cosine waves of the angular frequency W _O are generated respectively, and the fixed reference signal is fixed based on the primary reference currents i ^* _1α , i ^* _1β , In the vector control method of an induction motor comprising generating a coordinate primary voltage e _1d , e _iq and driving the induction motor based on the calculated primary reference voltages e _1d , e _iq . The step of (a) detecting the rotational direction of the motor and up or corresponding to that direction Generating an operating signal and, (b) up or in response to a down adding or subtracting counts the clock signal is synchronized with the supply voltage signal of the angular frequency W _O, and (c) in response to a count signal to the sample data on the cosine wave Output continuously, (d) sine wave sample data in response to a count signal, (e) convert the sample data of the cosine wave signal into an analog signal, and (f) convert the sample data of the sine wave into an analog signal. To convert,

Where r ₁ is the primary resistance, L ₁ is the primary inductance, L _σ is the equivalent leakage inductance (L ₁ L ₂ -M ² ) / L ₂ , M is the mutual inductance, and S is the Laplace operator. Calculate the dq coordinate primary voltages e _1d and e _1q ,

And calculating the three-phase primary reference voltages e ^* _a , e ^* _b , e ^* _c based on the primary reference voltages e _1d and e _1q calculated for driving the three-phase induction motor according to the equation And generates a triangular wave signal having an angular frequency (W _O ) synchronized with the sin / cos wave signal, and calculates the triphase primary voltages e ^* _a , e ^* _b , e ^* _c and the generated triangular wave signal. And in response to generating three pulse-modulated control signals, wherein the triangular wave generating step comprises: (a) a signal whose voltage level is proportional to the magnitude of the supply voltage angular frequency W _O indicates a direction of rotation; part of occurrence, regardless of the sign, and (b) VF converted into a signal of a frequency proportional to the voltage level of the generated signal, and dividing the frequency of (C) signals continuously (d) supply voltage gaksu frequency W _O A plurality of switching signals are generated corresponding to the magnitude, and (e) when the supply voltage frequency is low, the frequency of division is small. When the signal is high, the VF-converted signal is selectively outputted so that a signal having a high frequency of division is output, (f) the VF-converted signal is repeatedly added and subtracted in response to the carrier signal, and (g) digitally counting. The vector control method of the induction motor, characterized in that it further comprises the step of analog conversion. The method of claim 6, wherein the calculating of the fixed dq coordinate primary voltages e _1d , e _1q comprises: (a) calculating a _{first product of the} sinusoidal signal and (r ₁ + L _σ S), and (b) the cosine wave signal. Calculate the second product of and (r ₁ + L ₁ S), (c) calculate the third product of the sinusoidal signal and (r ₁ + L _σ S), and (d) the sinusoidal signal and (r ₁ + L ₁ S ), And (e) multiplies the first product (r ₁ + L _σ S) sin W _O t by the primary reference current i ^* 1 _β , and (f) the second product (r ₁ + L Multiply ₁ S) cos W _O t by the primary reference current i ^* _1α , (g) multiply the third (r ₁ + L _α S) cos W _O t by the primary reference current i ^* _1β , h) The fourth product (r ₁ + L ₁ S) sin W _O t is multiplied by the primary reference current i ^* _1α , and (i) the first multiplication result (r ₁ + L _σ S) sin W _O t · i ^* _{1β is obtained} from the second multiplication result (r ₁ + L ₁ S) cos W _O t · i ^* _{1α to} obtain a fixed dq coordinate voltage e _1d , and (j) the third multiplication result (r ₁ + L _σ S) cos W _O t · i ^* _1β to the fourth multiplied result (r ₁ + L ₁ S) by adding a sin W _O t · i ^* _1α to obtain the fixed dq coordinate voltage e _1q The vector control method of the induction motor comprising a step. The method of claim 6, wherein calculating the primary reference voltages e ^* _a , e ^* _b , e ^* _c includes (a) calculating (-l / 2 e _1d ) based on the primary voltage _e1d , b) based on the primary voltage e _1d (

e _1q ), and (c) the calculated (

e _1q ) into (-

e _1q ) and (d) the calculated value (-l / 2 e _1d ) to obtain the reference voltage e ^* _b (-

e _1q ) and (e) calculate the calculated value (-l / e _1q ) to obtain the reference voltage e ^* _c (

e _1q ) to the vector control method of the induction motor, characterized in that it comprises a step. The method of claim 6, wherein the calculating of the fixed dq coordinate primary voltages e _1d , e _1q comprises: (a) integrating the signal sin W _O t as the primary delay, and (b) the signal cos W _{O as the} primary delay. integrating t and removing the high frequency components superimposed on the positive and cosine wave signals so that the delay time is sufficiently small compared to the supply voltage angular frequency W _O. Way. 7. An induction motor according to claim 6, comprising outputting a pulse signal of a predetermined frequency to apply an initial secondary magnetic flux when the supply voltage angular frequency is substantially zero or the motor is stopped. Vector control method.

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