JPH11205261A - Space diversity type receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide the space diversity type receiver with a simple circuit configuration. SOLUTION: Antennas 1a, 1b receive a frequency multiplexed signal from blocks of a plurality of channels, and a processing section 20a applies a frequency conversion to the 1st frequency multiplexed signal of a plurality of channels received by the antenna 1a to generate a block so that its spectrum is placed at a positive frequency region, while a processing section 20b applies a frequency conversion to the 2nd frequency multiplex signal of a plurality of channels received by the antenna 1b to generate a block so that the spectrum is placed at a negative frequency region. A synthesizer 22 synthesizes two received conversion signals, while a channel separation filter bank 10 use different circuits to process two signals corresponding to the antennas 1a, 1b for the signals after the synthesis to separate channels and each multiplexer 21 selects a signal of two same channels which is larger.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention separates a frequency multiplexed signal of a plurality of channels received by a space diversity type antenna apparatus having at least two antennas into a signal of each channel. And a space diversity type receiving device for receiving using a channel separating filter device.

[0002]

2. Description of the Related Art A first conventional personal handy phone system (hereinafter referred to as a PHS system) which is currently used and is called a so-called simplified portable telephone system.
Uses a band that uses one carrier for one base station, as shown in FIG.
Here, the time division multiple access-time division duplex method (hereinafter, referred to as
It is called TDMA-TDD system. ) Is used and 1
A maximum of three mobile terminals can be used simultaneously for one base station. Since an actual PHS system requires a channel for line control, so-called ping-pong transmission is performed using four channels in one direction, as shown in FIG.

In a second prior art PHS system,
In order to increase the number of terminals in one service area, as shown in FIGS. 19A and 19B, a plurality of carriers CA1, CA2, and CA3 are used. In other words, one carrier is used. The number of base stations is increased to more than one within one service area. Here, the frequency of the carrier wave used by the base station is a predetermined 7 when the base station or the mobile terminal station calls.
An empty channel among the seven channels is used. In the conventional PHS system, a band-pass filter is used to separate each channel, and
/ 4 shift QPSK (Quadrature Phase Shift Keyin)
The differential detection method is widely used for demodulating the modulated signal of g).

However, in the first and second conventional radio systems using frequency multiplexing channels, when a bandpass filter is used to separate each channel, the circuit is large because an inductance circuit is used in the bandpass filter. In addition, the manufacturing cost is relatively high, and it is difficult to adjust the center frequency of the band pass band. Further, when the delay detection method is used for demodulation of the π / 4 shift QPSK modulation signal, there is a problem that the circuit becomes relatively complicated.

[0005] In order to solve this problem, the present applicant has filed a patent application of Japanese Patent Application Laid-Open No. 9-200565.
(A) A channel separating filter device which has a smaller and lighter circuit than the first and second conventional examples, can reduce the manufacturing cost, and can accurately adjust the center frequency of the bandpass band. (B) a PSK demodulator whose circuit is simpler than that of the first and second conventional examples, and (c) a first PSK demodulator.
And a PSK receiving apparatus whose circuit is simpler than that of the second conventional example. Hereinafter, these third conventional examples will be described in detail.

FIG. 20 is a block diagram showing a third conventional π / 4 shift QPSK receiving apparatus including a filter bank 10 for channel separation and a demodulator 11 using the filter bank. 21 and FIG.
FIG. 23 is a block diagram showing the demodulator 11 of FIG. 20. The receiving device of the third conventional example includes a positive pass filter (Positive Pass Filter; hereinafter, referred to as PPF) and a negative pass filter (Negat).
live Pass Filter; hereinafter referred to as NPF. ) And down-sampling processor (Down-sam)
Hereafter, it is called DSC. ), And a demodulator 11 using a filter bank configured using PPF, NPF, and DSC. First, the operating principle and theoretical analysis of PPF, NPF and DSC will be described.

<Operating Principle and Theoretical Analysis of PPF and NPF> In order to separate a plurality of frequency-multiplexed signals,
In general, a low-pass filter (hereinafter, LPF) having a transfer function H _L (z) and a high-pass filter (hereinafter, HPF) having a transfer function H _H (z) are used. It is separated into low frequency components and high frequency components. Third
Uses a complex filter that can divide a signal into two at a positive frequency and a negative frequency. Here, a PPF having only a positive frequency component and having a transfer function H _P (z) is used. And an NPF having a transfer function H _N (z) that passes only the component. These PPF and NPF are finite impulse response type digital filters (hereinafter referred to as F
This is called an IR digital filter. ).

FIG. 29 is a circuit diagram showing a configuration of an FIR digital filter. As shown in FIG. 29, input signals are input to an adder 503 via an amplifier 502-0 having a predetermined amplification degree h (0), and are connected in cascade with each other having a predetermined same delay amount. Multiple (M
-1) delay circuits 501-1 to 501- (M-1)
And an amplifier 502 having a predetermined amplification degree h (M-1)
-Is input to the adder 503 via (M-1). The signal output from the delay circuit 501-1 is input to the adder 503 via the amplifier 502-1 having a predetermined amplification degree h (1), and the signal output from the delay circuit 501-2 is output at a predetermined level. Amplifier 502-2 having amplification degree h (2)
, And similarly, the signal output from the delay circuit 501-m has a predetermined amplification factor h.
Adder 503 via amplifier 502-m having (m)
, Where m is 1, 2, 3,..., M−1. The adder 503 adds the plurality of M input signals and outputs a signal of the addition result as an output signal. Here, the amplifier 502-m has an amplification degree h for the input signal.
It can be considered as a multiplier for multiplying (m).

Next, the impulse response of the LPF realized by the FIR type digital filter shown in FIG. 29 is represented by the following equation by a filter coefficient h (m) which is an amplification degree.

[0010]

(Equation 1)

Here, since the filter coefficients h (m) are all real numbers, the sampling signal after the impulse response processing of the LPF viewed on the time axis is as shown in FIG. next,
As shown in FIG. 30B, the transfer function H _N (z) of the NPF obtained by rotating the filter coefficient h (m) by −π / 2 on the complex plane each time m advances by one is given by the following equation. Can be represented by

[0012]

(Equation 2)

From the above equation 2, the transfer function H of the LPF is obtained.
The relationship between _L (z) and the transfer function H _N (z) of the NPF can be expressed as follows.

[0014]

(Equation 3)

Further, the transfer function H _L (z) of the LPF,
The amplitude frequency and amplitude characteristics of the transfer function H _N (z) of the NPF are related as in the following equation.

[0016]

| H _N (e ^jω ) | = | H _L (e ^{jπ / 2} · e ^jω )
| = | H _L (e ^{j (ω + π / 2)} ) |

As can be seen from Equation 4, NPF is LP
It can be seen that the amplitude frequency characteristic of F (FIG. 31) is shifted on the frequency axis in the direction of negative frequency by π / 2 on the frequency axis (FIG. 32). Therefore, FIG.
The NPF has a 3dB passband with a bandwidth of π in the negative normal frequency range from 0 to -π. Similarly, the transfer coefficient H of the PPF generated by rotating the filter coefficient h (m) by π / 2 on the complex plane each time m advances by one
_P (z) can be obtained by shifting the transfer function H _L (z) of the LPF by π / 2 on the frequency axis in the direction of the positive frequency, and is shown in FIG. Therefore, as shown in FIG. 33, the PPF has a 3 dB pass band with a bandwidth π in a positive normal frequency range from 0 to π.

Tables 1 and 2 show design examples of filter coefficients when PPFs and NPFs with M = 12 stages are configured using the FIR digital filter of FIG.

[0019]

[Table 1] PPF filter coefficient h (M) ─────────────────────────────────── M Re ( h (M)) Im (h (M)) ０ 0 -3 0.896990 × 10 ⁻³ 0.0 1 0.0 1.885590 × 10 ⁻² 2 2.7103260 × 10 ⁻³ 0.0 3 0.0 8.4699540 × 10 ⁻² 4 8.846920 × 10 ^{−2 0.0 5 0.0 4.8438940 × 10 -1 6} -4.843894 × 10 -1 0.0 7 0.0 -8.8469920 × 10 -2 8 -8.4695940 × 10 -2 0.0 ^{9 0.0 -2.7103260 × 10 -3 10 -1.8856590} × 10 -2 0.0 11 0.0 3.8096990 × 10 -3 ────── ────────────────────────────

[0020]

[Table 2] NPF filter coefficient h (M) Ｍ M Re ( h (M)) Im (h (M)) ０ 0 -3 ^{.8096990 × 10 -3 0.0 1 0.0 -1.8856590} × 10 -2 2 2.7103260 × 10 -3 0.0 3 0.0 -8.4695940 × 10 -2 4 8.8469920 × 10 ^{-2 0.0 5 0.0 -4.8438940 × 10 -1} 6 -4.843894 × 10 -1 0.0 7 0.0 8.8469920 × 10 -2 8 -8.4695940 × 10 -2 0 0.09 0.0 2.7103260 × 10 ⁻³ 10 -1.8885590 × 10 ⁻² 0.0 11 0.0 −3.80969990 × 10 ⁻³ } ─────────────────────────────────

<Theory of DSC> Downsampling is also called decimation, and DSC is a circuit for lowering the sampling speed. Decimation coefficient Md
Represents the rate of decimating (ie, thinning) of the sampling signal, and is also referred to as level Md. M
d is a natural number of 2 or more. Also, if the input signal sequence is x
(N) and the output signal sequence is y (n), the relationship between the input signal sequence x (n) and the output signal sequence y (n) is expressed by the following equation.

[0022]

Y (n) = x (Md · n), n = 1, 2, 3,.

FIGS. 34 (a) and 34 (b) show the downsampling process when Md = 2. FIG. 34 (a)
And (b), the input signal sequence x (n) is 2
By sampling every other, i.e., skipping every other, the down-sampled output signal sequence y
(N) can be obtained. 34A and 34B, ts is a sampling period of the input signal sequence x (n).

<Amplitude Frequency Characteristics by Downsampling> Each amplitude frequency characteristic X (e ^jω ), Y (e ^jω ) of the input signal sequence x (n) and the output signal sequence y (n) of the DSC.
Is determined below. In order to derive this relational expression, the signal sequence x '(n) is defined as follows.

[0025]

X ′ (n) = x (n), n = 0, Md, 2Md, 3Md,... = 0, otherwise

The signal sequence x '(n) corresponds to a product of x (n) and an impulse train having a period Md. That is, the signal sequence x '(n) is represented by the following equation.

[0027]

(Equation 7)

In [7] on the right side of the above equation 7, the period Md
Is a Fourier series representation of the impulse train of FIG. Here, for clarity of notation, l = l (lowercase L). Let Q be [] on the right side of Equation 7. This can be confirmed as follows. When n is not an integral multiple of the period Md, Q is represented by the following equation.

[0029]

(Equation 8)

On the other hand, when n is an integer multiple of Md, Q is represented by the following equation.

[0031]

(Equation 9)

Under the above preparation, the output signal sequence y (m)
Is represented by the following equation by z-conversion.

[0033]

Since y (m) = x (Mdm) = x ′ (Mdm),

[Equation 11]

Here, since x '(m) is 0 except for an integral multiple of the period Md, it can be written as the following equation.

[0035]

(Equation 12)

The above equations can be summarized as follows.

[0037]

(Equation 13) However,

## ^EQU14 ## W = e ^{-j2π / Md}

The downsampling process generally has the characteristics as shown in the above equation (13), but for simplicity, Md = 2
Consider the characteristics in the case of. Substituting 2 for Md in Equation 13 gives the following equation.

[0039]

(Equation 15)

The right side of Equation 15 is divided into two terms, each of which has an important meaning. That is, Equation 15
The amplitude frequency characteristic of {(1/2) X (z ^1/2 )}, which is the first term on the right side of, has an amplitude 1/2 that of the original X (z) and doubles the bandwidth. It has become. Further, the frequency amplitude characteristic of {1/2 (e− ^jπ · z ^1/2 )} of the second term on the right side of ^Equation 15 is obtained by repeatedly shifting the equation of the first term by ± π on the frequency axis. It can be seen that ideally it has an infinitely repeated return characteristic. Considering the above, the amplitude frequency characteristic of Y (z) after down-sampling X (z) shown in FIG. 35 (a) with Md = 2 is the characteristic shown in FIG. 35 (b). . Note that the range of the spectrum is replaced by [0, 2π] to [−π, π]. In other words, the original X
Compared with (z), the amplitude is halved, the bandwidth is doubled, and it has a folded repetition characteristic that is repeatedly shifted by ± π on the frequency axis. Here, when the condition of the frequency band of the input signal is | ω _a | ≦ π / 4, the respective bands of the return repetition characteristics do not overlap even after the down-sampling process.

The channel separation filter bank 10 shown in FIGS. 20 to 22 is constituted by using the PPF, NPF and DSC described above.

Next, the demodulator 11 of FIG. 23 is connected to the PPF, N
A new principle that can be implemented using PF and DSC is described below. FIG. 36 shows a signal space diagram of π / 4 shift QPSK used in a PHS system, for example. As is clear from FIG. 36, there are eight signal points P1 to P8 shifted by π / 4 on the same amplitude circle.

FIG. 37 shows the spectral characteristics of the π / 4 shift QPSK modulation signal. As is clear from FIG.
It can be seen that the energy peak of the modulated signal changes according to the four modulated codes of the QPSK, and the code can be uniquely determined by detecting the frequency of the peak where the signal exists. This is schematically shown in FIG. 38 (the amplitude value may not be accurate in FIG. 38). As shown in FIG. 38, on the normalized ω / 2π,
_Within the range of normalized frequencies from −π to + π, the peak of code 00 is at the positive normalized frequency ω _c00 ,
The peak at 1 is at the positive normalized frequency ω _c01 (> ω _c00 ), the peak at 10 is at the negative normalized frequency ω _c10 , and the peak at 11 is the negative normalized frequency ω
_It can be seen that it is at the time of _c11 (<ω _c10 ). Here, only one peak of each code exists at a certain time.
That is, if the normalized frequency of the peak of each code is detected, the QPSK modulated signal can be demodulated. In the third prior art example, the PPF, NPF, and DSC are used as the demodulator 11 as in the case of the channel separation filter bank 10.
It is configured using.

<Configuration and Operation of QPSK Receiver> FIG.
0, a third conventional filter bank 10 for channel separation and a demodulator 1 using the filter bank.
The configuration and operation of the π / 4-shift QPSK receiving apparatus having 1 will be described. In the third conventional example, FIG.
(A), the in the radio systems, communication is performed using eight radio signal channels CH1 through CH8 of juxtaposed on the frequency axis at channel spacing of 2f ₀ from each other. That is, the signals of the eight channels are a frequency-division multiplexed signal group, and the eight channel signals do not always have to be all present. Here, the carrier frequency of channel CH1 is f _cc1 ,
Similarly, the carrier frequency of the channel CHm is set to f _ccm
And In FIG. 24A and the following drawings, the band between one band and the band of a channel adjacent thereto is set to 0. However, when actually configuring a wireless system, a margin frequency is set. You.

In FIG. 20, the transmission radio signal of the π / 4 shift QPSK modulation signal transmitted from the antenna of the transmitting apparatus of the other party is received by antenna 1 and then only the signals of all the bands of eight channels CH1 to CH8 are transmitted. The signal is output to the distributor 3 via the high-frequency amplifier 2 that passes and amplifies. Here, the QPSK modulated signal is obtained by converting a carrier signal into a Q signal according to a parallel 2-bit data signal input in the transmitting apparatus.
This is a PSK-modulated signal. The splitter 3 splits the input signal into two, outputs one split signal to the mixer 4a, and outputs the other split signal to the mixer 4b.
Output to The local oscillator 5 has a local oscillation frequency f _LO =
_{(F cc4 + f cc5) /} 2 and outputs to the mixer 4a generates a local oscillation signal having an output an input signal to the mixer 4b via the [pi / 2 phase shifter 6 shifts the phase [pi / 2 I do. Mixer 4a mixes by multiplying the two signals to be inputted, the signal after mixing and outputs it to the A / D converter 8a through LPF7a passing a frequency 8f ₀ or less. On the other hand, the mixer 4b similarly mixed by multiplying the two signals to be inputted, the signal after mixing and outputs it to the A / D converter 8b via LPF7b passing a frequency 8f ₀ or less. Therefore, as shown in FIG. 24 (b), the signal divided into two by the divider 3 is converted by the frequency converter 12 composed of the mixers 4a and 4b and the LPFs 7a and 7b, as shown in FIG.
Carrier frequency f _{cc4 of} channel CH4 and channel C
The channel CH1 is set so that the average frequency of the carrier frequency _{fcc5 of} H5 becomes zero, that is, the channel CH1 is set to a positive frequency.
The frequency conversion is performed so that the signal components of the channels CH5 to CH8 exist at the negative frequency while the signal components of the channels CH4 to CH4 exist. In FIG. 20, the configuration from the high frequency amplifier 2 to the A / D converters 8a and 8b is hereinafter referred to as a high frequency and intermediate frequency signal processing unit 20.

The A / D converters 8a and 8b A / D convert the input analog signals into digital signal data at a predetermined sampling rate, and separate the orthogonal I-channel data signals and Q-channel data signals into channels. To the filter bank 10. As described in detail later, the filter bank 10 for channel separation
By frequency-separating the 8-channel data signal for each channel, that is, channel-separating, the I-channel data signal and the Q-channel data signal of the channel CH1 are output to the demodulator 11-1. The channel data signal and the Q channel data signal are output to the demodulator 11-m (m = 2,
3, ..., 8). In response, demodulators 11-m (where m = 1, 2,..., M; hereinafter, demodulators 11-1 to 11-1)
1-8 are collectively referred to as 11. ) QPSK demodulates the input signal into a demodulated data signal and outputs the demodulated data signal.

<Structure of Filter Bank for Channel Separation> The structure of the filter bank 10 for channel separation in FIG. 20 will be described with reference to FIGS. In FIGS. 21 to 23, the symbol 2 on the signal line indicates that the signal flowing through the signal line is a complex signal, and the I-channel data signal of the real part and the Q-channel data of the imaginary part are orthogonal to each other. Signal. The channel separation filter bank 10 includes first to third stages.
It is composed of filter banks 10a, 10b and 10c at the stage. Each stage of the filter bank has one or more sets of P
It is composed of a pair of a PF and an NPF, a DSC for down-sampling the output of the PPF, and a DSC for down-sampling the output of the NPF.

Here, each PPF 101, 201, 22
1,301,321,341,361 have the same standard, that is, have the same positive pass band from 0 to 8f ₀ , and convert the input complex number data signal into the positive pass band of the positive pass band. Only the signal component is passed through the band and output. In addition, each NPF 111, 211, 231, 311,
331, 351 and 371 have the same standard, that is, have the same negative pass band from 0 to -8f ₀ , and convert an input complex data signal into a signal component of the negative pass band. Only band pass and output. Furthermore, each DSC 102, 112, 202, 212, 22
2,232,302,312,322,332,34
2, 352, 362, and 372 are configured according to the same standard, and perform an Md = 2 downsampling process on an input complex data signal to output a processed signal. Here, in the downsampling process of Md = 2, as described in detail later, the input data signal is set to 1
By downsampling every other signal, the output data signal has an amplitude value that is か つ compared to the input data signal, the bandwidth of each channel is doubled, and ± 16f ₀ It has a shifted and repeated spectrum.

As shown in FIG. 21, the I-channel data signal output from the A / D converter 8a is input to the PPF 101 and the NPF 111 of the first-stage filter bank 10a, while being output from the A / D converter 8b. The Q channel data signal to be output is the P-channel data of the first-stage filter bank 10a.
It is input to PF101 and NPF111.

In the first stage filter bank, PP
The data signal of the complex number output from F101 is DSC1
02 through the PPF of the second stage filter bank 10b
201 and NPF 211 while NPF 11
The complex data signal output from 1 is sent to the PPF 221 of the second-stage filter bank 10b via the DSC 112.
And NPF 231.

[0051] Therefore, the output data signal from the PPF 101, as shown in FIG. 25 (a), it becomes a band signal component of the channel CH1 through CH4 having a positive frequency band of 0 to 8f _0, the next DSC102 The output data signal from 0 to 16f as shown in FIG.
Band signal component of the channel CH1 through CH4 having a positive frequency band of ₀ has a spectrum repeated Kaee by shifted by ± 16f _0. On the other hand, NPF111
As shown in FIG. 25 (b), the output data signal from
Becomes a band signal component of the channel CH5 to CH8 having a negative frequency band of 0 to -8F _0, next DS
As shown in FIG. 25D, the output data signal from C112 has a band signal component of channels CH5 to CH8 having a negative frequency band of 0 to −16f ₀ ± 16.
is shifted by f ₀ has a spectrum that has been repeated Kaee in.

In the second-stage filter bank 10b, the complex data signal output from the PPF 201 is passed through the DSC 202 to the third-stage filter bank 10c.
NPF 311 and NPF 311
The complex data signal output from the PF 211 is a DSC
The PP of the filter bank 10c of the third stage through 212
It is input to F321 and NPF331. Also, PPF
221 outputs a complex data signal.
2 through the PPF3 of the third stage filter bank 10c.
41 and NPF 351 while NPF 231
Are input to the PPF 361 and the NPF 371 of the third-stage filter bank 10c via the DSC 232.

Therefore, for example, the output data signal from the PPF 201 is 0 to 8f as shown in FIG.
_As shown in FIG. 26C, the band signal components of the channels CH3 and CH4 having a positive frequency band up to ₀ are output from the DSC 202 at the next stage.
The band signal components of the channels CH3 and CH4 having a positive frequency band from ｆ to 16f ₀ have a spectrum repeated while being shifted by ± 16f ₀ . Also, for example, the output data signal from the NPF 211 is as shown in FIG.
(B), the will-band signal component of the channel CH1 and CH2 having a negative frequency band of 0 to -8F _0, the output data signal from the next stage DSC212, so that shown in FIG. 26 (d) the channel CH1 and CH2 having a negative frequency band of 0 to -16F ₀
Has a spectrum repeated by shifting the band signal components by ± 16 f ₀ .

In the third-stage filter bank 10c, the complex data signal output from the PPF 301 is output via the DSC 302, so that only the data signal of the channel CH4 is band-filtered and extracted, and then demodulated. Output to the unit 11-4. Also, NPF311
Is output through the DSC 312, so that only the data signal of the channel CH3 is band-filtered and extracted, and then demodulated by the demodulator 11-3.
Is output to The complex data signal output from the PPF 321 is output via the DSC 322,
After only the data signal of channel CH2 is band-filtered and extracted, it is output to demodulator 11-2. Also, NP
The complex data signal output from F331 is DSC3
By outputting the data signal via the channel 32, only the data signal of the channel CH1 is band-filtered and extracted, and then output to the demodulator 11-1. The complex data signal output from the PPF 341 is output via the DSC 342, so that only the data signal of the channel CH8 is band-filtered and extracted, and then output to the demodulator 11-8. The complex data signal output from the NPF 351 is output via the DSC 352, so that only the data signal of the channel CH7 is band-filtered and extracted, and then output to the demodulator 11-7. The complex data signal output from the PPF 361 is output via the DSC 362, so that only the data signal of the channel CH6 is band-filtered and extracted, and then output to the demodulator 11-6. The complex data signal output from the NPF 371 is output through the DSC 372,
After only the data signal of channel CH5 is band-filtered and extracted, it is output to demodulator 11-5.

Therefore, for example, the output data signal from the PPF 321 is 0 to 8f as shown in FIG.
It becomes a band signal component of the channel CH2 having a positive frequency band up to _0, and the output data signal from the DSC 322 at the next stage is from 0 to 16f _{0 as} shown in FIG.
Band signal component of the channel CH2 having a positive frequency band up to have a spectrum Kaee repeated by shifting one by ± 16f _0. Further, for example, as shown in FIG. 27B, the output data signal from the NPF 331 is a channel C having a negative frequency band from 0 to −8f _0.
Becomes H1 of-band signal components, the output data signal from the next stage DSC332, as shown in FIG. 27 (d), the channel C having a negative frequency band of 0 to -16F ₀
It has a spectrum in which the band signal component of H1 is shifted by ± 16f ₀ and repeated. The components of the data signals of the other channels are processed in the same manner, and the respective DSCs 302, 31
2,322,332,342,352,362,372
Therefore, only the band signal component of each channel CHm is output to the demodulator 11-m (m = 1, 2,..., 8).

In FIG. 22, each level detector 103,
113, 203, 213, 223, 233, 303, 3
13,323,333,343,353,363,37
Numeral 3 has the same configuration, calculates the amplitude value of the input complex number signal, and outputs the calculation result as a signal level signal. Each DSC 102, 112, 202, 212, 22
2,232,302,312,322,332,34
2, 352, 362, and 372 are output from the level detectors 103, 113, 203, and 2 respectively.
13,223,233,303,313,323,33
3, 343, 353, 363, and 373, and each level detector 103, 113, 203, 213, 223,
233,303,313,323,333,343,3
Each signal level signal output from 53, 363, 373 is input to the filter bank operation controller 100. In response to this, the filter bank operation controller 100 detects the signal level signal when the signal level signal becomes 0 (that is, when there is no signal and the signal level is off).
The operation in the circuits subsequent to the SC is disabled and stopped by the non-operation control signal, and the corresponding demodulator 1 is disabled.
The code of the demodulated data signal output from 1 is set to 00 (this 00 may be another predetermined code). By the operation based on this non-operation control signal, the DS which detects it is detected.
The current consumption can be reduced by setting the supply power supply current to 0 after the circuit C.

The above-described channel separation filter bank 1
0, a three-stage filter bank is described, but the present invention is not limited thereto, and a one-stage, two-stage, or four-stage or more filter bank may be configured. PPF
When configured with filter pair of NPF and filter bank having n stages, which can be the separation of ^the 2 ⁿ channels, by omitting one of the PPF or NPF in the second and subsequent stages, the two other ⁿ pieces It can be configured to be able to separate any multiple channels.

<Configuration of Demodulator 11> Referring to FIG.
The configuration of the demodulators 11-1 to 11-8 in FIG. 20 will be described. Since the configurations of the demodulators 11-1 to 11-8 are the same, one demodulator 11 will be described below. As shown in FIG. 23, the demodulator 11 includes a frequency conversion unit 11a and a signal demodulation unit 11b.

The frequency converter 11a includes two mixers 40
1a, 401b, local oscillator local oscillator 402 for generating a local oscillation signal of a frequency 8f _0, [pi / 2 phase shifter 403, two LPF404a for low-pass-filters the frequency component of each from 0 to ± 8f _0, 404b FIG. 2
As shown in FIGS. 28A and 28B, the center of the band signal component of each channel in the data signal output from the input data signal is the frequency 0 as in the case of the frequency conversion unit 12 of 0. as, namely, frequency conversion using a local oscillation frequency 8f _0. The complex data signal after the frequency conversion is output from the LPFs 404a and 404b to the DSC 411 of the signal demodulation unit 11b.

The part (NPF4) in the preceding stage of the signal demodulation unit 11b
16, PPF 426, NPF 436 or PPF 446) are circuits for separating frequency band components, as can be seen from the configuration of FIG. 21, and for separating channels CH3 to CH6 in the circuit of FIG. Corresponding to the circuit. Here, each PPF 422, 4
24, 426, 446 are the same standard, that is,
It is configured to have the same positive pass band from 0 to π on the frequency f = ω / 2π axis shown in FIG. 38, and the input complex data signal is band-limited only to the signal component of the positive pass band. Pass and output. In addition, each NPF 412, 41
4,416,436 have the same standard, that is, have the same negative passband from 0 to -π on the frequency f = ω / 2π axis shown in FIG. The data signal is output after passing only the signal component of the positive pass band. Furthermore, each DSC 411, 41
3, 415, 423, and 425 are configured according to the same standard, and execute an Md = 2 downsampling process on an input complex data signal to output a processed signal. Here, in the downsampling process of Md = 2, as described in detail later, the input data signal is set to 1
By down-sampling every other signal, the output data signal has an amplitude value that is か つ compared to the input data signal and the bandwidth of each channel is doubled, and is shifted by ± π. And have a repeated spectrum.

In FIG. 23, the peak of the π / 4 shift QPSK modulation signal is from −π / 2 to + π as shown in FIG.
/ 2, the DSC 411 performs a down-sampling process of Md = 2 on the input data signal and outputs the processed data signal to the NPF 412 and the PPF 422 in order to double the frequency on the frequency axis. . And N
The data signal output from the PF 412 is a DSC 413,
NPF 416 via NPF 414 and DSC 415
And PPF 426. On the other hand, the data signals output from the PPF 422 are DSC 423, PPF 424
And NPF 436 and PPF 44 via DSC 425
6 is output. Here, PPF422 and PPF42
4 is used to separate and extract the peak of the signal component of code 00 located on the positive normalized frequency side and the peak of the signal component of code 01, while the NPFs 412 and 414 are used for extracting the peak located on the negative normalized frequency side. It is used to separate and extract the peak of the signal component 11 and the peak of the signal component 10. Here, two-stage PPF 422 and PPF 42
4 or NPFs 412 and 414 are used because four peaks exist near the normalized frequency 0 as shown in FIG. And NPF 416, P
The peaks of the signal components denoted by reference numerals 11, 10, 01, and 00 can be band-filtered by the PF 426, the NPF 436, and the PPF 446, respectively.

The complex number data signal output from NPF 416 is input to level detector 417, the amplitude value of which is calculated, and the signal level signal indicating the calculation result is input to the non-inverting input terminal of comparator 418. . In addition, PPF4
The complex data signal output from 26 is input to a level detector 427, whose amplitude value is calculated, and a signal level signal indicating the calculation result is input to a non-inverting input terminal of a comparator 428. Further, the complex data signal output from NPF 436 is input to level detector 437 and its amplitude value is calculated, and then a signal level signal indicating the calculation result is input to the non-inverting input terminal of comparator 438. Further, the complex number data signal output from PPF 446 is input to level detector 447, whose amplitude value is calculated, and a signal level signal indicating the calculation result is output from comparator 447.
It is input to 48 non-inverting input terminals.

Here, each of the comparators 418, 428, 43
8, 448 are grounded, that is, a data signal of 0 is input, and therefore, each comparator 418, 42
8, 438 and 448 detect whether or not there is a peak indicating each code. If detected, an H level detection signal is output to the code discriminator 450. If not detected, an L level detection signal is output. Output to the discriminator 450. The signal level signals output from the level detectors 417, 427, 438, and 447 are input to the adder 451 and added, and a signal indicating the addition result is supplied to the reference clock recovery circuit 452.
Is input to In response, the reference clock reproducing circuit 452 reproduces a reference clock indicating the time position where the peak exists at the rising edge of the reference clock based on the input signal of the addition result, and outputs the reference clock to the code discriminator 450. . The code discriminator 450 outputs a demodulated data signal of code 11 when an H-level detection signal is input from the comparator 418 at the rise of the reference clock.
When an H-level detection signal is input from 8 to 10
When the H-level detection signal is input from the comparator 438, a demodulated data signal of code 01 is output. When the H-level detection signal is input from the comparator 448, code 00 is output. And outputs the demodulated data signal.

In the demodulator 11 described above, the DSC 413
And 423, the level detectors 461 and 4 are respectively provided similarly to the channel separation filter bank 10 of FIG.
When a signal is not present at the output of the DSC 413 by the signal demodulation unit operation controller 470, the operation of the subsequent circuit is stopped from the NPF 414 using the non-operation control signal, while no signal is present at the output of the DSC 423. When the PP is set using the non-operation control signal
The operation of the subsequent circuit is stopped from F424.

FIG. 39 is a block diagram showing the configuration of a fourth conventional example of a space diversity type receiving apparatus constructed using the third conventional example of the channel separation filter bank 10. As shown in FIG. This fourth conventional example is a configuration of a receiving apparatus when the frequency multiplexed transmission signal of FIG. 24 of the third conventional example is received using two antennas 1a and 1b having a space diversity configuration.

In FIG. 39, the frequency multiplexed transmission signal received by the antenna 1a is divided into a high frequency and intermediate frequency signal processing unit 20a having the same configuration as the high frequency and intermediate frequency signal processing unit 20 in FIG. After being processed by the channel separation filter bank 10a having the same configuration as the channel separation filter bank 10, the signals of the respective channels are input to the respective input terminals A of the multiplexers 21-1 to 21-8. On the other hand, the frequency-multiplexed transmission signal received by the antenna 1b is divided into a high-frequency and intermediate-frequency signal processing unit 20b having the same configuration as the high-frequency and intermediate-frequency signal processing unit 20 in FIG. After being processed by the channel separation filter bank 10b having the same configuration as that of 10, the signals of the respective channels are input to the respective input terminals B of the multiplexers 21-1 to 21-8. In response, each of the multiplexers 21-1 to 21-2
Reference numeral 1-8 outputs, from the output terminal C, a signal having a large input signal among the signals input to the input terminals A and B from the output terminal C to the demodulators 11-1 to 11-8 for each channel.
The demodulators 11-1 to 11-8 demodulate the input signals, for example, by PSK demodulation, and output demodulated demodulated data signals for each channel.

[0067]

However, in the fourth conventional example, it is necessary to provide two filter banks 10a and 10b for channel separation corresponding to the two antennas 1a and 1b, and the circuit configuration is large. There was a problem of becoming.

An object of the present invention is to solve the above problems and to provide a space diversity type receiving apparatus having a simple circuit configuration as compared with the fourth conventional example.

[0069]

According to a first aspect of the present invention, there is provided a space diversity type receiver having a plurality of channels having different carrier frequencies, and the plurality of channels having a block shape. At least a first receiving each of the frequency multiplexed signals comprising
And a space diversity antenna device having a second antenna, and a first frequency multiplexed signal having a plurality of channels received by the first antenna, such that its spectrum is located in a positive frequency region. First frequency conversion means for performing frequency conversion to have a block shape and outputting a first frequency multiplexed signal after frequency conversion;
The second frequency multiplexed signal having a plurality of channels received by the second antenna is frequency-converted so that its spectrum has a block shape positioned in a negative frequency region, and is subjected to frequency conversion. A second frequency converter for outputting a second frequency multiplexed signal, a first frequency multiplexed signal output from the first frequency converter, and a second frequency converter output from the second frequency converter. (2) combining the frequency multiplexed signals of the first and second frequency multiplexed signals and outputting the synthesized frequency multiplexed signal;
The two signals corresponding to the two antennas are processed by different circuits to separate the frequency multiplexed signal into only the frequency components of the signals of the same two channels corresponding to the at least two antennas. A channel separation unit; and a selection output unit that selects and outputs, for each channel, a signal having higher power among the two signals of the same channel output from the channel separation unit. I do.

A space diversity type receiving apparatus according to a second aspect of the present invention has a plurality of channels having different carrier wave frequencies, and the plurality of adjacent channels each correspond to one channel by one channel. At least a first receiving means for receiving a frequency multiplexed signal arranged so as to have a comb-shape only
And a space diversity antenna device having a second antenna, and a first frequency multiplexed signal having a plurality of channels received by the first antenna, the spectrum of which is divided into positive and negative frequency regions. Each two adjacent
The frequency conversion is performed so as to have a comb shape that is located in a predetermined first frequency region where one channel is separated from each other by one channel, and the first
And a second frequency multiplexed signal having a plurality of channels received by the second antenna, the spectrum of which is divided into a positive and a negative frequency domain. The two adjacent channels are separated by one channel from each other and frequency-converted so as to have a comb shape positioned in a predetermined second frequency region different from the first frequency region. Second frequency conversion means for outputting the converted second frequency multiplexed signal, first frequency multiplexed signal output from the first frequency conversion means, and output from the second frequency conversion means Combining means for combining the second frequency multiplexed signal to be output and the combined frequency multiplexed signal output from the combining means. The two signals corresponding to at least two antennas are processed by different circuits, so that the frequency multiplexed signal is converted into only the frequency components of the signals of the same two channels corresponding to the at least two antennas. Channel separation means for separating the channels, and selection output means for selecting and outputting, for each channel, a signal having a higher power among the two signals of the same channel output from the channel separation means. It is characterized by.

A space diversity receiver according to a third aspect of the present invention is the space diversity receiver according to the first or second aspect, wherein the channel separation means includes at least one filter bank. A first positive-pass filter for band-filtering only a signal component of a predetermined positive frequency out of a plurality of channel signals output from the frequency conversion means and outputting a signal after the band filtering. A filter, and first processing means for performing a down-sampling process on the signal output from the first positive-pass filter and outputting the processed signal as a signal including a signal of at least one channel. ,
A first negative-pass filter that band-filters only a signal component having a predetermined negative frequency out of a plurality of channel signals output from the frequency conversion unit and outputs a signal after band-filtering; A second processing means for performing a down-sampling process on a signal output from the first negative-pass filter and outputting the processed signal as a signal including a signal of at least one channel; The multiplexed signals of the plurality of channels are separated into signals of at least two channels.

The space diversity receiver according to a fourth aspect of the present invention is the space diversity receiver according to the third aspect, wherein the channel separation means includes at least two stages of filter banks, The bank includes a second positive-pass filter that band-filters only a signal component having a predetermined positive frequency in the signal output from the first processing unit and outputs a signal after band-filtering, A third processing unit that performs downsampling processing on a signal output from the second positive-pass filter and outputs the processed signal as a signal including a signal of at least one channel; A second negative-pass filter for band-filtering only a signal component having a predetermined negative frequency out of the signals output from the processing means, and outputting a signal after band-filtering; Fourth processing means for performing a down-sampling process on a signal output from the second negative-pass filter and outputting the processed signal as a signal including a signal of at least one channel; A third positive-pass filter for band-filtering only a signal component having a predetermined positive frequency in the signal output from the processing means and outputting a signal after the band-filtering; Fifth processing means for performing downsampling processing on the signal output from the filter and outputting the processed signal as a signal including at least one channel signal, and output from the second processing means A third negative-pass filter for band-filtering only a signal component having a predetermined negative frequency in the signal, and outputting a signal after band-filtering;
And a sixth processing means for performing a downsampling process on a signal output from the negative-pass filter and outputting the processed signal as a signal including a signal of at least one channel. It is characterized in that the channelized signals of a plurality of channels are separated into at least four channel signals.

A space diversity receiver according to a fifth aspect of the present invention is the space diversity receiver according to any one of the first to fourth aspects, wherein the signal of each channel of the frequency multiplexed signal is provided. Is P
It is an SK modulation signal.

[0074]

Embodiments of the present invention will be described below with reference to the drawings.

<First Embodiment> FIG. 1 is a block diagram showing a configuration of a space diversity type receiving apparatus according to a first embodiment of the present invention. In the first embodiment, the frequency multiplexed signal transmitted from the transmission device has a plurality of channels having different carrier frequencies and is arranged such that the plurality of channels have a block shape. A signal output from a high-frequency and intermediate-frequency signal processing unit (hereinafter, also referred to as a processing unit) 20a having the same configuration as the processing unit 20 of FIG. A synthesizer 22 for synthesizing a signal output from the frequency signal processing unit 20b and outputting the synthesized signal, and a channel separation process for the signal output from the synthesizer 22 having the same configuration as the third conventional example. A channel separation filter bank 10 that performs
Each 2 output from the channel separation filter bank 10
And a multiplexer 21 for selecting a large signal from the same channel. here,
The first embodiment uses the circuits shown in FIGS.

The processing section 20a has a local oscillation frequency f _LO =
Has a _{(f cc4 + f cc5) /} 2 + 8 f0, the frequency multiplexed signal having a plurality of channels CH1 to CH4, which are received by the antenna 1a (see FIG. 3 (a).), Its spectrum, the positive frequency domain The frequency conversion is performed so as to have a block-shaped spectrum (b-block) such that the frequency-multiplexed signal is output to the synthesizer 22 after the frequency conversion. On the other hand, the processing unit 20b outputs the local oscillation frequency f _LO =
Has a _{(f cc4 + f cc5) /} 2, the frequency multiplexed signal having a plurality of channels CH1 to CH4, which are received by the antenna 1b (see FIG. 3 (a).), Its spectrum,
The frequency conversion is performed so as to have a block-shaped spectrum (a block) positioned in the negative frequency region, and the frequency-multiplexed signal after the frequency conversion is output to the synthesizer 22.
The combiner 22 combines the two signals from the processing units 20a and 20b and outputs the combined signal to the channel separation filter bank 10. The channel separation filter bank 10 has a configuration similar to that of the third conventional example, executes channel separation processing, and outputs each signal as follows. (A) The signal of the channel CH1a is output from the DSC 372 to the input terminal A of the multiplexer 21-1. (B) The signal of the channel CH1b is output from the DSC 332 to the input terminal B of the multiplexer 21-1. (C) The signal of the channel CH2a is output from the DSC 362 to the input terminal A of the multiplexer 21-2. (D) The signal of the channel CH2b is output from the DSC 322 to the input terminal B of the multiplexer 21-2. (E) The signal of the channel CH3a is output from the DSC 352 to the input terminal A of the multiplexer 21-3. (F) The signal of the channel CH3b is output from the DSC 312 to the input terminal B of the multiplexer 21-3. (G) The signal of the channel CH4a is output from the DSC 342 to the input terminal A of the multiplexer 21-4. (H) The signal of the channel CH4b is output from the DSC 302 to the input terminal B of the multiplexer 21-4.

In response, each multiplexer 21-
1 to 21-8 output, from the output terminal C, among the signals input to the input terminals A and B, the signal having the larger power to the demodulators 11-1 to 11-8 for each channel. The demodulators 11-1 to 11-8 demodulate the input signals, for example, by PSK demodulation, and output demodulated demodulated data signals for each channel.

Hereinafter, amplitude frequency characteristics showing the operation of the space diversity type receiver of FIG. 1 will be described. (A) FIG.
20A is an amplitude frequency characteristic diagram of the output signal of the high-frequency amplifier 2 of FIG. 20, and FIG. 3B is a diagram of the NPFs 7a and 7 of FIG.
FIG. 6B is an amplitude frequency characteristic diagram of each output signal of FIG. (B) FIG.
FIG. 4A is an amplitude frequency characteristic diagram of the output signal of the PPF 101 in FIG. 21, and FIG. 4B is an amplitude frequency characteristic diagram of the output signal of the NPF 111 in FIG. (C) FIG. 5A is an amplitude frequency characteristic diagram of the output signal of the DSC 102 in FIG. 21, and FIG. 5B is an amplitude frequency characteristic diagram of the output signal of the DSC 112 in FIG. (D) FIG. 6A is an amplitude frequency characteristic diagram of the output signal of the PPF 201 of FIG. 21, and FIG. 6B is an amplitude frequency characteristic diagram of the output signal of the PPF 221 of FIG. (E) FIG. 7A shows DSC2 of FIG.
FIG. 7B is an amplitude frequency characteristic diagram of the output signal of FIG.
22 is an amplitude frequency characteristic diagram of the output signal of the DSC 222 in FIG. (F) FIG. 8A is an amplitude frequency characteristic diagram of the output signal of the PPF 301 in FIG. 21, and FIG.
3 is an amplitude frequency characteristic diagram of an output signal of the PPF 321. FIG.
(G) FIG. 9A is an amplitude frequency characteristic diagram of the output signal of the DSC 302 of FIG. 21, and FIG.
342 is an amplitude frequency characteristic diagram of the output signal of FIG. (H) FIG. 10A shows the amplitude of the output signal output from each LPF 404a, 404b of the demodulator 11-4 after being input from the DSC 302 of FIG. 21 via the multiplexer 21-4 to the demodulator 11-4. FIG. 10 (b) is a frequency characteristic diagram. FIG. 10 (b) shows the output from each of the LPFs 404a and 404b of the demodulator 11-4 after being input from the DSC 342 of FIG. 21 via the multiplexer 21-4 to the demodulator 11-4. FIG. 3 is a diagram illustrating amplitude frequency characteristics of a signal.

As described above, according to the first embodiment, the frequency multiplexed signal composed of four channels shown in FIG. 3A is compared with the third conventional example by one channel. Since the channel separation processing can be performed using only the separation filter bank 10, the circuit configuration can be simplified as compared with the third conventional example, thereby greatly reducing the manufacturing cost of the device. And power consumption can be reduced by almost half.

<Second Embodiment> FIG. 2 is a block diagram showing a configuration of a space diversity receiver according to a second embodiment of the present invention. The second embodiment is the first
The following configuration is different from the first embodiment.

As shown in FIG. 11 (a), the frequency multiplexed signal transmitted from the transmitting apparatus has a plurality of channels of mutually different carrier frequencies, and the two adjacent channels are mutually connected. They are arranged so as to have a comb shape separated by one channel, whereby the frequency-multiplexed signal after frequency conversion and synthesis becomes as shown in FIG. 11B. Here, the processing unit 20a determines the local oscillation frequency f _LO = (f _cc4 + f
has _cc5) / 2 + f _0, the processing unit 20b has a local oscillation frequency _{f LO = (f cc4 + f} cc5) / 2. Processing unit 20a
Represents a plurality of channels CH1 received by the antenna 1a.
b to CH4b, the spectrum of which is divided into two adjacent frequencies in the positive and negative frequency regions.
The frequency conversion is performed so as to have a comb shape that is located in a predetermined first frequency region where one channel is separated from each other by one channel, and the first
While the processing unit 20b outputs
A frequency multiplexed signal having a plurality of channels CH1a to CH4a received by the antenna 1a is divided into two adjacent channels in the positive and negative frequency regions by one channel from each other, and Frequency conversion is performed so as to have a comb shape positioned in a predetermined second frequency region different from the first frequency region, and a frequency-multiplexed signal after frequency conversion is output.

The channel separation filter bank 10 has a configuration similar to that of the third conventional example, executes a channel separation process, and outputs each signal as follows. (A) The signal of the channel CH1a is output from the DSC 322 to the input terminal A of the multiplexer 21-1. (B) The signal of the channel CH1b is output from the DSC 332 to the input terminal B of the multiplexer 21-1. (C) The signal of the channel CH2a is output from the DSC 302 to the input terminal A of the multiplexer 21-2. (D) The signal of the channel CH2b is output from the DSC 312 to the input terminal B of the multiplexer 21-2. (E) The signal of the channel CH3a is output from the DSC 362 to the input terminal A of the multiplexer 21-3. (F) The signal of the channel CH3b is output from the DSC 372 to the input terminal B of the multiplexer 21-3. (G) The signal of the channel CH4a is output from the DSC 342 to the input terminal A of the multiplexer 21-4. (H) The signal of the channel CH4b is output from the DSC 352 to the input terminal B of the multiplexer 21-4.

The amplitude frequency characteristics showing the operation of the space diversity receiver of FIG. 2 will be described below. (A) FIG.
1 (a) is an amplitude frequency characteristic diagram of an output signal of the high frequency amplifier 2 of FIG. 20, and FIG. 11 (b) is an NPF 7 of FIG.
It is an amplitude frequency characteristic figure of each output signal of a and 7b.
(B) FIG. 12A is an amplitude frequency characteristic diagram of the output signal of the PPF 101 in FIG. 21, and FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PF111.
(C) FIG. 13A is an amplitude frequency characteristic diagram of the output signal of the DSC 102 in FIG. 21, and FIG.
It is an amplitude frequency characteristic figure of the output signal of SC112.
(D) FIG. 14A is an amplitude frequency characteristic diagram of the output signal of the PPF 201 in FIG. 21, and FIG.
FIG. 9 is an amplitude frequency characteristic diagram of an output signal of the SC 202.
(E) FIG. 15A is an amplitude frequency characteristic diagram of the output signal of the PPF 301 in FIG. 21, and FIG.
FIG. 9 is an amplitude frequency characteristic diagram of an output signal of the PF 311.
(F) FIG. 16A is an amplitude frequency characteristic diagram of the output signal of the DSC 302 of FIG. 21, and FIG.
It is an amplitude frequency characteristic figure of the output signal of SC312.
(G) FIG. 17A shows the amplitude of the output signal output from each LPF 404a, 404b of the demodulator 11-2 after being input from the DSC 302 of FIG. 21 via the multiplexer 21-2 to the demodulator 11-2. FIG. 17 is a frequency characteristic diagram, and FIG.
(B) shows the configuration of the multiplexer 21 from the DSC 312 in FIG.
-2 is input to the demodulator 11-2 via the demodulator 1-2.
FIG. 3 is an amplitude frequency characteristic diagram of output signals output from each of the LPFs 404a and 404b of 1-2.

As described above, according to the second embodiment, the frequency multiplexed signal composed of four channels shown in FIG. 11A is compared with the third conventional example by one channel. Since the channel separation processing can be performed using only the separation filter bank 10, the circuit configuration can be simplified as compared with the third conventional example.
The manufacturing cost of the device can be significantly reduced, and the power consumption can be almost halved.

<Modification> Although the channel separation filter bank 10 for the QPSK modulated signal has been described in the channel separation filter bank 10 of the above embodiment, the present invention is not limited to this, and the FSK or QAM
Can be used to separate the channels of unmodulated signals such as other modulated signals or carrier signals.

The comparators 418 and 4 shown in FIG.
In 28, 438 and 448, 0 level is used as a threshold for detecting the presence of a signal, but the present invention is not limited to this, and a positive number close to 0 may be used.

In the demodulator 11 used in the embodiment, N
Although a two-stage filter bank configuration from PF 412 to DSC 415 and a two-stage filter bank configuration from PPF 422 to DSC 425 are used, the present invention is not limited to this, and a one-stage filter bank configuration may be used for each. In this case, a level detector is provided after the last DSC of the one-stage filter bank, and when there is no signal at the output of the DSC, the operation of the circuits after the DSC is stopped. Further, the DSC 411 may be omitted when unnecessary according to the spectrum characteristic of the PSK modulation signal.

In this embodiment, the π / 4 shift QPSK modulation method is used.
In the case of K, 16-valued PSK, 256-valued PSK, or other PSK modulation methods that do not perform the shift operation or the differential operation, the peak frequency changes according to the code in the same manner as in the present embodiment, so that BPSK (Binary Phase Shift Keying) is used. ), QPS
The demodulator can be configured in the same manner as the demodulator 11 in other PSK modulation schemes such as K, 16-level PSK, and 256-level PSK.

In the above embodiment, a wireless system having four channels has been described. However, the present invention is not limited to this, and can be applied to a wireless system having two or more channels.

In the above embodiment, the demodulator 11 using the filter bank is provided. However, the present invention is not limited to this, and a conventional delay detection type demodulator may be used. In the above embodiment, PPF, NPF and DS
C, the demodulator 11 is used.
The code identification may be performed by detecting the peak of the energy of PSK using T.

[0091]

As described above in detail, according to the space diversity type receiver according to the first aspect of the present invention, a plurality of channels having mutually different carrier frequencies are provided, and the plurality of channels have a block shape. Diversity antenna device having at least first and second antennas respectively receiving frequency multiplexed signals arranged as described above, and first frequency multiplexing having a plurality of channels received by the first antenna Frequency conversion means for converting the frequency-converted signal into a block shape such that its spectrum is located in the positive frequency domain, and outputting a frequency-converted first frequency-multiplexed signal; The second frequency multiplexed signal having a plurality of channels received by the second antenna is converted to a signal having a negative frequency. A second frequency conversion means for outputting a second frequency-multiplexed signal after frequency conversion by the frequency conversion so as to have a block shape as located in the region,
The first frequency multiplexed signal output from the first frequency conversion means and the second frequency multiplexed signal output from the second frequency conversion means are combined, and the combined frequency multiplexing is performed. A signal combining means for outputting a signal and a combined frequency multiplexed signal output from the combining means are processed by different circuits for each of the two signals corresponding to the at least two antennas. The signal is divided into two each corresponding to the at least two antennas.
Channel separation means for separating the channel into only frequency components of the same channel signal, and a signal having a larger power among the two same channel signals output from the channel separation means is selected for each channel. Output means for selecting the output. Therefore, for example, the frequency division multiplexed signal composed of four channels shown in FIG. 3A is subjected to the channel separation processing using only one channel separation filter bank 10 as compared with the third conventional example. Therefore, the circuit configuration can be simplified as compared with the third conventional example, whereby the manufacturing cost of the device can be significantly reduced and the power consumption can be reduced by almost half.

Further, according to the space diversity type receiver according to the second aspect of the present invention, each of the two channels having a plurality of channels having different carrier wave frequencies and adjacent to each other is connected to one another. A space diversity antenna device having at least first and second antennas for receiving frequency multiplexed signals arranged so as to have a comb shape separated by the number of channels, and received by the first antenna; A first frequency multiplexed signal having a plurality of channels is defined by a predetermined first frequency whose spectrum separates each two adjacent channels of the positive and negative frequency domains by one channel from each other. Frequency-converted so as to have a comb-like shape positioned in the region, and the first frequency multiplied after frequency conversion. Frequency conversion means for outputting a multiplexed signal and a second frequency multiplexed signal having a plurality of channels received by the second antenna. Are separated by one channel from each other, and are frequency-converted so as to have a comb shape that is located in a predetermined second frequency region different from the first frequency region. A second frequency converter for outputting a second frequency multiplexed signal, a first frequency multiplexed signal output from the first frequency converter, and a second frequency converter output from the second frequency converter. (2) combining the frequency multiplexed signal of the first and second frequency multiplexed signals and outputting the combined frequency multiplexed signal; and the combined frequency multiplexed signal output from the combining means. The two signals corresponding to the two antennas are processed by different circuits to separate the frequency multiplexed signal into only the frequency components of the signals of the same two channels corresponding to the at least two antennas. Channel separating means, and each of the two output from the channel separating means.
Selection output means for selecting and outputting a signal having higher power among the signals of the same channel for each channel. Therefore, for example, the frequency division multiplexed signal composed of four channels shown in FIG. 11A is subjected to the channel separation processing using only one channel separation filter bank 10 as compared with the third conventional example. Therefore, the circuit configuration can be simplified as compared with the third conventional example, whereby the manufacturing cost of the device can be significantly reduced and the power consumption can be reduced by almost half.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a configuration of a space diversity receiver according to a first embodiment of the present invention.

FIG. 2 is a block diagram illustrating a configuration of a space diversity receiver according to a second embodiment of the present invention.

3A and 3B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 1; FIG. 3A is an amplitude frequency characteristic diagram of an output signal of the high frequency amplifier 2 of FIG. 20;
(B) is an amplitude frequency characteristic diagram of each output signal of the NPFs 7a and 7b in FIG.

4A and 4B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 1; FIG. 4A is an amplitude frequency characteristic diagram of an output signal of a PPF 101 in FIG. 21;
(B) is an amplitude frequency characteristic diagram of the output signal of the NPF 111 in FIG.

5A and 5B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver in FIG. 1, and FIG. 5A is an amplitude frequency characteristic diagram of an output signal of the DSC 102 in FIG. 21;
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 112 in FIG.

6A and 6B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 1; FIG. 6A is an amplitude frequency characteristic diagram of an output signal of the PPF 201 in FIG. 21;
(B) is an amplitude frequency characteristic diagram of the output signal of the PPF 221 of FIG.

7A and 7B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 1; FIG. 7A is an amplitude frequency characteristic diagram of an output signal of the DSC 202 of FIG. 21;
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 222 in FIG.

8A and 8B are amplitude frequency characteristic diagrams showing the operation of the space diversity receiver of FIG. 1, and FIG. 8A is an amplitude frequency characteristic diagram of the output signal of the PPF 301 in FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the PPF 321 in FIG.

9A and 9B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 1; FIG. 9A is an amplitude frequency characteristic diagram of an output signal of the DSC 302 of FIG. 21;
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 342 of FIG.

10A and 10B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity receiver of FIG. 1, and FIG.
Is input to the demodulator 11-4 via the multiplexer 21-4 from the DSC 302 of the
21A and 21B are amplitude frequency characteristic diagrams of output signals output from F404a and F404b. FIG. 21B shows the amplitude frequency characteristics of the output signals from the DSC 342 in FIG. 21 after being input to the demodulator 11-4 via the multiplexer 21-4. Each LPF 404a, 404
FIG. 6 is an amplitude frequency characteristic diagram of an output signal output from FIG.

11A and 11B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 2; FIG.
21 is an amplitude frequency characteristic diagram of an output signal of the high frequency amplifier 2 of FIG. 20, and FIG. 21B is an amplitude frequency characteristic diagram of each output signal of the NPFs 7a and 7b in FIG.

12A and 12B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity receiver of FIG. 2, wherein FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 101 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the NPF 111 in FIG.

13A and 13B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity receiver of FIG. 2, wherein FIG.
FIG. 3 is an amplitude frequency characteristic diagram of an output signal of the DSC 102 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 112 in FIG.

14A and 14B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity type reception device in FIG. 2, wherein FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 201 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 202 in FIG.

15A and 15B are amplitude frequency characteristic diagrams illustrating the operation of the space diversity receiver of FIG. 2; FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 301 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the NPF 311 in FIG.

16A and 16B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity receiver of FIG. 2, wherein FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the DSC 302 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the DSC 312 in FIG.

17A and 17B are amplitude frequency characteristic diagrams illustrating an operation of the space diversity type receiving device in FIG. 2, wherein FIG.
Are input to the demodulator 11-2 via the multiplexer 21-2 from the DSC 302 of the
21 is an amplitude frequency characteristic diagram of output signals output from F404a and F404a, and FIG. 21 (b) is a diagram illustrating an example in which the signal is input from the DSC 312 in FIG. Each LPF 404a, 404
FIG. 6 is an amplitude frequency characteristic diagram of an output signal output from FIG.

FIG. 18 (a) is a TDMA-TD of the first conventional example.
It is a frequency characteristic figure of D system, and (b) is a timing chart which shows the time division multiplex format of the TDMA-TDD system of the 1st prior art example.

FIG. 19A is an external view showing a system configuration of a PHS system of a second conventional example, and FIG. 19B is an amplitude frequency characteristic diagram showing a frequency arrangement of the PHS system of the second conventional example. is there.

FIG. 20 shows a third conventional example, a filter bank 10 for channel separation and a demodulator 1 using the filter bank.
FIG. 2 is a block diagram showing a π / 4 shift QPSK receiving apparatus provided with 1;

21 is a block diagram showing a first part of the filter bank for channel separation of FIG. 20.

FIG. 22 is a block diagram showing a second part of the channel separation filter bank of FIG. 20;

FIG. 23 is a block diagram showing the demodulator 11 of FIG.

FIG. 24 is an amplitude frequency characteristic diagram showing an operation of the π / 4 shift QPSK receiving apparatus in FIG. 20, wherein FIG.
21 is an amplitude-frequency characteristic diagram of the output signal of the high-frequency amplifier 2 of FIG.

FIG. 25 is an amplitude frequency characteristic diagram showing an operation of the π / 4 shift QPSK receiving apparatus of FIG. 20, where (a) is FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 101 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the NPF 111 in FIG. 21, (c) is an amplitude frequency characteristic diagram of the output signal of the DSC 102 in FIG. 21, and (d) is the DSC in FIG.
FIG. 11 is an amplitude frequency characteristic diagram of an output signal of the output signal 112;

26A and 26B are amplitude frequency characteristic diagrams illustrating the operation of the π / 4 shift QPSK receiving apparatus in FIG. 20, where FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 201 of FIG.
(B) is an amplitude frequency characteristic diagram of the output signal of the NPF 211 in FIG. 21, (c) is an amplitude frequency characteristic diagram of the output signal of the DSC 202 in FIG. 21, and (d) is the DSC in FIG.
FIG. 12 is an amplitude frequency characteristic diagram of an output signal of a signal 212;

27A and 27B are amplitude frequency characteristic diagrams illustrating the operation of the π / 4 shift QPSK receiving apparatus in FIG. 20; FIG.
FIG. 4 is an amplitude frequency characteristic diagram of an output signal of the PPF 321 of FIG.
21B is an amplitude frequency characteristic diagram of the output signal of the NPF 331 of FIG. 21, FIG. 21C is an amplitude frequency characteristic diagram of the output signal of the DSC 322 of FIG. 21, and FIG.
FIG. 332 is an amplitude frequency characteristic diagram of the output signal of FIG.

FIG. 28 is an amplitude frequency characteristic diagram showing an operation of the π / 4 shift QPSK receiving apparatus in FIG. 20, where (a) is FIG.
23 is an amplitude frequency characteristic diagram of the output signals of the LPFs 404a and 404b of the demodulator 11-1 of FIG. 23, and FIG. 23B is an amplitude frequency characteristic diagram of the output signals of the LPFs 404a and 404b of the demodulator 11-1 of FIG.

FIG. 29 is a block diagram showing an FIR digital filter used to configure the PPF and NPF of FIGS. 21 and 22.

30A is a timing chart showing a sampling signal after impulse response processing of the LPF of the FIR digital filter of FIG. 29 on a time axis with respect to a complex plane, and FIG. 30B is a timing chart of the FIR digital filter of FIG. 5 is a timing chart showing a sampling signal after impulse response processing of an NPF on a time axis with respect to a complex plane.

FIG. 31 shows a third conventional low-pass filter (LP
It is a figure which shows the amplitude frequency characteristic of F).

FIG. 32 is a diagram showing amplitude frequency characteristics of a negative pass filter (NPF) used in the circuits of FIGS. 21 and 22.

FIG. 33 is a diagram showing amplitude frequency characteristics of a positive-pass filter (PPF) used in the circuits of FIGS. 21 and 22.

34 (a) is a timing chart showing a sampling signal before processing by the down-sampling processor of FIGS. 21 and 22 on a time axis, and FIG. 34 (b) is a down-sampling processor of FIGS. 21 and 22; 5 is a timing chart showing, on a time axis, a sampling signal after the processing of FIG.

35 (a) is an amplitude-frequency characteristic diagram of a signal before being processed by the down-sampling processor of FIGS. 21 and 22, and FIG. 35 (b) is a down-sampling processor (Md = It is an amplitude frequency characteristic figure of the signal after the process of 2).

36 shows a π / 4 shift Q used in the receiving apparatus of FIG.
It is a figure which shows the signal space diagram of PSK.

FIG. 37 shows a π / 4 shift Q used in the receiving apparatus of FIG.
FIG. 5 is a diagram illustrating the spectral characteristics of a PSK modulation signal.

FIG. 38 is an amplitude frequency characteristic diagram showing a code identification process in the demodulator 11 of FIG. 22.

FIG. 39 is a block diagram showing a configuration of a fourth conventional example of a space diversity type receiving apparatus configured using the channel separation filter bank 10 of the third conventional example.

[Explanation of symbols]

1a, 1b antenna, 2 high frequency amplifier, 3 distributor, 4a, 4b mixer, 5 local oscillator, 6 π / 2 phase shifter, 7a, 7b low-pass filter (LPF), 8a , 8b A / D converter, 10 filter bank for channel separation, 11, 11-1 to 11-8 demodulator, 11a frequency converter, 11b signal demodulator, 20a, 20b high frequency and intermediate frequency Signal processing unit, 21-1 to 21-4: multiplexer, 22: synthesizer, 100: filter bank operation controller, 101, 201, 221, 301, 321, 341, 3
61: Positive pass filter (PPF), 111, 211, 231, 311, 331, 351, 3
71 ... Negative pass filter (NPF) 102, 112, 202, 212, 222, 232, 3
02, 312, 322, 332, 342, 352, 36
2,372 ... downsampling processor (DSC), 103,113,203,213,223,233,3
03,313,323,333,343,353,36
3,373: Level detector, 401a, 401b: Mixer, 402: Local oscillator, 403: π / 2 phase shifter, 404a, 404b: Low-pass filter (LPF), 411, 413, 415, 423, 425 ... downsampling processor (DSC), 412, 414, 416, 436 ... negative pass filter (NPF), 422, 424, 426, 446 ... positive pass filter (PPF), 417, 427, 437, 447, 461 , 462, a level detector, 418, 428, 438, 448, a comparator, 450, a code discriminator, 451, an adder, 452, a reference clock recovery circuit, 470, a signal demodulation unit operation controller, 501, 501-1 through 501-1 501- (M-1): delay circuit; 502-0 to 502- (M-1): amplifier; 503: adder.

Claims (5)

[Claims] 1. At least first and second antennas each having a plurality of channels having mutually different carrier frequencies and receiving a frequency multiplexed signal, wherein the plurality of channels are arranged so as to have a block shape. A space diversity antenna device, and a first frequency multiplexed signal having a plurality of channels received by the first antenna is frequency-divided such that its spectrum has a block shape such that the spectrum is located in a positive frequency region. First frequency conversion means for converting and outputting a frequency-converted first frequency-multiplexed signal; and converting the second frequency-multiplexed signal having a plurality of channels received by the second antenna into a spectrum. Is frequency-converted so as to have a block shape positioned in a negative frequency region, and the second frequency after the frequency conversion is obtained. A second frequency converter for outputting a number multiplexed signal; a first frequency multiplexed signal output from the first frequency converter; and a second frequency output from the second frequency converter. Synthesizing means for synthesizing the multiplexed signal and outputting a synthesized frequency multiplexed signal; and for the synthesized frequency multiplexed signal output from the synthesizing means, two signals corresponding to the at least two antennas respectively. By processing signals with different circuits,
Channel separating means for separating the frequency-multiplexed signal into only frequency components of signals of two identical channels corresponding to the at least two antennas; and two identical identical signals output from the channel separating means. And a selection output unit for selecting and outputting a signal having a higher power among the channel signals for each channel. 2. A method according to claim 1, wherein said plurality of channels have different carrier frequencies and said plurality of channels are adjacent to each other.
A space diversity type antenna device having at least first and second antennas for receiving frequency multiplexed signals which are arranged so as to have a comb shape by separating one channel from each other by one channel; The first frequency multiplexed signal having a plurality of channels received by the antennas is separated such that its spectrum separates each two adjacent channels of the positive and negative frequency regions by one channel from each other. A first frequency conversion means for performing frequency conversion so as to have a comb shape positioned in a predetermined first frequency region and outputting a frequency-converted first frequency multiplexed signal; The received second frequency multiplexed signal having a plurality of channels is divided into a spectrum having positive and negative frequency ranges. Each two channels in contact separated by one channels each other and the first
A second frequency converting means for performing frequency conversion so as to have a comb-like shape positioned in a predetermined second frequency region different from the frequency region and outputting a frequency-converted second frequency multiplexed signal; The first frequency multiplexed signal output from the first frequency conversion means and the second frequency multiplexed signal output from the second frequency conversion means are combined, and the combined frequency multiplexing is performed. A combining unit that outputs a signal; and a frequency-multiplexed signal after combining output from the combining unit, wherein two signals corresponding to the at least two antennas are processed by different circuits.
Channel separating means for separating the frequency-multiplexed signal into only frequency components of signals of two identical channels corresponding to the at least two antennas; and two identical identical signals output from the channel separating means. And a selection output unit for selecting and outputting a signal having a higher power among the channel signals for each channel. 3. The space diversity receiver according to claim 1, wherein said channel separation means includes at least one filter bank, and said first filter bank is output from said frequency conversion means. A first positive-pass filter for band-filtering only a signal component having a predetermined positive frequency out of a plurality of channel signals and outputting a signal after the band-filtering; Perform downsampling processing on the output signal and process the processed signal,
First processing means for outputting as a signal including at least one channel signal, and band filtering by filtering only a signal component of a predetermined negative frequency out of a plurality of channel signals output from the frequency conversion means A first negative-pass filter that outputs a filtered signal; and a signal that has been processed by performing a downsampling process on a signal output from the first negative-pass filter.
Second processing means for outputting as a signal including a signal of at least one channel, wherein the frequency-multiplexed signals of the plurality of channels are separated into at least two channel signals. Type receiving device. 4. The space diversity receiver according to claim 3, wherein said channel separation means includes at least two stages of filter banks, and a second stage filter bank is output from said first processing means. A second positive-pass filter for band-filtering only a signal component of a predetermined positive frequency in the signal and outputting a signal after the band-filtering, and a signal output from the second positive-pass filter The down-sampling process is performed on the
Third processing means for outputting as a signal including a signal of at least one channel, and band-filtering by band-filtering only a signal component having a predetermined negative frequency in the signal output from the first processing means A second negative-pass filter that outputs a post-signal, and a signal that has been processed by performing a down-sampling process on the signal output from the second negative-pass filter,
Fourth processing means for outputting as a signal including a signal of at least one channel, and band-filtering by band-filtering only a signal component having a predetermined positive frequency in the signal output from the second processing means A third positive-pass filter that outputs a post-signal, and a signal that has been processed by performing a down-sampling process on the signal output from the third positive-pass filter,
Fifth processing means for outputting as a signal including at least one channel signal, and band-filtering by band-filtering only a signal component having a predetermined negative frequency in the signal output from the second processing means. A third negative-pass filter that outputs a subsequent signal, and a signal that has been processed by performing a down-sampling process on the signal output from the third negative-pass filter,
Sixth processing means for outputting as a signal including a signal of at least one channel, wherein the frequency-multiplexed signals of the plurality of channels are separated into at least four channel signals. Type receiving device. 5. The space diversity receiver according to claim 1, wherein a signal of each channel of the frequency multiplexed signal is PSK.
A space diversity receiving device characterized by being a modulated signal.