JPH08179037A - Radar equipment - Google Patents

Abstract

(57) [Abstract] [Purpose] To obtain a radar device having a high limit of integration time even when measuring a received signal having a bandwidth depending on its acceleration and magnitude with respect to a high-speed moving target. [Structure] It is composed of a reference signal generating means, a pulse compressing means, and a pulse Doppler processing means, performs integration processing corresponding to a moving target, and performs range bin / Doppler bin
A moving target corresponding coherent integrator that outputs a three-dimensional signal, and a signal that exceeds the set threshold value of a three-dimensional signal of range bin, Doppler bin, and time, which is obtained by repeating this process on a signal input in time series, is integrated. A non-coherent integrating means corresponding to the moving target for specifying and outputting the moving target value is provided.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a radar device for observing a high-speed moving target.

[0002]

2. Description of the Related Art A conventional radar device for detecting a moving target is disclosed in Japanese Patent Laid-Open No. 215845/1993. Also, as a way to detect a specific point using space
Paul VC Hough et al US Patent 3,069,654, ■ Met
Hod and Means for Recognizing Complex Patterns ■ and those applied to radar equipment by BD Carlson et al. IEEE Transactions on Aeros
pace and Electronic Systems Vol.30, No.1 (1994)
■ Serch Radar Detection and Track with
There is one written in the Hough Transform Part1: System Concept.

First, as a conventional example 1, the one disclosed in Japanese Patent Application Laid-Open No. 5-215845 filed by the same applicant will be described. FIG. 23 is a configuration block diagram showing a first conventional example of a pulse Doppler radar device in which a pulse compression method is a linear frequency modulation method (hereinafter referred to as a chirp method).
FIG. 24 is a configuration block diagram showing a first conventional example of a pulse Doppler radar device in which the pulse compression method is a code modulation method. As will be described later, the pulse Doppler radar device shown in FIGS. 23 and 24 described above, with respect to a high-speed moving target that moves the range bin during observation, changes in phase due to the Doppler effect of the target and moves the range bin of the target. This is a pulse Doppler radar device in which the distance measurement performance and the speed measurement performance for a high-speed moving target are improved by correcting the influence of the above and performing pulse compression and pulse Doppler processing. In FIG. 23 and FIG. 24, 1 is a transmission means for generating and modulating a transmission pulse to widen a frequency band, 2 is a transmission / reception switch, 3 is an antenna, 4 is a receiving means, 5 is a coherent integrating means corresponding to a moving target, and 6 is A detector, 9 is a display.

Next, an outline of the operation will be described. In FIG. 23, a transmission pulse having a pulse width τ is generated from the transmission pulse generator 1a of the transmission means 1, and as shown in FIG. 26 (a), the pulse stretcher 1b produces a pulse width T (T >> τ), A chirp (linear frequency modulation) pulse having a frequency bandwidth Δf (= 1 / τ) is generated. On the other hand, in FIG. 24, a transmission pulse having a pulse width τ is generated from the transmission pulse generator 1a of the transmission means 1, and in the example of 7-bit Barker coded phase modulation, the delay shown in FIG. Element 1d1
And the encoding phase modulator 1d including the adder 1d2
A coded phase modulated pulse having a pulse width T shown in 7 (b) is generated. In any of the chirp method and the code modulation method of FIGS. 23 and 24, the modulated transmission pulse is emitted to the target as a transmission radio wave through the transmission / reception switch 2 and the antenna 3, and the radio wave reflected by the target is transmitted to the antenna 3. Received by
The received signal is converted into a digital complex video signal by the receiving means 4 through the transmission / reception switch 2.

In the receiving means 4, as shown in FIG. 28, the received signal is multiplied by the output of the local oscillator 4b in the mixer 4a and converted into an intermediate frequency signal. Mixer 4a
Output is amplified by IF (intermediate frequency) amplifier 4c and then 2
The signals are distributed and input to the phase detector 4d. In the phase detector 4d, the product of the output signal of the coherent oscillator 4e and the phase of the output signal of the coherent oscillator 4e are set to 90
The phase shifter 4f multiplies the signals sent by 90 ° and the signals are phase-detected. The output of each phase detector is as the real part (I) and the imaginary part (Q) of the received complex video signal,
After being held by the sample holder 4g, it is converted into a digital complex video signal by the A / D converter 4h.

The output data of the receiving means 4 is two-dimensional data of range and pulse hit. Here, the range is a unit of relative distance from the transceiver, and represents half of the distance traveled by light during one sampling time, and the pulse hit is a unit of the number of transmitted pulses. Consider this range direction data for several pulse hits. Here, the number of data points in the range direction (total number of range bins) is N, and the number of data points in the pulse hit direction (this is the number of Fourier transform points and the total number of Doppler bins after Fourier transform) is M. The output of the reception means 4 is input to the moving target coherent integration means 5.

The moving target coherent integrator means 5 has the following structure as shown in FIGS. 23 and 24. Pulse compression means 51 and reference signal generation means 53 for generating a reference signal used for pulse compression in which speed compensation corresponding to the moving target is thinned out.
And pulse Doppler processing means 52 for integrating the signal in the pulse hit direction. The pulse compression means 51 has the following configuration as shown in FIGS. 23 and 24. N in which the data of the buffer memory 51a for storing the output of the receiving means 4 is Fourier-transformed in the range direction by N points
Point FFT calculator 51b, two-dimensional memory 51c that stores the output of N-point FFT calculator 51b, phase fluctuation due to the Doppler effect for each target relative velocity to be detected, and range bin movement of the target from the start of observation A three-dimensional reference memory 51e for storing a reference signal that has been compensated for, a complex multiplier 51d for complex-multiplying the output of the reference memory 51e and the output of the two-dimensional memory 51c, and a buffer memory 51f for storing the complex multiplication result. Inverse Fourier transform in the range direction via N
The point IFFT calculator 51g is provided.

Next, the pulse Doppler processing means 52
Shows the configuration of. A two-dimensional memory 52a for inputting and storing the output of the pulse compression processing means 51, and M for performing Fourier transform by reading data from the two-dimensional memory in the pulse hit direction.
A point FFT calculator 52b and data for a required Doppler frequency of the output of the M point FFT calculator 2 are stored.
The dimensional memory 52c is provided.

In the conventional example shown in FIGS. 23 and 24, in order to eliminate the following effects on the pulse compression due to the Doppler effect, the reference signal generating means 53 has a thinning Doppler after the reference signal generator 53a. A compensator 53b is added. Now suppose a moving target within a range bin has a relative velocity v that produces a Doppler frequency fd. In the case of the chirp method, the time-frequency characteristic (shown by the dotted line in FIG. 29) of the received pulse reflected by the moving target having the relative speed v is as follows. Frequency characteristics (Fig. 2
(Indicated by the solid line 9) by a frequency fd. Since the delay time vs. frequency characteristic of the pulse compression means of the conventional pulse Doppler radar device corresponds to the case where the target relative speed is zero, the compressed portion of the received pulse reflected by the moving target is shortened, In the 29 example,
The pulse compression is performed only in the region between the frequencies f1 and f2 of the characteristic shown by the dotted line, and the pulse compression means output × (Δ
loss of td / T) occurs. Further, in the pulse compression means, a delay of Δtd = (T / Δf) · fd occurs, which corresponds to a decrease in the display range of the (Δtd / τ) range bin.

In the case of the code modulation system, the Doppler frequency f
Due to the frequency shift of d, the phase rotation of the received pulse occurs, the phase characteristics of the pulse before pulse compression and the reference signal become inconsistent, the range sidelobe level increases, the compression ratio decreases, the position shift and separation of the compressed pulse, etc. Due to the deterioration of the pulse compression characteristic of, the distance measurement error of the high-speed moving target or the measurement failure is accompanied. For example, when a Barker code is used for the code sequence, if a phase rotation of 2π occurs in a pulse having a pulse width T before compression due to Doppler shift, the reference signal and the reference signal are completely out of phase. FIG. 30 shows an example of a pulse compression waveform when the 13-bit Barker code is used. In the figure, the sample point represents a range bin, and the sample point 13 represents a range bin in which a target exists. The output is the value after pulse compression. When a phase rotation of 2π occurs in a pulse having a pulse width T before compression, the following equation is obtained. 2πfd · T = 2π [rad] (1) Therefore, fd = 2v / λ (λ: radar transmission wavelength)
To the above equation, the relative velocity v of the moving target is v
= Λ / 2T, the output of the sample point 13 where the target exists becomes zero, and compression becomes impossible.

Furthermore, when the relative velocity of the target has a magnitude that causes range bin movement during the observation time, the pulse is compressed to a range in which the target exists at each time point, so that the target exists at a certain time within the observation time. Focusing on the range bin, the observation time is shortened as compared with the case where the target range bin movement does not occur. Therefore, the integration effect and the Doppler resolution are reduced in the pulse Doppler processing, and the target speed measurement performance is deteriorated.

Therefore, although the Doppler compensation is performed, the required Doppler resolution (1 Doppler bin width) is smaller than the required fineness of the Doppler compensation. Therefore, the thinning-out type Doppler compensator for thinning the Doppler compensation value is used. 53
Use b. For the relative velocity of each target that produces the Doppler frequency compensated by the thinning Doppler compensator 53b, the reference that compensates the distance traveled by the target from the start of observation at the time of transmitting each pulse for the time it takes for the radio wave to propagate. The signal is prepared. for that reason,
Range bin compensator 5 after thinning Doppler compensator 53b
3c is added. In this way, the reference signal compensated at each Doppler frequency and the received signal including all Doppler frequency components are complex-multiplied, and then inverse Fourier transform is performed in the range bin direction to perform pulse compression.

Further, in the conventional example 1, the N-point FFT calculator 51b is used.
After the above, a two-dimensional memory 51c is provided, which is used for complex multiplication of the output obtained by Fourier transforming each pulse in the observation time in the range bin direction by the N-point FFT calculator 51b by the reference signal, The signal is read from the dimensional memory 51c. In the reference memory 51e, the output of the reference signal generator 53a (one-dimensional data) is Doppler-compensated (two-dimensional data) by the thinning Doppler compensator 53b, and the range bin-compensated output of the range bin compensator 53c (three-dimensional data). 3D memory is used to store).

Next, an outline of operation will be described. First, the reference signal generation which is generated by the reference signal generating means 53 of FIGS. 23 and 24 and which is stored in the reference memory 51e of the pulse compression means 51 in advance will be described.
The reference signal generating means 53 includes a reference signal generator 53a, a thinning Doppler compensator 53b, a range bin compensator 53c, and an N-point FFT calculator 53d.
The reference signal output from the point FFT calculator 53d is input to the reference memory 51e. Then, the signal processing mode of the pulse Doppler radar device is changed, and reference signal generation parameters (number of range bins N, transmission pulse width T,
As long as the carrier frequency f0 and the sampling time ts) do not change, the same reference signal is used.
The reference signal generating means 53 is disconnected once the reference signal is input to the reference memory 51e. Then, when the mode is changed to the next signal processing mode, the reference signal generating means 53 is connected again, and a new reference signal is re-input to the reference memory 51e. The reference signal generator 53a is a transmission pulse generated by the transmission means 1 (in the case of the chirp system and the code modulation system, FIG.
27 (b) is sampled at a sampling time ts, and the signal is represented by TR (r), a reference signal R (r) represented by the following equation is generated. (In the case of the chirp system and the code modulation system, FIG.
(B), It shows in FIG.27 (c). Here, r represents a range bin, and time is divided by a sampling time ts (r = t / ts).

[0015]

[Equation 1]

Next, the reference signal R (r) is input to the thinning-out type Doppler compensator 53b, and a reference signal Rp (r, fd) having a time-frequency characteristic in which the Doppler frequency is shifted by fd is obtained. Here, a plurality of Doppler frequencies fd are assumed because the target relative velocity is unknown. When the peak level of the compressed pulse during the observation time is within one range bin, it is not necessary to perform Doppler compensation and range bin movement compensation. Therefore, the Doppler frequency required for Doppler compensation is determined in rough increments so that the range bin movement amount during the observation time is within 0.5 range bins. The step of Doppler frequency required for compensation is set as in the following equation (3). Δfd ≦ (0.5 · f0 · ts) / (PRI · M) (3) (f0: carrier frequency, ts: sampling time,
(PRI: pulse repetition period)

From this, the Doppler frequency used for compensation is, for example, as follows. fd = i · (M / 2) · Δfd (4) (i = 1, 2, ... L, L: any positive number)

The output Rp (r, fd) of the thinned-out type Doppler compensator 53b can be expressed by the following equation using the fd thus obtained. Rp (r, fd) = R (r) .exp (-2.pi.fd.r.ts) (5) The output Rp (r, r of the thinned Doppler compensator 53b
fd) is input to the range bin compensator 53c.

In the range bin compensator 53c, for each target velocity assumed in the thinning-out type Doppler compensator 53b, the time required for the pulse to propagate the distance traveled by the target from the start of observation is calculated, and the reference signal is calculated. I moved it only for that time. The target relative velocity vd with respect to the Doppler frequency fd used in the thinning Doppler compensator 53b is expressed by the following equation. vd = fd · C / 2f0 (6) (f0: carrier frequency, C: speed of light) Then, at the target where the Doppler frequency is fd,
The time required for the pulse to propagate the distance traveled by the target when the p pulse hits from the start of observation is sampled at the sampling time ts, t (f
d, p) is given by the following equation. t (fd, p) = vd · p / (PRF · C · ts) = fd · p / (PRF · 2f0 · ts) (7)

In order to cancel this t (fd, p), a reference signal which is moved by t (fd, p) on the time axis is also used. This reference signal Rpp (r, f
d, p) is shown by the following equation. Rpp (r, fd, p) = Rp (r, fd) .u [N- (rt (fd, p))] (8) where u (r) = 1,0≤r <(T / ts) u (r) = 0, (T / ts) ≦ r <N

Next, the output Rpp of the range compensator 53c
In order to convert (r, fd, p) into spatial frequency domain data, Fourier transform is performed in the range direction by the N-point FFT calculator 53d, and the spectrum RRpp (fr, fd of the reference signal shown by the following equation is obtained. , P) is obtained. Here, the data subjected to the Fourier transform in the range direction is the data in the spatial frequency domain. RRpp (fr, fd, p) = FR [(r, fd, p)] (9) (FR: Fourier transform in the range direction) The spectrum of the reference signal RRpp (fr,
fd, p) are stored in the reference memory 51e in advance. Once stored, the reference signal generating means 53
Separated from 51e. The method of generating the spectrum RRpp (fr, fd, p) of the reference signal is the same for both the chirp method and the code modulation method.

Next, an outline of the operation of the pulse compression means 51 and the pulse integration (pulse Doppler) processing means 52 will be described with reference to FIG.
This will be described with reference to the flowchart in FIG. In FIG. 25, in step S1, the output data S of the receiving means 2 is output.
In order to convert (r) into data in the spatial frequency domain, N
Fourier transform is performed in the range direction by the point FFT calculator 51b.
The output of this N-point FFT calculator can be expressed by the following equation. SR (fr) = FR [S (r)] (10) (fr: spatial frequency, FR: Fourier transform in the range direction) This is repeated for pulse hits and SR (fr,
p) and input to the two-dimensional memory 51c.

In step S2, the output S of step S1 is output.
R (fr, p) and the reference signal RRpp (fr,
fd, p) is subjected to complex multiplication to obtain a spectral component whose phase is constant over all frequencies. This result is URp
Let p (fr, fd, p). URpp (fr, fd, p) = SR (fr, p) × RRpp (fr, fd, p) (11) This is repeated for the spatial frequency fr and stored in the buffer memory 51f.

In step S3, the complex multiplication result URpp
In order to convert (fr, fd, p) into data in the time domain, inverse Fourier transform is performed in the spatial frequency direction, and U shown in the following equation is obtained.
Let pp (r, fd, p). Upp (r, fd, p) = Ffr-1 [URpp (fr, fd, p)] (12) (Ffr-1: Inverse Fourier transform in frequency direction)

In step S4, the processes of steps S2 and S3 are repeated for the pulse hit p,
It is stored in the two-dimensional memory 52a. This inverse Fourier transform result is read from the two-dimensional memory 52a, and the M point FFT calculator 52b is read.
Performs M-point Fourier transform in the pulse hit direction and performs pulse Doppler processing. By the above process, the center of the Doppler frequency fd2 is fd for each range bin.
Then, (fd−PRF / 2) ≦ fd2 <(fd + PRF /
2), (where PRF: pulse repetition frequency) range is decomposed into Doppler frequency components that change in steps of Δfd2 = PRF / M, and the signal power is integrated into the Doppler bin corresponding to the target relative velocity. Thus, the target relative speed can be detected. The output of the M-point FFT calculator can be expressed by the following equation. Up (r, fd, fd2) = FP [Upp (r, fd, p)] (13) (FP: Fourier transform in pulse hit direction)

This Fourier transform in the pulse hit direction is repeatedly stored in the two-dimensional memory 52c for the range bin r. At this time, only the data in the effective range of fd2 is stored in the two-dimensional memory 52c. This effective range is (fd−Δfd / 2) ≦ fd2 <(fd + Δfd /, using the rough step Δfd of the Doppler frequency required for correction shown in equation (3).
It can be represented by 2). Here, the number of fd2 is K = L × (Δf
d / Δfd2). The above step S4 is repeated for the Doppler frequency fd to obtain N of Up (r, fd2).
It is stored in the memory 52c as two-dimensional data of × K. By performing Fourier transform and pulse Doppler processing as described above, even if the Doppler frequencies used for Doppler compensation and range bin shift compensation are thinned out and reduced, the Doppler resolution does not deteriorate.

Further, here, the pulse-Doppler processing means 52 uses an M-point FFT calculator 52b, and a two-dimensional memory 52 is used.
Only the data within the effective range of the output of the M-point FFT calculator is stored in c, but M is used instead of the M-point FFT calculator.
You may make it calculate only the data of an effective range using a point DFT calculator.

The above radar device integrates a coherent signal having phase information. Coherent integration has high integration efficiency, but if the signal has a bandwidth due to the target having acceleration or magnitude, the integration time will be limited. That is, if the integration time is extended, the Doppler resolution Δf
Although d becomes small and the signal concentrates at one point to improve the integration performance, if the bandwidth of the target signal exceeds Δfd, the performance improvement due to integration cannot be obtained even if the integration time is made longer. In such a radar device that performs coherent signal processing, there is a limit in improving performance by integrating signals.

Next, as a second conventional example, a method of easily obtaining a straight line trajectory by plane conversion will be described. As an example of this,
Paul VC Hough et al US Patent 3,069,654, ■ Met
The ones listed in hodand Means for Recognizing Complex Patterns ■ are listed. This relates to the recognition of complex patterns in photographs and other photographic representations, more specific methods of complex straight lines and machine recognition means. Here, the study of the path of fine molecules passing through the observation region will be described, but it can be applied to the radar display.

FIG. 31 is an explanatory diagram of plane conversion of a straight line segment. FIG. 32 is a configuration block diagram of a straight line detection device from a camera image for explaining the second conventional example. FIG.
FIG. 33 is a detailed configuration block diagram of the electrical plane conversion circuit in Conventional Example 2 shown in FIG. 32. In FIG. 31, 205
Is a frame on the space, 202, 203, 204, are straight line segments drawn in the frame 205, 200 is a conversion diagram, 202a,
203a and 204a are spatial transformations of 202, 203 and 204, respectively. The geometric structure of the plane transformation is performed according to the following rules.

Rule (1). One point on the line segment of the frame 205 is represented by a straight line in the conversion diagram. Rule (2). The point on the straight line on the upper side of the frame 205 is a straight line on the conversion plane and is inclined 45 degrees to the right. A point on a straight line located in the middle of the frame 205 with respect to the horizontal direction becomes a vertical straight line on the conversion plane. The point on the straight line on the lower side of the frame 205 is a straight line on the conversion plane and is inclined to the left by 45 degrees. In general, the lines on the transformation plane are frame 20.
An angle proportional to the length is given to a perpendicular line drawn from the horizontal middle line 206 of 5 to a point on the line segment. Rule (3). Each line on the conversion plane intersects the horizontal middle line of the conversion diagram 200 at coordinates equal to the horizontal coordinates of each point on the line segment of the frame 205.

According to the above rules, 207a in the plane transformation 202a is drawn for a given point 207 on the line segment 202. The point 207 is located at the top of the frame 205 and in the middle of the horizontal intermediate line 206, so the line 207a is inclined to the right by about 45 degrees. The intersection of the horizontal middle line 201 and the line 207a in the conversion diagram 200 is
The distance from the left frame is equal to the horizontal coordinate of the point 207 of the line segment 202.

As a certain rule, if a series of points in the frame lie on a straight line segment, the plane-transformed line is a knot 20.
Intersect at a single point, represented as 8. The Cartesian coordinates of the knot 208 on the transformation diagram 200 represent the following properties. Characteristic (1). The horizontal coordinate of the knot 208 is equal to the horizontal coordinate of the frame 205 where the horizontal middle line 206 of the frame 205 and the line segments 202, 203, 204 intersect. Characteristic (2). The vertical coordinate of the knot 208 with respect to the horizontal middle line 201 in the conversion diagram 200 is proportional to the magnitude of the tangent angle between the vertical line and the line segments 202, 203, and 204. Plane conversion like this
The coordinates of the knot 208 at 202a, 203a, and 204a are the frame 205.
The intercepts and slopes of the line segments 202, 203, 204 of are given.

Next, the behavior of an electrical or similar device will be described. FIG. 33 is a configuration block diagram of an apparatus for explaining the second conventional example. In the figure, 210 is an input photo frame, 211 is a television camera, 212 is an electric plane conversion circuit, 213 is an oscilloscope, 214 is a television camera, 215 is a magnetic tape recorder, and 216 is a computer.

In FIG. 32, a picture containing a complex pattern, such as a bubble-like photograph, is divided into hundreds of rectangular areas or frames. The height of each frame is set so small that the pattern in the frame becomes a straight line and large enough so that the line segment can be surely recognized from the random bubbles in the background. The width of the frame is set according to the accuracy required to measure the position of the line segment within the frame.

A television camera 211, for example of the image pickup tube type, scans a frame 210 which contains one or more line segments of bubbles. An output pulse is produced as the scanning beam of television camera 211 passes over the bubbles on the scan line.
For each output pulse of the television camera 211, the electrical plane conversion circuit 212 draws a plane converted line on the display of the oscilloscope 213 according to the geometrical rules set forth in FIG. In this way, the coordinates of the knot of the plane-converted display on the display of the oscilloscope 213 indicate the inclination and the intercept of the line segment of the frame 210 as shown in FIG.

The second television camera 214 scans the plane-converted display of the oscilloscope 213 with an image pickup tube, for example, and detects the coordinate data of the knot. This data, which is the output of the second television camera 214, is input to and stored in the magnetic tape recorder 215. The magnetic tape is placed in the calculator 216 to calculate the coordinates of each line segment and recognize the complex pattern of the original picture.

When an image pickup tube type television camera scans the appearance of a bubble room, the bubbles appear within the scan line as a narrow range where the video output voltage is well below the voltage on either side of the background. The background video signal has fluctuations, and even in this fluctuating background, unnecessary parts in the room are omitted,
Means for recognizing bubbles must be provided. Video pulses must meet the following two basic criteria to be recognized as bubbles. That is, (a) the standard of narrowness.
The bubble marking the path must be narrow and of fairly constant width. Only pulses with that width are detected (with some tolerance). The wider and more obscure is ignored. (B) Difference threshold standard. The light intensity difference between the dark path and the light background on both sides must be greater than some minimum value. This threshold can be easily adjusted with system parameters. It is set to obtain the most reliable path detection and to improve the background noise removal performance.

Referring now to FIG. 33, there is shown a detailed diagram of the circuit 212 which converts the pulses from the television camera 211 displaying bubbles in the line of sight into a more efficient plane conversion pattern. For the sake of clarity, here, only one bubble on the line segment of the frame 210 is taken out, but the other bubbles are the same.

The video signal from the first television camera 211 is provided as the first input of the differential amplifier 221 without delay, and 0.4 times as the second input of the differential amplifier 221.
[Μs] Delayed. The difference in amplitude between the two outputs of the difference amplifier 221 represents the light intensity difference at two points along the scan line of the first television camera 211 divided by half the bubble width of the line segment of the frame 210. The output of the differential amplifier 221 corresponding to the input at 0.4 μs is given to the first Garwin simultaneous generation circuit 224 with a delay of 0.1 μs delay line. The other output of the difference amplifier approximately delays the other input of the Garwin circuit by 0.5 μs so that the two signals arrive at the coincidence circuit at the same time. 2 of bubble width of line segment of frame 210
If more than double and ambiguous, the Garwin circuit 224 cannot be triggered and is ignored. The output pulse amplitude of the Garwin simultaneous generation circuit 224 is due to the difference in luminous intensity between the bubbles of the line segment and the general background. If the output pulse of the Garwin simultaneous generation circuit 224 is small, the intensity of the general background changes. These are removed by inputting the output of the Garwin simultaneous generation circuit 224 to the 0.5 μs monostable multivibrator 225. In the 0.5 μs monostable multivibrator 225, the trigger bias is set so that the pulse from the bubble in the line segment of frame 210 has sufficient amplitude to trigger the multivibrator 225. In this way, a monopulse output is obtained from the multivibrator 225 when the scanning beam of the first television camera 211 passes through the bubble of the line segment of the frame 210.

The output pulse of the multivibrator 225 is
Trigger the 2 μs monostable multivibrator trigger circuit 226. Monostable multivibrator trigger circuit
One output of 226 delays the pulse by 0.3 μs: 1.
Cut and amplify with a length of 4 μs, oscilloscope 213
Move the unblanking amplifier 227 that fits the cathode of the cathode ray tube. This pulse is thus about 1.5 μs, which is sufficient to describe the corresponding line transformation to scan the bubble.
During this period, the cathode ray tube beam of the oscilloscope 213 is turned on. The second output of the multivibrator trigger circuit 226 drives a clipper 228 which provides a pulse output of 0.3 μs at the tip of the output pulse of the monostable multivibrator trigger circuit 226.

The output of the clipper 228 is input to the set pulse amplifier 229, amplified and fixed at 15 V for 0.3 μs, and then input to the fixed line generator 230. Two
The μs output pulse is output from the clipper 228 in the same manner as the 2 μs output pulse of the monostable multivibrator trigger circuit 226. The 2 μs output pulse from the clipper 228 is input to the reset amplifier 231 and is amplified and inverted.
Both the inverted 2 μs pulse from the reset amplifier 231 and the 15v output pulse from the set pulse amplifier 229 are simultaneously input to the fixed line generator 230.
The 15v output pulse input to the fixed line generator 230 is linearly attenuated to -15v. The 2 μs reversal pulse from the reset amplifier 231 gates the decay of the 15v pulse from the set pulse amplifier 229,
Set to -15v. The resulting fixed line generator 23
The 2 μs linear damping waveform output from 0 is amplified by the amplifier 232 and applied to the vertical deflection plate of the oscilloscope 213.

The 0.3 μs pulse from the clipper 228 is input to the set pulse modulation amplifier 233 and modulated. The modulation is provided by a vertical sawtooth wave generator 234 which is synchronized with the vertical deflection of the television camera 211. The modulation is such that when the vertical deflection of the TV camera 211 is at the top of the TV field of view, the amplitude of the 0.3 μs pulse is 50v, and the TV camera 2
The pulse amplitude is linearly reduced to 10 V when the vertical deflection of 11 is at the earliest. The 0.3 μs pulse from the set pulse modulation amplifier 233 is input to the variable line generator 235. Therefore, the variable amplitude of the set pulse is set to 25v for a while when the vertical deflection of the TV camera 211 is at the top of the TV field of view, and set to 5v when the vertical deflection comes to the end, and linearly attenuates during that time. To do. The variable line generator 235 has a planned attenuation factor, while vertical deflection is at the top of the television field of view −
While it is at the bottom at 25v, it attenuates to a linear waveform of -5v. The 2 μs inversion pulse from the reset amplifier 231 is input to the variable line generator 235 at the same time as the 0.3 μs set pulse from the set pulse modulation amplifier 233, and the set pulse is controlled to be cut off by the negative voltage. As a result, the 2 μs variable amplitude linear decay pulse from variable line generator 235 is provided to the input of adder 236.

2 μs from fixed line generator 230
The linear damping pulse is inverted by the inversion circuit 237 and input to the adder 236. The output is obtained from the horizontal deflection circuit of the television camera 211, amplified by the horizontal deflection amplifier 238 and added to the adder 236. The adder 236 operates with three inputs as follows. If the trigger is applied when the vertical deflection of the TV camera 211 is at the top of the TV field of view, the 2 μs output pulse of the variable line generator 235 starts at 25v. The fixed line generator 230 2 s reverse pulse always begins at -15 v. Adder 236 adds these two pulses for linear attenuation in the range 10v to -10v. If the 2 μs pulse of the variable line generator 235 is triggered at the bottom of the television field of view of the television camera 211, it will increase linearly in the range −10 v to 10 v. If the 2 μs pulse of the variable line generator 235 is triggered in the middle of the TV field of view of the TV camera 211, it will have a zero output. The output of the horizontal deflection amplifier 238 is the variable line generator 235.
And the variable linear amplitude of the fixed line generator 230, amplified by the amplifier 232 and applied to the horizontal deflection plate of the oscilloscope 213.

In this way, one line is drawn in the plane conversion for each bubble in the line segment of the frame 210. Oscilloscope 213
Fixed line generator 230 given to the vertical deflection plate of the
The linear sweep output of the 2 μs reverse linear decay from the fixed line generator 230 produces a straight line in the plane transform with an angle commensurate with the vertical displacement of the detected bubble path of the line segment of frame 210. The pulse and 2 μs variable amplitude linear attenuated output from the variable line generator 235 works with a variable amplitude linear sweep generated by applying pulses. If the detected bubble is at the top of frame 210,
The horizontal deflection applied to the horizontal deflection plate of the oscilloscope 213 is large and clear at first, and then attenuates linearly. If the detected bubble occurs in the middle of frame 210, the horizontal deflection is zero, and if it is below the middle, it decays linearly with a large negative polarity initially. The output from horizontal deflection amplifier 238 causes the focus on the display of oscilloscope 213 to follow the horizontal scan beam of television camera 211. When the horizontal scan beam intersects the detected bubble, the focus of oscilloscope 213 is at the horizontal position of the detected bubble and the video pulse at this moment causes the line transform to be drawn as described above. The time required to draw one line in the transformation is 1.5 μs. The delayed unblanking pulse of unblanking pulse delay amplifier 227 gates the oscilloscope during this time. The set and reset of line generators 230, 235 are not found within this conversion.

The above-described process is repeated when the scanning beam of the television camera 211 intersects the bubble of the line segment of the frame 210, and the plane transformation is drawn as shown in FIG. Although we have described the display of one frame above, it is possible to represent several frames at the same time. Here, the device for investigating the path of fine molecules passing through the observation region of the bubble room has been described.
It can also be applied to radar displays and map reading.

The above-mentioned conventional example 2 is an example using a circuit for performing plane conversion of a signal by controlling the voltage. An example in which the same conversion as this is performed by digital calculation using a computer will be shown. As Conventional Example 3, an example in which plane conversion is incorporated in a radar device will be described. This is because B. D. IEEE Transactions on Aerospace and Electronic Sy by Carlson et al.
Vol.30, No.1 (1994) of stems ■ Serch Ra
dar Detection and Track with the Hough Transform P
It is shown in art 1: System Concept ■.

FIG. 34 is a block diagram showing the structure of a radar device of the third conventional example. In the figure, 1 is a transmitting means, 2 is a transmission / reception switch, 3 is an antenna, 4 is a receiving means, 8 is a non-coherent integrating means corresponding to a moving target, and 9 is a display.

Next, an outline of the operation will be described. In FIG. 34, a transmission pulse having a pulse width τ is generated from the transmission pulse generator 1a of the transmission means 1, amplified by the amplifier 1c, and the amplified transmission pulse is emitted to the target as a transmission radio wave through the duplexer 2 and the antenna 3. The radio wave reflected by the target is received by the antenna 3, and the received signal is converted into a digital video signal by the receiving means 4 via the transmission / reception switch 2.

In the receiving means 4, as shown in FIG. 35, the received signal is multiplied by the output of the local oscillator 4b by the mixer 4a and converted into an intermediate frequency signal. Mixer 4a
Is amplified by an IF (intermediate frequency) amplifier 4c, held by a sample holder 4g, and then converted into a digital video signal by an A / D converter 4h.

The output data of the receiving means 4 is two-dimensional data of range pulse hit (or range scan). Here, the range is a unit of the relative distance from the transceiver, and represents half of the distance that light travels during one sampling time. The pulse hit is the unit of the number of transmitted pulses. The scan is the beam scanning within the scanning range. Then, it is a unit of the number of times of returning to the same area, and either pulse hit or scan is used depending on the situation. Scan is used here for the sake of explanation. Consider this range direction data for several scans. Here, the number of data points in the range direction (the number of all range bins) is n, and the number of data points in the scan direction is m. The output of the receiving means 4 is input to the moving target correspondence non-coherent integrating means 8.

The non-coherent integrating means 8 corresponding to the moving target has the following structure as shown in FIG. A memory 8a for storing the output of the receiving means 4, and a memory
The first intensity detector 8b that compares the data (amplitude value) of 8a with the first threshold and detects that the threshold is exceeded.
A memory 8c for storing the amplitude value detected by the first intensity detector 8b, a memory 8d for storing the array element number detected by the first intensity detector 8b, and a target route for the target velocity v and the target velocity v. A conversion matrix memory 8e for storing a conversion matrix for converting the current position R0 as shown by the distance ρ from the origin of the observation space of the distance channel and its angle θ.
And a matrix multiplier 8f that reads data from the memory 8d and the conversion matrix memory 8e and performs matrix multiplication.
An address calculator 8g that calculates which address corresponds to the result obtained by the matrix multiplier 8f, a memory 8h that stores the address obtained by the address calculator 8g, and a memory
The result of the intensity integrator 8i that integrates the amplitude value read from the memory 8c to the address read from 8h and the result of the intensity integrator 8i are compared with the second threshold, and the integrated value of the address and amplitude that exceeds the second threshold is detected. The second intensity detector 8j that does
A θ-ρ; v-Ro converter 8k for converting to -R0 and a display 9 for displaying the detection result are provided.

The θ-ρ conversion used in Conventional Example 3 will be described. That is, the trajectory conversion is facilitated by the plane conversion of Conventional Example 2. FIG. 36 is a diagram showing a target route on a distance-time plane (xy plane). Optional goal 1 shown
2 of the straight line α which is the route of and the straight line β which is the route of target 2.
An arbitrary point on one straight line is transformed on the θ-ρ plane by the following transformation formula.

Ρ = x · cos θ + y · sin θ (14)

Any point on the xy plane is converted into a curve on the θ-ρ plane shown in FIG. Each of these curves intersects at a point on the θ-ρ plane that indicates the length (ρ) and the angle (θ) of a perpendicular line drawn from the origin toward the straight line on the xy plane. Therefore, if the power at the point on the straight line is converted on the θ-ρ plane and integrated, the power is accumulated at the points P and Q indicating the straight line. That is, the point detection is performed rather than the straight line detection, and the detection accuracy is improved.

Next, a specific description will be given. Input signal memory
8a is read out, compared with the first threshold value T1 in the first intensity detector 8b, an array D listing the array numbers of points that are T1 or more, and an array S of signal intensity data are created,
The array S is stored in the memory 8c, and the array D is stored in the memory 8d. Array D is the range bin number r (1 at the point beyond the threshold.
≦ r ≦ N) and scan numbers t (1 ≦ t ≦ J) are arranged (they may be in any order) and are shown in the equation (15). Array S
Is the signal intensity at the point corresponding to array D and is shown in equation (16).

[0057]

[Equation 2]

The matrix multiplier 8f converts the array D into (1
The transformation is performed on the θ-ρ plane according to the equation (4), and the transformation matrix H used for the transformation is shown in the equation (17). The conversion matrix H is obtained in advance and stored in the conversion matrix memory 8e. In matrix multiplier 8f,
Matrix multiplication is performed according to the conversion equation (14) to obtain a matrix R indicating a place where the signal is integrated. The matrix R is shown in equation (18).

[0059]

(Equation 3)

Next, in the address calculator 8g, ρi, j of the matrix R is ρ for each θj and data i of numbers 1 to n.
find k that is closest to k (k = 1 to l)
And stored in memory 8h. In the intensity integrator 8i,
The array S is read from the memory 8c and ρi, j is read from the memory 8h, and the i-th data of the array S is integrated into X (j, ρi, j). In the second intensity detector 8j, the integration result is compared with the second threshold value T2, and θj, ρ of the point that is equal to or greater than T2.
Find i, j. The θ-ρ; v-R0 converter 8k converts θ-ρ detected by the second intensity detector 8j into v-R0. The target route can be expressed by using the current position R0 of the target and the velocity v as shown in equation (19). By comparing equation (19) with modified equation (20), θ, ρ
Can be used to obtain R0 and v as in equations (21) and (22).

R = R0−t · v (19) r = (ρ / cos θ) −t · (sin θ / cos θ) (20) R0 = ρ / cos θ (21) v = sin θ / cos θ (twenty two)

The target speed and the current position are obtained as described above and displayed on the display unit 9. In the non-coherent integration using only the amplitude information as in the conventional examples 2 and 3, within the range in which the target route is represented by a straight line, the limit of the integration time does not occur unlike the coherent integration in the conventional example 1. However, the integration efficiency is lower than that of coherent integration. As described above, conventionally, the conventional examples 1 and 3 are individually performed, and there is a limit to the S / N improvement effect by integration.

[0063]

The conventional radar device is constructed as described above, and the coherent integrator alone has a high efficiency, but there is a limit that it does not improve beyond the resolution even if the integration time is increased. However, the coherent integrator only has a high limit value but low efficiency.

The present invention has been made in order to solve the above problems, and is not affected by the phase fluctuation due to the Doppler effect of the target and the range bin movement of the target with respect to the high-speed moving target accompanied by the range bin movement during observation. It is an object of the present invention to obtain a radar device in which the limit of integration time is increased even when the received signal has a bandwidth depending on the target acceleration and magnitude.

[0065]

A radar apparatus according to the present invention comprises a reference signal generating means, a pulse compressing means, and a pulse Doppler processing means, performs integration processing corresponding to a moving target, and performs a range bin / doppler bin operation. A moving target corresponding coherent integrator that outputs a dimensional signal,
Doppler bin selecting means for selecting a candidate Doppler bin from the amplitude or power of the two-dimensional signal output of the range bin / Doppler bin, and a range bin / time two-dimensional signal obtained by repeating this processing in time series input signals. A non-coherent integrating means corresponding to the moving target for integrating the address of the signal exceeding the threshold value into the address group obtained by converting using the conversion matrix and specifying and outputting the moving target value is provided.

Alternatively, a moving target coherent integration which is composed of a reference signal generating means, a pulse compressing means, and a pulse Doppler processing means, performs integration processing corresponding to the moving target, and outputs a two-dimensional signal of a range bin / Doppler bin. Means and range bin, Doppler bin, and time that are obtained by repeating this process on a signal input in time series.
A non-coherent integrating means for moving target is provided, which spatially transforms the signals exceeding the set threshold value of the dimension signal and then integrates them to specify and output the moving target value.

Furthermore, after integrating the signals for each address, an output adjuster for determining the significance of the integrated value is added to the moving target correspondence non-coherent integrating means.

Furthermore, the non-coherent integrator corresponding to the moving target detects the amplitude or power value of the input signal in the non-coherent integrator corresponding to the moving target, or the integrated value after the address conversion by a constant false alarm probability detector. A significant signal was set.

Furthermore, the moving target correspondence non-coherent integrator has an output memory for specifying and storing a moving target value, and a detected target signal deleting device for deleting an input signal corresponding to the specified moving target. Then, first, when the first moving target is specified, the corresponding input is deleted to further specify the remaining moving target.

Furthermore, the non-coherent integration means corresponding to the moving target has an output memory for specifying and storing the moving target value, a first conversion matrix memory having a coarse resolution and a second conversion matrix having a fine resolution for the spatial converter. A memory is added, and when the movement target is specified by the first conversion matrix memory, the movement target is further specified by the second conversion matrix memory.

The radar apparatus according to the present invention is composed of pulse compression means without pulse velocity compensation and pulse Doppler processing means, performs integration processing corresponding to a moving target, and performs range bin.
It is composed of a coherent integrator that outputs a two-dimensional signal of the Doppler bin, a reference signal generator, a pulse compressor, and a pulse Doppler processor.
A moving target corresponding coherent integrator that outputs a two-dimensional signal, and a two-dimensional range bin Doppler bin signal, which is the output of this coherent integrator or is switched to output the coherent integrator corresponding to the moving target, into a signal input in time series. Non-coherent integrator means for integrating a repeated three-dimensional signal of range bin, doppler bin, and time with respect to each of range bin and doppler bin to specify and output a target value, and this coherent integrator output or switching target A three-dimensional signal of range bin, doppler bin, and time obtained by repeating a two-dimensional signal of range bin, doppler bin, which is the output of the corresponding coherent integrator, into a signal input in time series is integrated with respect to time for each of range bin, doppler bin. To specify the movement target value Comprising a moving target corresponding noncoherent integration means for force, an integral number control means for controlling to the selection of connections between the integrated number of settings for each integrating means consider the conditions determined by the observed object.

[0072]

In the radar apparatus according to the present invention, first, the received signal is subjected to velocity compensation with respect to the moving target by the coherent integration means to obtain the signal distribution of the two-dimensional range bin Doppler bin signal and the Doppler data is obtained from the obtained data. Select a bin, scan multiple time series, accumulate,
Two-dimensional scan data is obtained and integrated by non-coherent integrator using plane conversion.

Alternatively, the received signal is first subjected to velocity compensation with respect to the moving target to obtain a signal distribution of a two-dimensional range bin Doppler bin signal, and the obtained data is spatially transformed during the scanning period. Are integrated by a non-coherent integrator that uses the above, and output.

Furthermore, when performing non-coherent integration on the output of the velocity-compensated coherent integrator, the integration efficiency is compensated by the output adjuster whose detection level is set in advance according to the current position and velocity.

Furthermore, when non-coherently integrating the output of the velocity-compensated coherent integrator, the constant false alarm probability circuit detects the target so that the false alarm probability becomes constant.

Furthermore, when noncoherently integrating the output of the velocity-compensated coherent integrator, if there are a plurality of detection targets, the input signal relating to the detected movement target is deleted from the memory, and the next movement target is deleted. Is integrated and detected, the input data corresponding to the target is also deleted from the memory, and the process is repeated until the target is no longer detected.

Furthermore, when non-coherently integrating the output of the velocity-compensated coherent integrator, the detection parameters are first spatially converted in coarse steps and then the detection parameters are spatially converted in fine steps. To detect the target.

The radar device according to the present invention comprises a coherent integrator without velocity compensation, a moving target coherent integrator, a non-coherent integrator for integrating output signals of any one of them, or a moving target non-coherent integrator. And are selected and connected according to the condition determined by the observation target, and the integration number of each integrating means is changed and set. Then, the moving target is detected based on the set integrated number and connection.

[0079]

【Example】

Example 1. Hereinafter, a radar device according to a first embodiment of the present invention will be described with reference to the drawings. 1 is a block diagram showing the configuration of a radar device according to a first embodiment of the present invention. In the figure,
1 is a transmitting means for generating and modulating a transmission pulse to widen a frequency band, 2 is a transmission / reception switch, 3 is an antenna, 4 is receiving means, 5 is a coherent integrating means for moving target, 6 is a detector, and 7 is a Doppler bin. The selection means, 8 is a non-coherent integration means corresponding to the moving target, and 9 is a display. Conventional example 1
The transmitting means 1, the transmission / reception switcher 2, the antenna 3, the receiving means 4, the moving target corresponding coherent integrating means 5 having the same configuration as the above, and the moving target corresponding noncoherent integrating means 8 having the same configuration as the conventional example 3, Since the display 9 has already been described, the description is omitted here.

2 of moving target coherent integrator 5
The range-Doppler N × K two-dimensional complex data Up (r, fd2) is read from the dimensional memory 52c, and the detector 6
Is converted into an amplitude (or power value) by P to be P (r, fd2), and is input to the Doppler bin selection means 7. In the Doppler bin selection means 7, the intensity detector 7a uses the P (r, fd).
2) is compared with the threshold Td, and P that exceeds Td
(R, fd2) is integrated with respect to fd2 by the Doppler integrator 7b to obtain Y (r). The moving target coherent integrator 5, the detector 6, and the Doppler bin selector 7
The above process is repeated a plurality of times for scanning, and the Y (r) is stored as Z (r, t) (t; scan number) in the memory 8a of the non-coherent integrating means 8 for moving target. That is, the Doppler bin is selected in each scan and range of the three-dimensional data of the range / Doppler scan, and is input to the noncoherent integrator 8 as the two-dimensional data. The moving target correspondence non-coherent integrator 8 obtains the target velocity and the current position using Z (r, t) as described in the conventional example, and displays the result on the display 9.

Next, an outline of the operation of the moving target corresponding coherent integrating means 5, the detector 6, the Doppler bin selecting means 7, and the moving target corresponding noncoherent integrating means 8 will be described with reference to FIG.
It will be described with reference to the flowchart of FIG. Figure 2
In step 1, output data S of the receiving means 4
(R) is read from the buffer memory 51a and is converted into data in the spatial frequency domain by an N-point FFT calculator 51b.
Fourier transform in the range direction to obtain SR (fr). This is repeated for pulse hits and SR (fr, p)
And input to the two-dimensional memory 51c.

In step 2, the output SR of step 1
(Fr, p) and the reference signal RRpp (fr, f
d, p) is subjected to complex multiplication to obtain a spectral component in which the phases are constant over all frequencies. The result is URpp
(Fr, fd, p), and this is repeated N times for the spatial frequency fr and stored in the buffer memory 51f.

In step 3, the complex multiplication result URpp
In order to convert (fr, fd, p) into data in the time domain, inverse Fourier transform is performed in the spatial frequency direction, and U shown in the following equation is obtained.
pp (r, fd, p), step 2 and this inverse Fourier transform are repeated for the pulse hit p and stored in the two-dimensional memory 52a.

In step 4, the result of the inverse Fourier transform is read from the two-dimensional memory 52a and the M-point FFT calculator 52 is read.
At point b, M point Fourier transform is performed in the pulse hit direction,
Doppler processing is performed to obtain Up (r, fd, fd2). This is repeated for the range bin r and the Doppler frequency fd, and only the data in the effective range of fd2 is stored in the memory 52.
Remember in c.

In step 5 of FIG. 3, the two-dimensional memory 52c
Up (r, fd2) is read out from, and converted to amplitude (or power value) by the detector 6 to be P (r, fd2), which is input to the Doppler bin selection means 7. In the Doppler bin selection means 7, the intensity detector 7a compares the above P (r, fd2) with the threshold Td, and P (r, fd exceeds Td.
2) is repeatedly integrated with respect to fd2 by the Doppler integrator 7b to obtain Y (r). This is repeated for the range bin r, and the Y (r) is stored as Z (r, t) (t; scan number) in the memory 8a of the non-coherent integrating means 8 corresponding to the moving target. Further, the non-coherent integration processing corresponding to the moving target, the detection processing, and the Doppler bin selection processing are repeated a plurality of times, and Z (r, t) is stored in the memory 8a.

In step 6, the memory 8a to Z (r,
t) is read and compared with the first threshold value T1 in the first intensity detector 8b to create an array D in which the array numbers of points that are T1 or more and an array S of signal intensity data are created,
The array S is stored in the memory 8c, and the array D is stored in the memory 8d. This is repeated for the range bin r and the scan t, and T1
The number of points above is counted and set as m.

In step 7 of FIG. 4, in the matrix multiplier 8f, the array D is read from the memory 8d, the conversion matrix H is read from the memory 8e, matrix multiplication is performed to obtain ρ i, j, and the address calculator 8g outputs the first A k having a close value of ρk (k = 1 to 1) is obtained and stored in the memory 8h. This is repeated for data number i and detection angle θj.

In step 8, the intensity integrator 8i stores the memory
The array S is read from 8c, the address ρi, j is read from the memory 8h, and the i-th data of the array S is X-divided for each θj of 1 to n.
Integrate to the place indicated by (j, ρi, j).

In step 9, X is detected by the second intensity detector 8j.
Comparing (j, ρi, j) with the second threshold value T2,
[Theta] j- [rho] k at a point that is equal to or greater than T2, and [theta]-[rho];
Converter 8k The above θ-ρ is converted into v-R0 and displayed on the display 9. As described above, the processes of steps 1 to 9 are repeated.

With the above-mentioned configuration, the intensity of the target signal is weak and integration must be performed for a long time.
When the coherent integration and the non-coherent integration must be combined because of the bandwidth due to the fluctuation and shape of the target, the signal integration can be performed even if the observation target moves beyond the range bin.

In the above example, the chirp method is used for pulse compression, but the same can be done when the code modulation method is used.

Example 2. Next, a radar device according to a second embodiment of the present invention will be described with reference to the drawings. FIG. 5 is a configuration block diagram showing a radar device according to a second embodiment of the present invention. In the figure, 1 is transmitting means for generating and modulating a transmission pulse to widen a frequency band, 2 is a duplexer, 3 is an antenna, and 4 is
Is a receiving means, 5 is a moving target coherent integrating means,
Reference numeral 6 is a wave detector, 8A is a non-coherent integrating means corresponding to a moving target, and 9 is a display. The transmitting means 1, the transmission / reception switcher 2, the antenna 3, the receiving means 4 having the same configuration as that of the first embodiment,
Since the moving target coherent integrator 5 and the display 9 have already been described, the description thereof will be omitted here.

The conversion used in the second embodiment will be described. FIG.
The following conversion is performed between two arbitrary points on the straight line α, which is the route of the target 1 showing the route on the distance r-time t plane (xy plane) of 6 and the straight line β which is the route of the target 2. The r (x) intercept representing the target current position R0 and the slope representing the target velocity v are converted on the v-R0 plane by the formula. That is, the following equation (23) is applied. R0 = r + v · t (23) Any point on a straight line in the r-t plane is transformed into a curve in the v-R0 plane, and each of these curves intersects at one point on the v-R0 plane. Therefore, the power at the point on the straight line is v−
If converted on the R0 plane and integrated, the electric power is accumulated at the point indicating the straight line.

In this v-R0 conversion, the conversion matrix H shown in the conventional example 3 and the address R to be integrated are replaced according to the equation (23). Transform matrix H to (2
The matrix R is shown in the equation 4) and the matrix R is shown in the equation (25).

[0095]

[Equation 4]

Next, the address for integrating the signal is Ri,
It is assumed that j is closest to R0k (k = 1 to 1, R0k; a distance set with an arbitrary resolution to obtain the target current position). This address is obtained, Ri, j is replaced with k, and the i-th data in the array S is integrated into X (j, ρi, j). The target current position R0 and the velocity v can be obtained by comparing the integration result with the second threshold value T2 and obtaining vj, Ri, j of the point that is T2 or more.

Here, the relationship with the output of the moving target coherent integrator is considered. The output of the coherent integrator corresponding to the moving target is Up (r, f) as described in Conventional Example 1.
d2) Range-Doppler two-dimensional data. This data is obtained by scanning a plurality of times, and range-scan-
It becomes Doppler three-dimensional data. Here, Doppler corresponds to the target speed. Therefore, Up (r, t,
For fd2), conversion using vj corresponding to fd2 may be performed, and integration may be performed on data of i = 1 to m for each vj. Here, the velocity vj is expressed by the following equation. vj = λ · fd2 / 2 (26) (λ; transmission wavelength) At this time, an array D listing the array numbers of points that are at or above the first threshold T1, an array S of signal strength data, and R indicating an address for integrating the signal Are as shown in equations (27), (28), and (29).

[0098]

(Equation 5)

Next, an outline of the operation of the moving target coherent integrator 5, the detector 6, and the moving target noncoherent integrator 8A will be described with reference to the flow charts of FIGS. 6 to 8. In FIG. 4, steps 1 to 4 are the same as those shown in the first embodiment, and therefore the description thereof is omitted here.

In step 15, Up (r, fd2) is read from the two-dimensional memory 52c, converted into an amplitude (or power value) by the detector 6 to be P (r, fd2), which is the range bin r and the Doppler frequency fd2. About,
It is stored in the memory 8a of the non-coherent integrating means 8 corresponding to the moving target. Here, since the non-coherent integration process corresponding to the moving target, the detection process, and the Doppler bin selection process are repeatedly stored for a plurality of scan times, the P (r, fd2) is three-dimensional data Z (r, t, fd2) (t; Number).

In step 16, the memory 8a to Z (r,
t, fd2) is read out and compared with the first threshold value T1 in the first intensity detector 8b to create an array D in which the array element numbers of points which are T1 or more and an array S of signal intensity data are created. The array D is stored in the memory 8c and the array D is stored in the memory 8d. This is repeated for the range bin r, the scan t, and the Doppler frequency fd2, and the number of points equal to or higher than T1 is counted and set as m.

In step 17, the address calculator 8g reads the array D from the memory 8d, and the speed V * i corresponding to the Doppler frequency fd2i of the i-th data and the number of v when the speed follows the speed resolution. Then, the current position Ri for integrating the signal according to the equation (29) and the R0 number when the current position follows the current position resolution are obtained and stored in the memory 8h. This is repeated for data number i.

In step 18, the intensity integrator 8i reads the array S from the memory 8c and the addresses V.sub.i, Ri from the memory 8h, and the i-th data of the array S is X (V.sub.i, Ri).
Integrate where indicated by.

In step 19, the second intensity detector 8j
X (j, k) is compared with the second threshold value T2, and T
The velocity v and the current position R0 of a point of 2 or more are obtained and displayed on the display unit 9. Above, Steps 1-4 and Step 15-
The processing of 19 is repeated.

With the above-mentioned structure, the intensity of the target signal is weak and integration must be performed for a long time.
When the coherent integration and the non-coherent integration must be combined because of the bandwidth due to the fluctuation and shape of the target, the signal integration can be performed even if the observation target moves beyond the range bin. In addition, non-coherent integration that compensates for Doppler frequency components after coherent integration can be performed without increasing the amount of calculation, and integration performance is improved. Furthermore, since the target velocity and the current position that are desired to be directly obtained are used as the conversion parameters, it is not necessary to perform the conversion at the final stage.

Here, in the above example, the chirp method is used for pulse compression, but the same applies when the code modulation method is used.

Example 3. Next, a radar device according to a third embodiment of the present invention will be described with reference to the drawings. FIG. 9 is a configuration block diagram showing a radar device according to a third embodiment of the present invention. In the figure, the transmitting means 1, the transmission / reception switch 2, the antenna 3, the receiving means 4, the moving target corresponding coherent integrating means 5, the wave detector 6, and the display 9 are the same as those in the first and second embodiments. Therefore,
The description is omitted here.

Noncoherent integrating means 8 for moving target
A memory 8a for storing the output of the B detector and a first intensity detector 8b
A memory 8c for storing an array S of signal intensities, a memory 8d for storing an array D showing data addresses, an address calculator 8g, a memory 8h for storing its output, and an intensity integrator.
8i and an output value adjuster 8l that adjusts the output after intensity integration,
It is composed of a second intensity detector 8j. The memory 8a, the first intensity detector 8b, and the memory, which have the same configuration as the second embodiment,
8c, memory 8d, address calculator 8g, memory 8h,
Description of the intensity integrator 8i and the second intensity detector 8j is omitted here.

When the non-coherent integrator for moving target is used to convert a straight line on the time-distance plane into a point on the current position-velocity plane, when the resolution of the current position or velocity changes depending on its value, or when the target is If it moves out of the target time-distance range or moves out of the time-distance range within the observation time, the integration efficiency changes depending on the detected current position and speed. Therefore, when the integration result is detected at an arbitrary second threshold, the detection probability and the false alarm probability change depending on the current position and speed at that time. In order to avoid such effects, output value regulator
The intensity integration result is multiplied by a constant with a value previously obtained for each current position and speed by 8l to make the signal-to-noise power value after integration constant.

Next, a concrete description will be given. First, a constant for adjusting the output value is obtained in advance for each detected current position or speed and stored in the adjustment value memory 8l2. The method for obtaining this constant value is shown below. Z (r, stored in the memory 8a
t, fd2) (r = 1 to N, t = 1 to J, Nfd2 = 1
(-K), all values are input as 1 here.
Next, the threshold value of the first intensity detector is set to 0 so that all data of N × J × K are detected and integrated. At this time, the output x (j,
The reciprocal of k) is used as the adjustment value, and a (j, k) (= 1 / x
(J, k)) is stored in the adjustment value memory 8l2.

When the adjustment value is input and normal processing is performed, the output x (j, k) of the intensity integrator 8i is multiplied by a (j, k) read from the adjustment value memory, and the multiplication result is Two
Input to intensity detector 8j. As described above, by providing the output value adjuster in the moving target-corresponding non-coherent integrator, the detection probability and the false alarm probability are constant regardless of the detected speed or the current position. Although the v-R0 conversion is used in the third embodiment, the same applies when the θ-ρ conversion is used.

Example 4. Next, a radar device according to a fourth embodiment of the present invention will be described with reference to the drawings. FIG. 10 is a configuration block diagram showing a radar device according to a fourth embodiment of the present invention.
In the figure, a transmission means 1, a duplexer 2, an antenna 3,
The receiving means 4, the moving target coherent integrating means 5, the detector 6, and the display 9 are the same as those in the other embodiments described above. Therefore, the description is omitted here.

Noncoherent integrating means 8 for moving target
C is a memory 8a for storing the detector output, and a CFAR (constant false alarm probability) for detecting a larger one by comparing the detector output with the first threshold value determined by averaging the intensities in the sliding window A first CFAR detector 8m using a circuit, a memory 8c for storing an array S of signal intensities, a memory 8d for storing an array D indicating a data address, an address calculator 8g, a memory 8h, and an intensity integrator 8i. And an output value adjuster 8l and a second CFAR detector 8n.

The above memory having the same configuration as that of the third embodiment.
8a, memory 8c, memory 8d, address calculator 8g,
Descriptions of the memory 8h, the intensity integrator 8i, and the output value adjuster 8l are omitted here.

In the received signal, the signal closer to the distance r has a larger reflected signal from the fixed clutter such as the ground than the signal far, and the intensity difference between the far and near clutter within the observation distance range is large. If there is a reflection from a clutter that has spread spatially due to clouds, etc. that fluctuates with time, detection with a fixed threshold increases clutter and increases the false alarm probability, or the target signal is detected. It may not be done. In order to avoid the above-mentioned phenomenon, a sliding window is used to obtain a threshold value by using an average of signal intensities around a target distance signal.
The detection method using the AR circuit is applied to the first CFAR detector 8m and the second CFAR detector 8n.

The configuration of the first and second CFAR detectors is shown in FIG.
This will be described using 1. In the figure, the CFAR detector comprises a delay element 8m1, an adder 8m2, a coefficient multiplier 8m3, and a comparator 8m4. Sliding by 8m1 delay element
Although a window is constructed, the order of reference cells (a, a; arbitrary constants), guard cells (b, b; arbitrary constants), target cells, guard cells (b), reference cells (a) To do. The values of the 2a reference cells are input to the adder 8m2 and summed. The output of the adder 8m2 is the coefficient K in the coefficient multiplier 8m3.
(K; arbitrary constant) is multiplied, then the coefficient multiplication result and the output of the cell of interest are input to the comparator 8m4 and compared in magnitude. When the value of the cell of interest is larger, the signal strength of the cell of interest and its array number Is output.

By configuring the CFAR detector as described above, when performing coherent integration and non-coherent integration corresponding to a moving target, it becomes possible to detect the target signal even if clutter exists. It is possible to reduce false alarms. In the fourth embodiment, v-R0
Although the conversion is used, the same is true when the θ-ρ conversion is used. Further, only one of CFAR may be used, and the other may be configured by an intensity detector.

Example 5. Next, a radar device according to a fifth embodiment of the present invention will be described with reference to the drawings. FIG. 12 is a configuration block diagram showing a radar device according to a fifth embodiment of the present invention.
In the figure, a transmission means 1, a duplexer 2, an antenna 3,
The receiving means 4, the moving target coherent integrating means 5, the detector 6, and the display 9 are the same as those in the other embodiments described above. Therefore, the description is omitted here.

Noncoherent integrating means 8 for moving target
D is a memory 8a for storing a detector output, a first intensity detector 8b, a memory 8c for storing an array S of signal intensities, a memory 8d for storing an array D indicating a data address, and an address calculator 8g. A memory 8h, an intensity integrator 8i, a second intensity detector 8j, and a memory 8o for storing the second intensity detection result,
When there is a detection in the processing of the second intensity detector immediately before, the next detection is repeated through the detected target signal remover,
When there is no detection, the switch 8p that switches to display the detection result stored in the memory 8o, and when the target is detected by the second strength detector, the second strength immediately before the signal stored in the memory 8a is detected. It is composed of a detected target signal remover 8q that deletes the signal along the target path found by the detection by the detector and inputs it into the memory 8a again.

Next, an outline of the operation of the moving target coherent integrator 5, the detector 6, and the moving target noncoherent integrator 8D will be described with reference to the flowcharts of FIGS. 13 to 16. In FIG. 13, step 1 to step 4 and step 15 to step 18 are the same as those shown in the second embodiment, and therefore the explanation is omitted here.

At step 29, the second intensity detector 8j
X (j, k) is compared with the second threshold value T2, and T
Calculate the velocity v and the current position R0 of the points that are 2 or more, and store
The data is recorded at 8o as the mi-th detection target number data from the start of the non-coherent integration means target detection corresponding to the moving target.
It should be noted that the number of target detections after the previous detected target signal eliminator operates is counted as mj. Next, switch
If mj ≠ 0 at 8p, the process proceeds to step 30, and if mj = 0, the process proceeds to step 31.

In step 30, the detected target signal canceller is detected.
8q, the target number mj detected immediately before by the second intensity detector 8j
Remove the minute signal. First, the target speed v (mk) detected from the memory 8o and the target current position R0 (mk) (m
k = mi-mj + i, i = 1 to mj) is read out, and the signal along the target path is set to 0 as shown below.
To memorize.

Z (r, t, fd) = 0 (30) where t = 0 to Jr = R0 (mk) -v (mk) × t fd = 2 × v (mk) / λ

After deleting mj target signals, steps 6 to 9 are repeated again.

At step 31, information on all the detected targets is displayed on the display 9. As described above, the processes of steps 1 to 31 are repeated.

With the above-mentioned configuration, the intensity of the target signal is weak and integration must be performed for a long time.
When the coherent integration and the non-coherent integration must be combined because of the bandwidth due to the fluctuation and shape of the target, the signal integration can be performed even if the observation target moves beyond the range bin. Further, when a plurality of targets are present in the observation region, it is possible to prevent a target signal having large reflected power from preventing detection of a target signal having small reflected power. In addition, in Example 5, v-R
The 0 conversion was used, but the same is done when the θ-ρ conversion is used.

Example 6. Next, a radar device according to a sixth embodiment of the present invention will be described with reference to the drawings. FIG. 17 is a configuration block diagram showing a radar device according to a sixth embodiment of the present invention.
In the figure, a transmission means 1, a duplexer 2, an antenna 3,
The receiving means 4, the moving target coherent integrating means 5, the detector 6, and the display 9 are the same as those in the other embodiments described above. Therefore, the description is omitted here.

Noncoherent integrating means 8 for moving target
E is a memory 8a for storing a detector output, a first intensity detector 8b, a memory 8c for storing an array S of signal intensities, a memory 8d for storing an array D indicating a data address, and θ in coarse steps. -The first conversion matrix memory 8e for storing the conversion matrix for performing ρ conversion, and θ-in small steps
A second conversion matrix memory 8r that stores a conversion matrix for performing ρ conversion, a switch 8s that switches between the first conversion matrix memory 8e and the second conversion matrix memory 8r, and a matrix multiplier 8f that multiplies the array D and the conversion matrix. An address calculator 8g, a memory 8h for storing the output of the address calculator, an intensity integrator 8i, and a second
Intensity detector 8j and memory 8o for storing the second intensity detection result
If the result of using the rough first conversion matrix memory 8e is returned to finely using the second conversion matrix memory 8r, the switch 8p is switched to display the result on the display if the result of fine conversion is performed. Composed of.

Next, an outline of the operation of the moving target corresponding coherent integrating means 5, the detector 6, the Doppler bin selecting means 7, and the moving target corresponding non-coherent integrating means 8E will be described with reference to the flowcharts of FIGS. 18 to 21. In FIG. 18, steps 1 to 6 are the same as those shown in the first embodiment, and therefore the description thereof is omitted here.

In step 37, when the counter a = 0, the conversion matrix H1 obtained by using the detected angles θ0j set in coarse steps is used as the first conversion matrix memory 8e.
Read from the memory 8d, read the array D from the memory 8d, perform matrix multiplication with the matrix multiplier 8f to obtain ρ0i, j, and set ρ0k (k = 1 to lo) in the address calculator 8g with the nearest coarse increment. Memory for k
Store at 8h. This is repeated for data number i and detected angle θ0 j. Further, when the counter a = 1, the conversion matrix H2 obtained by using the detection angles θ1 j set in fine increments near the detection angle θ of the target path detected when a = 0
Is read from the second conversion matrix memory 8r and array D
Is read from the memory 8d, matrix multiplication is performed by the matrix multiplier 8f to obtain ρ1i, j, and ρ1k (k is set in the address calculator 8g in finer increments of the nearest value.
= 1 to 11) and k is stored in the memory 8h. This is repeated for data number i and detected angle θ1j.

At step 38, the intensity integrator 8i reads the array S from the memory 8c and the address ρai, j from the memory 8h, and the i-th data of the array S is X-divided for each θa j of the 1st to the nth.
Integrate to the place indicated by (j, ρai, j).

At step 39, the second intensity detector 8j
X (j, k) is compared with a second threshold value Ta2 set for each count a, and θ and ρ at a point that is equal to or greater than Ta2 are obtained, and θ-ρ; v-R0 converter 8k. -R0 is converted and recorded in the memory 8o as the detection target number mjth data.

In step 40, the switch 9p sets a = 1 when a = 0, and the conversion matrix stored in the second conversion matrix memory 8r is set to a finely set angle near the detection angle detected when a = 0. And switch 8s to the second conversion matrix memory 8r side,
Repeat from step 7. If a = 1 with the switch 9p, the process proceeds to step 41.

At step 41, the information of all the detected targets is displayed on the display unit 9. As described above, the processes of steps 1 to 41 are repeated.

With the above-mentioned configuration, the intensity of the target signal is weak and integration must be performed for a long time.
When the coherent integration and the non-coherent integration must be combined because of the bandwidth due to the fluctuation and shape of the target, the signal integration can be performed even if the observation target moves beyond the range bin. Also, first the detection parameter is coarsely set, the second threshold is set high, the target is detected, then the detection parameter is finely set by narrowing down to the vicinity of the first detection, and the second threshold is lowered. Reduce false alarms and detect. As a result, the amount of calculation can be kept low within the range in which the required detection performance can be obtained. Although the θ-ρ conversion is used in the sixth embodiment, the same operation as that described in each of the above-described embodiments is performed when the v-R0 conversion is used.

Example 7. Next, a radar device according to a seventh embodiment of the present invention will be described with reference to the drawings. 22 is a configuration block diagram showing a radar device according to a seventh embodiment of the present invention.
In the figure, a transmission means 1, a duplexer 2, an antenna 3,
The receiving means 4, the moving target corresponding coherent integrating means 5, the detector 6, the moving target corresponding non-coherent integrating means 8 and the display 9 are the same as those in the above embodiment. Further, 54 is a coherent integrator, 82 is a non-coherent integrator,
Reference numeral 91 is an integral number controller, and 91a, 91b and 91c are switches. The coherent integrating means 5 corresponding to the moving target, the detector 6, the non-coherent integrating means corresponding to the moving target 8 and the like, which are the same constituent elements as those of the first to sixth embodiments, have already been described, and therefore the description thereof is omitted here. .

The coherent integration means 54 is composed of a pulse compression means 55 and a pulse Doppler processing means 56.
The pulse compression means 55 includes a buffer memory 55a and an N-point FFT.
A computing unit 55b, a two-dimensional memory 55c, a reference memory 55 for storing a reference signal without compensation of the influence of Doppler shift due to movement of a target and compensation of influence of movement of range bin
e, complex multiplier 55d, buffer memory 55f, and N point I
The FFT calculator 55g is provided to perform pulse compression without compensation for the influence of the target movement. Also, the pulse Doppler processing means 56
Includes a two-dimensional memory 56a, an M-point FFT calculator 56b, and a two-dimensional memory 56c, performs Fourier transform in the pulse hit direction, and decomposes the signal into Doppler components. Coherent integration that does not compensate for the effect of the moving target is performed by the pulse compression means and pulse Doppler processing.

The non-coherent integrator 82 is provided with a two-dimensional memory 82a and an intensity integrator 82i, and has a plurality of pulse hits (scans) for each range bin and each Doppler bin.
Integrate the signal across. By the above processing, non-coherent integration that does not compensate for the influence of the moving target is performed.

The integral number controller 91 examines the search range of the radar at the time of observation, the condition such as weather in the search range, the target size of the target, the motion performance, etc. with another sensor or the like,
The time allocation of the coherent integration time and the non-coherent integration time is determined, and whether the range bin movement of the target occurs during the coherent integration time is determined by the time allocation. Decide whether to perform coherent integration. Also, the distribution of the integration time is determined in consideration of the calculation time at that time. Depending on the decision result, the switches 91a, 91b, 9
Switch 1c.

The switch 91a is switched to the coherent integrator 54 when the coherent integration time cannot be lengthened due to the target of interest and other surrounding conditions and range bin movement does not occur within the range of target velocity of interest.
When range bin movement occurs during the integration time, such as when the coherent integration time is long or the target speed range is wide, the moving target coherent integration means 5
Switch to.

When it is assumed that the range bin movement does not occur during the coherent integration time, and the range bin movement does not occur even within the non-coherent time, the switch 91
b is switched to the noncoherent integrating means 82. When range bin movement is assumed during the coherent integration time, or when range bin movement does not occur during the coherent integration time but range bin movement occurs during the longer time, the switch 91c does not pass through the coherent integration means. Switch to the one connected to.

If range bin movement is assumed during the non-coherent integration time, the switch 91c is switched to non-coherent integration means corresponding to the movement target. When non-coherent integration is not performed with a long coherent integration time after determination of the distribution of integration times within a series of signal processing times,
When the non-coherent integration is performed but the target range bin movement during the integration time is not assumed, the switch 91c is directly switched to the display.

With the above-described structure, it is possible to obtain a radar device capable of optimally distributing the integration time and selecting the integration method depending on the situation.

[0144]

As described above, according to the present invention, since the configuration is such that the coherent integration in which the influence of the range bin movement and the Doppler shift is compensated, the Doppler bin selecting means and the non-coherent integration are combined, the moving target is observed. In this case, the integration time can be continued beyond the limit of the coherent integration time, and a radar device for determining a more accurate target can be obtained.

Since the coherent integration that compensates for the effects of range bin shift and Doppler shift and the non-coherent integration that processes the range scan data for each Doppler bin are combined, the Doppler output of the coherent integration output in the non-coherent integration is used. By compensating the range bin movement for each frequency, the signal can be efficiently integrated, and further, there is an effect that a radar device can be obtained in which the integration time is continued beyond the limit of the coherent integration time and an accurate target is fixed. .

Further, since the non-coherent integrating means is provided with the output adjuster for guaranteeing the integration efficiency, the integration time can be continued beyond the limit of the coherent integration time, and the value of the integration parameter in the non-coherent integration can be maintained. It is possible to obtain a radar device in which the nonuniformity of the integration efficiency due to is eliminated and the detection probability is made constant.

Further, since the constant false alarm probability detector is provided in the non-coherent integration means, the integration time can be continued beyond the limit of the coherent integration time, and
There is an effect that a radar device having a constant false alarm probability can be obtained.

Further, since the detected target signal canceller is provided in the non-coherent integrating means, the integration time can be continued beyond the limit of the coherent integration time, and
When there are multiple targets in the observation area, it is effective to detect target signals with small reflected power.

Further, since the non-coherent integrator is provided with a plane conversion matrix having steps of different densities, the integration time can be continued beyond the limit of the coherent integration time and the required detection performance can be obtained. This has the effect of keeping the amount of calculation low.

Further, coherent integration means for performing pulse compression and pulse Doppler processing without velocity compensation, coherent integration means for moving target, non-coherent integration means for integration not corresponding to moving target, and non-coherent integration for moving target. Means is used by switching by the control of the integral number controller, and the integral number controller sets and controls the number of integrals of each integration method depending on the situation. The effect is that you can set the target by selecting.

[Brief description of drawings]

FIG. 1 is a configuration block diagram showing a configuration of a radar device according to a first embodiment of the present invention.

FIG. 2 is a diagram showing a flowchart of signal processing of the radar device according to the first embodiment of the present invention.

FIG. 3 is a diagram showing a flowchart of signal processing of the radar device according to the first embodiment of the present invention.

FIG. 4 is a diagram showing a flowchart of signal processing of the radar device according to the first embodiment of the present invention.

FIG. 5 is a configuration block diagram showing a configuration of a radar device according to a second embodiment of the present invention.

FIG. 6 is a diagram showing a flowchart of signal processing of a radar device according to a second embodiment of the present invention.

FIG. 7 is a diagram showing a flowchart of signal processing of the radar device according to the second embodiment of the present invention.

FIG. 8 is a diagram showing a flowchart of signal processing of the radar device according to the second embodiment of the present invention.

FIG. 9 is a configuration block diagram showing a configuration of a radar device according to a third embodiment of the present invention.

FIG. 10 is a configuration block diagram showing a configuration of a radar device according to a fourth embodiment of the present invention.

FIG. 11 is an internal configuration diagram of a CFAR detector used in the radar device of the fourth embodiment.

FIG. 12 is a configuration block diagram showing a configuration of a radar device according to a fifth embodiment of the present invention.

FIG. 13 is a diagram showing a flowchart of signal processing of a radar device according to a fifth embodiment of the present invention.

FIG. 14 is a diagram showing a flowchart of signal processing of a radar device according to a fifth embodiment of the present invention.

FIG. 15 is a diagram showing a flowchart of signal processing of a radar device according to a fifth embodiment of the present invention.

FIG. 16 is a diagram showing a flowchart of signal processing of a radar device according to a fifth embodiment of the present invention.

FIG. 17 is a configuration block diagram showing a machine of the radar apparatus according to the sixth embodiment of the present invention.

FIG. 18 is a diagram showing a flowchart of signal processing of the radar device according to the sixth embodiment of the present invention.

FIG. 19 is a diagram showing a flowchart of signal processing of a radar device according to a sixth embodiment of the present invention.

FIG. 20 is a diagram showing a flowchart of signal processing of the radar device according to the sixth embodiment of the present invention.

FIG. 21 is a diagram showing a flowchart of signal processing of a radar device according to a sixth embodiment of the present invention.

FIG. 22 is a configuration block diagram showing a configuration of a radar device according to a seventh embodiment of the present invention.

FIG. 23 is a configuration diagram of a conventional radar device that performs chirp-type coherent integration.

FIG. 24 is a configuration diagram of a conventional radar device that performs coherent integration using a code modulation method.

FIG. 25 is a diagram showing a flowchart of signal processing in the pulse compression means and the pulse integration means shown in FIGS. 23 and 24.

FIG. 26 is a diagram illustrating each signal waveform of the radar device in FIG. 23.

FIG. 27 is a diagram for explaining each signal waveform of the radar device of FIG. 24.

28 is an internal configuration diagram of the receiving means shown in the radar device of FIGS. 23 and 24. FIG.

FIG. 29 is a diagram for explaining the influence on the pulse compression by the Doppler effect of the chirp type radar device.

FIG. 30 is a diagram for explaining the influence on pulse compression by the Doppler effect of a code modulation type radar device.

FIG. 31 is an explanatory diagram of plane conversion of a straight line segment.

FIG. 32 is a configuration diagram of a straight line detection device using a television camera.

FIG. 33 is a configuration diagram of the electrical plane conversion circuit shown in FIG. 32.

FIG. 34 is a block diagram of a conventional radar device that performs non-coherent integration.

35 is an internal configuration diagram of a receiving means in the radar device shown in FIG. 34.

FIG. 36: x for explaining plane conversion of a straight line segment
-Y is a top view.

FIG. 37: θ for explaining the plane conversion of a straight line segment
-Ρ plan view.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 transmission means, 2 transmission / reception switch, 3 antennas, 4 reception means, 5 coherent integration means for moving target, 6
Detector, 7 Doppler bin selection means, 8, 8A, 8
B, 8C, 8D, 8E Non-coherent integrating means for moving target, 8m first CFAR detector, 8n second CFA
R detector, 8o memory, 8p switch, 8q detected target signal remover, 8r second conversion matrix memory,
9 indicator, 54 coherent integration means, 81 output value adjuster, 82 non-coherent integration means, 91 integral number controller, 91a switch, 91b switch, 91c
switch.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Tomomasa Kondo 5-1, 1-1 Ofuna, Kamakura-shi Electronic Systems Research Center, Mitsubishi Electric Corporation

Claims (7)

[Claims] 1. A coherent integrating means for moving target, which comprises a reference signal generating means, a pulse compressing means, and a pulse Doppler processing means, performs integration processing corresponding to the moving target, and outputs a two-dimensional signal of range bin / Doppler bin. A Doppler bin selection means for selecting a candidate Doppler bin from the amplitude or power of the two-dimensional signal output of the range bin / Doppler bin, and a range bin / time two-dimensional signal obtained by repeating the above process on a signal input in time series Radar device including non-coherent integrator for moving target, which integrates the addresses of signals exceeding the threshold of (3) into a group of addresses obtained by conversion using a conversion matrix to specify and output a moving target value. 2. A coherent integration means for moving target, which comprises a reference signal generating means, a pulse compression means, and a pulse Doppler processing means, performs integration processing corresponding to the moving target, and outputs a two-dimensional signal of a range bin / Doppler bin. , A moving target that repeats the above processing on a signal input in time series and spatially converts a signal exceeding a set threshold value of a three-dimensional signal of range bin, Doppler bin, and time, and then integrates it to specify and output a moving target value. A radar device equipped with a corresponding non-coherent integrator. 3. The method according to claim 1 or 2, wherein after the signals for each address are integrated, an output adjuster for determining the significance of the integrated value is further added to the moving target correspondence non-coherent integration means. The described radar device. 4. The non-coherent integrator corresponding to the moving target detects a significant false alarm probability detector by detecting the value of the amplitude or power of the input signal in the non-coherent integrator corresponding to the moving target or its integrated value after address conversion. 4. The signal according to claim 1 is defined.
The described radar device. 5. The moving target correspondence non-coherent integrator further includes an output memory for specifying and storing a moving target value,
A detected target signal canceller for deleting an input signal corresponding to the specified moving target is added, and when the first moving target is specified, the corresponding input is deleted to further specify the remaining moving target. The radar device according to any one of claims 1 to 4, wherein: 6. The non-coherent integrating means corresponding to the moving target further includes an output memory for specifying and storing the moving target value,
When a first conversion matrix memory with a coarse resolution and a second conversion matrix memory with a fine resolution are added to the space converter and a movement target is specified by the first conversion matrix memory, the second conversion matrix memory The radar device according to claim 1, wherein the moving target is further specified. 7. A coherent integrator, which is composed of a pulse compression means without speed compensation and a pulse Doppler processing means, performs integration processing corresponding to a moving target and outputs a two-dimensional signal of a range bin / Doppler bin, and a reference signal generation. Means, a pulse compression means, and a pulse Doppler processing means, performs integration processing corresponding to a moving target, outputs coherent integrating means corresponding to a moving target for outputting a two-dimensional signal of a range bin / Doppler bin, and the coherent integrating means output or A three-dimensional signal of range bin, Doppler bin, and time, which is obtained by repeating the two-dimensional signal of range bin / Doppler bin, which is the output of the coherent integrator corresponding to the moving target and is input in time series, to each of the range bin / Doppler bin In contrast, the target value is calculated by integrating with respect to time. And a non-coherent integrator that outputs the two-dimensional range bin Doppler bin signal, which is the output of the coherent integrator or the output of the coherent integrator corresponding to the moving target, which is a range bin A non-coherent integrator means for moving target, which integrates three-dimensional signals of Doppler bin and time with respect to time for each of range bin and Doppler bin, and specifies and outputs a moving target value, and conditions under which the observation target is determined are examined. A radar device comprising integral number control means for controlling setting of the integrated number of each integrating means and selection of connection.