JPH04137999A - Headphone sound field listening device - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

DETAILED DESCRIPTION OF THE INVENTION Field of the Invention The present invention relates to a sound field listening device that makes it possible to listen to a sound field equivalent to that reproduced by speakers when listening to headphones.

BACKGROUND OF THE INVENTION In recent years, technological trends in the audio-visual field have been changing from conventional stereo playback to methods that dynamically control the sound field in accordance with the video. The first conventional technology is United States Patent No. 3,746,792, 3E3
There is a Dolby surround active matrix type sound field control device shown in No. 32886 and No. 3959590.

Hereinafter, a first conventional sound field control device will be described with reference to the drawings.

First, we will explain how to encode Dolby Surround.

FIG. 7 is a block diagram showing the configuration of a Dolby Surround encoder.

In FIG. 7, 701 is an L (left channel) signal input terminal, 702 is an R (right channel) signal input terminal, and 70
3 is the C (center channel) signal input terminal, 704 is the S (
surround channel) signal input terminal, 705 is an attenuator that attenuates the C signal by 3 [dB], 706 is an attenuator 705
707 is an attenuator 705 that adds the output of the adder to the L signal.
An adder 708 adds the output of the S signal to the R signal.
709 is a band-pass filter that passes the output of the attenuator 708 from 100 [Hz to 7 kHz; 710 is a modified B-type noise reduction encoder that encodes the output of the band-pass filter 709; 711 is a modified B-type noise reduction encoder 71
A phase shifter 712 creates a signal having a phase difference of +90[deg] with respect to the output of phase shifter 711.
] An adder that adds the output to the output of the adder 706, an adder 713 that adds the -90[deg] output of the phase shifter 711 to the output of the adder 707, and an adder 714 that adds the output of the adder 712 to the L
Lt output as t (encoder left channel) signal
A signal output terminal 715 is an Rt signal output terminal that outputs the output of the adder 713 as an Rt (encoder right channel) signal.

The operation of the Dolby Surround encoder configured as above will be explained.

The L (left channel) signal input to the Dolby surround encoder is the speaker placed in front of the left of the listening position in the listening room, the R (right channel) signal is input to the speaker placed in the front right, and the C (center channel) signal is the speaker placed in the front left of the listening position in the listening room. is a signal that has been mixed on the assumption that it will be reproduced by a speaker placed in the front, and the surround channel signal S will be reproduced by two speakers placed at the left and right in the rear. The C signal is attenuated by 3 [dB] by the attenuator 705, and added to the L signal by the adder 7O6 and to the R signal by the adder 707, respectively. The S signal is attenuated by 3 [dB] by an attenuator 708, and is further attenuated by 100 [Hz ~ 7 [k] by a bandpass filter 709.
Hz].

The output of the bandpass filter 709 is encoded by a modified B-type noise reduction encoder 710. This encoding will be discussed later. The output of the modified B-type noise reduction encoder 710 is +90 [
degl and is added to the output of adder 70E3 by adder 712.

The output of adder 712 becomes Dolby surround encoder output Lt. Similarly, the output of the modified B-type noise reduction encoder 710 is -90 [deg
The signal is phase-shifted and added to the output of the adder 707 by an adder 713 to become the Dolby surround encoder output Rt.

The above processing can be summarized as equations (1) and (2).

Lt=t, +o, 7G+0. 7js...(
1) Rt=R+o, 7C-0, 7js...(
2) Here, j represents (-1)"2, and the phase rotation is 9
This indicates that the value is 0[deg].

The modified B-type noise reduction encoder 710 has an amplitude frequency characteristic that changes depending on the level of the input signal. By decoding this encoded signal, it is possible to reduce the high frequency components of noise generated in the transmission media. Table 1 shows the amplitude frequency characteristics of the modified B-type noise reduction encoder using the input signal level as a parameter.

Next, a decoder of a Dolby surround active matrix type sound field control device will be explained.

Table 1 (Continued) FIG. 8 is a block diagram showing the configuration of a decoder of a Dolby surround active matrix sound field control device.

In FIG. 8, 801 is an input terminal for the encoder output Lt (left channel) signal, and 802 is the encoder output Rt
(Right channel) signal input terminal, 803 is Lt and Rt
an input balance control device 804 that adjusts the balance with the
805 is a level control device that adjusts the absolute level of the balance-adjusted signals Lt and Rt, and 805 is the signal Lt that has been adjusted in absolute level.

Rt to L (left channel) signal, R (right channel)
806 is a delay device that delays the S signal output from the adaptive matrix 805, and 807 is a 7 [kHz] delay device for the S signal outputted by the adaptive matrix 805.
z] low-pass filter that passes the following signals, 808
809 is a modified B-type noise reduction decoder that attenuates the noise of components below 7 kHz of the S signal, and 809 is a signal output by the adaptive matrix 805.

A master level control device that controls the levels of the R signal, the C signal, and the S signal output by the modified B-type noise reduction decoder 808; 810 is a listening room; 811 is located in the front right of the listening position in the listening room; A speaker 812 that reproduces the R signal output from the control device 809 is placed in front of the listening position in the listening room, and a speaker 813 that reproduces the C signal output from the master level control device 809 is the listening position in the listening room. A speaker 814 is placed on the left front side of the main unit and reproduces the signal output by the master level control device 809.
is located at the rear right of the listening position in the listening room.
A speaker 815 that reproduces the S signal output from the master level control device 809 is placed at the rear left of the listening position in the listening room, and is a speaker that reproduces the S signal output from the master level control device 809.

The operation of the Dolby surround active matrix type sound field control device decoder configured as described above will be explained.

Equation (1) is input to the Dolby Surround Active Matrix sound field control device decoder.

These are encoder outputs Lt and Rt expressed by equation (2).

The input balance control device 803 receives the input signal Lt.

Adjust the balance of Rt. Level control device 804 adjusts the absolute levels of input signals Lt and Rt.

In the adaptive matrix 805, the input signal Lt.

The four output signals R, C, and S are controlled according to the level difference of Rt. Therefore, the input signals Lt, R
It is necessary to balance t and adjust the absolute level. The processing of the adaptive matrix 805 will be described in detail later. The delay device 806 is an adaptive matrix 805
S (surround channel) signal of 15 to 30 [ms
] Delay.

The low-pass filter 807 filters the delayed S signal at 7 [kHz
] The following signals are passed. The modified B-type noise reduction decoder 808 reduces high frequency noise generated in the transmission medium included in the S signal. Table 2 shows the amplitude frequency characteristics using the input level of the modified B-type noise reduction decoder 808 as a parameter. The first characteristic of the decoder is
The characteristics are the opposite of those of the encoder shown in the table.

Table 2 (Continued) The master level control device 809 outputs the adaptive matrix 805 (left channel) signal, R (right channel) signal, C (center channel) signal, and modified B signal.
This is a quadruple volume that controls the level of the S (surround channel) signal output by the type noise reduction decoder 808.

R signal output by master level control device 809.

The C signal, L signal, and S signal are reproduced by speakers 811 to 815 arranged in the listening room.

Here, the adaptive matrix 805 will be explained.

FIG. 9 is a block diagram showing the configuration of the adaptive matrix 805. In FIG. 9, 901 is an Lt input terminal, 902 is an Rt input terminal, 903 is a bandpass filter that limits the signal bands of Lt and Rt, and 904 is L'
(band-limited Lt) and R” (band-limited Rt
) and C.

An adder 905 that creates a signal takes the difference between L′ and R′,
90E3 to 909 are full-wave rectifier circuits that perform full-wave rectification of L" PjZ CZ Sl, respectively; 910 is a logarithmic difference circuit that outputs the logarithmic difference DLR between R" and L; and 911 is a Dcs, the difference between the logarithms of So and C"
912 is a logarithmic difference circuit 910 that outputs
Alternatively, a threshold switch 913 determines whether the output of 911 is within a predetermined range, and 913 is a logarithmic difference circuit with a low-pass filter with a time constant of 22 [ms] or 44 B [msl] depending on the determination result of the threshold switch 912. A dual time constant circuit 914 processes the output DLR of the threshold switch 912 with a time constant of 22 [ms] or 448 [ms] depending on the judgment result of the threshold switch 912.
] with a low-pass filter, the output De of the logarithmic difference circuit 911
A bitemporal constant circuit 915 processes s, and a bitemporal constant circuit 91
EL, which is the output of 3 multiplied by a coefficient according to its polarity, is E
A polarity dividing circuit that outputs R, 916 is a bitemporal constant circuit 91
Ec + E, which is the output of 4 multiplied by a coefficient according to its polarity
917 is a polarity division circuit 91 that outputs s.
5 outputs EL and ER and the output E of the polarity dividing circuit 916
Control input signals Lt and Rt by c+Es to generate ELL
IE LR+ ERL+ E*, l+ ECLt
ECTo Esc+ Voltage control amplifier that outputs ESR, 918 is the output ELL of voltage control amplifier 917
IELRIERLIERRIECLIECRIESLI
This is a coupling network that multiplies the ESR and input signals Lt and Rt by a predetermined constant, adds them, and outputs L, R, C, and S.

The operation of the adaptive matrix configured as described above will be explained below.

The adaptive matrix 805 calculates the difference in the logarithms of the signal levels for the LR axis or the C8 axis, and detects from which direction the signal is dominant based on this difference. and,
By outputting signals in the dominant direction as they are and attenuating signals in other directions, the sense of direction of the reproduced sound is emphasized.

The bandpass filter 903 converts the input signals Lt and Rt to 100
Bandwidth limited to [Hz ~ 7 kHz].

The outputs L' and R' of the bandpass filter 903 are expressed by equation (1),
As shown in equation (2), the main components are the enhancer input signal and the R signal, respectively. Further, the outputs of the adder 904 and the subtracter 905 are each expressed by equation (3).

It is expressed by equation (4).

C' = C + 0.7 (L + R) ... (
3) S' =-j S+〇, 7 (L-R)
(4) From equations (1) and (2), it can be seen that c' and s' each have a C9S signal as their main component.

L” R1, C1, and Sl signals are each sent to a full-wave rectifier circuit 9.
Full-wave rectification is performed at 06 to 909. After full wave rectification,
The pair of L'R' and c's' is processed by logarithmic difference circuits 910 and 911, respectively, and the output D LRI D c
s is obtained. The processing of the logarithmic difference circuits 910 and 911 is expressed by equations (5) and (6), respectively.

DLR= l o ga (R"/L')
...(5) Dcs=l Ogs (S
'/C' ) "1B) DLR indicates which side of LR is dominant regarding the LR axis, and D
cs indicates which C8 is dominant regarding the C8 axis.

Threshold switch 912 is L,! : When the level difference between R1 or C and S is large, the output of the adaptive matrix 805 is changed, and the short time constant 22 of the bi-time constant circuits 913 and 914 is used to quickly change R, C, and S.
[msl is selected, and conversely, when the level difference is small, the constant 484 [ms] is selected, R, C,
Change the S signal slowly.

The threshold switch 912 is connected to the logarithmic difference circuit 910.
If the outputs DL*t and Dcs of 911 are both within the range of threshold level ±Lth, the dual time constant circuit 91
A time constant of 484 [ms] of 3.914 is selected, and if either one is outside the range, a time constant of 22 [ms] is selected. Table 3 shows the time constants ([ms]) selected by the threshold switches 9 and 12.

In order to realize Table 3, the threshold switch 91
2 is the output DLRI D of the logarithmic difference circuit 910.911
It consists of four comparators that compare es and the threshold level ±Ltht, and an AND circuit that takes the logical sum of the comparator outputs.

The dual time constant circuit 913 integrates the output DLR of the logarithmic difference circuit 910 with a time constant of 484 or 22 [ms] depending on the determination result of the threshold switch 912. The integration circuit requires an RC integrator or one with transient characteristics equivalent to it. The bitemporal constant circuit 914 and the logarithmic difference circuit 911
Similar processing is performed on the output Dcs of.

The polarity dividing circuit 915 generates control voltages EL and ER for the subsequent voltage control amplifier 917 according to the polarity of the DLR integrated by the bi-temporal constant circuit 913. control voltage EL,
Elf is expressed by equations (7) and (8).

EL=DLRDLR<0 0 DLII>O...(7) ER=ODLR<0 DLRDLR>O...(8) Similarly, the control voltages Ec, Es for the voltage control amplifier 917 generated by the polarity dividing circuit 916 is the formula (9).

It is expressed by the formula (lO).

Ec= Dcs Dcs<0 0 Dcs> O=(9) Es: ODcs<0 Dcs Dcs≧O・(10) The voltage control amplifier 917 outputs the output EL of the polarity dividing circuit 915
, ER and the output Ee+E$ of the polarity dividing circuit 916 to control the input signals Lt and Rt to generate E LLIELRI.
EllL* ERTo ECLI Ec*e
Output EsLt Es*. Here, the input signal L9 controlled by the output E of the polarity dividing circuit will be expressed as Exv. FIG. 10 shows the relationship between the control voltage and amplification factor of the voltage controlled amplifier.

Coupling network 918 connects the outputs of voltage controlled amplifier 917 ELLI ELRI ERLI EIIRI
Ect+Eell+ESLI EIR and input signals Lt, Rt are added at the ratio shown in Table 4, and L, R
, C, S signals are output.

Table 4 (continued) Table (continued) At the output stage of the adaptive matrix 805, 25 [dB
It is possible to ensure separation between channels of more than

As described above, the Dolby surround active matrix type sound field control device decoder detects from which direction the signal is dominant with respect to the LR axis or the C8 axis.

If a dominant direction is detected, the signal in that direction is output as is, and the signals in other directions are attenuated, thereby emphasizing the sense of direction of the reproduced sound. Therefore, sounds with a clear direction, such as dialogue, provide a clear sense of direction. On the other hand, if no dominant direction is detected, a normal stereo effect is obtained.

The above is the explanation of the first conventional technique.

Furthermore, in the field of acoustics, technological trends are changing from original sound reproduction to original sound field reproduction, and sound field control devices for reproducing sound fields such as concert halls are being developed.

As a second conventional technique, for example, Japanese Patent Application Laid-Open No. 61-257
There is an acoustic control device disclosed in Japanese Patent No. 099.

A second conventional sound field control device will be described below with reference to the drawings.

FIG. 11 is a block diagram showing the configuration of a conventional sound field control device. In FIG. 11, 1101 is an audio signal input terminal for inputting a source signal; 1102 is a reflected sound parameter storage means for storing reflected sound parameters;
03 is a reflected sound generating means 1104 and 1105 that adds reflected sound to the source signal and outputs the resultant signal based on each reflected sound parameter stored in the reflected sound parameter storage means 1102;
゜1106, 1107. is reflected sound generation means 110
A speaker 1108 reproduces the signal processed in step 3, and a listening room 1108 reproduces the signal.

Regarding the conventional sound field control device configured as above,
The operation will be explained below.

The audio signal input from the audio signal input terminal 1101 is added with reflected sound by the reflected sound generation means 1103 based on each reflected sound parameter stored in the reflected sound parameter storage means 1102. The details of the processing will be explained later. The signal processed by the reflected sound generation means 1103 is transmitted to a speaker 1103 placed at a predetermined position in the listening room 1107. 1104.
Reproduced by 1105.1108. The listener can listen to the reproduced sound field at a predetermined position in the listening room 1107.

The audio signal input from the audio signal input terminal 1101 usually has two channels. Further, in this conventional example, the signal processed by the reflected sound generating means 1103 has four channels. However, depending on the processing, the number of output channels may be more than four or less than four.

The processing performed by the reflected sound generation means 1103 will be explained.

The input signal is X + + X 2, the output signal is y + +
If Y2+ya, ya, the input signal and output signal are (I
I) There is a relationship of the formula.

Y=H*X...(11
) here, Y"()'+,Y2.'Y3.Ya)"
...(12)X= (x++ X2)
”...(I3)H=(H,,
H2) ... (14) H+=
(hIl, b+a, ha3. ha4)”
...(I5)H2" (h2+, hts,
hts, ha-) ” ...(16) However, ′*” indicates convolution, “T” indicates transposed matrix, (
hIl in equations 11) and (1B) represents an impulse response convolved with the i-channel signal output to the j-channel. The impulse response hB to be convolved is stored in the reflected sound parameter storage means 1102. This impulse response is determined from measurements taken at an actual concert hall or computer simulations based on the shape of the concert hall. In addition, when implementing this in hardware, it requires an enormous amount of hardware to convolve all the impulse responses. For this reason, for example, h
Divide Il into early reflected sound and reverberant sound, and divide the early reflected sound into F.
Processing may be performed using small-scale hardware by having an IR filter and a reverberation generator using an IIR all-pass filter handle reverberation sound.

The above is the explanation of the second conventional technique.

Problems to be Solved by the Invention However, the conventional configuration is premised on speaker reproduction. Therefore, if a sufficiently large listening room cannot be prepared, a sound field control device cannot be installed. There is also the problem of having to play the music at a low volume in consideration of nearby noise, such as late at night.

The present invention has been made in view of the problems of conventional sound field control devices, and aims to make it possible to listen to a sound field equivalent to the sound field reproduced by speakers while listening to headphones. Consider it a technical issue.

Means for Solving the Problems In order to solve the above problems, the headphone sound field listening device of the present invention includes a sound field control signal processing device that performs sound field control processing on an input signal on the premise of speaker reproduction. , a head-related transfer function compensation device that compensates the output of the sound field control signal processing device using a head-related transfer function, and headphones that reproduce the output of the head-related transfer function compensation device.

Effect: The headphone sound field listening device of the present invention has the above-mentioned configuration.
The sound field control signal processing device performs the same sound field control processing as in conventional speakers, and the head-related transfer function compensation device compensates for the transfer function from the speaker to the entrance of the listener's ear canal. To generate the same sound pressure at the entrance of an ear canal when listening to headphones as when playing through a speaker.

Embodiment Hereinafter, an embodiment of the headphone sound field listening device of the present invention will be described with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of an embodiment of the headphone sound field listening device of the present invention.

In FIG. 1, 101 is a signal input terminal, 102 is a sound field control signal processing device that performs sound field control processing on the premise of speaker reproduction on the input signal, and 103 is a sound field control signal processing device 102 that is connected to a plurality of A head-related transfer function compensation device 104 is a headphone that compensates a signal output to a speaker using a head-related transfer function.

The operation of the headphone sound field listening device of the present invention configured as described above will be explained below.

A sound field control signal processing device 102 is an input terminal for an encoder output Lt (left channel) signal, and an input terminal for an encoder output Rt (right channel) of the decoder of the Dolby surround active matrix sound field control device shown in the first conventional example. Perform the same processing as that performed by the signal input terminal to the master level control device, or perform the same processing as that performed by the reflected sound parameter storage means and reflected sound generation means shown in the second embodiment. I do.

Here, for the following explanation, each speaker of the Dolby surround active matrix type sound field control device shown in the first conventional example is numbered as follows. Speaker 1 is placed in the front left of the listening position in the listening room and plays the L signal, speaker 2 is placed in front and plays the C signal, speaker 3 is placed in the front right and plays the R signal, and speaker 3 is placed in the right rear. There are 4 speakers placed at the rear left side that play the S signal, and 5 speakers placed at the rear left side that play the S signal.
shall be. In the following explanation, the speaker of the Dolby surround active matrix type sound field control device shown in the first conventional example will be explained as an example. In the case of the sound field control device shown in the second conventional example, it is not necessary to particularly limit the number of speakers, but it is sufficient to similarly number each speaker.

Next, the processing contents of the head-related transfer function compensator 103 will be explained.

In the processing of the sound field control signal processing device 102, the positions of each speaker and the listener are assumed. With this positional relationship, the transfer functions from the input terminal of the i-th (i = 1 to 5) speaker to the left and right ear canal entrances of the listener are expressed as HIL and H, respectively.
lll (i = 1 to 5).

At this time, the signal S + (i =
1 to 5), the sound pressures p, , 5 at the listener's left and right external auditory canal entrances are expressed by equations (7) and (8), respectively.
pPe,l'' is generated. This relationship is shown in FIG.

PIL”=HIL”S+
...(17)P +*sp= HIR@S +
(+8) Also, let SL and SLl be the signals given to the left and right input terminals of the headphones, and let GL and G11 be the transfer functions from the input terminals to the left and right ear canal entrances of the listener, respectively. At this time, sound pressures P L"l P R'ρ expressed by equations (19) and (20) are generated at the left and right ear canal entrances of the listener. This relationship is shown in FIG. 3.

PL"=GL"SL...(
19) PR"=GR・SR...(20) Therefore, in order to obtain the same sound pressure as that reproduced by the i-th speaker through headphones, the input terminals of the headphones must be input with equations (21) and (22), respectively. What is necessary is to input the signal Sue SIR (i = 1 to 5) given by the formula.

5IL=GL-'・HILll SI
...(21) SIR=GR-'・HIR"SI
...(22) Here, GL
-'IGR-' is the inverse characteristic of the transfer function from the headphone input terminal to the left and right ear canal entrances of the listener. The inverse characteristic will be discussed later.

The transfer function from the speaker input terminal to the listener's left and right ear canal entrances and the transfer function from the headphone input terminal to the listener's left and right ear canal entrances will be described. The impulse response from the input terminal of the i-th speaker to the entrance of the listener's ear canal can be measured in an anechoic chamber using a speaker and a probe microphone. The impulse response of the i-th speaker becomes the transfer function HIL IHIR used in equations (21) and (22). In addition, the transfer functions GL, Gl+ from the headphone input terminal to the entrance of the ear canal of the listener are
is similarly obtained from measurements, and the inverse characteristic GL-IGR-' can be calculated from this transfer function. Figure 4 shows a waveform diagram showing a measurement example of the impulse response from the speaker input terminal to the listener's ear canal entrance, and Figure 5 shows an example of the inverse characteristic of the transfer function from the headphone input terminal to the listener's ear canal entrance. A waveform diagram is shown. The sampling frequency in FIGS. 4 and 5 is 50 kHz, and the number of sampling points is 512.

Here, a method of calculating the inverse characteristic will be explained. The inverse characteristic is
This is a characteristic in which the impulse response of the entire system is taken as a unit sample by connecting it in series with a system having a transfer function from the headphone input terminal to the entrance of the listener's ear canal. There are two possible inverse characteristics that are actually used: one obtained from discrete Fourier transform and one obtained by deconvolution. Since the inverse characteristic obtained from the discrete Fourier transform has a very long time length, in this embodiment, the inverse characteristic obtained by deconvolution is adopted.

Let g(k) be the impulse response from the headphone input terminal to the entrance of the listener's ear canal, g-'(k) be the impulse response with the desired inverse characteristic, δ(k) be the unit sample, and m be the number of samples with the inverse characteristic. For example, the linear convolution relationship of equation (23) holds between these.

=δ (n −n@)
(23) In the equation (23), n11 represents the time lag between input and output with inverse characteristics, and is called a spike point.

Now, since the number of samples of the inverse characteristic is limited to m, the left side of equation (23) is not exactly a unit sample.

Therefore, equation (23) becomes an impossible simultaneous linear equation in which the equality sign does not hold. Therefore, the inverse characteristic is estimated by using the difference between the left side of equation (23) and the unit sample as a residual, and minimizing the sum of squares of this residual. In other words, the inverse characteristic can be calculated by treating equation (23) as a simultaneous linear equation and solving it using the method of least squares.

In the above explanation, the output processing for the i-th speaker has been described, but the processing of the sound field control signal processing device 102 is performed on the assumption that a total of five speakers are used. Therefore, the signals SL given to the input terminals of the headphones by equations (24) and (25), respectively,
SR will be input.

As a result of the above, the head-related transfer function compensator 103 obtains (24)
A filter that executes equation (25) may be used.

FIG. 6 shows a signal processing flow of the head-related transfer function compensator 103. In FIG. 6, 601 to 603 are input terminals for the signal S1 processed by the sound field control signal processing device 102;
09 to 611 are the transfer functions Hat (i
= 1 to 5) into the processed signal S1 assuming the i-th speaker, 612 to 614
are the transfer functions HIR from the input terminal of the i-th speaker to the listener's right ear canal entrance (i = 1 to 5), respectively.
615 is an FIR filter 609 that convolves the signal into the processed signal S1 assuming the i-th speaker.
616 is an adder that adds the outputs of FIR filters 612 to 614, and 617 is an adder that adds the outputs of FIR filters 612 to 614. 617 is an adder that adds the outputs of FIR filters 612 to 614, and 617 is an adder that adds the outputs of FIR filters 612 to 614. 618 is an FIR filter that convolves the characteristic GL-' with the inverse characteristic G R-' of the transfer function from the headphone input terminal to the listener's right ear canal entrance on the output of the adder 616;
619 is an output terminal for the output signal of the FIR filter 617;
620 is an output terminal for the output signal of the FIR filter 618. The FIR filter 609 includes a shift register 604 that shifts an input signal in synchronization with the sampling frequency, multipliers 605 to 607 that multiply each tap output of this shift register 604 by a filter coefficient, and multipliers 605 to 607.
The output of the adder 608 is the output of the FIR filter 609.

The filter coefficients multiplied by the multipliers 605 to 807 are the transfer function Hat from the input terminal of the first speaker to the listener's left ear canal entrance expressed on the time axis hat (k) (
k=L 2+ ''m; m is the number of samples (in this example, m=512), and the multiplier 605 converts (1) into
Multiplier 806 outputs haL(2),... Multiplier 607
is multiplied by haL(512). Other FIR filters have similar configurations.

The head-related transfer function compensator 103 configured as described above performs signal processing given by equations (24) and (25), and outputs signals SL and SR.

Headphones 104 reproduce signals SL, SR processed by head related transfer function compensator 103.

The above is the operation of the headphone sound field listening device according to the embodiment of the present invention.

Note that, as stated in the middle of the example, this example is based on the first example.
This paper describes the speaker arrangement of a conventional Dolby surround active matrix sound field control device. Similar processing can be performed for the speaker arrangement of the second conventional sound field control device.

As described in detail, the headphone sound field listening device of the present invention has the following features:
Since the head-related transfer function compensator performs compensation processing on the output of the sound field control signal processing device, it is possible to hear the same sound field as when playing through speakers even though it is played through headphones. . Therefore, there are no architectural restrictions on the listening room, and there is no need to consider nearby noise.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing the configuration of an embodiment of the headphone sound field listening device of the present invention, FIG. 2 is a plan view showing the state of transmission when the i-th speaker reproduces the signal SI, and FIG.
Figure 4 is a front view showing how the signal S is transmitted when reproduced through headphones, Figure 4 is a waveform diagram showing an example of measurement of the transfer function from the input terminal of the speaker to the entrance of the listener's ear canal, and Figure 5
The figure is a waveform diagram showing an example of the inverse characteristic of the transfer function from the headphone input terminal to the entrance of the ear canal of the listener. Figure 6 is a block diagram showing the signal processing flow of the head-related transfer function compensation device.
FIG. 8 is a block diagram showing the configuration of a Dolby Surround encoder in the first conventional example, FIG. 8 is a block diagram showing the decoder configuration of a Dolby Surround active matrix sound field control device in the conventional example, and FIG. A block diagram showing the configuration of the adaptive matrix in the conventional example, FIG. 10 is a characteristic diagram showing the relationship between the control voltage and amplification factor of the voltage control amplifier in the conventional example, and FIG. 11 is a sound field control diagram in the second conventional example. FIG. 2 is a block diagram showing the configuration of the device. 101... Signal input terminal, 102... Sound field control signal processing device, 103... Head related transfer function compensation device, 104... Headphones, 601-603...
Input terminal, 604--Shift register, 60
5 to 607... Multiplier, 608, 815, 616
...Adder, 609-814,817.818...
FIR filter, 819, 820...output terminal. Name of agent: Patent attorney Akira Okaji et al. 28 Fig. fig. fig. l Ngong 6tq, lZt> - A photo of the output of the Le issue! 11←

Claims (3)

[Claims] (1) A sound field control signal processing device that performs sound field control processing on an input signal with the assumption that it will be played back by a speaker; A headphone sound field listening device comprising: a head-related transfer function compensator that performs compensation using a partial transfer function; and headphones that reproduce an output of the head-related transfer function compensator. (2) The sound field control signal processing device includes: an input balance control device that inputs two channel signals and adjusts the balance thereof; a level control device that adjusts the absolute level of the output of the balance control device; and the level control device. Based on the device output, it detects which direction the signal from the front left, front, right front, or rear of the listening position is dominant, and the signal from the dominant direction is output louder, and the signal from other directions is output. an adaptive matrix that generates a left channel signal, a right channel signal, a center channel signal, and a surround channel signal so as to attenuate the signals; a delay device that delays the surround channel signal outputted by the adaptive matrix; kHz] or lower; a modified B-type noise reduction decoder that attenuates the noise output from the low-pass filter; and a left channel signal, right channel signal, and center channel signal output by the adaptive matrix. The headphone sound field listening device according to claim 1, further comprising: a master level control device that controls the level of the signal output by the modified B-type noise reduction decoder. (3) The sound field control signal processing device reproduces the reflected sound in the acoustic space or a model space similar thereto using a plurality of speakers arranged around the listening point, based on the data of the reflected sound in the acoustic space. a reflected sound parameter storage means for storing parameters of reflected sound to be emitted by the plurality of speakers; generating reflected sound of the source signal based on each reflected sound parameter stored in the reflected sound parameter storage means; The headphone sound field listening device according to claim 1, further comprising reflected sound generating means for outputting reflected sound.