JP2026003350A - Motor control device for electric compressor - Google Patents

Abstract

A motor control device for an electric compressor is provided that can prevent inaccurate estimation of the position of a motor (rotor) caused by superimposing harmonic voltages on a voltage command.
[Solution] A motor control device (100) includes a PWM calculator (24) that generates a signal to control a motor (210) of an electric compressor (200) based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current to a drive voltage command value for the motor (210), a position error estimator (29) that calculates an estimated error Δθ of the rotor position using the harmonic current, a harmonic amplitude regulator (23) that adjusts the amplitude Vh of the harmonic voltage command value, and a position estimator (30) that estimates the rotor position using the estimated error Δθ calculated by the position error estimator (29). The position error estimator (29) calculates the estimated error Δθ using a γ-axis harmonic current and a δ-axis harmonic current. The harmonic amplitude regulator (23) increases the amplitude Vh as the input voltage (Vdc) increases.
[Selected figure] Figure 5

Description

This disclosure relates to a motor control device for an electric compressor.

For example, Japanese Patent Application Laid-Open No. 8-308286 (Patent Document 1) discloses detecting motor operation based on the induced electromotive force induced in the motor.

Japanese Patent Laid-Open Publication No. 7-245981 (Patent Document 2) discloses a system that detects the magnetic pole position of a motor using a d-axis voltage command value on which an alternating voltage is superimposed.

Japanese Patent Application Publication No. 8-308286 Japanese Patent Application Publication No. 7-245981

When driving a motor that drives a relatively large-capacity electric compressor, the motor may frequently switch between on and off (switching on and off) when the refrigerant flow rate of the electric compressor is low. In this case, there is a risk of the electric compressor being damaged due to an oil shortage or other problem. Therefore, in order to avoid this on and off switching, it is possible to control the motor at a relatively low speed.

However, because the induced electromotive force of a motor is small in the low-speed range, the method described in Patent Document 1 above has the disadvantage that the motor's position cannot be accurately estimated.

In contrast, the method described in Patent Document 2 makes it possible to estimate the motor (rotor) position relatively accurately, even in low-speed regions. However, by adding a high-frequency alternating voltage (harmonic voltage) to the voltage command, a high-frequency alternating current (harmonic current) is superimposed on the AC current flowing through the motor. This causes the timing at which the AC current switches between positive and negative polarities to become irregular, which can result in inaccurate dead-time correction. In this case, there is a risk that the motor's position will be estimated inaccurately.

The objective of this technology is to provide a motor control device for an electric compressor that can suppress inaccurate motor (rotor) position estimation caused by superimposing harmonic voltages on voltage commands.

A motor control device for an electric compressor according to one aspect of the present disclosure includes a signal generator that generates a signal to control the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate the rotor position of the motor to a drive voltage command value for driving the motor of the electric compressor; a position error estimator that calculates an estimation error of the rotor position using the harmonic current; a position estimator that estimates the rotor position using the estimation error calculated by the position error estimator; and a harmonic amplitude adjuster that adjusts the voltage amplitude, which is the amplitude of the harmonic voltage command value. The position error estimator calculates the estimation error using a γ-axis harmonic current, which is a harmonic current in the γ-axis, which is an estimation axis of the d-axis, and a δ-axis harmonic current, which is a harmonic current in the δ-axis, which is an estimation axis of the q-axis. The harmonic amplitude adjuster adjusts the voltage amplitude based on the input voltage to the harmonic amplitude adjuster, and makes the voltage amplitude when the input voltage is a first voltage value larger than the voltage amplitude when the input voltage is a second voltage value smaller than the first voltage value.

In a motor control device for an electric compressor according to one aspect of the present disclosure, as described above, the position error estimator calculates the position error using the γ-axis harmonic current and the δ-axis harmonic current. The harmonic amplitude adjuster increases the voltage amplitude when the input voltage is a first voltage value compared to the voltage amplitude when the input voltage is a second voltage value that is smaller than the first voltage value. When calculating the estimation error using the γ-axis harmonic current and the δ-axis harmonic current, the estimation error increases as the input voltage to the harmonic amplitude adjuster increases, and decreases as the voltage amplitude increases. Therefore, by increasing the voltage amplitude when the input voltage is a first voltage value compared to the voltage amplitude when the input voltage is a second voltage value that is smaller than the first voltage value, it is possible to prevent an increase in the estimation error caused by the input voltage increasing from the second voltage value to the first voltage value. This prevents inaccurate motor (rotor) position estimation when harmonic voltages are superimposed on the voltage command.

The position error estimator may calculate the estimation error based on the ratio between the amplitude of the γ-axis harmonic current and the amplitude of the δ-axis harmonic current. Here, the amplitudes of the γ-axis harmonic current and the δ-axis harmonic current are each proportional to the voltage amplitude. Therefore, the above ratio is a value independent of the voltage amplitude. Therefore, the estimation error can be calculated regardless of the voltage amplitude.

The harmonic amplitude adjuster may increase the voltage amplitude as the input voltage increases. As described above, the estimation error increases as the input voltage to the harmonic amplitude adjuster increases, and decreases as the voltage amplitude increases. Therefore, by increasing the voltage amplitude as the input voltage increases, it is possible to effectively suppress an increase in the estimation error.

The harmonic amplitude adjuster may execute a process to reduce the voltage amplitude when the absolute value of the AC current output from the power converter, which outputs an AC voltage to the motor and performs dead time correction, to the motor is smaller than a predetermined value. Here, reducing the voltage amplitude reduces the amplitude of the harmonic current. Therefore, by executing a process to reduce the voltage amplitude when the absolute value of the AC current output from the power converter to the motor is smaller than a predetermined value, it is possible to prevent harmonic currents with large amplitudes from being superimposed on the AC current when the AC current value is near zero. As a result, it is possible to prevent frequent positive and negative changes in the AC current on which harmonic currents are superimposed, thereby preventing inaccurate dead time correction.

The harmonic amplitude adjuster may perform a process to reduce the voltage amplitude when the amplitude of the AC current output to the motor from a power converter that outputs an AC voltage to the motor and performs dead time correction is smaller than a predetermined value. Here, reducing the voltage amplitude reduces the amplitude of the harmonic current. Therefore, it is possible to prevent harmonic currents with relatively large amplitudes from being superimposed on AC currents with relatively small amplitudes. As a result, it is possible to prevent frequent positive and negative switching of the AC current on which harmonic currents are superimposed, thereby preventing inaccurate dead time correction.

This technology can prevent inaccurate motor (rotor) position estimation caused by superimposing harmonic voltages on voltage commands.

1 is a block diagram showing a configuration of a motor system according to an embodiment; FIG. 2 is a diagram illustrating a detailed configuration of a motor system according to an embodiment. FIG. 1 is a diagram showing the d, q axes and the γ, δ axes. FIG. 2 is a block diagram showing a detailed configuration of a controller according to an embodiment. FIG. 10 is a diagram illustrating the relationship between an input voltage Vdc and an amplitude Vh according to an embodiment. FIG. 10 is a diagram illustrating the relationship between the amplitude Aδ of a harmonic current and the estimation error Δθ according to an embodiment. FIG. 2 illustrates an AC current with superimposed harmonic currents according to one embodiment. FIG. 10 is a diagram illustrating an alternating current on which a harmonic current is superimposed according to a modified example of an embodiment.

Embodiments of the present invention will be described in detail with reference to the drawings. Note that identical or corresponding parts in the drawings will be designated by the same reference numerals and their description will not be repeated.

Figure 1 shows the overall configuration of a motor system 1 including a motor control device 100 for an electric compressor 200 according to this embodiment. The motor system 1 is mounted, for example, on an electric vehicle. However, the use of the motor system 1 is not limited to vehicles. The motor system 1 may also be used in a stationary system (for example, an air conditioning system). The motor system 1 includes the motor control device 100, a motor 210 for the electric compressor 200, a power source 300, and a main controller 400.

The power source 300 supplies power to the motor control device 100. The power source 300 is, for example, a DC power source (DC system) such as a storage battery or a solar cell. The power source 300 may also be an AC power source (AC system). In the case of an AC power source, a rectifier must be provided to convert AC to DC.

The motor control device 100 drives (controls) the motor 210. The motor control device 100 includes a power conversion unit 10 and a controller 20. The power conversion unit 10 performs power conversion operations on the power supplied from the power source 300. The controller 20 controls the power conversion unit 10 in accordance with control commands from the main controller 400. The control commands from the main controller 400 to the controller 20 include a speed command Fm* (a command related to the angular acceleration of the motor 210). The power conversion unit 10 is an example of a "power converter" in this disclosure.

The motor 210 is a three-phase AC rotating electric machine or a three-phase brushless DC rotating electric machine, such as an IPM (Interior Permanent Magnet) motor. The motor 210 is not provided with a position sensor (resolver) that detects the position of the rotor 211 (described below, Figure 3). Therefore, the motor control device 100 performs sensorless control of the motor 210.

Figure 2 is a diagram showing an example of the configuration of the motor system 1. However, the main controller 400 (Figure 1) is not shown in Figure 2.

In this example, the power source 300 is a storage battery. The power source 300 outputs DC power to the power conversion unit 10 via the DC terminals Tp and Tn of the power conversion unit 10. The power source 300 is provided with a monitoring unit 310 (including a voltage sensor, a current sensor, etc.) that monitors the state of the power source 300. The monitoring unit 310 outputs the monitored voltage, current, etc. to the controller 20.

In accordance with control commands from the controller 20, the power conversion unit 10 converts DC power (DC voltage) from the power source 300 into AC power (AC voltage) and outputs the AC power (AC voltage) to the motor 210. More specifically, the power conversion unit 10 includes, for example, a voltage sensor 12 and an inverter 13.

The voltage sensor 12 detects the voltage between the power line PL and the power line NL and outputs the detected voltage to the controller 20.

Inverter 13 is, for example, a two-level, three-phase full-bridge circuit. In accordance with control commands from controller 20, inverter 13 converts DC power between power lines PL and NL into AC power and outputs the AC power (AC voltage) to AC terminals Tu, Tv, and Tw. In this example, inverter 13 includes six switching elements Q1 to Q6 and six freewheel diodes D1 to D6. Each switching element Q1 to Q6 is a metal-oxide-semiconductor field-effect transistor (MOSFET), an insulated gate bipolar transistor (IGBT), a bipolar transistor, or the like. Freewheel diodes D1 to D6 are connected in antiparallel to switching elements Q1 to Q6, respectively. Switching elements Q1 and Q2 are connected in series to form the U-phase arm of the full-bridge circuit. Switching elements Q3 and Q4 are connected in series to form the V-phase arm of the full-bridge circuit. Switching elements Q5 and Q6 are connected in series to each other to form the W-phase arm of a full-bridge circuit. The U-phase arm, V-phase arm, and W-phase arm are connected to AC terminals Tu, Tv, and Tw, respectively. Each phase arm is connected between power line PL and power line NL. Note that when MOSFETs are used as switching elements Q1 to Q6, the parasitic diodes of the MOSFETs replace the freewheel diodes D1 to D6.

The motor 210 includes a rotor 211 (Figure 3) with a permanent magnet and a stator 212 with coils wound around it. In this example, the stator 212 has a U-phase coil, a V-phase coil, and a W-phase coil. One end of each phase coil is star-connected to the neutral point. The other end of each phase coil is connected to the connection point of the switching elements of each phase arm of the inverter 13.

The motor 210 is provided with current sensors 213 and 214. The current sensor 213 detects the V-phase current Iv flowing through the motor 210. The current sensor 214 detects the W-phase current Iw flowing through the motor 210. Each current sensor 213 and 214 outputs the detected current to the controller 20. Note that the U-phase current Iu and the V-phase current Iv, or the U-phase current Iu and the W-phase current Iw, may also be output to the controller 20.

The controller 20 controls the inverter 13 based on the speed command Fm* from the main controller 400 (Figure 1) and the detection results from various sensors (monitoring unit 310, voltage sensor 12, current sensors 213 and 214, etc.). For example, the controller 20 outputs a switching signal SW to each of the six switching elements Q1 to Q6 included in the inverter 13. The switching signal SW (Figure 1) is typically a PWM (Pulse Width Modulation) signal.

The controller 20 includes, as its main components, a processor 201 and a memory 202. The processor 201 includes processing circuitry such as a CPU (Central Processing Unit) and an MPU (Micro Processing Unit). The memory 202 includes volatile storage devices such as DRAM (Dynamic Random Access Memory) and SRAM (Static Random Access Memory), and non-volatile storage devices such as HDDs (Hard Disk Drives), SSDs (Solid State Drives), and flash memory. The memory 202 stores system programs including an OS (Operating System), control programs including computer-readable code, and various parameters for controlling the power conversion operation of the power conversion unit 10. The processor 201 performs various arithmetic operations by reading the system programs, control programs, and parameters, expanding them into the memory 202, and executing them. Arithmetic operations performed by the controller 20 may be performed using an ASIC (Application Specific Integrated Circuit), FPGA (Field Programmable Gate Array), or the like.

Inverter 13 has a dead time to prevent the upper arm switching elements (Q1, Q3, Q5) and the lower arm switching elements (Q2, Q4, Q6) in each phase from being turned ON simultaneously. Furthermore, dead time correction is performed in inverter 13 to prevent voltage errors (voltage errors between the PWM command and the inverter output voltage) caused by the dead time. If dead time correction is incorrect, an unintended excessive or insufficient voltage will be applied to motor 210. This voltage error causes ripples in the current flowing through motor 210, and this current ripple also causes ripples in the estimated position of rotor 211 (estimated position ripple).

Note that it is not necessary for the controller 20 and main controller 400 of the motor control device 100 to be provided separately. The controller 20 may also be configured to calculate the speed command Fm* by itself.

<Calculation of angle difference>
3 is a diagram illustrating the relationship between the magnetic pole position of the rotor 211 and the coordinate axes during operation of the motor 210. As shown in FIG. 3, the d-axis is an axis extending from the rotation axis C of the rotor 211 toward the north pole of the rotor 211. The d-axis rotates counterclockwise at the angular velocity ω of the rotor 211. The q-axis is an axis orthogonal to the d-axis (an axis extending in a direction 90 degrees electrical angle ahead of the d-axis).

When performing sensorless control of the motor 210, it is difficult for the controller 20 to accurately determine the d-axis and q-axis of the rotor 211. Therefore, a γ-δ rotating coordinate system is used instead of the d-q rotating coordinate system defined by the d-axis and q-axis. The γ-δ rotating coordinate system is defined by the γ-axis and δ-axis, which are estimated from the d-axis and q-axis. The γ-axis is the axis extending from the rotation axis C toward the estimated north pole of the rotor 211. The δ-axis is an axis perpendicular to the γ-axis (an axis extending 90 electrical degrees ahead of the γ-axis).

The γ-axis current and δ-axis current in the γ-δ rotating coordinate system are written as Iγ and Iδ, respectively. The γ-axis current command and δ-axis current command required to generate torque in motor 210 are written as Iγ* and Iδ*, respectively. The γ-axis current Iγ is the current used to generate a magnetic field in motor 210. The δ-axis current Iδ is the current corresponding to the torque of motor 210. By setting the δ-axis current command Iδ* to zero and the γ-axis current command Iγ* to a variable value, controller 20 prevents motor 210 from generating torque and generates a magnetic field at the specified position.

Hereinafter, the angular difference between the d-axis and the γ-axis (γ-δ rotating coordinate system relative to the d-q rotating coordinate system) will be referred to as the "estimation error Δθ." The d-axis self-inductance and q-axis self-inductance of the coil of stator 212 will be referred to as Ld and Lq, respectively.

<Function block>
4 is a functional block diagram of the controller 20 in this embodiment. The controller 20 includes a speed control unit 21, a current control unit 22, a harmonic amplitude regulator 23, a PWM calculator 24, a coordinate converter 25, a band stop filter (BSF) 26, a band pass filter (BPF) 27, an amplitude estimator 28, a position error estimator 29, and a position estimator 30. The controller 20 also includes subtractors 31 to 33, a multiplier 34, and an adder 35. The PWM calculator 24 is an example of the "signal generation unit" of the present disclosure.

The subtractor 31 calculates the angular velocity error by subtracting the current estimated velocity (angular velocity) Fm output from the position estimator 30 from the velocity command Fm* input to the motor control device 100 from, for example, the main controller 400 (Figure 1).

Speed control unit 21 generates a torque command value that brings the angular velocity error input from subtractor 31 closer to zero, and generates a γ-axis current command Iγ* and a δ-axis current command Iδ* according to PI control to generate the generated torque command value. Speed control unit 21 outputs the γ-axis current command Iγ* and the δ-axis current command Iδ* to subtractors 32 and 33, respectively.

The subtractor 32 calculates the γ-axis current deviation ΔIγ (= Iγ* - Iγ), which is the deviation between the γ-axis current Iγ from the coordinate converter 25, from which harmonic components have been removed by passing through the BSF 26, and the γ-axis current command Iγ* from the speed control unit 21, and outputs the γ-axis current deviation ΔIγ to the current control unit 22.

The subtractor 33 calculates the δ-axis current deviation ΔIδ (= Iδ* - Iδ), which is the deviation between the δ-axis current Iδ from the coordinate converter 25, from which harmonic components have been removed by passing through the BSF 26, and the δ-axis current command Iδ* from the speed control unit 21, and outputs the δ-axis current deviation ΔIδ to the current control unit 22.

The current control unit 22 performs a proportional-plus-integral (PI) calculation on the γ-axis current deviation ΔIγ from the subtractor 32, and outputs the calculation result to the adder 35 as the γ-axis voltage command Vγ*. The current control unit 22 performs a proportional-plus-integral (PI) calculation on the δ-axis current deviation ΔIδ from the subtractor 33, and outputs the calculation result to the PWM calculator 24 as the δ-axis voltage command Vδ*. Note that the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* are each referred to as the "drive voltage command value for driving the motor" in this disclosure.

The voltage of the power source 300 is input to the harmonic amplitude regulator 23 as the input voltage Vdc, for example, from the monitoring unit 310 (Figure 2). The harmonic amplitude regulator 23 calculates (adjusts) the amplitude Vh of the harmonic voltage command value based on the input voltage Vdc. The harmonic voltage command value is used to generate a harmonic current that is used to estimate the rotor position of the motor 210. Note that the amplitude Vh is an example of the "voltage amplitude" in this disclosure.

The multiplier 34 multiplies the amplitude Vh input from the harmonic amplitude regulator 23 by the cosine value (cos _ωht ) of the product ( _ωht ) of the harmonic angular frequency _ωh and time t. The multiplied value calculated by the multiplier 34 is added to the γ-axis voltage command Vγ* by the adder 35. This makes it possible to suppress torque pulsation caused by adding the multiplied value (harmonic component) to the γ-axis voltage command that is unrelated to the torque of the motor 210. The added value (Vγ*+ _Vhcosωht ) calculated by the adder 35 is input to the PWM calculator 24. Note that the multiplied value may be added to the δ-axis voltage command Vδ*.

The PWM calculator 24 converts the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* on the dq(γδ) two-phase coordinate system into a U-phase voltage command, a V-phase voltage command, and a W-phase voltage command on the UVW three-phase coordinate system according to a known coordinate transformation formula (dq two-phase → UVW three-phase transformation formula (inverse Park transformation formula)) using the estimated position Hm of the rotor 211 input from the position estimator 30. The PWM calculator 24 then generates a switching signal SW from the voltage commands for the three phases. More specifically, the PWM calculator 24 generates a PWM signal as the switching signal SW based on a comparison of the voltage commands for each phase with a predetermined carrier wave. The PWM calculator 24 outputs the generated switching signal SW to the power conversion unit 10 (inverter 13, Figure 2).

The coordinate converter 25 calculates the γ-axis current Iγ and the δ-axis current Iδ based on the V-phase current Iv detected by the current sensor 213 and the W-phase current Iw detected by the current sensor 214. The coordinate converter 25 calculates the γ-axis current Iγ and the δ-axis current Iδ according to a known coordinate transformation formula (UVW three-phase to dq two-phase transformation formula (Park transformation formula)) using the estimated position Hm of the rotor 211 input from the position estimator 30. The coordinate converter 25 outputs the γ-axis current Iγ and the δ-axis current Iδ to the BSF 26 and the BPF 27, respectively.

The BPF 27 removes sub-harmonic components from each of the input γ-axis current Iγ and δ-axis current Iδ. The BPF 27 extracts the γ-axis harmonic current iγ' and δ-axis harmonic current iδ' from which the sub-harmonic components have been removed, and outputs the γ-axis harmonic current iγ' and δ-axis harmonic current iδ' to the amplitude estimator 28.

Here, the γ-axis harmonic current iγ' and the δ-axis harmonic current iδ' are expressed by the following equations (1) and (2), respectively. Note that _L0 is (Ld+Lq)/2, and _L1 is (Lq-Ld)/2. Also, Δθ represents the difference (estimation error) between the estimated angle of the rotor 211 and the actual angle of the rotor 211.
iγ'=Vh(L ₀ +L ₁ cos2Δθ) sin ω _h t/ω _h (L ₀ ² -L ₁ ² )...(1)
iδ'=Vh(L ₁ sin2Δθ) sinω _h t/ω _h (L ₀ ² −L ₁ ² )...(2)

The amplitude estimator 28 estimates (calculates) the amplitude Aγ of the γ-axis harmonic current iγ' from the input γ-axis harmonic current iγ'. The amplitude estimator 28 estimates (calculates) the amplitude Aδ of the δ-axis harmonic current iδ' from the input δ-axis harmonic current iδ'. The amplitude estimator 28 outputs the amplitude Aγ and the amplitude Aδ to the position error estimator 29.

The position error estimator 29 calculates the estimated error Δθ using the amplitudes Aγ and Aδ input from the amplitude estimator 28. Specifically, the position error estimator 29 calculates the estimated error Δθ using the following equations (3) and (4). Note that equation (4) is an approximation equation when it is assumed that Δθ can be approximated to 0. The position error estimator 29 outputs the calculated estimated error Δθ to the position estimator 30. Note that Δθ can be approximated to 0 means that it is possible to assume that the absolute value of Δθ is sufficiently small.
Aδ/Aγ=(L ₁ sin2Δθ)/(L ₀ +L ₁ cos2Δθ) (3)
tan ^-1 [{(L ₀ +L ₁ )/L ₁ }(Aδ/Aγ)]/2≒Δθ...(4)

The position estimator 30 uses the estimated error Δθ input from the position error estimator 29 to calculate the current estimated speed Fm of the rotor 211 and the current estimated position Hm of the rotor 211. Specifically, the position estimator 30 calculates the estimated speed Fm and estimated position Hm that converge the estimated error Δθ to zero using PI control. The position estimator 30 outputs the estimated speed Fm to the subtractor 31. The position estimator 30 outputs the estimated position Hm to each of the PWM calculator 24 and the coordinate converter 25.

However, in conventional motor control devices, the timing at which the AC current switches between positive and negative becomes irregular due to harmonic currents being superimposed on the AC current, which can result in inaccurate dead time correction.

In this embodiment, as shown in FIG. 5, the harmonic amplitude regulator 23 increases the amplitude Vh as the input voltage Vdc increases. For example, the harmonic amplitude regulator 23 linearly increases the amplitude Vh in proportion to the input voltage Vdc. Note that the amplitude Vh may also be increased exponentially in proportion to the input voltage Vdc.

As can be seen from equations (1) and (2) above, the amplitude Aγ of the γ-axis harmonic current iγ' and the amplitude Aδ of the δ-axis harmonic current iδ' each increase as the amplitude Vh increases. When Δθ can be approximated to 0, the amplitude Aγ becomes sufficiently larger than the amplitude Aδ (Aγ >> Aδ). Therefore, when the amplitude Aγ and the amplitude Aδ increase by increasing the amplitude Vh, the slope of the estimated error Δθ relative to the amplitude Aδ decreases based on equation (4) above (see Figure 6). In other words, the fluctuation in the estimated error Δθ (estimated position ripple) relative to the fluctuation in the amplitude Aδ caused by the δ-axis current ripple (current ripple due to dead-time correction failure) decreases. Note that Δθ can be approximated to 0 means that Δθ can be assumed to be sufficiently small.

Furthermore, the voltage error caused by a dead time correction failure is proportional to the pulse height (input to inverter 13) based on the input voltage Vdc. Therefore, the voltage error is proportional to the input voltage Vdc. Therefore, the estimated position ripple increases in proportion to the input voltage Vdc.

As described above, the estimated position ripple increases in proportion to the input voltage Vdc and decreases in proportion to the amplitude Vh. Therefore, as shown in Figure 5, by controlling the amplitude Vh to increase as the input voltage Vdc increases, it is possible to prevent the estimated position ripple from increasing (by the amount that the estimated position ripple caused by the amplitude Vh becomes smaller). Note that in areas where the input voltage Vdc is relatively low, the estimated position ripple caused by the input voltage Vdc is relatively small, so even if the amplitude Vh is set relatively small, the estimated position ripple is prevented from becoming excessively large.

Furthermore, in the region where the input voltage Vdc is relatively low, the upper limit of the voltage that the inverter 13 can output is relatively small. Therefore, by making the amplitude Vh relatively small in the region where the input voltage Vdc is relatively low, it is possible to prevent the harmonic superimposed voltage from being limited to the above upper limit.

Furthermore, since the amplitude Vh is relatively small in the region where the input voltage Vdc is relatively low, it is possible to reduce noise caused by harmonic currents and prevent the efficiency of the motor 210 from deteriorating due to harmonic currents.

Furthermore, as shown in FIG. 7, the harmonic amplitude regulator 23 performs a process to reduce the amplitude Vh when the absolute value of the AC current (U-phase current Iu is shown as a representative in FIG. 7) output from the inverter 13 (FIG. 2) to the motor 210 is smaller than the threshold value Ith1. FIG. 7 illustrates how the amplitude of the harmonic current superimposed on the AC current is reduced by reducing the amplitude Vh. As a result, as shown in FIG. 7, the amplitude of the harmonic current superimposed on the AC current is reduced. As a result, it is possible to suppress the occurrence of zero crossings of harmonics. Note that the threshold value Ith1 is an example of a "predetermined value" in the present disclosure. The threshold value Ith1 is a value close to 0. For example, the threshold value Ith1 may be a value approximately equal to the amplitude (maximum value) of the harmonic current before the amplitude Vh reduction process.

The harmonic amplitude regulator 23 may also reduce the amplitude Vh by a preset percentage (for example, 50%) when the absolute value of the AC current becomes smaller than the threshold value Ith1. The harmonic amplitude regulator 23 may also determine the percentage using a map or table depending on the current amplitude Vh, etc. While Figure 7 has been described based on the U-phase current Iu, the same applies to the W-phase current Iw and the V-phase current Iv.

The harmonic amplitude adjuster 23 may perform the processing shown in FIG. 7 based on the γ-axis harmonic current iγ and δ-axis harmonic current iδ output from the coordinate converter 25 and the estimated position Hm output from the position estimator 30. The harmonic amplitude adjuster 23 may also perform the processing shown in FIG. 7 based on the V-phase current Iv detected by the current sensor 213 and the W-phase current Iw detected by the current sensor 214. The processing to reduce the amplitude Vh shown in FIG. 7 is not required.

As described above, in this embodiment, the harmonic amplitude regulator 23 increases the amplitude Vh as the input voltage Vdc to the harmonic amplitude regulator 23 increases. As a result, while the ripple in the estimation error Δθ increases due to an increase in the input voltage Vdc, the ripple in the estimation error Δθ decreases as the amplitude Vh increases, thereby preventing the estimation error Δθ (total) from increasing. As a result, it is possible to prevent torque pulsation from increasing due to an increase in the estimation error Δθ (estimated position ripple).

Furthermore, in this embodiment, the estimated error Δθ is a value that is independent of the amplitude Vh based on the above equation (4), so the estimated error Δθ can be accurately calculated regardless of how the amplitude Vh is changed.

<Modification>
In the above embodiment, an example was described in which the amplitude Vh was reduced when the absolute value of the AC current output to the motor 210 was smaller than the threshold value Ith1, but the present disclosure is not limited to this. For example, as shown in FIG. 8 , the harmonic amplitude regulator 23 may perform a process to reduce the amplitude Vh when the amplitude of the AC current (U-phase current Iu in FIG. 8 ) output from the inverter 13 ( FIG. 2 ) to the motor 210 is smaller than the threshold value Ith2. For example, the threshold value Ith2 may be a value substantially equal to (the maximum value of) the amplitude of the harmonic current before the process to reduce the amplitude Vh. Note that the threshold value Ith2 is an example of a "predetermined value" in the present disclosure.

As a result, when the motor 210 is under low load and the amplitude of the AC current is small, the amplitude Vh is reduced, thereby reducing the amplitude of the harmonic current, making it possible to prevent the occurrence of zero crossings of harmonics. Figure 8 shows that when the amplitude of the AC current is less than threshold value Ith2 (solid line), the amplitude of the harmonic current superimposed on the AC current is smaller due to the reduction in amplitude Vh than when the amplitude of the AC current is equal to or greater than threshold value Ith2 (dashed line).

For example, the harmonic amplitude regulator 23 may reduce the amplitude Vh by a preset percentage (e.g., 50%) when the amplitude of the AC current becomes smaller than the threshold value Ith2. The harmonic amplitude regulator 23 may determine this percentage using a map, table, or the like, depending on the current amplitude Vh, etc. While FIG. 8 has been described based on the U-phase current Iu, the same applies to the W-phase current Iw and the V-phase current Iv. The processing in FIG. 8 may be performed based on the detection values of the current sensors 213, 214, or based on the γ-axis current command Iγ* and the δ-axis current command Iδ*, etc.

In the above embodiment, an example was shown in which the amplitude Vh increases as the input voltage Vdc increases, but the present disclosure is not limited to this. For example, the amplitude Vh may be constant at a first voltage when the input voltage Vdc is equal to or less than a threshold, and may be constant at a second voltage higher than the first voltage when the input voltage Vdc is greater than the threshold.

The configurations described in the above embodiments and the various modifications described above may be implemented in any combination.

The embodiments disclosed herein should be considered in all respects to be illustrative and not restrictive. The scope of the present disclosure is indicated by the claims, not by the description of the above embodiments, and is intended to include all modifications within the meaning and scope of the claims.

10 Power conversion unit (power converter), 23 Harmonic amplitude regulator, 24 PWM calculator (signal generator), 29 Position error estimator, 30 Position estimator, 100 Motor control device, 200 Electric compressor, 210 Motor, Aγ amplitude (amplitude of γ-axis harmonic current), Aδ amplitude (amplitude of δ-axis harmonic current), Ith1 threshold (predetermined value), Ith2 threshold (predetermined value), Vh amplitude (voltage amplitude).

Claims (5)

a signal generating unit that generates a signal to control the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate a rotor position of the motor to a driving voltage command value for driving the motor of the electric compressor;
a position error estimator that calculates an estimated error in the rotor position using the harmonic current;
a position estimator that estimates the rotor position using the estimated error calculated by the position error estimator;
a harmonic amplitude adjuster that adjusts a voltage amplitude that is the amplitude of the harmonic voltage command value,
the position error estimator calculates the estimation error using a γ-axis harmonic current, which is the harmonic current in a γ-axis that is an estimated axis of the d-axis, and a δ-axis harmonic current, which is the harmonic current in a δ-axis that is an estimated axis of the q-axis;
The harmonic amplitude adjuster
adjusting the voltage amplitude based on an input voltage to the harmonic amplitude adjuster;
A motor control device for an electric compressor, wherein the voltage amplitude when the input voltage is a first voltage value is made larger than the voltage amplitude when the input voltage is a second voltage value smaller than the first voltage value. The motor control device for an electric compressor according to claim 1, wherein the position error estimator calculates the estimation error based on the ratio between the amplitude of the γ-axis harmonic current and the amplitude of the δ-axis harmonic current. A motor control device for an electric compressor according to claim 1 or 2, wherein the harmonic amplitude regulator increases the voltage amplitude as the input voltage increases. a power converter that outputs an AC voltage to the motor performs dead time correction;
3. The motor control device for an electric compressor according to claim 1, wherein the harmonic amplitude adjuster executes a process of reducing the voltage amplitude when an absolute value of the AC current output from the power converter to the motor is smaller than a predetermined value. a power converter that outputs an AC voltage to the motor performs dead time correction;
3. The motor control device for an electric compressor according to claim 1, wherein the harmonic amplitude adjuster executes a process to reduce the voltage amplitude when the amplitude of the AC current output from the power converter to the motor is smaller than a predetermined value.