JP2020065392A - Motor control device, motor control method, motor module, electric power steering device and traction motor unit - Google Patents

Abstract

To provide a motor control device capable of reducing acoustic noise that may be generated in direct torque control.SOLUTION: A motor control device 100 according to one embodiment includes: a first hysteresis comparator (140) that generates a quantized flux signal based on a difference between a reference flux and a real flux; a second hysteresis comparator (130) for generating a quantized torque signal based on a difference between a reference torque and a real torque; a noise comparison unit (150) for calculating cumulative noise by accumulating noise levels including switching noise within a specific frequency bandwidth, and changing the bandwidth of the hysteresis of at least one of the first hysteresis comparator and the second hysteresis comparator according to a difference between reference noise and the calculated cumulative noise; and a switching selection unit (160) for determining an output voltage vector based on the quantized magnetic flux signal, the quantized torque signal, and a phase angle of the flux.SELECTED DRAWING: Figure 3

Description

  The present disclosure relates to a motor control device, a motor control method, a motor module, an electric power steering device, and a traction motor unit, and particularly to a motor control device and a motor control method that directly perform torque control.

  Direct torque control (DTC) is known as one of motor control methods. The direct torque control is a method of controlling the interlinkage magnetic flux and the instantaneous torque to select the output voltage vector of the inverter. International Publication No. 2011/065406 discloses a motor control device that selects an output voltage vector of an inverter based on a torque error between a reference torque and an estimated torque, a magnetic flux error between a reference magnetic flux and an estimated magnetic flux, a phase angle of a magnetic flux, and the like. is doing.

International Publication No. 2011/065406

  In the direct torque control, noise (referred to as "switching noise") due to the switching pattern of the switch element of the inverter may occur. The noise is acoustic noise and may cause noise.

  Embodiments of the present disclosure are capable of reducing acoustic noise that may occur in direct torque control, a novel motor control device, a motor control method, a motor module and a traction motor unit including the motor control device, and An electric power steering apparatus including the motor module is provided.

  The motor control device of the present disclosure, in the exemplary embodiment, based on the difference between the reference torque and the actual torque, and the first hysteresis comparator that generates the quantized magnetic flux signal based on the difference between the reference magnetic flux and the actual magnetic flux. A second hysteresis comparator that generates a quantized torque signal, and a noise level including switching noise in a specific frequency band are accumulated to calculate cumulative noise, and the cumulative noise is calculated according to a difference between the reference noise and the calculated cumulative noise. A noise comparison unit that changes the bandwidth of at least one hysteresis of the first hysteresis comparator and the second hysteresis comparator, and an inverter of the inverter based on the quantized magnetic flux signal, the quantized torque signal, and the phase angle of the magnetic flux. Determine output voltage vector associated with switching mode That includes a switching selection unit.

  In the exemplary embodiment, the motor control method of the present disclosure uses a first hysteresis comparator to generate a quantized magnetic flux signal based on a difference between a reference magnetic flux and an actual magnetic flux, and uses a second hysteresis comparator. Then, a quantized torque signal is generated based on the difference between the reference torque and the actual torque, cumulative noise is calculated by accumulating noise levels including switching noise within a specific frequency band, and reference noise and Changing the bandwidth of at least one hysteresis of the first hysteresis comparator and the second hysteresis comparator according to the calculated difference from the cumulative noise, and the quantized magnetic flux signal, the quantized torque signal, and the magnetic flux. Based on the phase angle, the output power associated with the inverter switching mode. Comprising determining a vector, a, and controlling the motor based on the output voltage vector.

  According to an exemplary embodiment of the present disclosure, a novel motor control device capable of reducing acoustic noise that may occur in direct torque control, a motor control method, a motor module including the motor control device, and a traction motor. Provided is a unit and an electric power steering device including the motor module.

FIG. 1 is a block diagram schematically showing hardware blocks of a motor module 1000 according to an exemplary embodiment 1. FIG. 2 is a block diagram showing a hardware configuration of the inverter 300 in the motor module 1000 according to the first exemplary embodiment. FIG. 3 is a functional block diagram showing functional blocks of the motor control device 100 according to the first embodiment. FIG. 4 is a diagram showing the interlinkage magnetic flux Ψ _αβ and its phase angle ρ in the αβ fixed coordinate system. FIG. 5 is a functional block diagram showing the main functional blocks of the noise comparison unit 150 in more detail. FIG. 6 is a diagram showing a magnetic flux locus representing a locus of the interlinkage magnetic flux vector Ψαβ that moves or stops with time. FIG. 7 is a diagram showing regions R (1) to R (6) obtained by dividing the coordinate plane of the αβ fixed coordinate system into six equal parts based on the phase angle ρ of the magnetic flux. FIG. 8 is a schematic diagram showing Y-connected three-phase windings La, Lb, and Lc. FIG. 9 is a schematic diagram showing the output voltage vectors V0 to V7. FIG. 10 is a schematic diagram showing a typical configuration of the EPS system 2000 according to the exemplary embodiment 2. FIG. 11 is a schematic diagram showing the configuration of the traction motor unit 3000 according to the third exemplary embodiment.

  Hereinafter, embodiments of a motor control device, a motor control method, a motor module, an electric power steering device, and a traction motor unit of the present disclosure will be described in detail with reference to the accompanying drawings. However, in order to avoid unnecessary redundancy in the following description and facilitate understanding by those skilled in the art, detailed description may be omitted more than necessary. For example, detailed description of well-known matters and repeated description of substantially the same configuration may be omitted.

(Embodiment 1)
[Configuration of Motor Module 1000]
FIG. 1 is a block diagram schematically showing a hardware block of a motor module 1000 according to this embodiment.

  The motor module 1000 typically includes a motor M, a motor control device 100, a drive circuit 200, an inverter (also referred to as an “inverter circuit”) 300, a plurality of current sensors 400, and an analog-digital conversion circuit. (Hereinafter, referred to as “AD converter”.) 500, a ROM (Read Only Memory) 600, and an angle sensor 700. However, as described later, the angle sensor 700 is not an essential component of the motor module 1000. The motor module 1000 can be modularized and can be manufactured and sold, for example, as a motor module having a motor, a sensor, a driver, and a motor controller, such as an electromechanical integrated motor.

  The motor M is, for example, a surface magnet type synchronous motor (SPMSM) or a switched reluctance motor (SRM). The motor M has, for example, three-phase (A-phase, B-phase, and C-phase) windings (not shown). The three-phase winding is electrically connected to the inverter 300. Not limited to a three-phase motor, a multi-phase motor such as a five-phase or a seven-phase motor is also within the scope of the present disclosure. The motor connection may be either Y connection or Δ connection. In this specification, an embodiment of the present disclosure will be described by taking a motor control device that controls a three-phase motor as an example.

  Motor controller 100 is typically a processor. The motor controller 100 can be, for example, an integrated circuit (IC) chip such as a digital signal processor (DSP) or a micro control unit (MCU). Alternatively, the motor control device 100 can also be realized by, for example, a field programmable gate array (FPGA) in which a core of a central processing unit (CPU) is incorporated.

  The motor control device 100 controls the motor M using direct torque control (hereinafter referred to as “DTC”). DTC is a control method that adjusts the output voltage vector of the inverter, which is the manipulated variable, based on the controlled variables of the interlinkage magnetic flux and the instantaneous torque. The motor control device 100 outputs a signal representing the output voltage vector (denoted as “output voltage vector signal”) to the drive circuit 200.

  The drive circuit 200 is typically a gate driver (or pre-driver). The drive circuit 200 generates a control signal for controlling ON / OFF of the switch element in the inverter 300 according to the output voltage vector signal output from the motor control device 100. The switch element is, for example, a field effect transistor (typically MOSFET) described later. The drive circuit 200 gives a gate control signal to the gate of each switch element. When the drive target is a motor that can be driven with a low voltage, the drive circuit 200 may not be necessarily required. In that case, the function of the gate driver may be implemented in the motor control device 100.

  The inverter 300 converts DC power supplied from a DC power supply (not shown) into AC power, for example. For example, the inverter 300 converts DC power into three-phase AC power that is a pseudo sine wave of A phase, B phase, and C phase based on the gate control signal output from the drive circuit 200. The motor M is driven by the converted three-phase AC power.

  The plurality of current sensors 400 include at least two current sensors that detect at least two phase currents flowing in the A-phase, B-phase, and C-phase windings of the motor M. In the present embodiment, the plurality of current sensors 400 have two current sensors 400A and 400B (see FIG. 2) that detect the currents flowing in the A phase and the B phase. Of course, the plurality of current sensors 400 may include three current sensors that detect three phase currents flowing in the A-phase, B-phase, and C-phase windings. You may have two current sensors which detect the phase current which flows, or the phase current which flows in C phase and A phase. The current sensor has, for example, a shunt resistor and a current detection circuit (not shown) that detects a current flowing through the shunt resistor.

  The AD converter 500 samples the analog signals output from the plurality of current sensors 400, converts the analog signals into digital signals, and outputs the converted digital signals to the motor control device 100. However, the motor control device 100 may perform AD conversion. In that case, the plurality of current sensors 400 directly output an analog signal to the motor control device 100.

  The ROM 600 is, for example, a writable memory (for example, PROM), a rewritable memory (for example, flash memory), or a read-only memory. The ROM 600 stores a control program having a group of instructions for causing the motor control device 100 to control the motor M. For example, the control program is once expanded in the RAM (not shown) at the time of booting. The ROM 600 does not need to be externally attached to the motor control device 100 and may be mounted in the motor control device 100. The motor control device 100 equipped with the ROM 600 can be, for example, the MCU described above.

  The angle sensor 700 is, for example, a resolver or a Hall IC. Alternatively, the angle sensor 700 is also realized by a combination of an MR sensor having a magnetoresistive (MR) element and a sensor magnet. The angle sensor 700 detects the rotation angle of the rotor (hereinafter referred to as “rotor angle”) and outputs the rotor angle to the motor control device 100.

  In the sensor control that requires the angle sensor shown in FIG. 1, the rotor angle is calculated based on the measured value of the angle sensor. On the other hand, in the sensorless control that does not require the angle sensor shown in FIG. 2, the rotor angle is estimated based on the current measured by the current sensor and the like. When performing sensorless control, the angle sensor 700 is not essential.

  The hardware configuration of the inverter 300 will be described in more detail with reference to FIG.

  FIG. 2 is a block diagram showing the hardware configuration of the inverter 300 in the motor module 1000 according to this embodiment.

  The inverter 300 is a half-bridge circuit and has three low-side switch elements and three high-side switch elements. The illustrated switch elements SW_L1, SW_L2 and SW_L3 are low side switch elements, and the switch elements SW_H1, SW_H2 and SW_H3 are high side switch elements. As the switch element, for example, a semiconductor switch element such as a MOSFET or an insulated gate bipolar transistor (IGBT) can be used.

  FIG. 2 shows the shunt resistance Rs of the two current sensors 400A and 400B that detect the phase currents flowing in the A phase and the B phase. As illustrated, for example, the shunt resistor Rs can be electrically connected between the low side switch element and the ground. Alternatively, for example, the shunt resistor Rs can be electrically connected between the high side switch element and the power supply.

  The motor control device 100 can drive the motor M by performing control by three-phase energization using direct torque control (hereinafter, referred to as “three-phase energization control”). For example, the motor control device 100 generates an output voltage vector signal for performing three-phase energization control, and outputs the signal to the drive circuit 200. The drive circuit 200 produces | generates the gate control signal which controls the switching operation of each FET in the inverter 300 based on an output voltage vector signal, and gives it to the gate of each FET.

  The motor module 1000 may have a speed sensor or an acceleration sensor instead of the angle sensor 700, for example. When the speed sensor is used as the angle sensor, the motor control device 100 can calculate the rotation angle, that is, the rotor angle by performing integration processing or the like on the rotation speed signal or the angular speed signal. The rotation speed is represented by a rotation speed (rpm) at which the rotor rotates in a unit time (for example, 1 minute) or a rotation speed (rps) at which the rotor rotates in a unit time (for example, 1 second). The angular velocity is represented by the angle (rad / s) that the rotor rotates in 1 second. When the acceleration sensor is used as the angle sensor, the motor control device 100 can calculate the rotor angle by performing an integration process or the like on the angular acceleration signal.

  The motor controller of the present disclosure is implemented in non-limiting exemplary embodiments only in hardware, such as an application specific integrated circuit (ASIC) or FPGA. Alternatively, it may be realized by a memory that stores a program that defines a series of processes required for motor control, and a processor that sequentially executes each process according to the program. Further, it can be realized by combining hardware and software.

  The motor control method of the present disclosure can be executed using, for example, an apparatus included in the motor module shown in FIG. 1 or 2. The motor control method of the present disclosure can be widely used for controlling various types of motors such as SPMSM or SRM.

  Please refer to FIG. 3 to FIG. Hereinafter, the function / operation of the motor control device 100 that performs DTC will be described in more detail.

[Functional Block of Motor Control Device 100]
FIG. 3 is a functional block diagram showing functional blocks of the motor control device 100 according to this embodiment.

  In this specification, each block in the functional block diagram is shown in a functional block unit rather than in a hardware unit. The software for controlling the motor may be, for example, a module forming a computer program for executing a specific process corresponding to each functional block. Such a computer program is stored in the ROM 600, for example.

  The motor control device 100 includes an estimator 110, differencers 120A and 120B, a torque hysteresis comparator 130, a magnetic flux hysteresis comparator 140, a noise comparison unit 150, and a switching selection unit 160. Motor control device 100 determines the output voltage vector associated with the switching mode of inverter 300 based on the magnetic flux error, the torque error, and the phase angle of the magnetic flux. In this specification, the functional block may be referred to as a unit for convenience of description. However, such notation should not be interpreted with the intention of limiting each functional block to hardware or software.

When each functional block is implemented in the motor control device 100 as software, the execution body of the software may be a processor, for example. The motor control device of the present disclosure includes, in a non-limiting and exemplary embodiment, a processor and a memory that stores a program that controls the operation of the processor. The processor executes the following processing according to the program.
(1) Using the magnetic flux hysteresis comparator, a quantized magnetic flux signal is generated based on the difference between the reference magnetic flux and the actual magnetic flux.
(2) A quantized torque signal is generated based on the difference between the reference torque and the actual torque using the torque hysteresis comparator.
(3) Cumulative noise is calculated by accumulating noise levels including switching noise within a specific frequency band.
(4) The bandwidth of at least one hysteresis of the magnetic flux hysteresis comparator and the torque hysteresis comparator is changed according to the difference between the reference noise and the calculated cumulative noise.
(5) The output voltage vector associated with the switching mode of the inverter is determined based on the quantized magnetic flux signal, the quantized torque signal, and the phase angle of the magnetic flux.
(6) Control the motor based on the output voltage vector.

  The motor control device 100 can also be realized by an FPGA. In that case, all or some of the functional blocks may be implemented by hardware. By distributing the processing using a plurality of FPGAs, it is possible to distribute the calculation load of a specific computer. In that case, all or some of the functional blocks illustrated in FIG. 3 may be distributed and implemented in the plurality of FPGAs. A plurality of FPGAs are communicably connected to each other, for example, by a vehicle-mounted control area network (CAN), and data is transmitted and received between them.

The estimator 110 acquires the phase currents Ia and Ib measured by the current sensor 400. The estimator 110 calculates the phase current Ic from the phase currents Ia and Ib. The estimator 110 estimates the flux linkage Ψ _αβ , the torque T _e, and the phase angle ρ of the flux linkage based on the phase currents Ia, Ib, and Ic.

FIG. 4 is a diagram showing the interlinkage magnetic flux Ψ _αβ and its phase angle ρ in the αβ fixed coordinate system.

The estimated interlinkage magnetic flux Ψ _αβ is represented by a phasor display in the αβ fixed coordinate system, and is calculated based on the equation (1). Here, v _αβ is a stator voltage vector in the αβ fixed coordinate system. R _s is the armature resistance. i _αβ is the stator current vector in the αβ fixed coordinate system. All of these variables are represented in phasor notation. ∫ is an integral operator.
Ψ _αβ = ∫ (v _αβ −R _s · i _αβ ) dt Equation (1)

The estimated torque T _e is calculated based on the equation (2). The estimated phase angle ρ of the interlinkage magnetic flux is calculated based on the equation (3). In this specification, the estimated interlinkage magnetic flux Ψ _αβ estimated by the estimator 110 will be referred to as “actual magnetic flux Ψ _αβ ”, and the estimated torque T _e will be referred to as “actual torque T _e ”. Actual flux [psi _.alpha..beta and actual torque T _e is the estimated value calculated based on the phase current Ia is determined by the current sensor 400, to Ib and Ic.
T _e = Ψ _αβ × i _αβ Equation (2)
ρ = arctan (Ψ _β / Ψ _α ) Formula (3)

In the motor module 1000 including the angle sensor 700 shown in FIG. 1, the estimator 110 may further acquire the rotor angle θ _r from the angle sensor 700. The estimator 110 can estimate the interlinkage flux Ψ _αβ , the torque T _e, and the phase angle ρ of the interlinkage flux based on the phase currents Ia, Ib, Ic, and the rotor angle θ _r . For example, the estimator 110 acquires the estimated phase angle ρ by adding the rotor angle θe to the estimated torque angle δ shown in FIG. 4, which is estimated based on the phase currents Ia, Ib, and Ic.

The differentiator 120A calculates a signed difference ΔT between the reference torque T ^* and the actual torque T _e and outputs it to the torque hysteresis comparator 130. The torque error between the reference torque T ^* and the actual torque T _e is defined by the difference ΔT. The differentiator 120B calculates a signed difference ΔΨ between the reference magnetic flux Ψ ^* and the actual magnetic flux Ψ _αβ and outputs it to the magnetic flux hysteresis comparator 140. The magnetic flux error between the reference magnetic flux Ψ ^* and the actual magnetic flux Ψ _αβ is defined by the difference ΔΨ.

The reference torque T ^* and the reference magnetic flux Ψ ^* are calculated by, for example, an upstream controller (not shown) and given to the difference units 120A and 120B, respectively. The reference torque T ^* represents, for example, a torque required to bring the rotation speed of the rotor to a target (target) speed.

The torque hysteresis comparator 130 is a hysteresis comparator that generates the quantized torque signal τ based on the torque error ΔT. The torque hysteresis comparator 130 generates, for example, a ternary quantized torque signal τ. When the reference torque T ^* is larger than the actual torque T _e , the torque error ΔT has a positive value, and when the reference torque T ^* is smaller than the actual torque T _e , the torque error ΔT has a negative value. For example, when the torque error ΔT is larger than the predetermined positive threshold value, the torque hysteresis comparator 130 outputs “1”, and when the torque error is smaller than the predetermined negative threshold value, the torque hysteresis comparator 130 outputs “−”. 1 ”is output. When the torque error ΔT is greater than or equal to the negative threshold value and less than or equal to the positive threshold value, the torque hysteresis comparator 130 outputs “0”.

  The predetermined positive threshold value corresponds to the upper limit value of the hysteresis bandwidth of the torque hysteresis comparator 130, and the predetermined negative threshold value corresponds to the lower limit value of the hysteresis bandwidth. In other words, the hysteresis bandwidth of the torque hysteresis comparator 130 represents the difference between the upper limit of the positive threshold and the lower limit of the negative threshold.

The magnetic flux hysteresis comparator 140 is a hysteresis comparator that generates the quantized magnetic flux signal ψ based on the magnetic flux error ΔΨ. The magnetic flux hysteresis comparator 140 generates, for example, a binary quantized magnetic flux signal ψ. When the reference magnetic flux Ψ ^* is larger than the actual magnetic flux Ψ _αβ , the magnetic flux error ΔΨ has a positive value, and when the reference magnetic flux Ψ ^* is smaller than the actual magnetic flux Ψ _αβ , the magnetic flux error ΔΨ has a negative value. When the magnetic flux error ΔΨ is larger than the predetermined positive threshold value, the magnetic flux hysteresis comparator 140 outputs “1”. When the magnetic flux error ΔΨ is smaller than the predetermined negative threshold value, the magnetic flux hysteresis comparator 140 outputs “−1”. Is output. When the magnetic flux error ΔΨ is greater than or equal to the negative threshold value and less than or equal to the positive threshold value, the magnetic flux hysteresis comparator 140 maintains the output of “−1” or “1” in the immediately previous state.

  Similar to the torque hysteresis comparator 130, the predetermined positive threshold value corresponds to the upper limit value of the hysteresis bandwidth of the magnetic flux hysteresis comparator 140, and the predetermined negative threshold value corresponds to the lower limit value of the hysteresis bandwidth. In other words, the hysteresis bandwidth of the magnetic flux hysteresis comparator 140 represents the difference between the positive threshold value that is the upper limit thereof and the negative threshold value that is the lower limit thereof.

  FIG. 5 is a functional block diagram showing the main functional blocks of the noise comparison unit 150 in more detail.

  The noise comparison unit 150 has a cumulative noise calculation unit 151, a difference unit 152, a PI controller 153, and a limiter 154. The noise comparison unit 150 accumulates noise levels including switching noise within a specific frequency band to calculate cumulative noise. Further, the noise comparison unit 150 changes the bandwidth of at least one hysteresis of the torque hysteresis comparator 130 and the magnetic flux hysteresis comparator 140 according to the difference between the reference noise and the calculated accumulated noise.

  In the present embodiment, the noise comparison unit 150 accumulates noise levels in a specific frequency band included in the phase currents Ia, Ib, and Ic to calculate cumulative noise. However, the noise comparison unit 150 does not have to calculate the cumulative noise for all the three-phase currents, and may calculate the cumulative noise for at least one of the three phases. The noise comparison unit 150 changes only the hysteresis bandwidth of the magnetic flux hysteresis comparator 140 based on the noise error defined by the difference between the reference noise and the accumulated noise. The noise comparison unit 150 may change the hysteresis bandwidth of the torque hysteresis comparator 130. However, changing its bandwidth can cause torque ripple and increase mechanical noise. Therefore, it is preferable to change only the hysteresis bandwidth of the magnetic flux hysteresis comparator 140 while keeping the hysteresis bandwidth of the torque hysteresis comparator 130 fixed.

  The cumulative noise calculation unit 151 receives the phase currents Ia, Ib and Ic, and calculates the cumulative noise for each phase based on them. Hereinafter, a specific example of calculating the A-phase cumulative noise will be described. The accumulated noise for the B and C phases can be calculated in the same manner as for the A phase.

The cumulative noise calculation unit 151 calculates the I _a (S) by performing _a fast Fourier transform (FFT) of the A-phase instantaneous phase current ia (t) based on the equation (4). Here, Ts is defined as the sample time (time resolution) of the FFT simulation. t is a time interval of FFT calculation. n is the number of samples represented by t / T _s (the number of data of the instantaneous values of current included in the time interval t).
I _a (S) = FFT (ia (t)) / n Equation (4)

The cumulative noise calculation unit 151 calculates the power spectrum of I _a (S) based on the equation (5). Here, Z is a temporary value of the armature impedance.
Power ₀ (S) = 10 log (Z · I _a (S) ² ) Formula (5)

  According to the knowledge of the inventor, unlike vector control, the switching frequency is not constant when DTC is performed, and thus it becomes difficult to reduce audible acoustic noise. For example, a vehicle-mounted traction motor unit is desired to operate as quietly as possible. Therefore, in controlling a vehicle motor, how to reduce acoustic noise that may occur due to a switching operation of a switching element of an inverter becomes a problem.

  For example, if the switching frequency can be set to 20 kHz or higher, it leads to reduction of audible noise. This is because human hearing is said to be most sensitive to the frequency band from 500 Hz to 8 kHz. For low frequencies, the sensitivity gradually decreases when the frequency becomes 1 kHz or less, and almost disappears when the frequency falls below 20 Hz. Similarly, for high frequencies, the sensitivity gradually decreases as the frequency increases, and becomes almost zero when the frequency generally exceeds 20 kHz. However, in a high power drive system that requires high power such as a traction motor, when the switch element is switched at a switching frequency of 20 kHz, a high power loss may occur in the switch element. This should be avoided.

  For example, in certain traction applications, the switching frequency is on the order of 7 kHz to 10 kHz. However, since these frequencies include audible frequencies, audible acoustic noise may occur due to switching noise when the motor is driven. Furthermore, when performing DTC, the switching frequency is not constant, which makes it more difficult to reduce audible acoustic noise.

  In view of the above problems, the noise comparison unit 150 performs weighting by using the frequency weighting characteristic A that takes human auditory perception characteristics into consideration, which is defined by the International Electrotechnical Commission (IEC) standard: IEC 61672-2013. It is preferable to correct the cumulative noise level and obtain the corrected cumulative noise level. Considering that the noise of the electric motor is less than 100 dB, the frequency weighting characteristic A is more suitable for correcting the cumulative noise level than the frequency weighting characteristic C.

The noise comparison unit 150 interpolates the data into the A characteristic curve H (S) using the sampling number n to calculate the power spectrum Power _x (S). The noise comparison unit 150 calculates the noise power NoisePower (S) by adding the power spectrum Power _x (S) to the power spectrum Power ₀ (S) of the phase current A based on the equation (6).
NoisePower (S) = Power ₀ (S) + Power _x (S) Formula (6)

（式７） The noise comparing unit 150 calculates a corrected cumulative noise level Cumulative noise by integrating NoisePower (S) given by Expression (6) based on Expression (7) in a specific frequency section or frequency band. To do. Here, fstart and fstop represent the lower end and the upper end of a specific frequency section, respectively. Cumulative noise is a sound pressure level after being weighted by the A characteristic, and its unit is represented by [dBA]. The specific frequency section can be set for each motor application. The section is set in the range from 1 kHz (fstart) to 5 kHz (fstop), for example. For example, fstop can be at frequencies up to 20 kHz.

(Equation 7)

  The differencer 152 of the noise comparison unit 150 calculates a noise error defined by the difference between the reference noise and the corrected cumulative noise level. The noise comparison unit 150 changes the hysteresis bandwidth bw of the magnetic flux hysteresis comparator 140 based on the noise error. More specifically, the noise comparison unit 150 determines the bandwidth bw of the hysteresis of the magnetic flux hysteresis comparator 140 by proportionally integrating the noise error using the PI controller 153 and determining its limit value using the limiter 154. To do.

  The noise comparison unit 150 sets the calculated hysteresis bandwidth bw in the magnetic flux hysteresis comparator 140. The calculation of the hysteresis bandwidth bw is repeatedly executed, for example, in synchronization with a cycle in which each phase current is measured by the current sensor 400, that is, an AD conversion cycle of the AD converter 500. Therefore, the hysteresis bandwidth bw set in the magnetic flux hysteresis comparator 140 can change according to the integrated noise level and can be updated in synchronization with the AD conversion cycle.

As described above, by correcting the power spectrum of the accumulated noise using the frequency weighting characteristic A, it becomes possible to appropriately reduce audible acoustic noise including switching noise. However, the differencer 152 of the noise comparison unit 150 may calculate the noise error defined by the difference between the reference noise and the original cumulative noise level before correction, without correcting the power spectrum of the cumulative noise. The original cumulative noise level is obtained by substituting Power ₀ (S) given by equation (5) into equation (7) instead of NoisePower (S).

  The outline of the DTC will be described with reference to FIGS. 6 to 9.

FIG. 6 is a diagram showing a magnetic flux locus representing a locus of the interlinkage magnetic flux vector Ψ _αβ that moves or stops in time according to the switching mode.

The DTC optimally limits the magnetic flux error ΔΨ and the torque error ΔT within a small allowable range and rotates the interlinkage magnetic flux vector Ψ _αβ in the magnetic flux locus shown in FIG. 6 clockwise or counterclockwise. The output voltage vector Vn is determined based on the quantized magnetic flux signal ψ, the quantized torque signal τ, and the magnetic flux phase angle ρ. The minute allowable range is defined by the hysteresis bandwidth bw of the torque hysteresis comparator 130 or the magnetic flux hysteresis comparator 140 described above.

The inner circle C1 of the two circles C1 and C2 shown in FIG. 6 represents the lower limit of the hysteresis bandwidth bw, which means the allowable range, and the outer circle C2 represents the upper limit of the hysteresis bandwidth bw. Is expressed. The difference (gap) in radius between the inner and outer circles C1 and C2 is the hysteresis bandwidth bw. The interlinkage magnetic flux vector Ψ _αβ is controlled so as to move within the region of the hysteresis bandwidth bw in the magnetic flux locus. According to the conventional method, the bandwidth bw of hysteresis is always fixed. On the other hand, according to the method of the present disclosure, the bandwidth bw of the hysteresis may change according to the cumulative noise level calculated by the noise comparison unit 150.

  In the DTC, the switching frequency can be lowered by expanding the hysteresis bandwidth bw of the magnetic flux hysteresis comparator 140. On the other hand, it is possible to increase the switching frequency by reducing the hysteresis bandwidth bw.

  The switching selection unit 160 determines the output voltage vector Vn (A, B, C) associated with the switching mode of the inverter 300 based on the quantized magnetic flux signal ψ, the quantized torque signal τ and the magnetic flux phase angle ρ. . More specifically, the switching selection unit 160 refers to the look-up table LUT that associates the switching mode with the output voltage vector Vn (A, B, C) to determine the quantized magnetic flux signal ψ, the quantized torque signal τ and the magnetic flux. The output voltage vector Vn (A, B, C) is determined based on the phase angle ρ.

Table 1 shows the table used to select the switching mode. The “↑” in the column of the torque T indicates that the quantized torque signal τ output from the torque hysteresis comparator 130 is “1”. This indicates that the actual torque _Te is less than the reference torque T ^* . In that case, it is necessary to increase the switching frequency of the inverter to increase the torque as quickly as possible. That is, “↑” indicates that, for example, when the rotor is rotating counterclockwise, the interlinkage magnetic flux vector Ψ _αβ is rotated counterclockwise in the magnetic flux locus of FIG. 6, thereby increasing the torque. .

"↓" in the column of torque T indicates that the quantized torque signal τ is "-1". This indicates that the actual torque _Te has reached the reference torque T ^* . In that case, it is necessary to reduce the switching frequency of the inverter to reduce the torque as gently as possible. That is, “↓” indicates that, for example, when the rotor is rotating counterclockwise, the interlinking magnetic flux vector Ψ _αβ is rotated clockwise in the magnetic flux locus of FIG. 6, thereby reducing the torque.

  “→” in the column of the torque T indicates that the quantized torque signal τ is “−1”. This indicates that the torque error is within the allowable range defined by the hysteresis bandwidth. In that case, the torque is maintained. However, in reality, the torque gradually decreases due to the gradual decrease of the magnetic flux.

“↑” in the column of the magnetic flux Ψ indicates that the quantized magnetic flux signal ψ output from the magnetic flux hysteresis comparator 140 is “1”. This indicates that the actual magnetic flux Ψ _αβ is less than the reference magnetic flux Ψ ^* . "↑" means that the magnetic flux needs to be increased. “↓” indicates that the quantized magnetic flux signal ψ is “−1”. This indicates that the actual magnetic flux Ψ _αβ has reached the reference magnetic flux Ψ ^* . "↓" means that the magnetic flux needs to be reduced. “−” Means to ignore the quantized magnetic flux signal ψ.

  FIG. 7 is a diagram showing regions R (1) to R (6) obtained by dividing the coordinate plane of the αβ fixed coordinate system into six equal parts based on the phase angle ρ of the magnetic flux.

  In Table 1, n indicates a switching mode to be selected from switching modes 0 to 7. By substituting the value indicating the range of the phase angle ρ of the magnetic flux into K, n is determined. Each range of the regions R (1) to R (6) indicates a range of the phase angle ρ corresponding to π / 3. For example, the range of the region R (1) indicates the range of the magnetic flux phase angle ρ from −π / 6 to π / 6. K is a value of any of area numbers 1 to 6.

For example, when the torque T indicates “↑” and the magnetic flux Ψ indicates “↓”, K is 2, that is, in the coordinate plane of the αβ fixed coordinate system, the interlinkage magnetic flux vector Ψ _αβ is the region R. When located in (2), the switching selection unit 160 selects 4 as the n value. In other words, the switching selection unit 160 selects the switching mode 4. When the torque T indicates “↓” and the magnetic flux Ψ indicates “↑”, K is 4, that is, the magnetic flux vector is located in the region R (4) on the coordinate plane of the αβ fixed coordinate system. The switching selection unit 160 selects 3 as the n value. In other words, the switching selection unit 160 selects the switching mode 3. However, if the value of K + 2 or K + 1 exceeds 6, the value obtained by subtracting 6 from those values is selected as the n value. When the value of K-2 or K-1 is less than 1, a value obtained by adding 6 to those values is selected as the n value.

  FIG. 8 is a schematic diagram showing Y-connected three-phase windings La, Lb, and Lc. FIG. 9 is a schematic diagram showing the output voltage vectors V0 to V7. FIG. 7 shows the relationship between the six divided regions R (1) to R (6) and the output voltage vectors V0 to V7.

  Since the leg of the half bridge of each phase has two switch elements, there are eight types of inverter switching patterns. The output voltage vector Vn (A, B, C) corresponding to the switching pattern can be defined. n is an integer of 0 to 7. Each of A, B and C of the output voltage vector Vn can take a value of "-1" or "1" depending on the value of n.

  As shown in FIG. 9, the output voltage vector Vn (A, B, C) is, for example, two types of zero voltage vectors V0 (-1, -1, -1) and V7 (1, 1) in the αβ fixed coordinate system. , 1), six types of voltage vectors V1 (1, -1, -1), V2 (1,1, -1), V3 (-1) having the same magnitude and having a phase difference of π / 3 with each other. , 1, -1), V4 (-1,1,1), V5 (-1, -1,1) and V6 (1, -1,1).

  In the output voltage vector Vn, “1” means that a positive voltage is applied to the coil end of each phase shown in FIG. 8, and “−1” means that a negative voltage is applied to the coil end of each phase. means. For example, the output voltage vector V1 (1, -1, -1) gives a positive voltage to the coil end a and a negative voltage to the coil ends b and c. The output voltage vector V2 (1,1, -1) gives a positive voltage to the coil ends a and b and a negative voltage to the coil end c.

  Table 2 shows a lookup table LUT that associates the switching mode with the output voltage vector Vn. Switching modes 0-7 are associated one-to-one with output voltage vectors V0-V7. The switching selection unit 160 refers to the reference table LUT to select the output voltage vector Vn corresponding to the switching mode selected based on the table shown in Table 1, and to select the output voltage vector Vn in the latter stage inverter 300, more specifically, the drive circuit 200. Output the output voltage vector signal.

  The relationship between the output voltage vector Vn (A, B, C) and the switching pattern will be briefly described. When the output value is "1", the high side switch element in the leg of the phase is turned on and the low side switch element is turned off. On the contrary, when the output value is “−1”, the high side switch element in the leg of the phase is turned off and the low side switch element is turned on.

  The drive circuit 200 generates a gate control signal to be given to each switch element of the inverter based on the output voltage vector signal input from the motor control device 100, and outputs the gate control signal to the inverter 300. For example, when the drive circuit 200 receives an output voltage vector signal indicating the output voltage vector V3, the drive circuit 200 turns off the A-side and C-phase high-side switching elements of the inverter 300 and turns on the low-side switching elements, and the B-phase high side switching elements. A gate control signal for turning on the side switching element and turning off the low side switching element is generated.

  When a current is caused to flow through a coil or winding having an inductance as a basic element, noise contained in the current may be a main cause of acoustic noise. On the other hand, when a voltage is applied to a capacitor or the like having a capacitance as a basic element, noise contained in the voltage may be a main cause of acoustic noise.

In the present embodiment, the cumulative noise calculation unit 151 has described one mode of calculating the cumulative noise by accumulating the noise levels in the specific frequency band included in the phase current, but the present disclosure is not limited to this. The noise comparison unit 150 may calculate the cumulative noise by accumulating the noise levels in the specific frequency band included in the phase voltage. Specifically, the noise comparison unit 150 calculates the power spectrum Power ₀ (S) of V _a (S) for the A phase based on the equations (8) and (9). Here, va (t) is the instantaneous value of the phase voltage of the A phase. n indicates the number of samplings. Z is a prerequisite value for the armature impedance. The power spectra of the B and C phases are calculated similarly to the A phase.
_{V a (S) = FFT (} va (t)) / n (8)
Power ₀ (S) = 10 log (V _a (S) ² / Z) Formula (9)

  According to this modification, it is possible to suppress acoustic noise that may occur due to the capacitance of the bypass capacitor provided in the power supply line of the inverter 300, for example.

  Below, the result of verifying the adequacy of the motor control method of the DTC according to the present disclosure using Matlab / Simlink of MathWorks is shown. For this verification, a model of a three-phase, eight-pole, twelve-slot permanent magnet type synchronous motor controlled by the DTC was used. Table 3 shows the values of various system parameters at the time of verification.

  When the maximum switching frequency is set to 7 kHz or 10 kHz, the cumulative noise in the frequency range from 100 Hz to 20 kHz and the frequency range from 1 kHz to 5 kHz are obtained under the condition that the hysteresis hysteresis and the magnetic flux hysteresis comparator have different hysteresis bandwidths. The cumulative noise at was measured. The accumulated noise is a measurement value obtained by accumulating the noise levels in the above frequency range included in the phase current.

The hysteresis bandwidth of the torque hysteresis comparator was set to 1e-06 N · m, 2.5% and 5%. The hysteresis bandwidth of the magnetic flux hysteresis comparator was set to 1e-06Wb (or Vs), 2.5% and 5%. Here, the% of the bandwidth of the torque hysteresis indicates the ratio of the bandwidth to the reference torque T ^* (7.5 N.m). The percentage of the bandwidth of the magnetic flux hysteresis is the ratio of the bandwidth to the reference magnetic flux Ψ ^* (0.0150 V · s).

  Table 4 shows a simulation result of accumulated noise when the maximum switching frequency is set to 7 kHz. Table 5 shows a simulation result of accumulated noise when the maximum switching frequency is set to 10 kHz.

  From the simulation result, it can be seen that the cumulative noise changes by adjusting the bandwidth of the hysteresis. When the maximum switching frequency is set to 7 kHz and 10 kHz and the hysteresis bandwidth of the torque hysteresis comparator is fixed (1e-06, 2.5% or 5% is set), the hysteresis band of the magnetic flux hysteresis comparator is set. It was confirmed that the cumulative noise in the audible frequency band can be appropriately reduced by reducing the width. This is because the switching frequency is increased by reducing the hysteresis bandwidth, and as a result, the power spectrum of the phase current includes a higher frequency component. In other words, this is because the component of the power spectrum is shifted to the high frequency side. Thus, it can be seen that the DTC motor control method according to the present disclosure is effective in reducing audible acoustic noise.

(Embodiment 2)
FIG. 10 is a schematic diagram showing a typical configuration of the EPS system 2000 according to this embodiment.

  Vehicles such as automobiles typically have EPS systems. The EPS system 2000 according to this embodiment includes a steering system 520 and an auxiliary torque mechanism 540 that generates an auxiliary torque. The EPS system 2000 generates an auxiliary torque that assists the steering torque of the steering system generated by the driver operating the steering wheel. The auxiliary torque reduces the driver's operational burden.

  The steering system 520 includes, for example, a steering handle 521, a steering shaft 522, universal joints 523A and 523B, a rotary shaft 524, a rack and pinion mechanism 525, a rack shaft 526, left and right ball joints 552A and 552B, tie rods 527A and 527B, and knuckles. 528A, 528B and left and right steering wheels 529A, 529B.

  The auxiliary torque mechanism 540 includes, for example, a steering torque sensor 541, a vehicle electronic control unit (ECU) 542, a motor 543, and a speed reduction mechanism 544. The steering torque sensor 541 detects the steering torque in the steering system 520. The ECU 542 generates a drive signal based on the detection signal of the steering torque sensor 541. The motor 543 generates an auxiliary torque according to the steering torque based on the drive signal. The motor 543 transmits the generated auxiliary torque to the steering system 520 via the speed reduction mechanism 544.

  The ECU 542 includes, for example, a motor control device and a drive circuit. In automobiles, an electronic control system centered on the ECU is built. In the EPS system 2000, for example, a motor control system is constructed by the ECU 542, the motor 543, and the inverter 545. The motor module 1000 according to the first embodiment can be preferably used in the motor control system. By appropriately adjusting the bandwidth of the hysteresis, it is possible to appropriately reduce acoustic noise that may occur when the motor is driven at a low speed, for example.

(Embodiment 3)
FIG. 11 is a schematic diagram showing the configuration of the traction motor unit 3000 according to this embodiment.

  The traction motor unit 3000 includes a motor 910, a gear box GB, a housing 940, an inverter unit 950, and an inverter housing 960. The traction motor unit 3000 is mounted on a vehicle having a motor as a power source, such as a hybrid vehicle (HEV), a plug-in hybrid vehicle (PHV), or an electric vehicle (EV), and is used as the power source.

  The motor 910 is a so-called traction motor, for example, an SRM.

  The gearbox GB has a speed reducer 920 and a differential gear unit 930, and is housed in a housing 940. The gearbox GB is connected to a shaft 912 extending parallel to the motor axis J1. The torque output from the motor 910 is transmitted to the differential device 930 via the speed reducer 920.

  The speed reducer 920 is connected to the rotor of the motor 910 via the shaft 912. The speed reducer 920 reduces the rotation speed of the motor 910 and increases the torque output from the motor 910 according to the speed reduction ratio.

  The speed reducer 920 includes a first gear (intermediate drive gear) 911, a second gear (intermediate gear) 921, a third gear (filenal drive gear) 922, and an intermediate shaft (not shown). Have. The torque output from the motor 910 is transmitted to the ring gear (gear) 931 of the differential gear unit 930 via the shaft 912 of the motor 910, the first gear 911, the second gear 921, the intermediate shaft and the third gear 922. To be done. The gear ratio of each gear, the number of gears, and the like can be changed according to the required reduction ratio. The speed reducer 920 is, for example, a parallel shaft gear type speed reducer in which the axes of the gears are arranged in parallel with each other.

  The differential device 930 is connected to the motor 910 via the speed reducer 920. The differential device 930 is a device that transmits the torque output from the motor 910 to the wheels of the vehicle. The differential device 930 transmits the same torque to the axles of the left and right wheels while absorbing the speed difference between the left and right wheels when the vehicle turns.

  The housing 940 is a metal member formed of, for example, an aluminum alloy. The housing 940 may be configured by combining a plurality of members.

  The inverter unit 950 has an inverter and a motor control device. The motor control device 100 according to the first embodiment can be preferably used as the motor control device. This provides a traction motor unit capable of appropriately reducing acoustic noise including switching noise that may occur due to switching of the inverter.

  INDUSTRIAL APPLICABILITY The motor control device and the motor control method of the present disclosure are suitably used for a motor control system such as X-by-wire such as shift-by-wire, steering-by-wire or brake-by-wire, which requires reduction of acoustic noise. For example, the motor control device 100 according to the first embodiment may be mounted on an autonomous vehicle that corresponds to levels 0 to 5 (standard of automation) defined by the Government of Japan and the Department of Road Traffic Safety of the United States (NHTSA).

  INDUSTRIAL APPLICABILITY The motor control device, the motor control method, and the motor module of the present disclosure can be widely used for various devices having various motors such as a vacuum cleaner, a dryer, a ceiling fan, a washing machine, a refrigerator, and an electric power steering system.

  100 motor control device, 200 drive circuit, 300 inverter, 400 current sensor, 500 AD converter, 600 ROM, 700 angle sensor, 1000 motor module

Claims (15)

A first hysteresis comparator that generates a quantized magnetic flux signal based on the difference between the reference magnetic flux and the actual magnetic flux;
A second hysteresis comparator that generates a quantized torque signal based on the difference between the reference torque and the actual torque;
Cumulative noise is calculated by accumulating noise levels including switching noise within a specific frequency band, and the first hysteresis comparator and the second hysteresis comparator are calculated according to the difference between the reference noise and the calculated cumulative noise. A noise comparison unit that modifies at least one hysteresis bandwidth of
A switching selection unit that determines an output voltage vector associated with the switching mode of the inverter based on the quantized magnetic flux signal, the quantized torque signal, and the phase angle of the magnetic flux;
And a motor control device.   The noise comparison unit changes only the bandwidth of the hysteresis of the first hysteresis comparator of the first hysteresis comparator and the second hysteresis comparator according to the difference between the reference noise and the calculated cumulative noise. The motor control device according to item 1.   The noise comparison unit accumulates the noise levels in the specific frequency band included in the phase current to calculate the accumulated noise, and a noise error defined by a difference between the reference noise and the accumulated noise. The motor control device according to claim 1, wherein the hysteresis bandwidth of the first hysteresis comparator is changed based on the above.   The noise comparison unit accumulates the noise levels in the specific frequency band included in the phase voltage to calculate the accumulated noise, and a noise error defined by a difference between the reference noise and the accumulated noise. The motor control device according to claim 1, wherein the hysteresis bandwidth of the first hysteresis comparator is changed based on the above.   The noise comparison unit corrects the level of the cumulative noise by performing weighting using the frequency weighting characteristic A defined by the IEC 61672 standard, and acquires the corrected cumulative noise level, The motor control device according to claim 3, wherein the hysteresis bandwidth of the first hysteresis comparator is changed based on a noise error defined by a difference from the corrected cumulative noise level.   The noise comparison unit proportionally integrates the noise error to calculate the bandwidth of the hysteresis, and sets the calculated bandwidth of the hysteresis in the first hysteresis comparator. Motor controller. A motor,
An inverter that converts electric power from the outside into electric power supplied to the motor,
A motor control device according to any one of claims 1 to 6,
A motor module including.   An electric power steering apparatus comprising the motor module according to claim 7. A motor,
An inverter that converts electric power from the outside into electric power supplied to the motor,
A motor control device according to any one of claims 1 to 6,
Reducer,
Traction motor unit. Generating a quantized magnetic flux signal based on the difference between the reference magnetic flux and the actual magnetic flux using the first hysteresis comparator;
Generating a quantized torque signal based on the difference between the reference torque and the actual torque using the second hysteresis comparator;
Calculating cumulative noise by accumulating noise levels including switching noise within a specific frequency band,
Changing a bandwidth of at least one hysteresis of the first hysteresis comparator and the second hysteresis comparator according to a difference between the reference noise and the calculated cumulative noise;
Determining an output voltage vector associated with a switching mode of the inverter based on the quantized magnetic flux signal, the quantized torque signal, and the phase angle of the magnetic flux;
Controlling the motor based on the output voltage vector;
A motor control method including the following.   The bandwidth of the hysteresis is changed by changing only the bandwidth of the hysteresis of the first hysteresis comparator among the first hysteresis comparator and the second hysteresis comparator according to the difference between the reference noise and the calculated cumulative noise. The motor control method according to claim 10, further comprising: Changing the bandwidth of the hysteresis is
Calculating the cumulative noise by accumulating the noise level in the specific frequency band, which the phase current includes,
Changing the bandwidth of the hysteresis of the first hysteresis comparator based on a noise error defined by the difference between the reference noise and the accumulated noise;
The motor control method according to claim 10, further comprising: Changing the bandwidth of the hysteresis is
Calculating the accumulated noise by accumulating the noise level in the specific frequency band, which the phase voltage includes,
Changing the bandwidth of the hysteresis of the first hysteresis comparator based on a noise error defined by the difference between the reference noise and the accumulated noise;
The motor control method according to claim 10, further comprising: Changing the bandwidth of the hysteresis is
Correcting the cumulative noise level by weighting using the frequency weighting characteristic A defined by the IEC 61672 standard, and obtaining the corrected cumulative noise level;
Changing a hysteresis bandwidth of the first hysteresis comparator based on a noise error defined by a difference between the reference noise and the corrected cumulative noise level;
The motor control method according to claim 12, further comprising: Changing the bandwidth of the hysteresis is
Calculating the hysteresis bandwidth by proportionally integrating the noise error;
Setting the calculated bandwidth of the hysteresis in the first hysteresis comparator;
The motor control method according to claim 12, further comprising:

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