JP2018148689A - Power converter control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power converter control device for suppressing element deterioration due to resonance generated by imbalance of emitter potential in a configuration in which a plurality of power elements are connected in parallel. Each switch function unit of a power converter is configured by connecting a plurality of power elements in parallel. The potential difference detection circuit 70 detects the emitter potential difference ΔVke, which is the difference between the emitter potentials Vke_1 and Vke_2, for the two controlled elements 16 and 26 connected in parallel. The drive circuit 561 relates to a gate signal that outputs to at least one control target element in a direction in which the emitter potentials of the two control target elements 16 and 26 are brought closer to each other based on the emitter potential difference ΔVke detected by the potential difference detection circuit 70. The gate command value (current Ig_1, Ig_2) is controlled. [Selection diagram] Fig. 2

Description

  The present invention relates to a power converter control device that controls a power converter including a plurality of power elements connected in parallel.

2. Description of the Related Art Conventionally, in a power converter in which a large current flows in a current path, for example, a technology is known in which a switch of each phase upper and lower arms of an AC inverter is configured by connecting a plurality of low-rated power elements in parallel. In such a configuration, it is required to ensure a balance with respect to variations in characteristics of power elements.
For example, the apparatus disclosed in Patent Document 1 relaxes the current imbalance for each power element by adjusting the drive signal based on the result obtained by performing voltage conversion on the sense emitter current of the power element (IGBT) and calculating it.

JP 2013-17092 A

  An inductance component is included in a wiring member such as a bus bar used as a transmission path for a large current in a circuit in which an inverter is configured. When the variation in the emitter potential of the plurality of power elements connected in parallel is large, resonance occurs due to the inductance component of the wiring member and the capacitance component of the power element. As a result, the durability of the power element may be reduced.

However, in the sense voltage detection method using the sense terminal as in the prior art of Patent Document 1, the potential difference between the emitter terminals cannot be detected. Therefore, it is impossible to prevent element degradation due to resonance that occurs due to variations in the emitter potential between a plurality of power elements.
The present invention was created in view of the above points, and an object of the present invention is to suppress the element degradation caused by resonance caused by the unbalance of the emitter potential in a configuration in which a plurality of power elements are connected in parallel. It is to provide a converter control device.

The present invention relates to a control device for a power converter (100) that converts power by switching operations of a plurality of power elements and energizes a load (80).
The one or more switch function units (101-106) that can switch between energization or interruption of the power current in each current path of the power converter have a plurality of power elements (11-16, 21-26) connected in parallel. It is configured. The plurality of power elements are accompanied by diodes (18, 28) that allow energization from the emitter side to the collector side.

This power converter control device includes a potential difference detection circuit (70) and drive circuits (561, 562).
The potential difference detection circuit detects an emitter potential difference (ΔVke) for two control target elements selected from a plurality of power elements connected in parallel.
The drive circuit controls a gate command value related to a gate signal output to at least one control target element in a direction in which the emitter potentials of the two control target elements are brought close to each other based on the emitter potential difference detected by the potential difference detection circuit. .

  The present invention does not detect the sense voltage as in the prior art, but detects the emitter potential difference between two control target elements connected in parallel and gates the emitter potentials of the two control target elements close to each other. Control the command value. As a result, it is possible to suppress element degradation due to resonance that occurs due to an imbalance of the emitter potential between the two controlled elements. Therefore, the reliability of the power converter is improved.

As a method for controlling the gate command value, the drive circuit has one or more positive threshold values (Vref +) and one or more negative threshold values (Vref−) with respect to the emitter potential difference, and the emitter potential difference is any positive threshold value. Or the gate command value may be changed when it falls below any negative threshold.
Alternatively, the drive circuit may have information that preliminarily defines the relationship between the emitter potential difference and the gate command value, and may set the gate command value according to the detected emitter potential difference.

1 is an overall view of an MG drive system to which a power converter control device according to a first embodiment is applied. The schematic diagram of the electric potential difference detection circuit and drive circuit by 1st Embodiment. The figure explaining the recovery current which flows into a power element. The model figure which shows the inductance component which the electric current path of a power converter has. FIG. 5 is a diagram (1) for explaining a mechanism for decreasing element durability in the model diagram of FIG. FIG. 5 is a diagram (2) for explaining a mechanism for decreasing element durability in the model diagram of FIG. 4 is a time chart showing adjustment of gate current by the drive circuit of the first embodiment. The relationship between the magnitude of the power element current and the recovery current, The figure which shows the relationship between vehicle travel mode and a power element electric current. The schematic diagram of the electric potential difference detection circuit and drive circuit by 2nd Embodiment. The schematic diagram of the drive circuit by 3rd Embodiment. The schematic diagram of the drive circuit by 4th Embodiment. The schematic diagram of the drive circuit by 5th Embodiment. The schematic diagram of the drive circuit by 6th Embodiment. FIG. 10 is a schematic diagram of a potential difference detection circuit and a drive circuit according to a seventh embodiment. The figure which shows the relationship between an emitter electric potential difference and a gate command value ((a) gate current, (b) gate voltage, (c) gate resistance).

  Hereinafter, a plurality of embodiments of a power converter control device will be described based on the drawings. In the plurality of embodiments, substantially the same configuration is denoted by the same reference numeral, and description thereof is omitted. The power converter control device of each embodiment includes, for example, a plurality of inverters serving as “power converters” in a system that drives a motor generator (hereinafter “MG”) that is a power source of a hybrid vehicle or an electric vehicle. Controls switching of power elements.

(First embodiment)
A first embodiment will be described with reference to FIGS.
[System configuration]
First, an overall configuration of an MG drive system to which the power converter control device of the first embodiment is applied will be described with reference to FIG. FIG. 1 illustrates an MG drive system 90 including one MG, but the present invention can be similarly applied to an MG drive system including two or more MGs.

Inverter 100 converts the DC power of battery 91, which is a chargeable / dischargeable secondary battery, into three-phase AC power and supplies it to MG 80 as a load. A smoothing capacitor 92 is provided at the input of the inverter 100.
The MG 80 is, for example, a permanent magnet type synchronous three-phase AC motor. The MG 80 has a function as an electric motor that generates torque for driving the driving wheels of the vehicle and a function as a generator that generates electric power using torque transmitted from the driving wheels and the engine of the hybrid vehicle.

A current sensor for detecting a phase current is provided in a current path connected to the two-phase winding among the three-phase windings 81, 82, and 83 of the MG 80. In the example of FIG. 1, current sensors 87 and 88 for detecting phase currents Iv and Iw are provided in current paths connected to the V-phase winding 82 and the W-phase winding 83, respectively, and the remaining U-phase current Iu is estimated based on Kirchhoff's law. In other embodiments, any two-phase current may be detected, and a three-phase current may be detected. Or you may employ | adopt the technique which estimates the other two-phase electric current based on the electric current detection value of one phase.
In the present specification, description regarding detection of the rotor electrical angle is omitted.

Inverter 100 includes a plurality of switch function units 101-106 that can switch energization or interruption of power current in each current path. Specifically, the switch function units 101, 102, and 103 have upper arm switch functions of the U phase, V phase, and W phase, respectively, and the switch function units 104, 105, and 106 have U phase, V phase, and W phase, respectively. It has a switch function for the lower arm of the phase. Each switch function unit 101-106 is composed of “a plurality of power elements connected in parallel”.
In the example of FIG. 1, the switch function unit 101 of the U-phase upper arm is composed of two power elements 11 and 21 connected in parallel. Similarly, the other five switch function units 102-106 are configured by connecting in parallel a power element 12-16 and a power element 22-26 that have the same reference numeral.

The power elements 11-16 and 21-26 are, for example, IGBTs (Insulated Gate Bipolar Transistors), and basically have the same electrical performance. In particular, in the inverter 100 that requires a large current, it is advantageous to standardize the design by using a plurality of standard power elements connected in parallel instead of using a dedicated power element having a large rating.
Each power element 11-16, 21-26 is accompanied by a flywheel diode (or a freewheeling diode) as a diode that allows energization from the emitter side on the low potential side to the collector side on the high potential side. This flywheel diode will be described later with reference to FIG.

A gate signal is commanded from the corresponding drive circuit 51-56 to the gate of the power element 11-16, 21-26 of each switch function unit 101-106. In this specification, the base terminal in the IGBT is also referred to as “gate” in accordance with the gate terminal of the FET.
The emitter terminal is referred to as a “Kelvin emitter terminal” in order to clearly distinguish it from a sense emitter terminal used for current detection in the prior art such as Japanese Patent Application Laid-Open No. 2013-17092. A current that is proportional to the collector-emitter current that is the output current of the power element and smaller than the output current flows through the sense emitter terminal. The voltage across the sense resistor connected to the sense emitter terminal is detected as a sense voltage.
On the other hand, a collector-emitter current flows between the collector terminal and the Kelvin emitter terminal. Here, the potential of the Kelvin emitter terminal detected directly with respect to the reference potential (0 V) is defined as “emitter potential”. In FIG. 1, the illustration of the configuration relating to the detection of the emitter potential is omitted.

The microcomputer 50 includes a CPU, a ROM, an I / O (not shown), a bus line that connects these components, and the like, and software processing by executing a program stored in advance by the CPU, and a dedicated electronic circuit The control by the hardware processing by is executed.
In addition to the information on the phase currents Iv and Iw from the current sensors 87 and 88 shown in FIG. 1, the microcomputer 50 receives information on the electrical angle of the MG 80, a torque command from the host ECU, and the like.

The microcomputer 50 calculates a voltage command value to the inverter 100 by current feedback control using vector control based on these commands and feedback information, and outputs the voltage command value to each drive circuit 51-56. Since the technique of motor control by the microcomputer 50 is a well-known technique, detailed description is omitted.
In the present embodiment, a group of control circuits including the microcomputer 50 and the drive circuits 51 to 56 constitute a “power converter control device”.

  Next, referring to FIG. 2, a configuration for detecting an emitter potential, which is not shown in FIG. 1, will be described. In FIG. 2, among the six switch function units 101 to 106 of the inverter 100, the symbol of the switch function unit 106 of the W-phase lower arm configured by the power elements 16 and 26 is used as a representative. The configuration shown in FIG. 2 is the same for the other switch function units 101-105.

  By the way, in an actual circuit mounting, the current path of the inverter 100 is configured by a power card, a bus bar, a substrate pattern, and the like. For example, as indicated by a two-dot chain circle, a portion directly connected to the collector terminal and the emitter terminal of the power elements 16 and 26 is constituted by a power card. As indicated by the two-dot chain rhombus frame, the place connecting the two power cards is constituted by a bus bar. As indicated by a two-dot chain line square frame, the wiring from the drive circuit 561 is configured by a substrate pattern.

  The potential difference detection circuit 70 is configured by a differential amplifier. The potential difference detection circuit 70 receives the emitter potentials Vke_1 and Vke_2 of the two power elements 16 and 26, and outputs an emitter potential difference ΔVke that is the difference between them. The emitter potential difference ΔVke takes a positive or negative value depending on the magnitudes of the emitter potentials Vke_1 and Vke_2.

The comparator 77 of the drive circuit 561 compares the emitter potential difference ΔVke with the positive threshold value Vref +, and the comparator 78 compares the emitter potential difference ΔVke with the negative threshold value Vref−.
The positive threshold value Vref + and the negative threshold value Vref− are positive and negative values having the same absolute value. When the emitter potential difference ΔVke is positive, the comparison with the positive threshold value Vref + by the comparator 77 is significant, and when the emitter potential difference ΔVke is negative, the comparison with the negative threshold value Vref− by the comparator 78 is significant. Other description regarding the drive circuit 561 will be described later.

Further, the power elements 16 and 26 are accompanied by flywheel diodes 18 and 28 whose reference numerals are omitted in FIG. The flywheel diodes 18 and 28 allow energization from the emitter side to the collector side. As the flywheel diodes 18 and 28, general rectifier diodes or fast rectifier diodes (FRD) are used.
Here, the recovery current flowing through the flywheel diodes 18 and 28 will be described with reference to FIG. The vertical axis in FIG. 3 represents the power element current Ice.

  During the return of the flywheel diodes 18 and 28 of the power elements 16 and 26 of the W-phase lower arm, the power elements 13 and 23 are turned on from the OFF state at time t0. Then, the collector side becomes a higher potential than the emitter side, and the voltage polarity applied to the flywheel diodes 18 and 28 changes to a reverse bias. However, there is a period of energization in the reverse direction from time t0 to time t1. This period is referred to as a recovery period Trc (or reverse recovery period), and the current in the reverse flow direction flowing through the flywheel diodes 18 and 28 at this time is referred to as a recovery current (or reverse recovery current).

  That is, during the period from time t0 to time t1, a current obtained by adding the recovery current to the current flowing through the main bodies of the power elements 13 and 23 flows, and thus a peak value appears in the power element current Ice. From time t1 when the recovery is completed to time t2 when the power elements 13 and 23 are turned off, only the current flowing through the main body of the power elements 13 and 23 becomes the power element current Ice.

  By the way, the two power elements 16 and 26 connected in parallel and the accompanying flywheel diodes 18 and 28 basically have the same electrical performance. However, the on-state current may be unbalanced due to variations in component characteristics and board mounting. For example, the power element current Ice_1 indicated by a solid line is larger than the power element current Ice_2 indicated by a broken line, and a current difference ΔIce is generated. If this state continues, the power element through which a larger current flows may be rapidly deteriorated.

Next, problems to be solved by the present embodiment will be described with reference to FIGS.
FIG. 4 schematically shows the inductance component of the current path of each part of the circuit for the switch function part of the upper and lower arms of one phase. As reference numerals in FIG. 4, the power elements 13, 23, 16, and 26 and the drive circuits 53 and 56 of the switch function units 103 and 106 of the W-phase upper and lower arms in FIG. 1 are illustrated. The collectors of the power elements 13 and 23 in the upper arm are connected to the high potential line P, and the emitters of the power elements 16 and 26 in the lower arm are connected to the low potential line N.
As in FIG. 2, in FIG. 4, a two-dot chain line round frame, rhombus frame, and square frame indicate portions constituted by a power card, a bus bar, and a substrate pattern, respectively. These power cards, bus bars, and substrate patterns have inductance components. This inductance component may cause a resonance phenomenon.

Next, referring to FIG. 5 and FIG. 6 for the mechanism of element durability reduction in an inverter in which two power elements are connected in parallel. It is considered that the durability of the device is reduced by the following mechanism due to current imbalance.
Solid arrows at timing 1 in FIG. 5 and timings 3 and 5 in FIG. 6 indicate currents flowing through the power element 16 or 26. The block arrows at timing 2 in FIG. 5 and timing 4 in FIG. 6 indicate the current flowing through the low potential side bus bar due to the potential difference generated between the emitters of the power elements 16 and 26.

(1) When the upper arm power elements 13 and 23 are turned on, a recovery current flows through the flywheel diodes 18 and 28 associated with the lower arm power elements 16 and 26.
(2) Assuming that the second flywheel diode 28 is earlier than the first flywheel diode 18 due to variations in characteristics of the flywheel diodes 18, 28, the recovery timing at this time is assumed. The current concentrates on the second flywheel diode 28 that is early (timing 1 in FIG. 5).
(3) Upon completion of the recovery in (2), the emitter potential Vke_2 of the second element 26 becomes lower than the emitter potential Vke_1 of the first element 16 (Vke_1> Vke_2), and a potential difference ΔVke occurs between the emitters (FIG. 5 timing 2).

(4) The current concentrates on the first flywheel diode 18 whose recovery timing is late (timing 3 in FIG. 6).
(5) When the recovery in (4) is completed, the emitter potential Vke_1 of the first element 16 becomes lower than the emitter potential Vke_2 of the second element 26 (Vke_1 <Vke_2), and a potential difference ΔVke is generated between the emitters (FIG. 6 timing 4).
(6) Due to the generation of the emitter potential difference ΔVke, a current flows through the bus bar and the emitter of the substrate. Then, resonance occurs due to the capacitance component (C) due to the electric charge accumulated in the power element and the inductance component (L) of the power card, bus bar, substrate, etc. constituting the current path of the inverter. This resonance leads to a decrease in durability of the power element (timing 5 in FIG. 6).

With respect to this problem, an object of the present embodiment is to suppress element degradation due to resonance caused by imbalance of recovery currents between power elements connected in parallel.
As a solution, the potential difference detection circuit 70 detects the emitter potential difference ΔVke between the two power elements 16 and 26 connected in parallel at the timing when the recovery current flows. When the comparators 77 and 78 detect that the emitter potential difference ΔVke exceeds the positive threshold value Vref + or falls below the negative threshold value Vref−, the drive circuit 561 always balances the two power elements 16 and 26. Control to adjust.
Therefore, the two power elements 16 and 26 connected in parallel are appropriately referred to as “control target elements 16 and 26”. When the two control target elements 16 and 26 are described separately, they are referred to as “first target element 16” and “second target element 26”.

  In a configuration in which three or more power elements are connected in parallel as in the seventh embodiment described later, two power elements selected from three or more are controlled objects in one balance adjustment process. It becomes an element. That is, in one balance adjustment process, specific two of the three or more power elements are treated as “control target elements”, and the other power elements are excluded from the control target. On the other hand, in a configuration in which two power elements are connected in parallel, there is only one combination for selecting two of the two elements, so the “control target element” is always fixed.

Next, the configuration and operation of the drive circuit 561 will be described. Here, the reference numeral “56” of the drive circuit corresponding to the switch function unit 106 in FIG. 1 is set to “561” in the first embodiment shown in FIG. 2, and the second embodiment shown in FIG. The drive circuit sign is “562”.
The drive circuit 561 controls the “gate command value” related to the gate signal output to the control target elements 16 and 26 so that the emitter potentials Vke_1 and Vke_2 of the control target elements 16 and 26 are close to each other.
By the way, the most preferable process in the balance adjustment is to make the emitter potentials Vke_1 and Vke_2 of the control target elements 16 and 26 equal. However, there is a limit to realizing strict equalization due to restrictions on the number of parts and the amount of control calculation. Therefore, the drive circuit 561 may control the gate command value so that at least the emitter potentials Vke_1 and Vke_2 of the control target elements 16 and 26 are close to each other before the balance adjustment.

The constant current drive type drive circuit 561 according to the first embodiment includes comparators 77 and 78 and a current adjustment circuit 601. The comparator 77 outputs an unbalance signal Sub to the current adjustment circuit 601 when the positive emitter potential difference ΔVke exceeds the positive threshold value Vref +. The comparator 78 outputs the unbalance signal Sub to the current adjustment circuit 601 when the negative emitter potential difference ΔVke falls below the negative threshold value Vref−.
In addition, the current adjustment circuit 601 acquires the emitter potential difference ΔVke, and grasps which of the emitter potentials Vke_1 and Vke_2 of the control target elements 16 and 26 is greater. The acquired emitter potential difference ΔVke may be used for abnormality determination in addition to balance adjustment.

  The current adjustment circuit 601 supplies the gate current Ig_1 of the first path 611 connected to the gate G1 of the first target element 16 via the gate resistor 17 and the gate G2 of the second target element 26 via the gate resistance 27. The gate current Ig_2 of the connected second path 612 can be variably adjusted. When the unbalance signal Sub is input from the comparators 77 and 78, the current adjustment circuit 601 changes the gate currents Ig_1 and Ig_2 in a direction in which the output currents of the control target elements 16 and 26 are brought closer to each other. Thus, the drive circuit 561 of the first embodiment controls the gate currents Ig_1 and Ig_2 as gate command values.

  Next, a specific example of balance adjustment processing by the drive circuit 561 will be described with reference to the time chart of FIG. Here, in general, the “n-th” and “(n + 1) -th” switching cycles (“SW cycle” in the figure) are referred to as “current” and “next” switching cycles for convenience. FIG. 7 shows changes in the gate currents Ig_1 and Ig_2, the power element currents Ice_1 and Ice_2, and the emitter potential difference ΔVke in the current and next switching cycles. Here, a value obtained by subtracting the emitter potential Vke_1 of the control target element 16 from the emitter potential Vke_2 of the second target element 26 is defined as an emitter potential difference ΔVke (= Vke_2−Vke_1).

  The gate currents Ig_1 and Ig_2 flow in a direction to turn on the control target elements 16 and 26 from time t10 to time t11 of the current switching cycle and from time t20 to time t21 of the next switching cycle. Further, the gate currents Ig_1 and Ig_2 flow in a direction to turn off the control target elements 16 and 26 from time t12 to time t13 of the current switching cycle and from time t22 to time t23 of the next switching cycle.

  The currents Ig_1 and Ig_2 flowing at the turn-on of the switching period are the same. At this time, the emitter potential difference ΔVke is positive. That is, the emitter potential Vke_2 of the second target element 26 is higher than the emitter potential Vke_1 of the first target element 16, and the element current Ice_2 of the second target element 26 is smaller than the element current Ice_1 of the first target element 16. In addition, since the emitter potential difference ΔVke exceeds the positive threshold value Vref +, the comparator 77 outputs the unbalance signal Sub to the current adjustment circuit 601.

Here, it is difficult in terms of response speed to switch the gate command value within the same pulse period from time t10 to time t11 when the unbalance signal Sub is output. Therefore, the drive circuit 561 has a gate current value Ig_1, which is a gate command value, when the controlled elements 16 and 26 are turned on in the “switching cycle after the next” of the switching cycle in which the unbalance signal Sub is output by the comparators 77 and 78. Change Ig_2.
The example shown in FIG. 7 realizes quick balance adjustment by changing the gate command value in the next switching cycle. However, the drive circuit 561 may change the gate command value to a switching cycle several cycles after the switching cycle in which the unbalance signal Sub is output, for example.

The current adjustment circuit 601 relatively increases the gate current Ig_2 that is passed through the second target element 26 having a relatively high emitter potential Vke_2. In other words, the current adjustment circuit 601 relatively reduces the gate current Ig_1 that is passed through the first target element 16 having a relatively low emitter potential Vke_1.
As a result, the rising slope of the element current Ice_2 of the second target element 26 becomes steep as indicated by a two-dot chain line. That is, the switching speed when the second target element 26 is turned on becomes relatively high. On the other hand, the rising slope of the element current Ice_1 of the first target element 16 becomes gentle as indicated by the alternate long and short dash line. That is, the switching speed when the first target element 16 is turned on is relatively slow. Further, the emitter potential difference ΔVke approaches zero. The thin solid line and the thin broken line in the next switching cycle indicate the element currents Ice_1 and Ice_2 in the current switching cycle as a reference.

  Note that the gate currents Ig_1 and Ig_2 of the two control target elements 16 and 26 are not both changed, and only the gate current of one of the control target elements is changed, and the gate current of the other control target element is changed. May be maintained. For example, the gate current Ig_1 of the first target element 16 only needs to be “relatively” decreased by greatly changing only the gate current Ig_2 of the second target element 26.

(effect)
In the configuration in which a plurality of power elements are connected in parallel, the conventional technique of Patent Document 1 that adjusts the drive signal based on the result of voltage conversion and calculation of the sense emitter current of the power element detects the potential difference between the Kelvin emitter terminals. I can't. Therefore, it is impossible to prevent element degradation due to resonance that occurs due to variations in the emitter potential between a plurality of power elements. On the other hand, in this embodiment, the emitter potential difference ΔVke between the two control target elements 16 and 26 connected in parallel is detected, and the gate command value is controlled in the direction in which the emitter potential Vke is brought closer to each other. As a result, it is possible to suppress deterioration of the control target elements 16 and 26 due to resonance generated by imbalance of the emitter potential Vke.

Further, the timing when the drive circuit 561 changes the gate command value is when the switching cycle after the next switching cycle in which it is detected that the emitter potential difference ΔVke exceeds the positive threshold value Vref + or falls below the negative threshold value Vref− is turned on. is there. Thereby, it can be set as the structure which can be controlled realistically from the point of response speed.
Furthermore, in the first embodiment, the emitter potential difference ΔVke is compared with the positive / negative thresholds Vref + and vref−, and the gate command value is switched stepwise depending on the magnitude relationship. By performing the process of switching a finite number of command values, it is possible to reduce the load of control calculation.

Next, supplementary items for implementing the first embodiment will be listed.
(Change of gate command value based on detection result of emitter potential difference)
In addition to changing the gate command value every time based on the emitter potential difference ΔVke for each detection period of the emitter potential Vke, the detection result of the emitter potential difference ΔVke for a plurality of times is stored, and the gate command value is stored based on the information for a plurality of times. The value may be changed. For example, it is conceivable to use information such as an average value or a maximum value of the emitter potential difference ΔVke multiple times. Based on multiple times of information, it is possible to reduce the amount of control calculation and realize efficient balance adjustment.

(Detection timing of emitter potential difference)
As shown in FIG. 8A, the recovery current and the current difference ΔIrc after the completion of recovery are larger at low current than at high current. Further, as shown in FIG. 6B, there is a proportional relationship between the power element current Ice and the emitter potential difference ΔVke. Therefore, it is preferable to detect the emitter potentials Vke_1 and Vke_2 at the timing when the recovery current flows when the output currents Ice_1 and Ice_2 of the power elements 16 and 26 are relatively small. The determination result may be maintained for a predetermined period after the emitter potential difference ΔVke is detected when the power element current Ice is low, regardless of the magnitude of the element current Ice.

Further, assuming a system that drives MG 80, which is a power source of an electric vehicle or a hybrid vehicle, FIG. 9 shows a relationship between the vehicle travel mode and the power element current Ice. From the lowest current, the order is “vehicle start”, “normal driving”, “acceleration, climbing”, and “abnormal”.
Among these, the time suitable for detecting the emitter potential difference ΔVke is when the power element current Ice is lower than that during acceleration and climbing.

  Therefore, the potential difference detection circuit 70 acquires information on the output current Ice of the power element energized to the MG 80, and detects the emitter potential difference ΔVke when the output current Ice of the power element is lower than the current value energized when the vehicle is accelerated. To do. Specifically, the potential difference detection circuit 70 detects the emitter potential difference ΔVke when the output current Ice of the power element is in a current value range that is energized when the vehicle is started or during normal running. Thereby, the emitter potential difference ΔVke caused by the recovery current can be detected more effectively.

(Threshold setting)
The positive / negative threshold values Vref + and Vref− of the emitter potential difference ΔVke are set so that the absolute value becomes a potential difference equal to or less than the breakdown point of the power element. Further, the positive / negative thresholds Vref + and Vref− may be adjusted according to environmental conditions such as voltage, current, and temperature.
Furthermore, a plurality of positive threshold values and a plurality of negative threshold values may be set and switched according to conditions. In this case, it is preferable that the plurality of positive thresholds and the plurality of negative thresholds are a plurality of sets of values having the same absolute value. For example, a first level threshold value having a relatively large absolute value related to element deterioration and a second level threshold value having a relatively small absolute value related to loss deterioration are set. The determination based on the first level threshold may be performed all the time, and the determination based on the second level threshold may be performed only during normal driving.

Here, “during normal travel” is defined based on the following rules, for example.
(1) JC08 mode regulations stipulated by Japanese laws and regulations (2) WLTP regulations, an internationally defined exhaust gas test method (3) Power elements, motors, power element control boards, etc. The related parts are operating normally without failure. “Operating normally without failure” means that a large current is not flowing due to a failure such as an overcurrent of a power element, a short circuit, or a short circuit of a motor.

Next, variations relating to the configuration of the drive circuit will be described as second to sixth embodiments.
(Second Embodiment)
The drive circuit 562 of the second embodiment shown in FIG. 10 is a constant voltage drive type drive circuit compared to the constant current drive type drive circuit 561 shown in FIG.
The drive circuit 562 includes drive power sources 621 and 622 and a voltage adjustment circuit 602. The drive power supplies 621 and 622 are connected to the gates G1 and G2 of the control target elements 16 and 26 via the gate resistors 17 and 27, respectively. Gate voltages Vg_1 and Vg_2, which are voltages of the drive power sources 621 and 622, are applied to the gates G1 and G2, respectively.

The drive circuit 562 adjusts the gate voltages Vg_1 and Vg_2 as “gate command values”. That is, when the unbalance signal Sub is input from the comparators 77 and 78, the voltage adjustment circuit 602 variably adjusts the gate voltages Vg_1 and Vg_2 in a direction in which the output currents of the controlled elements 16 and 26 are close to each other.
According to the example of FIG. 7, the voltage adjustment circuit 602 turns on the second target element 26 by relatively increasing the gate voltage Vg_2 applied to the second target element 26 having a relatively high emitter potential Vke_2. When switching speed is relatively high. In other words, the voltage adjustment circuit 602 relatively reduces the switching speed at turn-on of the first target element 16 by relatively reducing the gate voltage Vg_1 applied to the first target element 16 having a relatively low emitter potential Vke_1. Slow down. The effect by this is the same as that of 1st Embodiment.

(Third and fourth embodiments)
FIGS. 11 and 12 show third and fourth embodiments relating to the detailed configuration of the drive circuit. In the third and fourth embodiments, the gate resistance connected to the controlled elements 16 and 26 is variable in the configurations of the constant current driving method and the constant voltage driving method, respectively.
The constant current drive type drive circuit of the third embodiment is obtained by replacing the portion of the broken line frame XA shown in FIG. 11 with the portion of the broken line frame XA of the drive circuit 561 shown in FIG.
The gate signal generated by the drive circuit is output to the gate G1 of the first target element 16 and the gate G2 of the second target element 26. The same applies to FIGS. 12 to 14 below.

In the current adjustment circuit 603, a plurality of sub-circuits 651 and 652 connected to the gate G1 are provided in parallel, and a plurality of sub-circuits 653 and 654 connected to the gate G2 are provided in parallel. Each of the sub-circuits 651, 652, 653, and 654 includes a reference voltage unit 66, a comparator 67, an FET 68, and a gate resistor Rg_n for current adjustment.
Here, “n” of the gate resistance Rg_n means a number of 1 to 4 that is the same as the number of the third digit of the sub circuit. The resistance value of the gate resistance Rg_1 of the sub circuit 651 is different from the resistance value of Rg_2 of the sub circuit 652. The resistance value of the gate resistance Rg_3 of the sub circuit 653 and the resistance value of Rg_4 of the sub circuit 654 are different from each other.

  A terminal voltage Vom is applied to the high potential side of the reference voltage unit 66 and one end of the gate resistor Rg_n. The − input terminal of the comparator 67 is connected between the gate resistance Rg_n and the drain of the FET 68, and the + input terminal is connected to the low potential side of the reference voltage unit 66. The output terminal of the comparator 67 is connected to the gate of the FET 68. When the FET 68 is turned on, the current flowing through the gates G1 and G2 is adjusted by the resistance value of the gate resistance Rg_n.

When the unbalance signal Sub is input, the current adjustment circuit 603 switches the sub circuit used for generating the gate signal in the direction in which the emitter potentials Vke of the control target elements 16 and 26 are close to each other, whereby the gate of the signal output path The resistor Rg_n is switched. This operation by the current adjustment circuit 603 is simply referred to as “switching the gate resistance”.
The voltage adjustment circuit 603 relatively reduces the gate resistance Rg_n connected to the control target element having a relatively high emitter potential Vke, and sets the gate resistance Rg_n connected to the control target element having a relatively low emitter potential Vke. Make it relatively large.

The constant voltage drive type drive circuit of the fourth embodiment is obtained by replacing the portion of the broken line frame YA shown in FIG. 12 with the portion of the broken line frame YA of the drive circuit 562 shown in FIG.
A plurality of gate resistors Rg_1 and Rg_2 having different resistance values are connected in parallel between the voltage adjustment circuit 604 and the gate G1. A plurality of gate resistors Rg_3 and Rg_4 having different resistance values are connected in parallel between the voltage adjustment circuit 604 and the gate G2.

When the unbalance signal Sub is input, the voltage adjustment circuit 604 switches the signal output path to which the gate resistors Rg_1 to Rg_4 are connected in a direction in which the output currents Ice of the control target elements 16 and 26 are close to each other. This operation by the voltage adjustment circuit 604 is simply referred to as “switching the gate resistance”.
A specific method for changing the gate resistance Rg_n is the same as in the third embodiment.

(Fifth and sixth embodiments)
FIGS. 13 and 14 show fifth and sixth embodiments relating to the detailed configuration of the drive circuit. In the fifth and sixth embodiments, a trimming resistor is used as a gate resistor connected to the control target elements 16 and 26 in the configurations of constant current drive and constant voltage drive, respectively. The fifth and sixth embodiments are suitable for initial adjustment at the manufacturing stage.
The constant current drive type drive circuit of the fifth embodiment is obtained by replacing the portion of the broken line frame XA shown in FIG. 13 with the portion of the broken line frame XA of the drive circuit 561 shown in FIG.

The current adjustment circuit 605 is provided with a sub circuit 655 connected to the gate G1 and a plurality of circuits 656 connected to the gate G2. Each of the sub-circuits 655 and 656 includes a reference voltage unit 66, a comparator 67, an FET 68, and a trimming resistor Rg_5 or Rg_6 for current adjustment.
The sub-circuits 655 and 656 are the same as in the third embodiment except that the gate resistance is constituted by a trimming resistor. When the FET 68 is turned on, the currents flowing through the gates G1 and G2 are variably adjusted by the trimming resistors Rg_5 and Rg_6.

For example, in the inspection at the manufacturing stage, when the absolute value of the emitter potential difference ΔVke of the control target elements 16 and 26 is larger than the absolute values of the positive and negative thresholds Vref + and Vref−, the sub-circuits 655, The trimming resistors Rg_5 and Rg_6 656 are adjusted.
As a result, in the fifth embodiment, the product can be shipped in a state where the balance of the output current at the time of turn-on of the plurality of power elements connected in parallel is good, and the reliability of the product can be improved.

The constant voltage drive type drive circuit of the sixth embodiment is obtained by replacing the portion of the broken line frame YA shown in FIG. 14 with the portion of the broken line frame YA of the drive circuit 562 shown in FIG.
A trimming resistor Rg_5 whose resistance value can be variably adjusted is connected between the voltage adjustment circuit 606 and the gate G1. A trimming resistor Rg_6 is connected between the voltage adjustment circuit 606 and the gate G2.
The effect of 6th Embodiment is the same as that of 5th Embodiment.

(Seventh embodiment)
Next, FIG. 15 shows an example of a drive circuit used in a power converter in which three power elements 16, 26, and 36 are connected in parallel to form one switch function unit as a seventh embodiment.
The signs and symbols of the third power element 36 such as the flywheel diode 38, the gate G3, the Kelvin emitter terminal KE3, and the emitter potential Vke_3 are the same as those of the power elements 16 and 26.
In addition, description of the code | symbol used only for 7th Embodiment is abbreviate | omitted to the reference code in the parenthesis in a claim.

The constant current drive type drive circuit 567 according to the first embodiment includes a current adjustment circuit 607 and comparators 771, 781, 772, 782, 773 and 783. In addition to the current adjustment circuit 601 of the first embodiment, the current adjustment circuit 607 can variably adjust the gate current Ig_3 of the third path 613 connected to the gate G3 of the power element 36 via the gate resistor 37.
It should be noted that a constant voltage driving method according to the second embodiment may be adopted as the driving circuit method.

In this embodiment, three sets of “control target elements” in which two of the three power elements are selected are generated as follows, and a balance adjustment process is performed on each of the three sets.
First set of controlled elements: power elements 16, 26
Second set of controlled elements: power elements 16, 36
Third set of controlled elements: power elements 26 and 36

The potential difference detection circuits 701, 702, and 703 detect the emitter potential differences ΔVke _1-2 , ΔVke _1-3 , and ΔVke _2-3 of the first set, the second set, and the third set of control target elements, respectively.
The comparators 771, 772, and 773 are respectively used when the emitter potential differences ΔVke _1-2 , ΔVke _1-3 , and ΔVke _{2-3 of} the first set, the second set, and the third set of control target elements exceed the positive threshold value Vref +. The unbalance signal Sub is output.
In the comparators 781, 782, and 783, the emitter potential differences ΔVke _1-2 , ΔVke _1-3 , and ΔVke _{2-3 of} the first set, the second set, and the third set of control target elements are less than the negative threshold value Vref−, respectively. The unbalance signal Sub is output.
The current adjustment circuit 607 changes the gate command value when the unbalance signal Sub is output for each set of control target elements.

The balance adjustment process for each set of control target elements may be performed by rotating in turn the same number of times. Alternatively, the number of processing times may be differentiated according to the past unbalance occurrence frequency or the like.
As described above, the balance adjustment in each of the above embodiments can be similarly performed on a power converter in which one switch function unit is configured by connecting three or more power elements in parallel.

(Eighth embodiment)
An eighth embodiment will be described with reference to FIG.
In the first to seventh embodiments, the emitter potential difference ΔVke detected by the potential difference detection circuit 70 is compared with the positive / negative thresholds Vref + and Vref−, and the emitter potential difference ΔVke exceeds the positive threshold Vref + or falls below the negative threshold Vref−. The gate command value is switched step by step.
On the other hand, the drive circuit of the eighth embodiment stores in advance information such as mathematical formulas and maps that define the relationship between the emitter potential difference ΔVke and the gate command value, and the emitter potential difference ΔVke acquired from the potential difference detection circuit 70 is stored in the drive circuit. Set the gate command value accordingly.

  FIGS. 16A, 16B, and 16C show the relationship between the emitter potential difference ΔVke and the gate current Ig that is the gate command value, the gate voltage Vg, and the gate resistance Rg. On the horizontal axis of FIG. 16, the emitter potential difference ΔVke is defined as a value obtained by subtracting the emitter potential Vke_1 of the first target element 16 from the emitter potential Vke_2 of the second target element 26 (Vke_2−Vke_1). That is, when the emitter potential difference ΔVke is positive, it means that the emitter potential Vke_2 of the second target element 26 is higher than the emitter potential Vke_1 of the first target element 16. When the emitter potential difference ΔVke is 0, the gate command values for the two controlled elements 16 and 26 are equal.

When the gate command value shown in FIG. 16A is the gate current Ig, the drive circuit decreases the gate current Ig_1 supplied to the first target element 16 as the emitter potential difference ΔVke increases, and the second target element 26 Is set so as to increase the gate current Ig_2 energized.
When the gate command value shown in FIG. 16B is the gate voltage Vg, the drive circuit decreases the gate voltage Vg_1 applied to the first target element 16 as the emitter potential difference ΔVke increases, and the second target element 26 Is set to increase the gate voltage Vg_2 applied to.
When the gate command value shown in FIG. 16C is the gate resistance Rg, the drive circuit increases the gate resistance Rg_1 connected to the first target element 16 as the emitter potential difference ΔVke increases, and the second target element 26 Is set to decrease the gate resistance Rg_2 connected to.
Thus, in the eighth embodiment, the gate command value can be set finely according to the emitter potential difference ΔVke.

(Other embodiments)
(A) The plurality of power elements connected in parallel may be SiC elements, GaN elements, or the like in addition to the IGBTs exemplified in the above embodiment. The diode associated with the power element is not limited to the flywheel diode, and may be any diode that allows a recovery current to flow, such as a Schottky barrier diode or a MOSFET built-in diode.
(B) The power converter to which the present invention is applied is not limited to the inverter, and is a boost converter that is connected between the battery and the inverter and boosts the DC voltage of the battery, or between the low voltage side and the high voltage side. A step-up / step-down converter or the like that can step up and down in both directions may be used. Moreover, in the case of an AC inverter, the present invention is not limited to three phases and can be similarly applied to a multiphase AC inverter having four or more phases.

(C) The load of the power converter is not limited to a motor generator used as a power source of a hybrid vehicle or an electric vehicle, but may be a motor for an auxiliary machine of a vehicle, a train other than a vehicle, an elevator, a general machine, etc. Alternatively, a load other than the motor may be used. If the power converter uses a plurality of power elements connected in parallel in order to cope with a large current, the effect of the present invention is similarly exhibited.
As mentioned above, this invention is not limited to the said embodiment at all, In the range which does not deviate from the meaning of invention, it can implement with a various form.

100 .. Inverter (power converter),
101-106 ... switch function part,
11-16, 21-26... Power element,
18, 28... Flywheel diode,
561, 562... Drive circuit,
70: Potential difference detection circuit,
80: Motor generator (load).

Claims (14)

A control device for a power converter (100) that converts power by switching operation of a plurality of power elements and energizes a load (80),
One or more switch function units (101-106) capable of switching energization or interruption of power current in each current path of the power converter are diodes (18, 28) that allow energization from the emitter side to the collector side. A plurality of power elements (11-16, 21-26) accompanied by are connected in parallel,
A potential difference detection circuit (70) for detecting an emitter potential difference (ΔVke) that is a difference in emitter potential for two control target elements selected from the plurality of power elements connected in parallel;
Based on the emitter potential difference detected by the potential difference detection circuit, a gate command value related to a gate signal output to at least one of the control target elements is controlled in a direction in which the emitter potentials of the two control target elements are close to each other. A drive circuit (561, 562);
A power converter control device comprising: The drive circuit has one or more positive thresholds (Vref +) and one or more negative thresholds (Vref−) for the emitter potential difference,
2. The power converter according to claim 1, wherein when the emitter potential difference exceeds any of the positive threshold values or falls below any of the negative threshold values, the gate command value for at least one of the control target elements is changed. Control device.   The power converter control device according to claim 2, wherein the one or more positive thresholds and the one or more negative thresholds are one or more sets of positive and negative values having the same absolute value.   The drive circuit, when the control target element is turned on in the switching period after the next switching period in which it is detected that the emitter potential difference exceeds any positive threshold or falls below any negative threshold, The power converter control device according to claim 2 or 3, wherein the gate command value is changed. The drive circuit, when the emitter potential difference exceeds any of the positive threshold, or below any of the negative threshold,
5. The power converter control device according to claim 2, wherein the gate command value is changed so as to relatively increase a switching speed at turn-on of the control target element having a relatively high emitter potential. 6. The gate command value is a gate current,
The drive circuit, when the emitter potential difference exceeds any of the positive threshold, or below any of the negative threshold,
The power converter control device according to claim 5, wherein the gate current is changed so as to relatively increase a gate current supplied to the control target element having a relatively high emitter potential. The gate command value is a gate voltage,
The drive circuit, when the emitter potential difference exceeds any of the positive threshold, or below any of the negative threshold,
The power converter control device according to claim 5, wherein the gate voltage is changed so as to relatively increase the gate voltage applied to the control target element having a relatively high emitter potential. The gate command value is a gate resistance,
The drive circuit, when the emitter potential difference exceeds any of the positive threshold, or below any of the negative threshold,
The power converter control device according to claim 5, wherein the gate resistance is changed so that the gate resistance connected to the control target element having a relatively high emitter potential is relatively small. The drive circuit is
2. The information according to claim 1, wherein the gate command value is set according to the detected emitter potential difference, having information that preliminarily defines a relationship between the emitter potential difference and the gate command value for the device to be controlled. Power converter control device.   The power converter control device according to claim 9, wherein the gate command value is a gate current passed through the control target element.   The power converter control device according to claim 9, wherein the gate command value is a gate voltage applied to the control target element.   The power converter control device according to claim 9, wherein the gate command value is a gate resistance connected to the control target element. As the load, used to control a power converter that energizes a motor generator that is a power source of a vehicle,
The potential difference detection circuit includes:
The information on the output current of the power element energized to the motor generator is acquired, and the emitter potential difference is detected when the output current of the power element is lower than a current value energized when the vehicle is accelerated. The power converter control device according to any one of the above. The potential difference detection circuit includes:
14. The power converter control device according to claim 13, wherein the emitter potential difference is detected when an output current of the power element is in a range of a current value that is energized when the vehicle is started or during normal running.

Images