JP2017034760A - Motor control device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To restrict the influence of interference voltage in a harmonic-wave coordinate system, by simple processing.SOLUTION: A control device 30 for controlling an SR motor 20 by adjusting motor current, comprises: a fundamental wave command voltage generating part 42a that sets a command value for fundamental wave voltage such that d-axis current and q-axis current have command values; a harmonic wave command voltage generating part 42b that sets a command voltage, which is a command value for harmonic wave voltage, such that dh-axis current and qh-axis current have command values; an estimation part that calculates an estimated voltage for harmonic wave voltage that does not include interference voltage caused between harmonic wave components, on the basis of a reverse model of the SR motor 20, which uses motor current as input and motor voltages as output, and on the basis of the actual value of a harmonic wave component, and that estimates interference voltage on the basis of the difference between the estimated voltage and the command voltage; a correction part that corrects the command voltage on the basis of the interference voltage; and an addition part 49 that converts the corrected command voltage into a fundamental-wave coordinate system and adds the result to the command value of the fundamental wave voltage.SELECTED DRAWING: Figure 3

Description

  The present invention relates to a motor control device that controls a motor by adjusting a motor current flowing in the motor.

  Mainly due to the distortion of the magnetic flux, a harmonic component having a frequency n times the fundamental wave component (n is a natural number of 2 or more) is generated in the motor current. Due to this harmonic, adverse effects such as torque ripple occur. Therefore, a motor control device is known in which the d-axis current and q-axis current, which are fundamental wave components, and the dh-axis current and qh-axis current, which are harmonic components, are independently controlled.

  Here, the dh-axis current and the dq-axis current, which are harmonic components, interfere with each other to generate an interference voltage, and the responsiveness of the harmonic component is deteriorated by the interference voltage. In order to suppress the deterioration of responsiveness due to the interference voltage, the interference voltage is estimated in a feed-forward manner using the command values of the dh-axis current and the dq-axis current and various parameters of the motor, and the estimated value is used. Thus, a method for decoupling the dh axis and the dq axis has been proposed (Patent Document 1). The various motor parameters include a motor resistance value, an inductance value, an induced voltage, and the like.

Japanese Patent No. 3928575

  Here, in order to estimate the interference voltage on the harmonic coordinates, it is necessary to perform a complicated calculation, and the practicality is low. Further, when an error occurs in the resistance value, inductance value, or induced voltage of the motor, the estimated value of the interference voltage also includes an error, and the control responsiveness deteriorates.

  The present invention has been made in view of the above problems, and has as its main object to suppress the influence of interference voltage in the harmonic coordinate system by simple processing.

  The first configuration is a motor control device (30) that controls the motor by adjusting the motor current flowing through the motor (20), and includes a d-axis current and q that are fundamental wave components of the motor current. A fundamental wave control unit (42a) for setting a command value of a fundamental wave voltage applied to the motor so that the shaft current becomes a predetermined command value; and n times (n is 2) the fundamental wave component of the motor current. A command that is a command value of a harmonic voltage applied to the motor so that the dh-axis current and the qh-axis current, which are harmonic components of the motor current having a frequency of the above natural number), have predetermined command values. A harmonic control unit (42b) for setting a voltage; an inverse model (53) of the motor having the motor current as an input and a motor voltage applied to the motor as an output; and a harmonic of the motor current Based on the actual value of the component Accordingly, an estimation unit that calculates an estimated voltage of the harmonic voltage not including an interference voltage generated between the harmonic components and estimates the interference voltage based on a deviation between the estimated voltage and the command voltage ( 60), a correction unit (56) for correcting the command voltage based on the interference voltage estimated by the estimation unit, and converting the command voltage corrected by the correction unit into a coordinate system of a fundamental wave. And an adding unit (49) for adding to the command value of the fundamental voltage.

  In the above configuration, an estimated voltage that can be easily calculated from the interference voltage is calculated based on the actual value of the harmonic component of the motor current and the inverse model of the motor. The interference voltage is estimated based on the deviation between the estimated voltage and the command voltage. Thus, by setting it as the structure which estimates an interference voltage indirectly, the influence by an interference voltage can be suppressed by simple processing.

Sectional drawing showing the structure of SR motor. The circuit diagram showing the electric constitution of the inverter which drives SR motor. The functional block diagram showing the control apparatus which controls SR motor. The functional block diagram showing a fundamental wave command voltage generation part. The functional block diagram showing a harmonic command voltage generation part. The timing chart showing the effect of this embodiment.

  Hereinafter, an embodiment that embodies an SR (Switched Reluctance) motor control device will be described with reference to the drawings. In the present embodiment, the SR motor is assumed to be a main motor mounted on a hybrid vehicle.

  First, the configuration of the SR motor 20 according to the present embodiment will be described with reference to FIG. The SR motor 20 includes a rotor 21 having four salient poles 21a projecting in the radial direction, a cylindrical stator 22 having six salient poles 22a facing the salient poles 21a, and a protrusion of the stator 22. It is configured as a three-phase motor having windings 23 to 25 wound around a pole 22a. The rotor 21 and the stator 22 are arranged coaxially. The windings 23, 24, and 25 constitute a U phase, a V phase, and a W phase, respectively.

  Next, the inverter 10 (power conversion circuit) used for controlling the SR motor 20 will be described with reference to FIG. The inverter 10 is a circuit that converts input power input from the DC power supply 15 having the voltage Vdc into three-phase power and supplies the three-phase power to the SR motor 20. The inverter 10 is configured by connecting a U-phase power conversion circuit, a V-phase power conversion circuit, and a W-phase power conversion circuit in parallel with each other. Since the power conversion circuit of each phase has the same configuration, the U-phase power conversion circuit will be described below as a representative.

  The U-phase power conversion circuit includes switching elements S1u and S2u and diodes D1u to D4u. IGBT, MOSFET, etc. are employable as switching element S1u and S2u. The switching element S1u is connected in series to the cathode of the diode D2u. The switching element S2u is connected in series to the anode of the diode D3u. Diodes D1u and D4u are connected in parallel to the switching elements S1u and S2u, respectively. A winding 23 is connected between a connection point between the switching element S1u and the diode D2u and a connection point between the switching element S2u and the diode D3u. That is, the U-phase power conversion circuit is a so-called H bridge circuit, specifically an asymmetric H bridge circuit.

  Similarly, the V-phase power conversion circuit and the W-phase power conversion circuit are also asymmetric H-bridge circuits. Switching elements S1v and S2v of the V-phase power conversion circuit correspond to switching elements S1u and S2u, and diodes D1v to D4v correspond to diodes D1u to D4u. The switching elements S1w and S2w of the W-phase power conversion circuit correspond to the switching elements S1u and S2u, and the diodes D1w to D4w correspond to the diodes D1u to D4u.

  Therefore, the inverter 10 is a circuit in which three H-bridge circuits, specifically, three asymmetric H-bridge circuits are connected in parallel to each other. A smoothing capacitor 16 is connected between the input terminals of the inverter 10, and a DC power supply 15 is connected in parallel to the smoothing capacitor 16. The DC power supply 15 is a high voltage battery such as a lithium secondary battery, for example, and the voltage Vdc of the DC power supply 15 becomes the input voltage of the inverter 10. As described above, since the inverter 10 is configured by connecting the H bridge circuits of each phase in parallel, the voltage and current of each phase of the SR motor 20 can be independently controlled using the inverter 10.

  The switching elements S1u to S1w and S2u to S2w are turned on or off by an operation signal transmitted from the control device 30. When the switching elements S1u and S2u are turned on, a positive voltage application mode is set. Specifically, current flows from the positive electrode side of the DC power supply 15 through a path passing through the switching element S1u, the winding 23, and the switching element S2u, and a positive voltage Vdc is applied to the winding 23. Further, when the switching element S1u is turned on and the switching element S2u is turned off, the zero voltage application mode is set. Specifically, the current circulates through the path of the switching element S1u, the winding 23, and the diode D3u, and the voltage applied to the winding 23 becomes zero. Similarly, when the switching element S1u is turned off and the switching element S2u is turned on, the zero voltage application mode is set. Further, when the switching elements S1u and S2u are turned off, a negative voltage application mode is set. Specifically, a current flows from the negative electrode side of the DC power supply 15 through a path of the diode D2u, the winding 23, and the diode D3u, and a negative voltage −Vdc is applied to the winding 23. The same applies to the V phase and the W phase.

  Next, the control device 30 will be described with reference to FIG. The control device controls the motor by adjusting the motor current flowing through the SR motor 20. The control device 30 is a microcomputer including a CPU, a ROM, a RAM, an I / O, and the like. The control device 30 transmits an operation signal to each switching element of the inverter 10 by the CPU executing various programs stored in the ROM. Further, the control device 30 takes in the detection value detected by the current sensor 91 (current detection means) and the detection value detected by the resolver 92 (position detection means). The current sensor 91 is a sensor that detects at least two phase currents among the actual currents iu, iv, iw flowing through the windings 23 to 25 of the SR motor 20. If the two-phase current is detected, the remaining phase current can be calculated. The resolver 92 is a sensor that detects the rotation angle θ of the rotor 21 of the SR motor 20.

  As shown in FIG. 3, the control device 30 includes functions of a command current generation unit 41, a command voltage generation unit 42, and a PWM processing unit 47, and performs feedback control of the actual currents id, iq, i0 flowing through the SR motor 20. carry out.

  The fundamental wave command current generation unit 41a of the command current generation unit 41 generates a rotation coordinate system command current id *, iq *, i0 * that commands the current flowing through the SR motor 20 based on the command torque Tr *. Specifically, the fundamental wave command current generation unit 41a generates command currents id *, iq *, i0 * using a map showing the correspondence between the command torque Tr * and the command currents id *, iq *, i0 *. To do.

  The fundamental wave command voltage generator 42a (fundamental wave controller) of the command voltage generator 42 generates current deviations Δid, Δiq, Δi0 between the command currents id *, iq *, i0 * and the actual currents id, iq, i0. Based on this, command voltages vd *, vq *, v0 * for the SR motor 20 are calculated. Then, the fundamental wave command voltage generation unit 42a converts the command voltages vd *, vq *, and v0 * into the three-phase command voltages vu *, vv *, and vw * (Vr *) using the rotation angle θ. ). Detailed control by the fundamental wave command voltage generation unit 42a will be described later.

  The actual currents id, iq, i0 are calculated by the coordinate conversion unit 43. The coordinate conversion unit 43 converts the real currents iu, iv, iw in the fixed coordinate system to the real currents id, iq, i0 in the rotary coordinate system using the rotation angle θ.

  The PWM processing unit 47 generates an operation signal for turning on / off the switching element of the inverter 10 by performing PWM control based on the calculated command voltages vu *, vv *, vw *.

  Next, the control by the fundamental wave command voltage generation unit 42a will be described with reference to FIG. A zero-phase voltage equation that is the d-axis, q-axis, and DC component of the SR motor 20 is expressed by Expression (1). R represents the winding resistance of the windings 23-25. The self-inductances L′ dd, L′ qq, and L′ 00 are defined by partial differentials of the interlinkage magnetic fluxes λd, λq, and λ0 with the actual currents id, iq, and i0, respectively. The mutual inductances Lqd and Ldq are defined by partial differentiation of the interlinkage magnetic fluxes λq and λd by the actual currents id and iq, respectively. The mutual inductances L′ 0q and L′ q0 are defined by partial differentiation of the linkage fluxes λ0 and λq with the actual currents iq and i0, respectively. The mutual inductances L′ d0 and L′ 0d are defined by partial differentiation of the linkage fluxes λd and λ0 with the actual currents i0 and id, respectively. Further, ω represents a rotational angular frequency calculated by differentiation of the rotational angle θ, and s represents a Laplace operator (differential operator). Further, the dots represent partial differentiation with respect to the rotation angle θ.

式（１）で示すように、ｄ軸とｑ軸間、ｄ軸及びｑ軸のそれぞれと零相間に干渉がある。そこで基本波指令電圧生成部４２ａは、図４に示すような非干渉制御を実施する。非干渉制御は、ｄ軸とｑ軸との間の互いの干渉、並びに、ｄ軸及びｑ軸のそれぞれと零相との間の磁気結合による干渉を打消し、ｄ軸側、ｑ軸側、及び零相側で、互いに独立した制御を行うことを可能とするものである。

As shown in Expression (1), there is interference between the d-axis and the q-axis, and between each of the d-axis and the q-axis and the zero phase. Therefore, the fundamental wave command voltage generation unit 42a performs non-interference control as shown in FIG. Non-interference control cancels the mutual interference between the d-axis and the q-axis and the interference due to the magnetic coupling between each of the d-axis and the q-axis and the zero phase, and the d-axis side, the q-axis side, In addition, independent control can be performed on the zero phase side.

  The block diagram shown in FIG. 4 is based on Expression (1). In the SR motor 20, as shown in a block 27, the voltage generated on the d-axis is an interference voltage −sL′dq · iq proportional to the actual current iq on the q-axis with respect to the command voltage vd * on the d-axis. The interference voltage −sL′d0 · i0 proportional to the zero-phase actual current i0 and the interference voltage −ω (−λq + λ′d) proportional to the rotation angular frequency ω are superimposed. The d-axis actual current id is obtained by dividing the voltage obtained by superimposing the interference voltage on the command voltage vd * by the d-axis impedance R + sL′dd.

  Similarly, as shown in block 28, the voltage generated on the q-axis is the interference voltage −sL′dq · id, −sL′d0 · i0, −ω (λd + λ′q) with respect to the command voltage vq *. Superposed voltage. Further, as shown in the block 29, the voltage generated in the zero phase is a voltage obtained by superimposing the interference voltages −sL′d0 · id, −sL′q0 · iq, and −ωλ0 on the command voltage v0 *.

  On the other hand, the fundamental wave command voltage generation unit 42a generates command voltages vd *, vq *, v0 * that cancel the interference voltage generated in the SR motor 20. In FIG. 4, the block 44 in the fundamental wave command voltage generator 42a calculates the d-axis command voltage vd *, the block 45 calculates the q-axis command voltage vq *, and the block 46 calculates the zero-phase command voltage v0 *. Shows control.

  As shown in block 44, the fundamental wave command voltage generation unit 42a performs a PI calculation based on the current deviation Δid between the command current id * and the actual current id to calculate a current idp *. Then, the command voltage generation unit 42 multiplies the voltage calculated by multiplying the current idp * and the impedance R + sLdd so as to cancel the interference voltage, and the voltages sL′dq · iqp *, sL′d0 · i0p *, ω (− λq + λ′d) is added to calculate the command voltage vd *. The current iqp * is a value calculated by performing the PI calculation based on the current deviation Δiq, and the current i0p * is a value calculated by performing the PI calculation based on the current deviation Δi0. The flux linkages λd, λq, λ0 are calculated based on the actual currents id, iq, i0 and the rotation angle θ. Similarly, the block 45 and the block 46 calculate the command voltage vq * and the command voltage v0 *.

  As described above, the control device 30 calculates the command currents id *, iq *, i0 * based on the command torque Tr *, and converts the actual currents id, iq, i0 flowing through the SR motor 20 into the command current id *, Feedback control is performed on iq * and i0 *. Furthermore, the control device 30 performs non-interference control.

  In the present embodiment, in addition to the above-described feedback control of the fundamental wave currents id, iq, i0, the harmonic currents idh, iqh, i0h (harmonic components) having a frequency n times the fundamental wave current are controlled. .

  Mainly due to the distortion of the magnetic flux, a harmonic component having a frequency n times the fundamental wave component (n is a natural number of 2 or more) is generated in the motor current. More specifically, when the self-inductance or mutual inductance of the SR motor 20 has m-order inductance with respect to the rotation angle θ, 2 (m + 1) -order and 2 (m-1) -order harmonic currents flow. . In particular, in a three-phase motor, adverse effects such as torque ripple occur due to components of the 6kth order (k is a natural number of 1 or more) in this harmonic current. Therefore, the d-axis current id and q-axis current iq that are fundamental wave components and the dh-axis current and qh-axis current that are harmonic components are controlled independently. Furthermore, since the SR motor 20 of the present embodiment is an SR motor, the control of the zero phase current i0h, which is a harmonic current corresponding to the zero phase current i0, is performed.

  That is, as shown in FIG. 3, the control device 30 performs the harmonic current control in addition to the fundamental wave current control described above. The command current generation unit 41 includes a harmonic command current generation unit 41b in addition to the fundamental wave command current generation unit 41a. The harmonic command current generator 41b calculates the harmonic command currents idh *, iqh *, i0h * using a map prepared in advance based on the command torque Tr *.

  The command voltage generation unit 42 includes a harmonic command voltage generation unit 42b (harmonic control unit) in addition to the fundamental wave command voltage generation unit 42a. The harmonic command voltage generation unit 42b has command voltages vdh *, vqh *, v0h, which are n-order harmonic components in reverse phase to the torque ripple, so as to cancel the n-th order torque ripple superimposed on the current flowing through the SR motor 20. * Is calculated.

  Then, the command voltage generator 42 adds the command voltage vdh *, vqh *, v0h * of the harmonic component to the command voltage vd *, vq *, v0 * of the fundamental wave and is calculated by the adder 49. The voltage is converted into a three-phase command voltage vu *, vv *, vw * in a fixed coordinate system. Thus, by implementing harmonic current control, torque ripple can be reduced and the efficiency of the SR motor 20 can be improved.

高調波電流において、式（３）に示すように、ｄｈ軸とｑｈ軸との間、ｄｈ軸及びｑｈ軸のそれぞれと、零相との間の干渉電圧（非対角成分）が生じる。なお、Ｌａｖｅ＝（Ｌｄｄ＋Ｌｑｑ）／２であり、θ’ｈ＝２θｈ＋δ（δは所定の位相）である。また、零相電流ｉ０ｈ及び零相電圧Ｖ０ｈは直流成分であるため、座標変換による影響を受けない。このため、零相電流ｉ０とｉ０ｈとは等しく、零相電圧ｖ０とｖ０ｈとは等しい。 Here, the voltage equation at the n-th harmonic of the SR motor 20 is expressed as Equation (3) by performing coordinate transformation on Equation (1) using the matrix represented by Equation (2). Can do.

In the harmonic current, as shown in Expression (3), interference voltages (non-diagonal components) are generated between the dh axis and the qh axis, between the dh axis and the qh axis, and the zero phase. Note that Lave = (Ldd + Lqq) / 2, and θ′h = 2θh + δ (δ is a predetermined phase). Further, since the zero-phase current i0h and the zero-phase voltage V0h are direct current components, they are not affected by coordinate conversion. For this reason, the zero-phase currents i0 and i0h are equal, and the zero-phase voltages v0 and v0h are equal.

  When performing non-interference in the dh-axis, qh-axis, and zero phase, the resistance component R, the self-inductances Ldd, Lqq, L00 in the harmonic component, the mutual inductances Ld0, Lq0 in the harmonic component, and the induced voltage If there is an error, it becomes difficult to make it non-interfering. Further, as shown in Expression (3), the interference terms between the dh axis and the qh axis and the zero phase include pulsation components, and therefore, based on the actual currents idh, iqh, i0h of the dh axis, the qh axis, and the zero phase. It is difficult to directly calculate the interference term. Therefore, in the present embodiment, the following non-interference control is performed in the harmonic command voltage generation unit 42b shown in FIG.

  As shown in FIG. 5, a high-pass filter 50 is applied to each of the d-axis actual current id, the q-axis actual current iq, and the zero-phase actual current i0. Here, the time constant of the high-pass filter 50 is set to a value that can acquire the real current idh of the dh-axis current, which is the nth-order harmonic, the real current iqh of the qh-axis, and the real current i0h of the zero-phase harmonic. ing. The coordinate converter 51 converts the actual current idh of the dh-axis current, the actual current iqh of the qh-axis, and the actual current i0h of the zero-phase harmonic output from the high-pass filter 50 from the dq0 coordinate system to the harmonic coordinate system. To do.

  The automatic current controller 52 generates harmonic command voltage basic values vdh ^, vqh ^, v0h ^ based on the harmonic actual currents idh, iqh, i0h and the harmonic command currents idh *, Iqh *, i0h *. Set. More specifically, based on the deviation between the harmonic actual currents idh, iqh, i0h and the harmonic command currents idh *, iqh *, i0h *, the PI calculation is performed so that the deviation decreases. The harmonic command voltage basic values vdh ^, vqh ^, v0h ^ are calculated. The correction unit 56 adds the interference voltage d2 ^ calculated by the estimation unit 60 to the harmonic command voltage basic values vdh ^, vqh ^, v0h ^ to make the harmonic current control non-interfering. .

  The estimation unit 60 includes a motor inverse model 53, a deviation calculation unit 54, and a low-pass filter 55. The motor inverse model 53 has a motor current (actual current) flowing through the SR motor 20 as an input and a motor voltage applied to the SR motor 20 as an output. In the harmonic control, the harmonic currents idh, iqh, i0h are input to the motor inverse model 53. Then, estimated voltages vdh, vqh, v0h that do not include interference terms are calculated from the motor inverse model 53. Specifically, the estimated voltages vdh, vqh, and v0h that do not include the interference term are calculated by causing the diagonal terms R + sLdd, R + sLqq, and R + sL00 to act on the harmonic actual currents idh, iqh, and i0h, respectively.

  The motor inverse model 53 is set based on the resistance component R of the SR motor 20 and the self-inductances Ldd, Lqq, and L00 (Ldq0) of the SR motor 20. Here, the self-inductances Ldd, Lqq, and L00 in the harmonic component change depending on the fundamental wave component of the motor current and the rotation angle θ of the motor. Therefore, based on the d-axis actual current id, the q-axis actual current iq, the zero-phase actual current i0, and the rotation angle θ of the SR motor 20, which are fundamental wave components of the motor current, a self-inductance is obtained using a map. Ldd, Lqq, and L00 are set.

  The deviation calculating unit 54 calculates the harmonic command voltage basic values vdh ^, vqh ^, v0h ^ (command voltage) to which the interference voltage d2 ^ is added, and the estimated voltages vdh, vqh, v0h of the harmonic voltage not including the interference term. The deviation from is calculated. Then, the interference voltage d2 ^ is calculated by applying a low-pass filter 55 (1 / (1 + sτ)) to the deviation calculated by the deviation calculating unit 54. Then, the correction unit 56 adds the interference voltage d2 ^ to the harmonic command voltage basic values vdh ^, vqh ^, v0h ^.

  Here, the crossing angular frequency (reciprocal of the time constant τ) of the low-pass filter 55 is set based on the rotational angular frequency ω of the SR motor 20. More specifically, when the m-th inductance exists, since the 2 (m + 1) -order and 2 (m-1) -order harmonic currents flow, the crossing angular frequency of the low-pass filter is the rotation angular frequency ω. It is set to 2 (m + 1) times or 2 (m-1) times and 6 k times the rotational angular frequency (k is a natural number of 1 or more).

  The non-interference term calculation unit 57 (feedforward control unit) calculates the non-interference voltage d1 ^ of the harmonic voltage based on the harmonic actual currents idh, iqh, i0h by feedforward control. Specifically, R + sLave, R + sLave, and R + sL00 that are diagonal terms are applied to the harmonic actual currents idh, iqh, and i0h, respectively, thereby calculating the non-interference voltage d1 ^ of the harmonic voltage.

  Then, the adding unit 58 includes harmonic command voltage basic values vdh ^, vqh ^, v0h ^ obtained by adding the non-interference voltage d1 ^ of the harmonic voltage calculated by the non-interference term calculating part 57 and the interference voltage d2 ^. Are added to calculate the harmonic command voltages vdh *, vqh *, v0h *. Furthermore, the coordinate conversion unit 59 converts the harmonic command voltages vdh *, vqh *, v0h * from the harmonic coordinate system to the fundamental wave coordinate system.

  FIG. 6 is a timing chart showing the operational effects according to the present embodiment. By the non-interference control in this embodiment, the followability to the actual current with respect to the command currents iq *, id *, i0 * is improved in the q-axis current iq, the d-axis current id, and the zero-phase current i0. Further, the distortion of the waveform of the phase currents iu, iv, iw is eliminated.

  The effects of this embodiment will be described below.

  Based on the actual value (actual current) of the harmonic component of the motor current and the inverse model of the SR motor 20, an estimated voltage that can be easily calculated compared to the interference voltage is calculated. Based on the deviation, the interference voltage d2 ^ is calculated. Thus, by setting it as the structure which calculates interference voltage d2 ^ indirectly, the influence by interference voltage d2 ^ can be suppressed by simple processing.

  Specifically, an estimated voltage that does not include an interference voltage is calculated based on the actual currents idh, iqh, i0h of the harmonic components and the motor inverse model 53 of the SR motor 20, and the deviation between the estimated voltage and the command voltage Is calculated. By applying a low-pass filter 55 to the deviation, the interference voltage d2 ^ is calculated. Here, the stability of control can be improved by applying the low-pass filter 55.

  The pulsation frequency θ′h of the interference voltage d2 ^ changes according to the rotational angular frequency ω of the motor. Therefore, in order to suppress the influence of the pulsating component of the interference voltage d2 ^, the time constant of the low-pass filter 55 is set according to the rotational angular frequency ω of the SR motor 20. More specifically, when the self-inductance or mutual inductance of the SR motor 20 has an m-order inductance (m is a natural number of 2 or more) with respect to the rotation angle θ, the 2 (m + 1) th order and 2 (m−1) ) The next harmonic current flows. In the three-phase motor, an adverse effect such as torque ripple occurs due to a component of the 6th order (k is a natural number of 1 or more) of the harmonic current. Therefore, the crossing angular frequency of the low-pass filter 55 is 2 (m + 1) times or 2 (m−1) times the rotational angular frequency ω, and is 6k times the rotational angular frequency (k is a natural number of 1 or more). The configuration is set.

  In addition, the motor inverse model 53 of the SR motor 20 of the present embodiment includes only the resistance component R of the SR motor 20 and the dh-axis, qh-axis, and zero-phase (harmonic component) self-inductances Ldd, Lqq, and L00 of the SR motor 20. It is the simple structure represented by these. For this reason, it becomes possible to simplify a process.

  The values of the self-inductances Ldd, Lqq, and L00 in the harmonic component change due to magnetic saturation caused by the motor current. In addition, the values of the self-inductances Ldd, Lqq, and L00 in the harmonic components change depending on the relative position between the rotor 21 and the stator 22, that is, the rotation angle θ of the SR motor 20. Therefore, the self-inductances Ldd, Lqq, and L00 of the harmonic components are set based on the fundamental wave component of the motor current and the rotation angle θ.

  Further, in the above configuration, the stability of the control can be further improved by performing the feedforward control in the harmonic current control as an auxiliary.

(Other embodiments)
-In the above-mentioned embodiment, although it was set as the structure provided only with one harmonic voltage generation part, it is good also as a structure provided with a some harmonic voltage generation part by changing this. In this case, the final command voltage is the sum of the fundamental voltage command value by the fundamental voltage generator and the harmonic voltage command values by the plurality of harmonic voltage generators.

  The low-pass filter 55 may be omitted.

  The non-interference term calculation unit 87 may be omitted.

  -Although the control apparatus in the said embodiment is targeting SR motor, you may change this. That is, it may be a control device applied to a field winding type synchronous motor or the like in which zero phase current does not flow. In this case, the control device controls the d-axis current and the q-axis current as the fundamental wave current, the dh-axis current and the qh-axis current as the harmonic current, and decouples the interference voltage between the dh-axis and the qh-axis.

  DESCRIPTION OF SYMBOLS 20 ... SR motor, 30 ... Control apparatus, 42a ... Fundamental wave command voltage generation part, 42b ... Harmonic command voltage generation part, 49 ... Addition part, 53 ... Motor reverse model, 56 ... Correction part, 60 ... Estimation part.

Claims (7)

A motor control device (30) for controlling the motor by adjusting a motor current flowing in the motor (20),
A fundamental wave control unit (42a) for setting a command value of a fundamental voltage applied to the motor so that a d-axis current and a q-axis current, which are fundamental wave components of the motor current, have predetermined command values;
The dh-axis current and the qh-axis current, which are harmonic components of the motor current having a frequency n times (n is a natural number of 2 or more) the fundamental wave component of the motor current, are set to predetermined command values. A harmonic control unit (42b) for setting a command voltage which is a command value of the harmonic voltage applied to the motor;
Based on the inverse model (53) of the motor having the motor current as an input and the motor voltage applied to the motor as an output, and the actual value of the harmonic component of the motor current, An estimation unit (60) that calculates an estimated voltage of the harmonic voltage that does not include the interference voltage that occurs at the same time, and estimates the interference voltage based on a deviation between the estimated voltage and the command voltage;
A correction unit (56) for correcting the command voltage based on the interference voltage estimated by the estimation unit;
An adder (49) for converting the command voltage corrected by the correction unit into a coordinate system of a fundamental wave and adding the command voltage to the command value of the fundamental wave voltage;
A control device comprising:   The control device according to claim 1, wherein the estimation unit estimates the interference voltage by applying a filter (55) to a deviation between the estimated voltage and the command voltage.   The control device according to claim 2, wherein the filter is a low-pass filter, and an intersection angular frequency of the low-pass filter is set according to a rotation angular frequency of the motor. The motor is a three-phase motor,
The crossing angular frequency is 2 (m + 1) times the rotational angular frequency when the self-inductance of the motor or the mutual inductance has an m-th inductance with respect to the rotational angle of the motor (m is a natural number of 2 or more), The control apparatus according to claim 3, wherein the control device is set to 2 (m−1) times and 6 k times the rotation angular frequency (k is a natural number of 1 or more).   5. The control device according to claim 1, wherein the inverse model is set based on a resistance component of the motor and a self-inductance of the motor.   The estimation unit sets a self-inductance of the motor based on a d-axis current, a q-axis current, a zero-phase current that is a DC component, and a rotation angle of the motor, which are fundamental wave components of the motor current. The control device according to claim 5, wherein:   Based on the dh-axis current, qh-axis current, which is a harmonic component of the motor current, and zero-phase current, which is a DC component, a non-interference voltage which is a voltage not including the interference voltage is calculated, and the non-interference voltage The control apparatus according to claim 1, further comprising: a feedforward control unit that corrects the command voltage by adding to the command voltage corrected by the correction unit.

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