JP2017005648A - Reflective phase shifter - Google Patents

Abstract

The present invention enables a large phase shift change. A reflection type phase shifter includes a 90 ° coupler having a first reflection terminal and a second reflection terminal, a first load circuit connected to the first reflection terminal, and a second reflection. And a second load circuit 13a connected to the terminal 12b. Each of the first load circuit 13 a and the second load circuit 13 b includes variable reactance elements 21 and 23, and is configured to have at least three resonance points in the variable range of the variable reactance elements 21 and 23. The at least three resonance points include two series resonance points and a parallel resonance point between the two series resonance points, or a series resonance between the two parallel resonance points and the two parallel resonance points And a point. [Selection] Figure 1

Description

  The present invention relates to a reflective phase shifter.

As shown in FIG. 6, the conventional reflection type phase shifter 101 is configured by connecting a variable capacitance element 103 to each of the two reflection terminals 102 a and 102 b of the 90 ° coupler 101. The signal input to the input terminal 104a of the 90 ° coupler 101 changes in phase and is output from the output terminal 104b. When the reflection coefficients at the reflection terminals 102a and 102b are Γ ₁ and Γ ₂ , a signal of (Γ ₁ + Γ ₂ ) / 2 is output from the output terminal 104b. The phase of the signal output from the output terminal 104 b is adjusted by changing the capacitance of the variable capacitance element 103. A reflection type phase shifter is disclosed in Patent Document 1, for example.

JP 2014-216936 A

  Since the conventional reflection type phase shifter does not have a large variable range of phase, it is necessary to connect a plurality of reflection type phase shifters in series in order to obtain a large phase change. When a plurality of reflection type phase shifters are required, the size and cost are increased.

  Therefore, a reflection type phase shifter capable of a large phase change is desired.

From a certain viewpoint, the present invention includes a 90 ° coupler having a first reflection terminal and a second reflection terminal, a first load circuit connected to the first reflection terminal, and a second load circuit connected to the second reflection terminal. When,
Is provided. Each of the first load circuit and the second load circuit includes a variable reactance element, and is configured to have at least three resonance points in a variable range of the variable reactance element. At least three of the resonance points include two series resonance points and a parallel resonance point between the two series resonance points. Alternatively, at least three of the resonance points may include two parallel resonance points and a series resonance point between the two parallel resonance points.

  According to the present invention, a large phase change is possible.

It is a circuit diagram of a reflection type phase shifter. It is a Smith chart. It is a figure which shows the impedance of a load circuit. It is a chart which shows the impedance of a load circuit, and the amount of phase shifts. It is explanatory drawing of the modification and impedance of a load circuit. It is a circuit diagram of the conventional reflection type phase shifter.

  Hereinafter, preferred embodiments of the present invention will be described with reference to the drawings.

[1. Outline of Embodiment]
(1) The reflection type phase shifter according to the embodiment is connected to the 90 ° coupler having the first reflection terminal and the second reflection terminal, the first load circuit connected to the first reflection terminal, and the second reflection terminal. A second load circuit. Each of the first load circuit and the second load circuit includes a variable reactance element, and is configured to have at least three resonance points in a variable range of the variable reactance element. The at least three resonance points include, for example, two series resonance points and a parallel resonance point between the two series resonance points. Further, the at least three resonance points may include, for example, two parallel resonance points and a series resonance point between the two parallel resonance points.

(2) The first load circuit is a plurality of series resonance circuits connected in parallel to the first reflection terminal, and the second load circuit is a plurality of series resonance circuits connected in parallel to the second reflection terminal. Is preferred. In this case, in the variable range of the variable reactance element, two series resonance points and a parallel resonance point between the two series resonance points can be obtained.

(3) It is preferable that the number of series resonant circuits included in the plurality of series resonant circuits is two. In this case, the size can be reduced.

(4) The first load circuit is a plurality of parallel resonance circuits connected in series to the first reflection terminal, and the second load circuit is a plurality of parallel resonance circuits connected in series to the second reflection terminal. Is preferred. In this case, two parallel resonance points and a series resonance point between the two parallel resonance points can be obtained in the variable range of the variable reactance element.

(5) The number of parallel resonant circuits included in the plurality of parallel resonant circuits is preferably two. In this case, the size can be reduced.

[2. Details of Embodiment]
A reflection type phase shifter 10 shown in FIG. 1 includes a 90 ° coupler 11 and load circuits 13a and 13b. The reflection type phase shifter 10 of the present embodiment has a variable amount of phase shift. The reflective phase shifter 10 is provided, for example, in a wireless communication device and is used to adjust the phase of a wireless signal. More specifically, the reflective phase shifter 10 can be used for phase control in an active array antenna system that controls the phase of a radio signal.

  The 90 ° coupler 11 includes a first reflection terminal 12a, a second reflection terminal 12b, an input terminal 14a, and an output terminal 14b. The 90 ° coupler 11 is also called a 90 ° hybrid coupler. A first load circuit 13a is connected to the first reflection terminal 12a. A second load circuit 13b is connected to the second reflection terminal 12b. The signal input to the input terminal 14a changes in phase and is output from the output terminal 14b. The phase of the signal output from the output terminal 14b depends on the impedance Z of the first load circuit 13a and the second load circuit 13b. The impedance Z of the first load circuit 13a and the second load circuit 13b is controlled by the control unit 30. By changing the impedance Z of the first load circuit 13a and the second load circuit 13b, the reflection coefficients Γ and Γ viewed from the reflection terminals 12a and 12b are changed, and the amount of phase shift by the phase shifter 10 can be changed. it can.

The first load circuit 13a and the second load circuit 13b are configured to include a lumped constant element.
The first load circuit 13a has a plurality of series resonance circuits 15a and 15b connected in parallel to the first reflection terminal 12a. The second load circuit 13b has the same configuration as the first load circuit 13a, and includes a plurality of series resonant circuits 15a and 15b connected in parallel to the second reflection terminal 12b. The first load circuit 13a and the second load circuit 13b in FIG. 1 each have two series resonance circuits 15a and 15b.

  The first series resonance circuit 15a includes a first capacitor 21 that is a reactance element, and a first coil 22 that is also a reactance element. The second series resonance circuit 15b includes a second capacitor 23 that is a reactance element and a second coil 24 that is a reactance element.

  The first series resonance circuit 15a and the second series resonance circuit 15b each have a variable reactance element. The impedance Z of the first load circuit 13a and the second load circuit 13b changes due to the change in reactance of the variable reactance element. In FIG. 1, the first capacitor 21 and the second capacitor 23 are variable capacitance elements, that is, variable reactance elements. The first capacitor 21 and the second capacitor 23 are each composed of a varactor diode. By controlling the voltage applied to the varactor diode by the control unit 30, the capacitance C of the varactor diodes 21 and 23 changes. In this embodiment, each load circuit 13a, 13b has only two variable reactance elements 21, 23, and can be downsized.

As described above, the phase of the signal output from the output terminal 14b is determined depending on the reflection coefficient Γ at the reflection terminals 12a and 12b. The reflection coefficient Γ viewed from the reflection terminals 12a and 12b depends on the impedance Z of the load circuits 13a and 13b as follows. In the following equation, Z ₀ is the characteristic impedance of the 90 ° coupler 11.

  Here, since the impedance Z is a complex number, as is apparent from the above equation, the reflection coefficient is also a complex number. If the impedance Z can be rotated 360 ° in the complex plane indicating the impedance Z, the phase of the signal output from the output terminal 14b is changed by 360 ° by rotating the reflection coefficient 360 ° in the complex plane indicating the reflection coefficient. Can be made. However, as shown in FIG. 6, simply having a variable capacitance element as a load circuit cannot rotate the impedance of the load circuit by 360 °, and therefore the amount of phase shift is less than 360 °.

  The present inventor has examined what load circuits 13a and 13b are suitable for realizing a phase shift in a larger range, specifically, a phase shift in a range of 360 °. First, in the Smith chart shown in FIG. 3, the reflection coefficient Γ at the right end (r = ∞) of the real axis r is 1, and the phase does not change. The reflection coefficient Γ at the left end (r = 0) of the real axis r is −1, and the phase is inverted by 180 °. Accordingly, the light is reflected counterclockwise from the right end (r = ∞) of the real axis r through the region (inductive region) above the real axis r to the left end (r = 0) of the real axis r. As the coefficient Γ changes, the phase can change from 0 ° to + 180 °. Further, the light is reflected clockwise from the right end (r = ∞) of the real axis r to the left end (r = 0) of the real axis r through a region (capacitive region) below the real axis r. As the coefficient Γ changes, the phase can change in the range of 0 ° to -180 °. By utilizing this point, phase shift in the range of 360 ° from + 180 ° to −180 ° becomes possible.

  Here, the left end (r = 0) of the real axis r corresponds to the case where the reflection terminals 12a and 12b are short, and the right end (r = ∞) of the real axis r is the reflection terminals 12a and 12b. This corresponds to the case of being open. Therefore, the reflection terminals 12a and 12b are opened from the state in which the reflection terminals 12a and 12b are short-circuited and the amount of phase shift is + 180 ° through the state where the inductive impedance is connected to the reflection terminals 12a and 12b. As a result, the phase shift amount reaches 0 °, and the reflection terminals 12a and 12b are short-circuited again via the state in which the capacitive impedance is connected to the reflection terminals 12a and 12b. By changing the impedance Z of the load circuits 13a and 13b so as to reach a state where the amount is −180 °, phase shift in the range of + 180 ° to −180 ° becomes possible.

  In the phase shifter 10 shown in FIG. 1, by changing the capacitance of the variable capacitance elements 21 and 23, the reflection terminals 12a and 12b are changed from short to inductive to open to capacitive to short and + 180 ° to The phase shift in the range of −180 ° can be performed. The first load circuit 13a of the present embodiment is obtained by connecting two series resonant circuits in parallel and has an angular frequency ω-impedance Z characteristic as shown in FIG. The same applies to the second load circuit 13b.

As shown in FIG. 3, the first load circuit 13a has at least three resonance frequencies. Similarly, the second load circuit 13b has at least three resonance frequencies. In the circuit of FIG. 1, at least three resonance frequencies are a first series resonance frequency ω _r1 , a second series resonance frequency ω _r2, and a parallel resonance frequency ω _m between the two series resonance frequencies ω _r1 and ω _r2. And including. Between the first series resonance frequency ω _r1 and the parallel resonance frequency ω _m is inductive, and between the parallel resonance frequency ω _m and the second series resonance frequency ω _r2 is capacitive. Note that ω = 2πf, and f is the frequency of the signal.

The impedance Z of the load circuits 13a and 13b shown in FIG. 1 is expressed by the following equation. Here, the capacitance of the first capacitor 21 and the second capacitor 23 included in the load circuits 13a and 13b is C, the inductance of the first coil 22 is L1, and the inductance of the second coil 24 is L2. Note that L ₁ > L ₂ .

Here, the resonance frequency is a frequency at which the resonance circuit takes an extreme value, the series resonance frequency is a frequency at which the absolute value of the impedance of the resonance circuit is a minimum value (zero), and the parallel resonance frequency is the resonance circuit. This is the frequency at which the absolute value of the impedance becomes the maximum value (+ ∞, −∞). Based on the above formula, the series resonance frequencies ω _r1 and ω _r2 are angular frequencies at which the denominator of X is 0, and the parallel resonance frequency ω _m is an angular frequency at which the numerator of X is 0.

The first series resonance frequency ω _r1 is expressed by the following equation. As is apparent from the following equation, the first series resonance frequency ω _r1 is equal to the series resonance frequency of the first series resonance circuit 15a.

The second series resonance frequency ω _r2 is expressed by the following equation. As is apparent from the following equation, the second series resonance frequency ω _r2 is equal to the series resonance frequency of the second series resonance circuit 15b.

The parallel resonance frequency ω _m is expressed by the following equation.

As described above, when L ₁ > L ₂ , ω _r1 <ω _r2 is satisfied, and ω _m is between ω _r1 and ω _r2 .

As shown in FIG. 3, the impedance Z is capacitive in a frequency range lower than the first series resonance frequency ω _r1 . In the frequency range that is higher than the first series resonance frequency ω _{r1 and} lower than the parallel resonance frequency ω _m , the impedance Z is inductive. In a frequency range that is higher than the number of parallel resonance frequencies ω _{m and} lower than the second series resonance frequency ω _r2 , the impedance Z is capacitive. In the frequency range higher than the second series resonance frequency _ωr2 , the impedance Z is inductive.

The reason why the inductive property and the capacitive property are switched as shown in FIG. 3 depending on the angular frequency ω is as follows. First, the series resonance frequency ω ₀ of a single LC series resonance circuit (for example, the series resonance circuit including the first capacitor 21 and the first coil 22 in FIG. ¹ ) is (√ (LC)) ⁻¹ . When the angular frequency of the signal applied to the LC series resonance frequency is ω, when ω <ω ₀ , the impedance of the LC series resonance circuit is capacitive, and when ω> ω ₀ , the impedance of the LC series resonance circuit. Becomes capacitive, and when ω = ω ₀ , the impedance of the LC series resonance circuit is zero.

When a plurality of series resonant circuits 15a and 15b are connected in parallel as in the load circuits 13a and 13b of FIG. 1, if the impedance of any of the series resonant circuits 15a and 15b becomes zero, the load circuits 13a and 13b The overall impedance is also zero. Therefore, in the load circuits 13a and 13b, the impedance Z becomes zero at each of the two series resonance frequencies ω _r1 and ω _r2 .

In the range of ω <ω _r1 , the series resonance circuits 15a and 15b are all capacitive, and the impedance Z of the entire load circuits 13a and 13b is also capacitive. In the range of ω> _ωr2 , since the plurality of series resonant circuits 15a and 15b are all inductive, the impedance Z of the entire load circuits 13a and 13b is also inductive.

The parallel resonance frequency ω ₀ of a single LC parallel resonance circuit is (√ (LC)) ⁻¹ . When the angular frequency of the signal given to the LC parallel resonance frequency is ω, the impedance of the LC parallel resonance circuit is inductive when ω <ω ₀ , and the impedance of the LC parallel resonance circuit when ω> ω _0. Becomes inductive, and when ω = ω ₀ , the impedance of the LC parallel resonant circuit becomes maximum (+ ∞, −∞).
A circuit in which a plurality of series resonant circuits 15a and 15b are connected in parallel like the load circuits 13a and 13b in FIG. 1 resonates in parallel at the number of parallel resonant frequencies ω _m . The operation is the same as that of a single LC parallel resonant circuit. Therefore, in the range of ω _r1 <ω _m , the impedance Z of the entire load circuits 13a and 13b is inductive, and in the range of ω _m <ω _r1 , the impedance Z of the entire load circuits 13a and 13b is capacitive.

In the load circuits 13a and 13b, the resonance frequencies ω _r1 , ω _r2 , and ω _m are determined by the capacitance C and the inductances L ₁ and L ₂ . The characteristics shown in FIG. 3 can be shifted left and right. Therefore, at a specific frequency ω _Target , the impedances of the load circuits 13a and 13b can be changed from zero → inductivity → + ∞, −∞ → capacitance → zero. In other words, the reflection terminals 12a and 12b can be changed from short circuit → inductivity → open → capacitance → short circuit. As a result, the phase shifter 10 is capable of phase shifting in the range of + 180 ° to −180 °.

FIG. 4 shows reactance X of the load circuits 13a and 13b when the capacitance C of the variable capacitance elements 21 and 23 shown in FIG. 1 is changed. Here, the inductance L ₁ = 0.8 [nH], L2 = 0.4 [nH], and C = x [pF] of the first coil 22. x is a variable range of the variable capacitance elements 21 and 23, and is 2 [pF] to 8 [pF].

As shown in FIG. 4A, when the capacitance C of the variable capacitance elements 21 and 23 is changed, the first series resonance occurs at C _r1 = 2.58472 [pF], and C _m = 3.4463 [pF ], A parallel resonance occurred, and a second series resonance occurred at C _r2 = 5.16945 [pF]. As described above, the load circuits 13a and 13b include the first series resonance point where the first series resonance occurs and the second series where the second series resonance occurs in the variable range of the variable reactance elements (variable capacitance elements) 21 and 23). A resonance point; and a parallel resonance point between the first series resonance point and the second series resonance point. As shown in FIG. 4A, X is positive and inductive between the first series resonance point and the parallel resonance point, and X is negative between the parallel resonance point and the second series resonance point. And is capacitive.

FIG. 4B shows the amount of phase shift by the phase shifter 10 when the capacitance C of the variable capacitance elements 21 and 23 of FIG. 1 is changed. At the point A ₁ corresponding to the first series resonance point, the phase shift amount is + 180 °, and at the point A ₂ corresponding to the parallel resonance point, the phase shift amount is 0 ° and the point A corresponding to the second series resonance point. ₃ , the amount of phase shift is −180 °. Between the point A1 and the point A3, the phase shift amount changes continuously. Thus, according to the reflective phase shifter 10 of FIG. 1, the phase shift in the range of + 180 ° to −180 ° is possible. That is, the shooter type phase shifter 10 functions as a 360 ° variable phase shifter. Therefore, it is possible to shift the phase in the range of 360 ° without connecting multiple reflection type phase shifters. As a result, it is possible to suppress an increase in the size of the device on which the reflective phase shifter 10 is mounted, and to prevent an increase in cost.

  FIG. 5 shows another example of load circuits 13a and 13b connected to the reflection terminals 12a and 12b. Here, in the load circuits 13a and 13b in FIG. 1, the inductive impedance is connected to the reflection terminals 12a and 12b from the state in which the reflection terminals 12a and 12b are short-circuited and the phase shift amount is + 180 °. The reflection terminals 12a and 12b are opened and the phase shift amount is 0 °, and further, the reflection terminals 12a and 12b are connected to the reflection terminals 12a and 12b via the capacitive impedance. The impedance Z changes so that 12a and 12b are short again and the phase shift amount is -180 °.

  In order to perform the phase shift of 360 °, the method of changing the impedance Z of the load circuits 13a and 13b is not limited to the above, and for example, the reflection terminals 12a and 12b are open and the amount of phase shift is 0 °. From a certain state, through the state where the capacitive impedance is connected to the reflection terminals 12a and 12b, the reflection terminals 12a and 12b are short-circuited, and the phase shift amount is + 180 ° and further −180 °. Through the state where the inductive impedance is connected to the reflection terminals 12a and 12b, the load terminals 13a and 12b are opened so that the reflection terminals 12a and 12b are opened again and the phase shift amount is 0 °. The impedance of 13b may change. Such a change in the impedance Z is obtained, for example, by the load circuits 13a and 13b shown in FIG.

  The load circuits 13a and 13b shown in FIG. 5A have a plurality of parallel resonant circuits 16a and 16b connected in series to the reflection terminals 12a and 12b of the 90 ° coupler 11. The first load circuit 13a and the second load circuit 13b have the same configuration. The first parallel resonant circuit 16a includes a first capacitor 21 that is a reactance element and a first coil 22 that is also a reactance element. The second parallel resonant circuit 16b includes a second capacitor 23 that is a reactance element and a second coil 24 that is a reactance element.

  The first parallel resonant circuit 16a and the second parallel resonant circuit 16b each have a variable reactance element. The impedance Z of the first load circuit 13a and the second load circuit 13b changes due to the change in reactance of the variable reactance element. In FIG. 5A, the first capacitor 21 and the second capacitor 23 are variable capacitance elements, that is, variable reactance elements. The first capacitor 21 and the second capacitor 23 are each composed of a varactor diode. By controlling the voltage applied to the varactor diode by the control unit 30, the capacitance C of the varactor diodes 21 and 23 changes. In this embodiment, each load circuit 13a, 13b has only two variable reactance elements 21, 23, and can be downsized.

  In the phase shifter 10 having the load circuits 13a and 13b shown in FIG. 5A, the reflection terminals 12a and 12b are opened → capacitance → short → inductive by changing the capacitance of the variable capacitance elements 21 and 23. → It is configured to be able to perform phase shift in the range of 0 ° to + 180 ° and −180 ° to 0 ° by changing to open.

The load circuits 13a and 13b in FIG. 5A have at least three resonance frequencies. In the circuit of FIG. 5 (a) b, as shown in FIG. 5 (b), at least three resonance frequencies include a first parallel resonance frequency ω _r1 , a second parallel resonance frequency ω _r2 , and two parallel resonance frequencies. series resonance frequency ω _m between ω _r1 and ω _r2 . The first parallel resonance frequency ω _r1 and the series resonance frequency ω _m are capacitive, and the series resonance frequency ω _m and the second parallel resonance frequency ω _r2 are inductive.

In the load circuits 13a and 13b, the resonance frequencies ω _r1 , ω _r2 , and ω _m are determined by the capacitance C and the inductances L ₁ and L ₂ . The characteristic shown in FIG. 5B can be shifted left and right. Therefore, at a specific frequency ω _Target , the impedance of the load circuits 13a and 13b can be changed as + ∞ → capacitance → 0 → inductivity → −∞. That is, the reflection terminals 12a and 12b can change from open → capacitance → short → inductivity → open. As a result, the phase shifter 10 can perform a phase shift of 360 °.

Load circuit 13a of FIG. 5 (a), the case of 13b, the first parallel resonance frequency omega _r1, the first parallel resonance occurs, in the second parallel resonance frequency omega _r2 occurs second parallel resonance, series resonance frequency Series resonance occurs at ω _m . The load circuit 13 (a) 13 (b) in FIG. 5A includes a first parallel resonance point where a first parallel resonance occurs in the variable range of the variable reactance elements (variable capacitance elements 21 and 23), and a second parallel resonance point. It has a second parallel resonance point where parallel resonance occurs, and a series resonance point between the first parallel resonance point and the second parallel resonance point. Between the first parallel resonance point and the series resonance point is capacitive, and between the series resonance point and the second parallel resonance point is inductive.

  The circuit of FIG. 5A is the same as the circuit of FIG.

  The specific circuit configuration of the load circuits 131 and 13b is not limited to the circuit shown in FIG. 1 or FIG.

[3. Addendum]
The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the meanings described above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

DESCRIPTION OF SYMBOLS 10 Reflective type phase shifter 11 90 degree coupler 12a 1st reflective terminal 12b 2nd reflective terminal 13a 1st load circuit 13b 2nd load circuit 21 Variable reactance element 23 Variable reactance element

Claims (5)

A 90 ° coupler having a first reflective terminal and a second reflective terminal;
A first load circuit connected to the first reflection terminal;
A second load circuit connected to the second reflective terminal;
With
Each of the first load circuit and the second load circuit includes a variable reactance element, and is configured to have at least three resonance points in a variable range of the variable reactance element;
At least three of the resonance points are
Including two series resonance points and a parallel resonance point between the two series resonance points, or including two parallel resonance points and a series resonance point between the two parallel resonance points ,
Reflective phase shifter. The first load circuit is a plurality of series resonant circuits connected in parallel to the first reflection terminal,
The reflection type phase shifter according to claim 1, wherein the second load circuit is a plurality of series resonant circuits connected in parallel to the second reflection terminal. The reflection type phase shifter according to claim 2, wherein the number of series resonance circuits included in the plurality of series resonance circuits is two. The first load circuit is a plurality of parallel resonant circuits connected in series to the first reflection terminal,
The reflection type phase shifter according to claim 1, wherein the second load circuit is a plurality of parallel resonance circuits connected in series to the second reflection terminal. The reflection type phase shifter according to claim 4, wherein the number of parallel resonant circuits included in the plurality of parallel resonant circuits is two.