JP2014057469A - Semiconductor integrated circuit and method of operating the same - Google Patents

Abstract

An object of the present invention is to reduce ripples in the output voltage when a switching operation is restarted from a forced stop period of the switching operation at a light load.
A hysteresis comparator circuit 13 controls to turn on a soft start control transistor Q2 in response to a decrease in a feedback signal FB below a low threshold voltage VthL due to the arrival of a maximum value of an output voltage _VOUT at a light load. The output signal CNT is generated, and the switching operation of the switch element Q1 is stopped. The comparison circuit 13 generates a control output signal CNT for turning off the element Q2 in response to the increase of the signal FB over the high threshold voltage VthH due to the switching stop. The soft start circuit 18, the PWM comparator 12, and the control drive circuits 16 and 17 generate a drive output signal GD that drives the element Q1 to the ON state when switching of the element Q1 in response to the signal CNT is resumed, and the pulse width of the signal GD Is controlled from narrow to gradually wide when restarting the switching operation.
[Selection] Figure 1

Description

  The present invention relates to a semiconductor integrated circuit used for a DC-DC converter of a switching regulator type and an operation method thereof, and more particularly, an output voltage fluctuation ripple when a switching operation is restarted from a forced stop period of a switching operation at a light load. The present invention relates to a technique effective for reducing the above.

  A power source is used in electronic devices such as computers, peripheral devices, televisions, audios, and mobile phones, but there are various power sources. The linear voltage regulator has the advantage that the circuit is simple and the output ripple is small, but has the disadvantage that the conversion efficiency is low and the power consumption is large. On the other hand, switching power supplies that do not have this drawback are widespread.

  In a step-down regulator as a kind of switching power supply, an input voltage is supplied to one end of an inductor via a main switch, a sub switch is connected between one end of the inductor and the ground voltage, and the other end of the inductor and the ground voltage are connected. A smoothing capacitor and a load are connected in parallel. When the capacitor is charged by controlling the main switch and the sub switch in the on state and the off state, respectively, and the output voltage rises to a specified voltage of a high level, the main switch and the sub switch are switched between the off state and the on state. The capacity is discharged by controlling each. Therefore, when the output voltage drops to a low level specified voltage due to the discharge of the capacitor, the capacitor is recharged by controlling the main switch and the sub switch in the on state and the off state, respectively.

  A flyback converter as another type of switching power supply is used when electrical insulation is required when obtaining a DC voltage directly from a commercial power supply of 110V / 220V. This flyback converter uses a transformer rather than an inductor. The primary and secondary windings of the transformer are connected in reverse polarity, the input voltage is supplied to one end of the primary winding, the other end of the primary winding is connected to the ground potential via the main switch, and the secondary winding One end of the winding is connected to the anode of the rectifying diode, and the cathode of the rectifying diode is connected to the other end of the secondary winding through a parallel connection of a smoothing capacitor and a load. While the main switch is on, the primary current of the primary winding of the transformer increases. However, the rectifier diode is reverse-biased in the secondary winding of the transformer and no secondary current flows. When the main switch is turned off, the magnetic field of the transformer tries to maintain the current, so that the secondary current flows and the smoothing capacitor is charged via the rectifier diode. The output voltage is determined by the input voltage, the on time and the off time of the main switch, and the turns ratio of the primary and secondary windings of the transformer.

  In the disclosure related to FIG. 1 and FIG. 2 of Patent Document 1 below, a soft start circuit having a large time constant at the start-up of the DC-DC converter is provided in order to suppress an overshoot voltage at the start-up of the DC-DC converter. It is described that the pulse width for the switching operation is controlled from being narrow to gradually wider by using it. This DC-DC converter is the flyback converter described above, and an input voltage is supplied to one end of the primary winding of the transformer, and the other end of the primary winding is connected to the ground potential via the main switch. One end of the line is connected to the anode of the rectifier diode, and the cathode of the rectifier diode is connected to the other end of the secondary winding through a parallel connection of the smoothing capacitor and the output voltage detection circuit. The output voltage detection circuit includes a light emitting portion of the photocoupler. When the output voltage is higher than the rated voltage, the current of the light emitting portion of the photocoupler of the output voltage detection circuit increases, and the light emission amount of the light emitting portion increases. The soft start circuit includes a photocoupler light-receiving unit optically coupled to the photocoupler light-emitting unit of the output voltage detection circuit, and the internal impedance of the photocoupler light-receiving unit responds to an increase in the amount of light emitted from the photocoupler of the output voltage detection circuit. It decreases and the voltage at the feedback terminal increases. As a result, the pulse width control IC controls the switching pulse width from wide to narrow in response to an increase in the voltage at the feedback terminal, so that the increase in the output voltage is compensated by the decrease in the output voltage. It is.

JP 2000-50624 A

  Prior to the present invention, the present inventor was engaged in the development of a semiconductor integrated circuit used in a switching regulator type DC-DC converter with improved power efficiency at light load.

  Conventionally, in a switching power supply, when the output voltage decreases due to an increase in load current at heavy load, the drive pulse width is increased by pulse width modulation (PWM) by feedback control to compensate for the decrease in output voltage. Is.

  Furthermore, conventionally, in a switching power supply, in order to prevent a decrease in power efficiency due to an increase in loss due to an increase in switching loss at light load, control of pulse frequency modulation (PFM) for decreasing the switching frequency at light load is performed. It is intended to move to. For this PFM control, conventionally, a control method called skip mode or burst mode has been adopted. In this skip mode or burst mode, if the feedback voltage increases above the high level threshold due to a decrease in the output voltage, the output voltage is increased by PFM control. When the feedback voltage falls below the low level threshold due to the increase in the output voltage, the switching operation of the switching power supply is stopped, the smoothing capacity of the switching power supply is discharged by the load current, and the output voltage is lowered linearly. By repeating this operation, the switching operation of the switching power supply is executed intermittently, so that the switching frequency can be effectively reduced at light loads, and the switching loss can be reduced at light loads. It becomes possible.

  FIG. 6 is a diagram showing a configuration of a semiconductor integrated circuit used in a switching regulator type DC-DC converter studied by the present inventors prior to the present invention.

  As shown in FIG. 6, the switching regulator type DC-DC converter includes a semiconductor integrated circuit 1 as a PWM control circuit, an N-channel power MOS field effect transistor Q1 as a switching element, a transformer 2, and a rectifying smoothing / output voltage detecting circuit. 3 is included. As shown in FIG. 6, since the primary winding and the secondary winding of the transformer 2 are connected in reverse polarity, the DC-DC converter of the switching regulator type shown in FIG. 6 is configured as the flyback converter described above. Has been.

  The semiconductor integrated circuit 1 that is a PWM control circuit includes a feedback circuit 11, a PWM comparator 12, a hysteresis comparison circuit 13, a triangular wave oscillator 14, an OR logic circuit 15, a control flip-flop 16, and a drive circuit 17.

The input voltage _VIN is supplied as the power supply voltage Vcc of the semiconductor integrated circuit 1 and is connected to the drain of the transistor Q1 through the primary winding of the transformer 2, and the source of the transistor Q1 is connected to the ground potential GND. The feedback circuit 11 includes a resistor R1, a capacitor C3, and a light receiving unit PC2 of a photocoupler. The resistor R1 is connected between the power supply voltage Vcc and the feedback terminal FB, and the capacitor C3 and the light receiving part PC2 of the photocoupler are connected in parallel between the feedback terminal FB and the ground potential GND. The feedback terminal FB is connected to the inverting input terminal − of the PWM comparator 12 and the inverting input terminal − of the hysteresis comparison circuit 13. The reference voltage _VBB is supplied to the non-inverting input terminal + of the hysteresis comparison circuit 13, and the triangular wave oscillator 14 is connected to the non-inverting input terminal + of the PWM comparator 12 and the set input terminal S of the control flip-flop 16. The output terminal of the PWM comparator 12 and the output terminal of the hysteresis comparison circuit 13 are respectively connected to one input terminal and the other input terminal of the OR logic circuit 15, and the output terminal of the OR logic circuit 15 is connected to the control flip-flop 16. Connected to the reset input terminal R. The data output terminal Q of the control flip-flop 16 is connected to the input terminal of the drive circuit 17, and the drive output signal GD generated from the output terminal of the drive circuit 17 is supplied to the gate of the transistor Q1.

  One end of the secondary winding of the transformer 2 is connected to the anode of the rectifier diode D1, the cathode of the rectifier diode D1 is connected to the cathode of the zener diode ZD1 and one end of the smoothing capacitor C2, and the anode of the zener diode ZD1 The other end of the smoothing capacitor C2 and the other end of the light emitting unit PC1 of the photocoupler are connected to the other end of the secondary winding of the transformer 2.

A decoupling capacitor C1 for reducing a ripple component of the input voltage _VIN is connected between the power supply terminal of the power supply voltage Vcc of the semiconductor integrated circuit 1 and the ground potential GND, and the smoothing of the rectification smoothing / output voltage detection circuit 3 is performed. An output voltage _VOUT is generated from between both ends of the capacitor C2. The photocoupler light receiving portion PC2 of the feedback circuit 11 and the photocoupler light emitting portion PC1 of the rectifying / smoothing / output voltage detecting circuit 3 are optically coupled.

  FIG. 7 is a diagram showing waveforms for explaining the operation of the switching regulator type DC-DC converter studied by the present inventors prior to the present invention shown in FIG.

The waveform of the output voltage _VOUT generated from both ends of the smoothing capacitor C2 of the rectifying / smoothing / output voltage detecting circuit 3 of the switching regulator type DC-DC converter shown in FIG. 6 is shown in the upper part of FIG. Yes.

  In the center of FIG. 7, the high level threshold VthH and the low level threshold VthL of the hysteresis comparison circuit 13 of the DC-DC converter of the switching regulator type shown in FIG. 6, and the feedback signal FB of the feedback terminal FB Each waveform with the triangular wave oscillation signal OSC of the triangular wave oscillator 14 is shown.

When the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the threshold voltage of the hysteresis comparison circuit 13 changes from the high level threshold VthH to the low level threshold VthL. A high level set input signal is supplied to the set input terminal S of the control flip-flop 16 in response to the falling of each triangular wave of the triangular wave oscillation signal OSC of the triangular wave oscillator 14, and the control flip-flop 16 is controlled to the set state. Therefore, the data output terminal Q of the control flip-flop 16 changes from the low level to the high level. As a result, in response to the falling of each triangular wave of the triangular wave oscillation signal OSC of the triangular wave oscillator 14, the drive output signal GD generated from the output terminal of the drive circuit 17 also changes from the low level to the high level. When the triangular wave oscillation signal OSC becomes higher than the feedback signal FB, the high-level output signal of the PWM comparator 12 is supplied to the reset input terminal R of the control flip-flop 16 via the OR logic circuit 15, and the control flip-flop 16 Are controlled to the reset state, and the data output terminal Q of the control flip-flop 16 changes from the high level to the low level. Therefore, the drive output signal GD generated from the output terminal of the drive circuit 17 changes from high level to low level. During the high level period of the drive output signal GD, the primary side current of the primary winding of the transformer 2 flows through the transistor Q1. Accordingly, the voltage level of the output voltage _VOUT generated from between both ends of the smoothing capacitor C2 of the rectifying / smoothing / output voltage detecting circuit 3 increases. As a result, the amount of light emitted from the light emitting portion PC1 of the photocoupler of the rectifying / smoothing / output voltage detecting circuit 3 is increased, so that the amount of light received by the light receiving portion PC2 of the photocoupler of the optically coupled feedback circuit 11 is also increased. Therefore, the current of the light receiving part PC of the photocoupler increases, the voltage drop of the resistor R1 of the feedback circuit 11 increases, and the voltage level of the feedback signal FB decreases. Thus, in the waveform shown in FIG. 7, the pulse width of the drive output signal GD gradually decreases due to the decrease in the voltage level of the feedback signal FB, and the voltage level of the output voltage _VOUT reaches the maximum value.

Contrary to the increase of the output voltage _VOUT , the voltage level of the feedback signal FB decreases. When the voltage level of the feedback signal FB decreases below the low level threshold value VthL of the hysteresis comparison circuit 13, the hysteresis comparison circuit 13 The threshold voltage changes from the low level threshold VthL to the high level threshold VthH. Further, when the voltage level of the feedback signal FB falls below the low level threshold value VthL of the hysteresis comparison circuit 13, the high level output signal generated from the hysteresis comparison circuit 13 is controlled via the OR logic circuit 15. 16 reset terminals R are supplied. Therefore, as shown in FIG. 7, the control flip-flop 16 is controlled to be in the reset state during the period T _RESET in which the threshold voltage of the hysteresis comparison circuit 13 is the high level threshold VthH. The data output terminal Q and the drive output signal GD of the drive circuit 17 are maintained at a low level. As a result, since the switching operation of the switching power supply is forcibly stopped during this period T _RESET , the smoothing capacitor C2 is discharged by the load current, and the output voltage _VOUT decreases linearly.

The voltage level of the feedback signal FB increases contrary to the decrease of the output voltage _{VOUT. When} the feedback signal FB rises above the high level threshold value VthH of the hysteresis comparison circuit 13, the threshold value of the hysteresis comparison circuit 13 is increased. The voltage changes from the high level threshold VthH to the low level threshold VthL. Accordingly, as shown in FIG. 7, the drive output signal GD that has been pulse width modulated by voltage comparison between the feedback signal FB and the triangular wave oscillation signal OSC is supplied to the gate of the transistor Q1, and the increase of the output voltage _VOUT is resumed. .

However, in the DC-DC converter shown in FIG. 6, when the switching operation is restarted from the forced stop period T _RESET of the switching operation at a light load, the drive output signal GD having the maximum pulse width is the gate of the transistor Q1. Therefore, the problem that the fluctuation ripple of the output voltage _VOUT is large has been clarified by the study by the present inventor prior to the present invention. As a result, in an extreme case, there arises a problem that the transistor Q1 is destroyed. Further, the problem that “sounding” occurs due to abrupt application of a drive signal having a large pulse width to an inductor or a capacitor has been clarified by examination by the inventors prior to the present invention.

  Means for solving such problems will be described below, but other problems and novel features will become apparent from the description of the present specification and the accompanying drawings.

  The outline of the typical embodiment disclosed in the present application will be briefly described as follows.

  That is, the semiconductor integrated circuit according to the representative embodiment includes a switch element (Q1), a feedback circuit (11), a PWM comparator (12), a hysteresis comparison circuit (13), a triangular wave oscillator (14), and a control drive circuit ( 16, 17), a soft start circuit (18), and a soft start control transistor (Q2).

The hysteresis comparison circuit (13) generates a low threshold voltage (VthL) and a high threshold voltage (VthH), and feeds back by reaching the maximum value of the output voltage (V _OUT ) at light load. In response to the signal (FB) falling below the low threshold voltage (VthL), a control output signal (CNT) for controlling the soft start control transistor (Q2) to be turned on is generated.

The PWM comparator (12) and the control drive circuits (16, 17) are controlled output signals (13) generated from the hysteresis comparison circuit (13) when the output voltage (V _OUT ) reaches the maximum value at light load. In response to CNT), the switching operation of the switch element (Q1) is stopped (see period T _{RESET in} FIG. 2).

The hysteresis comparison circuit (13) responds to the feedback signal (FB) rising above the high threshold voltage (VthH) due to a decrease in the output voltage (V _OUT ) due to the switching operation of the switching element (Q1) being stopped. Thus, a control output signal (CNT) for controlling the soft start control transistor (Q2) to be turned off is generated. The soft start circuit (18), the PWM comparator (12), and the control drive circuit (16, 17) are connected to the control output signal (CNT) of the hysteresis comparison circuit (13) after the switching operation of the switch element (Q1) is stopped. When the switching operation of the switch element (Q1) is resumed in response to the above, a drive output signal (GD) that drives the switch element (Q1) to the ON state is generated. The pulse width of the drive output signal (GD) is controlled from narrow to gradually wide when restarting the switching operation (see FIGS. 1 and 2).

  The following is a brief description of an effect obtained by the typical embodiment of the embodiments disclosed in the present application.

  In other words, according to the present semiconductor integrated circuit, it is possible to reduce the fluctuation ripple of the output voltage when the switching operation is restarted from the forced stop period of the switching operation at a light load of the DC-DC converter of the switching regulator system.

FIG. 1 is a diagram showing a configuration of a semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the first embodiment. FIG. 2 is a diagram showing waveforms for explaining the operation of the semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the first embodiment shown in FIG. FIG. 3 shows the switching regulator according to the first embodiment shown in FIG. 1 in which the change in the pulse width of the drive output signal GD is emphasized by expanding the time width of the time axis of the operation explanation waveform diagram shown in FIG. It is a figure which shows the waveform explaining operation | movement of the semiconductor integrated circuit used for the DC-DC converter of a system. FIG. 4 is a diagram showing a configuration of a semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the second embodiment. FIG. 5 is a diagram showing a configuration of a semiconductor integrated circuit used in the synchronous rectification switching regulator type DC-DC converter according to the third embodiment. FIG. 6 is a diagram showing a configuration of a semiconductor integrated circuit used in a switching regulator type DC-DC converter studied by the present inventors prior to the present invention. FIG. 7 is a diagram showing waveforms for explaining the operation of the switching regulator type DC-DC converter studied by the present inventors prior to the present invention shown in FIG.

1. First, an outline of a typical embodiment disclosed in the present application will be described. The reference numerals of the drawings referred to in parentheses in the outline description of the representative embodiment merely exemplify what is included in the concept of the component to which the reference numeral is attached.

  [1] A semiconductor integrated circuit according to a typical embodiment includes a switch element (Q1: M1, M2), a feedback circuit (11, 22), a PWM comparator (12), a hysteresis comparison circuit (13), and a triangular wave oscillator ( 14), a control drive circuit (16, 17, 24), a soft start circuit (18), and a soft start control transistor (Q2).

The switch elements (Q1: M1, M2) can be connected to an output circuit (3: L, C2), and the output circuit generates an output voltage (V _OUT ) in response to a switching operation of the switch element. .

  The feedback circuit (11, 22) generates a feedback signal (FB) in response to the output voltage generated from the output circuit. The feedback signal is compared with the PWM comparator (12) and the hysteresis comparison. It is supplied to the circuit (13).

  The soft start control transistor (Q2) is connected between the soft start terminal of the soft start circuit (18) and a ground potential (GND), and the ON state and the OFF state of the soft start control transistor (Q2) are The hysteresis comparison circuit (13) is controlled by an on control output signal and an off control output signal (CNT), respectively.

  The soft start circuit includes a charging circuit (CS1) connected between a power supply voltage (Vcc) and the soft start terminal, and a soft start capacitor is connected between the soft start terminal and the ground potential (GND). (C4) is connectable, and the soft start control transistor is controlled to be in an OFF state, thereby generating a soft start signal (SS) from the soft start terminal.

  The triangular wave oscillator (14) generates a triangular wave oscillation signal (OSC), and the triangular wave oscillation signal is supplied to the PWM comparator and the control drive circuit.

  The hysteresis comparison circuit (13) generates a low threshold voltage (VthL) and a high threshold voltage (VthH) that is higher than the low threshold voltage.

  The hysteresis comparison circuit (13) is responsive to the feedback signal falling below the low threshold voltage due to the maximum value of the output voltage at light load, in response to the soft start control transistor (Q2). ) Is generated to control the on state to the on state.

The PWM comparator (12) and the control drive circuit (16, 17, 24) are generated from the hysteresis comparison circuit (13) when the output voltage reaches the maximum value at the time of the light load. In response to the ON control output signal, the switching operation of the switch element is stopped (see period T _{RESET in} FIG. 2).

  The hysteresis comparison circuit (13) is responsive to the feedback signal rising above the high threshold voltage due to a decrease in the output voltage due to the switching operation of the switching element being stopped. The off control output signal for controlling (Q2) to the off state is generated.

  The soft start circuit, the PWM comparator, and the control drive circuit are configured to perform the switching operation of the switch element in response to the OFF control output signal of the hysteresis comparison circuit after the stop of the switching operation of the switch element. When the operation is resumed, a drive output signal (GD) for driving the switch element to the ON state is generated.

  The pulse width of the drive output signal (GD) is controlled from narrow to gradually wide when the switching operation is resumed (see FIGS. 1 and 2).

  According to the embodiment, it is possible to reduce the fluctuation ripple of the output voltage at the time of restarting the switching operation from the forced stop period of the switching operation at the time of light load.

  In a preferred embodiment, when the switching operation is resumed, the PWM comparator and the control drive circuit are generated from the falling edge of the triangular wave oscillation signal generated from the triangular wave oscillator and the hysteresis comparison circuit. In response to the OFF control output signal, the switch element is driven to the ON state.

  Upon restarting the switching operation, the PWM comparator selects a low level signal from the feedback signal of the feedback circuit and the soft start signal of the soft start circuit, and the selected low level signal Further, a comparison output signal for controlling the switch element to the OFF state is generated in response to the triangular wave oscillation signal becoming a high level.

  In response to the comparison output signal generated from the PWM comparator, the control drive circuit drives the switch element to the OFF state (see FIGS. 1 and 2).

  In another preferred embodiment, the switch element (Q1) can be connected to the primary winding of the transformer (2), and the secondary winding of the transformer (2) is connected to the output circuit (3). The output circuit includes a light emitting unit (PC1) of a photocoupler.

  The feedback circuit (11) is characterized in that the output circuit includes a light receiving part (PC2) of the photocoupler optically coupled to the light emitting part (PC1) of the photocoupler (see FIG. 1). ).

  In still another preferred embodiment, the output circuit further includes a rectifier diode (D1) and a smoothing capacitor (C2).

  The anode of the rectifier diode is connected to one end of the secondary winding of the transformer, and one end of the light receiving unit of the photocoupler and one end of the smoothing capacitor are connected to the cathode of the rectifier diode, and the second of the transformer The other end of the next winding is connected to the other end of the light receiving portion of the photocoupler and the other end of the smoothing capacitor (see FIG. 1).

  In a more preferred embodiment, the primary winding and the secondary winding of the transformer are connected in reverse polarity, and the semiconductor integrated circuit, the switch element, the transformer (2), and the output circuit (3 ) Is a feature of constituting a flyback converter (see FIG. 1).

  In another more preferred embodiment, the switch element is constituted by a semiconductor chip of a transistor.

  The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, and the soft start control transistor are integrated on one semiconductor chip of the semiconductor integrated circuit. The

  The semiconductor chip of the transistor constituting the switch element and the one semiconductor chip of the semiconductor integrated circuit are sealed in one package of a system-in-package (SIP). (See FIG. 1).

  In still another more preferred embodiment, the switch element is constituted by a transistor.

  The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, and the transistor constituting the switch element are monolithic semiconductor integrated circuits. It is characterized by being integrated on one chip.

  In another more preferred embodiment, the switch element includes a high side switch (M1) and a low side switch (M2).

  The high side switch and the low side switch are driven complementarily by the control drive circuit (24).

  A switching node (SW) to which the high side switch and the low side switch are connected can be connected to one end of an inductor (L).

The other end of the inductor can be connected to one end of a smoothing capacitor (C2), the other end of the smoothing capacitor is connected to the ground potential (GND), and the other end of the inductor and the one end of the smoothing capacitor An output voltage (V _OUT ) can be generated from the connection node (see FIG. 5).

In still another more preferred embodiment, the PWM comparator (12) and the control drive circuit (24) are configured to compare the hysteresis when the output voltage reaches the maximum value at the light load. In response to the ON control output signal generated from the circuit (13), the switching operation of the high-side switch (M1) is stopped (refer to the period T _{RESET in} FIG. 1).

  In a specific embodiment, the semiconductor integrated circuit generates a detection signal by detecting a voltage increase of the switching node (SW) due to a polarity reversal of a current flowing through the low-side switch (M2). A current detection circuit (25) is further provided.

The control drive circuit (24) responds to the detection signal of the reverse current detection circuit during a period (T _RESET ) during which the switching operation of the high side switch (M1) is stopped. ) Switching operation is stopped (see FIG. 5).

  In another specific embodiment, the semiconductor integrated circuit, the high-side switch, the low-side switch, the inductor, and the smoothing capacitor constitute a synchronous rectification switching regulator. (See FIG. 5).

  In a more specific embodiment, the high-side switch and the low-side switch are respectively configured by a first semiconductor chip of a first transistor and a second semiconductor chip of a second transistor.

  The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, and the reverse current detection circuit are included in one of the semiconductor integrated circuits. Integrated on a semiconductor chip.

  The first semiconductor chip, the second semiconductor chip, and the one semiconductor chip of the semiconductor integrated circuit are sealed in one package of a system-in-package (SIP). Yes (see Figure 1).

  In another more specific embodiment, the high-side switch and the low-side switch are constituted by a first transistor and a second transistor, respectively.

  The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, the reverse current detection circuit, the first transistor, and the second transistor. Is characterized by being integrated on one chip of a monolithic semiconductor integrated circuit.

  [2] A typical embodiment of another aspect is that the switch elements (Q1: M1, M2), the feedback circuit (11, 22), the PWM comparator (12), the hysteresis comparison circuit (13), and the triangular wave oscillator ( 14), a control driving circuit (16, 17, 24), a soft start circuit (18), and a soft start control transistor (Q2).

The switch elements (Q1: M1, M2) can be connected to an output circuit (3: L, C2), and the output circuit generates an output voltage (V _OUT ) in response to a switching operation of the switch element. .

  The feedback circuit (11, 22) generates a feedback signal (FB) in response to the output voltage generated from the output circuit. The feedback signal is compared with the PWM comparator (12) and the hysteresis comparison. It is supplied to the circuit (13).

  The soft start control transistor (Q2) is connected between the soft start terminal of the soft start circuit (18) and a ground potential (GND), and the ON state and the OFF state of the soft start control transistor (Q2) are The hysteresis comparison circuit (13) is controlled by an on control output signal and an off control output signal (CNT), respectively.

  The soft start circuit includes a charging circuit (CS1) connected between a power supply voltage (Vcc) and the soft start terminal, and a soft start capacitor is connected between the soft start terminal and the ground potential (GND). (C4) is connectable, and the soft start control transistor is controlled to be in an OFF state, thereby generating a soft start signal (SS) from the soft start terminal.

  The triangular wave oscillator (14) generates a triangular wave oscillation signal (OSC), and the triangular wave oscillation signal is supplied to the PWM comparator and the control drive circuit.

  The hysteresis comparison circuit (13) generates a low threshold voltage (VthL) and a high threshold voltage (VthH) that is higher than the low threshold voltage.

  The hysteresis comparison circuit (13) is responsive to the feedback signal falling below the low threshold voltage due to the maximum value of the output voltage at light load, in response to the soft start control transistor (Q2). ) Is generated to control the on state to the on state.

The PWM comparator (12) and the control drive circuit (16, 17, 24) are generated from the hysteresis comparison circuit (13) when the output voltage reaches the maximum value at the time of the light load. In response to the ON control output signal, the switching operation of the switch element is stopped (see period T _{RESET in} FIG. 2).

  The hysteresis comparison circuit (13) is responsive to the feedback signal rising above the high threshold voltage due to a decrease in the output voltage due to the switching operation of the switching element being stopped. The off control output signal for controlling (Q2) to the off state is generated.

  The soft start circuit, the PWM comparator, and the control drive circuit are configured to perform the switching operation of the switch element in response to the OFF control output signal of the hysteresis comparison circuit after the stop of the switching operation of the switch element. When the operation is resumed, a drive output signal (GD) for driving the switch element to the ON state is generated.

  The pulse width of the drive output signal (GD) is controlled from narrow to gradually wide when the switching operation is resumed (see FIGS. 1 and 2).

  According to the embodiment, it is possible to reduce the fluctuation ripple of the output voltage at the time of restarting the switching operation from the forced stop period of the switching operation at the time of light load.

2. Details of Embodiment Next, the embodiment will be described in more detail. In all the drawings for explaining the best mode for carrying out the invention, components having the same functions as those in the above-mentioned drawings are denoted by the same reference numerals, and repeated description thereof is omitted.

[Embodiment 1]
<< Configuration of Semiconductor Integrated Circuit Used in DC-DC Converter >>
FIG. 1 is a diagram showing a configuration of a semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the first embodiment.

  As shown in FIG. 1, a switching regulator type DC-DC converter includes a semiconductor integrated circuit 1 as a PWM control circuit, an N-channel power MOS field effect transistor Q1 as a switching element, a transformer 2, and a rectifying smoothing / output voltage detection circuit. 3 is included. As shown in FIG. 1, the primary winding and the secondary winding of the transformer 2 are connected in opposite polarities, so that the switching regulator type DC-DC converter shown in FIG. 1 is configured as the flyback converter described above. Is done. The semiconductor chip of the semiconductor integrated circuit 1 as the PWM control circuit and the semiconductor chip of the N-channel power MOS field effect transistor Q1 are semiconductor devices sealed in one resin package. This semiconductor device is a hybrid type semiconductor integrated circuit called a system in package (SIP) or a multi-chip module (MCP) in the semiconductor industry.

  The semiconductor integrated circuit 1 which is a PWM control circuit includes a feedback circuit 11, a PWM comparator 12, a hysteresis comparison circuit 13, a triangular wave oscillator 14, a control flip-flop 16, a drive circuit 17, and a soft start circuit 18.

The input voltage _VIN is supplied as the power supply voltage Vcc of the semiconductor integrated circuit 1 and is connected to the drain of the transistor Q1 through the primary winding of the transformer 2, and the source of the transistor Q1 is connected to the ground potential GND. The feedback circuit 11 includes a resistor R1, a capacitor C3, and a light receiving unit PC2 of a photocoupler. The resistor R1 is connected between the power supply voltage Vcc and the feedback terminal FB, and the capacitor C3 and the light receiving part PC2 of the photocoupler are connected in parallel between the feedback terminal FB and the ground potential GND. The feedback terminal FB is connected to the first inverting input terminal − of the PWM comparator 12 and the inverting input terminal − of the voltage comparator COMP1 of the hysteresis comparison circuit 13. In the hysteresis comparison circuit 13, a second constant current circuit CS2 is connected between the power supply voltage Vcc and the non-inverting input terminal + of the voltage comparator COMP1, and between the non-inverting input terminal + of the voltage comparator COMP1 and the ground potential GND. Is connected to a resistor R2 which is an external component. The constant current value of the second constant current circuit CS2 is variably set by the output control signal CNT of the voltage comparator COMP1. As an initial state, the output control signal CNT is set to a low level, so that the constant current value of the second constant current circuit CS2 is set to a small current, and the voltage across the resistor R2 is set to the low threshold voltage VthL. The When the feedback signal FB falls below the low threshold voltage VthL, the output control signal CNT of the voltage comparator COMP1 becomes high level, the constant current value of the second constant current circuit CS2 is set to a large current, and the terminal of the resistor R2 The inter-voltage is set to the high threshold voltage VthH. The two threshold values VthL and VthH described above of the voltage comparator COMP1 can be adjusted by the resistance value of the resistor R2. That is, the resistance value of the resistor R2 can be set to a desired burst mode pause operation (switching operation forced stop period T _RESET ) so as to correspond to the specifications of the DC-DC converter. Further, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the source and drain of the transistor Q2 are connected to the ground potential GND and the soft start circuit 18. Each is connected to a soft start terminal SS. The soft start circuit 18 includes a first constant current circuit CS1 connected between the power supply voltage Vcc and the soft start terminal SS, and a capacitor C4 connected between the soft start terminal SS and the ground potential GND. The soft start terminal SS of the soft start circuit 18 is connected to the second inverting input terminal − of the PWM comparator 12. The triangular wave oscillator 14 is connected to the non-inverting input terminal + of the PWM comparator 12 and the set input terminal S of the control flip-flop 16. The output terminal of the PWM comparator 12 is connected to the reset input terminal R of the control flip-flop 16, and the data output terminal Q of the control flip-flop 16 is connected to the input terminal of the drive circuit 17 and is generated from the output terminal of the drive circuit 17. The drive output signal GD is supplied to the gate of the transistor Q1.

  In the rectifying / smoothing / output voltage detecting circuit 3, one end of the secondary winding of the transformer 2 is connected to the anode of the rectifying diode D1, and the cathode of the rectifying diode D1 is connected to the cathode of the Zener diode ZD1 and one end of the smoothing capacitor C2. . The anode of the Zener diode ZD1 is connected to one end of the light emitting unit PC1 of the photocoupler, and the other end of the smoothing capacitor C2 and the other end of the light emitting unit PC1 of the photocoupler are connected to the other end of the secondary winding of the transformer 2.

A decoupling capacitor C1 for reducing a ripple component of the input voltage _VIN is connected between the power supply terminal of the power supply voltage Vcc of the semiconductor integrated circuit 1 and the ground potential GND, and the smoothing of the rectification smoothing / output voltage detection circuit 3 is performed. An output voltage _VOUT is generated from between both ends of the capacitor C2. The photocoupler light receiving portion PC2 of the feedback circuit 11 and the photocoupler light emitting portion PC1 of the rectifying / smoothing / output voltage detecting circuit 3 are optically coupled.

<Operation explanation waveform diagram>
FIG. 2 is a diagram showing waveforms for explaining the operation of the semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the first embodiment shown in FIG.

  As can be understood from FIG. 2, in order to facilitate the understanding of the operation of the DC-DC converter according to the first embodiment, compared with the frequency of the triangular wave oscillation signal OSC having the waveform shown in FIG. The triangular wave oscillation signal OSC shown in the waveform is set at a high frequency.

  FIG. 3 shows the switching regulator according to the first embodiment shown in FIG. 1 in which the change in the pulse width of the drive output signal GD is emphasized by expanding the time width of the time axis of the operation explanation waveform diagram shown in FIG. It is a figure which shows the waveform explaining operation | movement of the semiconductor integrated circuit used for the DC-DC converter of a system.

In the upper part of FIG. 2, the output voltage V _OUT generated from both ends of the smoothing capacitor C2 of the rectification smoothing / output voltage detection circuit 3 of the DC-DC converter of the switching regulator type according to the first embodiment shown in FIG. The waveform is shown. When the load is light, since the load current flowing into the load connected between both ends of the smoothing capacitor C2 is small, the increase rate of the output voltage _VOUT increases, and the output voltage _VOUT reaches the maximum value in a short time. . The maximum value of the output voltage _VOUT described here means the maximum value of the output voltage _VOUT in the waveform of FIG. The maximum value of the output voltage _VOUT during normal operation is generated at the timing when the feedback signal FB becomes lower than the low threshold voltage VthL and the pulse width of the high level pulse of the drive output signal GD becomes zero.

Before the output voltage _VOUT reaches the maximum value, the pulse of the drive output signal GD for driving the gate of the transistor Q1 is gradually widened from the narrow as shown in the lower part of FIG. Then, the width of the output voltage _VOUT is gradually decreased from the wide width to reduce the ripple of the output voltage _VOUT .

When the output voltage _VOUT reaches the maximum value, the feedback signal FB falls below the low level threshold value VthL of the hysteresis comparison circuit 13, and the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changes from the low level. By changing to the high level, the hysteresis comparison circuit 13 changes from the low level threshold value VthL to the high level threshold value VthH. Accordingly, the N-channel MOS field effect transistor Q2 for soft start control changes from the OFF state to the ON state, and the soft start signal SS at the soft start terminal SS of the soft start circuit 18 suddenly increases from the power supply voltage V _DD to the ground potential. Decreases to GND. Accordingly, in response to the ground potential GND of the soft start signal SS, a high level output signal is supplied from the output terminal of the PWM comparator 12 to the reset terminal R of the control flip-flop 16, so that the threshold value of the hysteresis comparison circuit 13 is increased. The control flip-flop 16 is controlled to be in the reset state during the period T _TRESET where the voltage is the high level threshold VthH. Therefore, the data output terminal Q of the control flip-flop 16 and the drive output signal GD of the drive circuit 17 are maintained at a low level. As a result, during this period T _RESET , the switching operation of the switching power supply is forcibly stopped. Therefore, the switching operation of the switching power supply is stopped, and the switching frequency at light load can be effectively reduced. During this period T _RESET , the smoothing capacitor C2 is discharged by the load current, and the output voltage _VOUT decreases linearly.

The voltage level of the feedback signal FB increases contrary to the decrease of the output voltage _{VOUT. When} the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the output control signal of the voltage comparator COMP1. CNT becomes low level, and the hysteresis comparison circuit 13 is set from the high level threshold value VthH to the low level threshold value VthL. Since the low level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, the transistor Q2 is controlled to be in the off state. The soft start terminal SS starts to rise from the ground potential GND toward the power supply voltage V _DD . Therefore, the PWM comparator 12 selects a low level signal from the feedback signal FB and the soft start signal SS, and outputs a drive output signal GD that has been pulse width modulated by voltage comparison between the selected low level signal and the triangular wave oscillation signal OSC. To be generated. As a result, as shown in the lower part of FIG. 2, the pulse of the drive output signal GD for driving the gate of the transistor Q1 is controlled from narrow to gradually wide, and then from wide to gradually narrow to output voltage V _OUT Ripple is reduced.

  In the center of FIG. 2, the high level threshold value VthH and the low level threshold value VthL of the hysteresis comparison circuit 13 of the switching regulator type DC-DC converter according to the first embodiment shown in FIG. 1, and the feedback terminal FB The waveforms of the feedback signal FB, the triangular wave oscillation signal OSC of the triangular wave oscillator 14 and the soft start signal SS of the soft start terminal SS are shown.

When the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changes from high level to low level. Therefore, since the constant current value of the second constant current circuit CS2 changes from a large current to a small current, the threshold voltage of the hysteresis comparison circuit 13 changes from the high level threshold VthH to the low level threshold VthL. Further, in response to the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changing from the high level to the low level, the N-channel MOS field effect transistor Q2 for soft start control changes from the on state to the off state. To do. Therefore, the soft start signal SS at the soft start terminal SS of the soft start circuit 18 starts to rise from the ground potential GND toward the power supply voltage V _DD .

  A high level set input signal is supplied to the set input terminal S of the control flip-flop 16 in response to the falling of each triangular wave of the triangular wave oscillation signal OSC of the triangular wave oscillator 14, and the control flip-flop 16 is controlled to the set state. Therefore, the data output terminal Q of the control flip-flop 16 changes from the low level to the high level. As a result, in response to the falling of each triangular wave of the triangular wave oscillation signal OSC of the triangular wave oscillator 14, the drive output signal GD generated from the output terminal of the drive circuit 17 also changes from the low level to the high level.

The PWM comparator 12 selects a low level signal from the feedback signal FB supplied to the first inverting input terminal − and the soft start signal SS supplied to the second inverting input terminal −. As shown in FIG. 2, since the soft start signal SS is a lower level signal than the feedback signal FB at first, the PWM comparator 12 first selects the soft start signal SS as a low level signal. Therefore, the PWM comparator 12 generates a high-level output signal of the PWM comparator 12 and resets the control flip-flop 16 when the triangular wave oscillation signal OSC becomes a high level from the soft start signal SS as the selected low-level signal. Supply to input terminal R. As a result, the control flip-flop 16 is controlled to the reset state, and the data output terminal Q of the control flip-flop 16 changes from the high level to the low level. Therefore, the drive output signal GD generated from the output terminal of the drive circuit 17 changes from high level to low level. During the high level period of the drive output signal GD, the primary current of the primary winding of the transformer 2 flows through the transistor Q1. Accordingly, the voltage level of the output voltage _VOUT generated from between both ends of the smoothing capacitor C2 of the rectifying / smoothing / output voltage detecting circuit 3 increases. During this time, the pulse width of the drive output signal GD gradually increases. As a result, the amount of light emitted from the light emitting portion PC1 of the photocoupler of the rectifying / smoothing / output voltage detecting circuit 3 is increased, so that the amount of light received by the light receiving portion PC2 of the photocoupler of the optically coupled feedback circuit 11 is also increased. Accordingly, the current of the light receiving part PC2 of the photocoupler increases, the voltage drop of the resistor R1 of the feedback circuit 11 increases, and the voltage level of the feedback signal FB decreases.

As a result, since the feedback signal FB is a lower level signal than the soft start signal SS, the PWM comparator 12 selects the feedback signal FB as a low level signal. Accordingly, when the triangular wave oscillation signal OSC becomes higher than the feedback signal FB as the selected low level signal, the PWM comparator 12 generates a high level output signal of the PWM comparator 12 and resets the control flip-flop 16. Supply to input terminal R. As a result, the control flip-flop 16 is controlled to the reset state, and the data output terminal Q of the control flip-flop 16 changes from the high level to the low level. Accordingly, the drive output signal GD generated from the output terminal of the drive circuit 17 changes from the high level to the low level, and the primary side of the primary winding of the transformer 2 is provided in the transistor Q1 during the high level period of the drive output signal GD. Current flows. As the voltage level of the feedback signal FB decreases, the pulse width of the drive output signal GD gradually decreases, and the voltage level of the output voltage _VOUT reaches the maximum value.

When the feedback signal FB falls below the low threshold voltage VthL, the output control signal CNT of the voltage comparator COMP1 becomes high level, the constant current value of the second constant current circuit CS2 is set to a large current, and the resistance R2 is connected between the terminals of the resistor R2. The voltage is set to the high threshold voltage VthH. Further, the high level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the transistor Q2 is maintained in the ON state. The soft start terminal SS of the circuit 18 is maintained at the ground potential GND. In response to the ground potential GND of the soft start terminal SS, a high level output signal is supplied from the PWM comparator 12 to the reset terminal R of the control flip-flop 16. Therefore, as shown in FIG. 2, the control flip-flop 16 is controlled to be in the reset state during the period _TRESET when the threshold voltage of the hysteresis comparison circuit 13 is the high level threshold VthH. The output terminal Q and the drive output signal GD of the drive circuit 17 are maintained at a low level. As a result, since the switching operation of the switching power supply is forcibly stopped during this period T _RESET , the smoothing capacitor C2 is discharged by the load current, and the output voltage _VOUT decreases linearly.

The voltage level of the feedback signal FB increases contrary to the decrease of the output voltage _{VOUT. When} the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the output control signal of the voltage comparator COMP1. CNT becomes low level, the constant current value of the second constant current circuit CS2 is set to a small current, and the voltage between the terminals of the resistor R2 is set to the low level threshold VthL. Further, the low level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the transistor Q2 is controlled to be in the off state. The 18 soft start terminals SS start to rise from the ground potential GND toward the power supply voltage V _DD . Therefore, the PWM comparator 12 selects a low level signal from the feedback signal FB and the soft start signal SS, and outputs a drive output signal GD that has been pulse width modulated by voltage comparison between the selected low level signal and the triangular wave oscillation signal OSC. To be generated.

<< Effects of Embodiment 1 >>
Therefore, according to the DC-DC converter according to the first embodiment shown in FIG. 1, when the switching operation is restarted (restarted) from the forced stop period T _RESET of the switching operation at a light load as shown in FIG. In addition, it becomes possible to supply the drive output signal GD, whose pulse width is controlled from narrow to gradually wide, to the gate of the transistor Q1. Therefore, according to the DC-DC converter according to the first embodiment shown in FIG. 1, it is possible to reduce the fluctuation ripple of the output voltage _VOUT . Furthermore, the DC-DC converter according to the first embodiment shown in FIG. 1 can solve the problem that the transistor Q1 is destroyed and the problem that noise is generated. .

<Operation of soft start circuit when power supply voltage is turned on>
In the DC-DC converter according to Embodiment 1 shown in FIG. 1, immediately after the power supply voltage V _DD is turned on, the voltage level of the feedback signal FB of the feedback circuit 11 is low because of the time constant of the resistor R1 and the capacitor C3. On the other hand, the high level threshold VthH is supplied to the non-inverting input terminal + of the voltage comparator COMP1. Therefore, the soft start control N-channel MOS field effect transistor Q2 is maintained in the ON state by the high level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13, and the soft start terminal SS of the soft start circuit 18 is softened. The start signal SS is maintained at the ground potential GND. Thereafter, the voltage level of the feedback signal FB of the feedback circuit 11 rises due to the time constant of the resistor R1 and the capacitor C3, so that the voltage level of the feedback signal FB is higher than the high level threshold value VthH in the hysteresis comparison circuit 13. It becomes. As a result, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changes from high level to low level, and the soft start control N-channel MOS field effect transistor Q2 changes from on to off. Therefore, the soft start signal SS at the soft start terminal SS of the soft start circuit 18 starts to rise from the ground potential GND toward the power supply voltage V _DD . As a result, it is possible to gradually increase the pulse width of the drive output signal GD immediately after turning on the power supply voltage. Therefore, the DC-DC converter according to the first embodiment shown in FIG. 1 can reduce the fluctuation ripple of the output voltage _VOUT when the power supply voltage is turned on. Furthermore, according to the DC-DC converter according to the first embodiment shown in FIG. 1, it is possible to solve the problem that the transistor Q1 is destroyed when the power supply voltage is turned on and the problem that noise is generated. It will be.

[Embodiment 2]
FIG. 4 is a diagram showing a configuration of a semiconductor integrated circuit used in the switching regulator type DC-DC converter according to the second embodiment.

  The semiconductor integrated circuit 1 according to the second embodiment shown in FIG. 4 is different from the semiconductor integrated circuit 1 according to the first embodiment shown in FIG. 1 in the internal circuit configuration of the hysteresis comparison circuit 13, and the other points are the same. It is.

That is, the hysteresis comparison circuit 13 of the semiconductor integrated circuit 1 according to the second embodiment shown in FIG. 4 includes the voltage comparator COMP1, the inverter Inv, the N channel MOS field effect transistors Q3 and Q6, the P channel MOS field effect transistors Q4 and Q5, A reference voltage generator _VBB and resistors R2, R3, and R4 are included.

The source of the transistor Q3 is connected through a resistor R2 to the ground potential GND, and the gate of the transistor Q3 is connected to the reference voltage generator V _BB, the drain of the transistor Q3 is connected to the drain of the transistor Q4. Since the sources of the transistors Q4 and Q5 are connected to the power supply voltage Vcc, the gate and drain of the transistor Q4 are connected, and are connected to both gates of both the transistors Q4 and Q5, the transistors Q4 and Q5 are current mirrors. Functions as an input transistor and an output transistor. As a result, an output constant current proportional to the input constant current flowing through the series path of the transistor Q4, transistor Q3, and resistor R2 flows through the source / drain current path of the transistor Q5.

  As an initial state, the output control signal CNT is set to a low level, so that the output signal of the inverter Inv is set to a high level, and the transistor Q6 is turned on. Therefore, the output constant current of the transistor Q5 The hysteresis comparison circuit 13 is set to the low threshold voltage VthL. After that, when the feedback signal FB falls below the low threshold voltage VthL, the output control signal CNT of the voltage comparator COMP1 becomes high level and the transistor Q6 is turned off, so that the output constant current of the transistor Q5 becomes resistance. The hysteresis comparison circuit 13 is set to the high threshold voltage VthH.

  The other configuration and operation of the semiconductor integrated circuit 1 according to the second embodiment shown in FIG. 4 are the same as those of the semiconductor integrated circuit 1 according to the first embodiment shown in FIG.

[Embodiment 3]
FIG. 5 is a diagram showing a configuration of a semiconductor integrated circuit used in the synchronous rectification switching regulator type DC-DC converter according to the third embodiment.

  A switching regulator type DC-DC converter according to Embodiment 3 shown in FIG. 5 includes a semiconductor integrated circuit 1 as a PWM control circuit, an N-channel power MOS field effect transistor M1 of a high-side switch, and an N-channel power MOS of a low-side switch. The field effect transistor M2, the inductor L, and the smoothing capacitor C2 are included. The semiconductor chip of the semiconductor integrated circuit 1 as the PWM control circuit, the semiconductor chip of the N-channel power MOS field effect transistor M1, and the semiconductor chip of the N-channel power MOS field effect transistor M2 are included in the system in package (SIP). It is sealed in one resin package. The PWM control circuit, the N-channel power MOS field effect transistor M1, and the N-channel power MOS field effect transistor M2 are integrated on one semiconductor chip, and the semiconductor chip is sealed in one resin package. It is also possible.

  The semiconductor integrated circuit 1 that is a PWM control circuit includes a PWM comparator 12, a hysteresis comparison circuit 13, a triangular wave oscillator 14, a soft start circuit 18, a feedback circuit 22, an error amplifier 23, a control drive circuit 24, and a reverse current detection circuit 25. Is included.

The input voltage _VIN is supplied to the drain of the transistor M1, the source of the transistor M1 and the drain of the transistor M2 are connected to the switching node SW, and the source of the transistor M2 is connected to the ground potential GND. The switching node SW is connected to one end of the inductor L, the other end of the inductor L is connected to one end of the smoothing capacitor C2, the other end of the smoothing capacitor C2 is connected to the ground potential GND, and the other end of the inductor L and the smoothing capacitor C2 An output voltage _VOUT is generated from a connection node to one end of the output voltage _VOUT .

The output voltage _VOUT is supplied to the feedback circuit 22, and the two voltage dividing resistors R5 and R6 of the feedback circuit 22 divide the output voltage _VOUT . The divided output voltage generated from the feedback circuit 22 is supplied to the inverting input terminal − of the error amplifier 23, the reference voltage Vref is supplied to the non-inverting input terminal + of the error amplifier 23, and the feedback signal is output from the output terminal of the error amplifier 23. FB is generated. A phase compensation circuit constituted by a series connection of a resistor R7 and a capacitor C5 is connected between the inverting input terminal − and the output terminal of the error amplifier 23, and also between the output terminal of the error amplifier 23 and the ground potential GND. The capacitor C6 constituting the phase compensation circuit is connected. The feedback signal FB generated from the output terminal of the error amplifier 23 is connected to the first inverting input terminal of the PWM comparator 12 and the inverting input terminal of the voltage comparator COMP1 of the hysteresis comparison circuit 13.

  In the hysteresis comparison circuit 13, a second constant current circuit CS2 is connected between the power supply voltage Vcc and the non-inverting input terminal + of the voltage comparator COMP1, and between the non-inverting input terminal + of the voltage comparator COMP1 and the ground potential GND. Is connected to a resistor R2 which is an external component. The constant current value of the second constant current circuit CS2 is variably set by the output control signal CNT of the voltage comparator COMP1. As an initial state, the output control signal CNT is set to a low level, so that the constant current value of the second constant current circuit CS2 is set to a small current, and the voltage across the resistor R2 is set to the low threshold voltage VthL. Is done. When the feedback signal FB falls below the low threshold voltage VthL, the output control signal CNT of the voltage comparator COMP1 becomes high level, the constant current value of the second constant current circuit CS2 is set to a large current, and the terminal of the resistor R2 The inter-voltage is set to the high threshold voltage VthH. Further, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the source and drain of the transistor Q2 are connected to the ground potential GND and the soft start circuit 18. To the soft start terminal SS. The soft start circuit 18 includes a first constant current circuit CS1 connected between the power supply voltage Vcc and the soft start terminal SS, and a capacitor C4 connected between the soft start terminal SS and the ground potential GND. The soft start terminal SS of the soft start circuit 18 is connected to the second inverting input terminal − of the PWM comparator 12. The triangular wave oscillator 14 is connected to the non-inverting input terminal + of the PWM comparator 12 and the control drive circuit 24. The output terminal of the PWM comparator 12 is connected to the control drive circuit 24. The control drive circuit 24 connects the gate of the N-channel power MOS field effect transistor M1 of the high side switch and the gate of the N channel power MOS field effect transistor M2 of the low side switch. To drive. The drive by the control drive circuit 24 causes the low side switch M2 to be off during the on period of the high side switch M1, and the control drive circuit 24 drives the high side switch M1 and the low side switch M2 in a complementary manner. Is. On the other hand, in the case of a light load, the control drive circuit 24 controls both the high side switch M1 and the low side switch M2 to be in an off state. The off control of both the switches is an operation mode in which the current of the inductor L is not continuous, and is therefore called a current discontinuous mode (DCM).

During the ON period of the high side switch M1, an inductor current flows from the input voltage _VIN of the drain of the high side switch M1 to the smoothing capacitor C2 through the inductor L. On the contrary, during the ON period of the low side switch M2, The inductor current flows from the ground potential GND of the source of the low-side switch M2 to the smoothing capacitor C2 through the inductor L. Therefore, the voltage level of the output voltage _VOUT is determined by the on period of the high side switch M1 and the on period of the low side switch M2.

  Even when the on period of the low side switch M2 is shorter than the on period of the high side switch M1, the energy stored in the inductor L is charged to the smoothing capacitor C2 through the body diode formed by the parasitic diode of the low side switch M2. Is performed. On the contrary, when the on period of the low side switch M2 is longer than the on period of the high side switch M1, a reverse current tends to flow through the inductor L after the discharge of the energy stored in the inductor L is completed. However, the reverse current detection circuit 25 detects the reverse current and performs an operation of controlling the low-side switch M2 from the on state to the off state.

<Operation explanation waveform diagram>
FIG. 2 is a diagram showing waveforms for explaining the operation of the semiconductor integrated circuit used in the synchronous rectification switching regulator type DC-DC converter according to the third embodiment shown in FIG.

2 shows the waveform of the output voltage _VOUT generated from both ends of the smoothing capacitor C2 of the DC-DC converter of the synchronous rectification type switching regulator system according to the third embodiment shown in FIG. Yes.

  2, the high level threshold VthH and the low level threshold VthL of the hysteresis comparison circuit 13 of the synchronous rectification type switching regulator DC-DC converter according to the third embodiment shown in FIG. Each waveform of the feedback signal FB, the triangular wave oscillation signal OSC of the triangular wave oscillator 14 and the soft start signal SS of the soft start terminal SS is shown.

When the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changes from high level to low level. Therefore, since the constant current value of the second constant current circuit CS2 changes from a large current to a small current, the threshold voltage of the hysteresis comparison circuit 13 changes from the high level threshold VthH to the low level threshold VthL. Further, in response to the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changing from the high level to the low level, the N-channel MOS field effect transistor Q2 for soft start control changes from the on state to the off state. To do. Therefore, the soft start signal SS at the soft start terminal SS of the soft start circuit 18 starts to rise from the ground potential GND toward the power supply voltage V _DD .

  The gate drive signal GD of the high side switch M1 generated from the control drive circuit 24 in response to the falling of each triangular wave of the triangular wave oscillation signal OSC of the triangular wave oscillator 14 changes from low level to high level.

The PWM comparator 12 selects a low level signal from the feedback signal FB supplied to the first inverting input terminal − and the soft start signal SS supplied to the second inverting input terminal −. As shown in FIG. 2, since the soft start signal SS is a lower level signal than the feedback signal FB at first, the PWM comparator 12 first selects the soft start signal SS as a low level signal. Therefore, when the triangular wave oscillation signal OSC becomes high level from the soft start signal SS as the selected low level signal, the PWM comparator 12 generates a high level output signal of the PWM comparator 12 and is generated from the control drive circuit 24. The gate drive signal GD of the high side switch M1 changes from a high level to a low level. According to the ratio of the on period of the high side switch M1 and the on period of the low side switch M2, the voltage level of the output voltage _VOUT generated from both ends of the smoothing capacitor C2 increases. On the contrary, the voltage level of the feedback signal FB at the output terminal of the error amplifier 23 decreases.

  As a result, since the feedback signal FB is a lower level signal than the soft start signal SS, the PWM comparator 12 selects the feedback signal FB as a low level signal. Therefore, the PWM comparator 12 generates a high-level output signal of the PWM comparator 12 and supplies it to the control drive circuit 24 when the triangular wave oscillation signal OSC becomes higher than the feedback signal FB as the selected low-level signal. To do.

As a result, the gate drive signal GD of the high side switch M1 generated from the control drive circuit 24 changes from the high level to the low level. According to the ratio between the on period of the high side switch M1 and the on period of the low side switch M2, the voltage level of the output voltage _VOUT generated between both ends of the smoothing capacitor C2 reaches the maximum value.

When the feedback signal FB falls below the low threshold voltage VthL, the output control signal CNT of the voltage comparator COMP1 becomes high level, the constant current value of the second constant current circuit CS2 is set to a large current, and the resistance R2 is connected between the terminals of the resistor R2. The voltage is set to the high threshold voltage VthH. Further, the high level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the transistor Q2 is maintained in the ON state. The soft start terminal SS of the circuit 18 is maintained at the ground potential GND. In response to the ground potential GND of the soft start terminal SS, a high level output signal is supplied from the PWM comparator 12 to the control drive circuit 24. Therefore, as shown in FIG. 2, during the period T _RESET in which the threshold voltage of the hysteresis comparison circuit 13 is the high level threshold VthH, the gate drive signal GD of the high side switch M1 generated from the control drive circuit 24 is , Maintained at a low level. On the other hand, during the on period of the low side switch M2, which is the off period of the high side switch M1, the inductor current flowing through the inductor L in the direction of the smoothing capacitor C2 from the ground potential GND of the source of the low side switch M2 has a negative current value. There is a possibility. That is, polarity inversion of the current flowing through the low-side switch M2 occurs, and the drain voltage of the switching node SW, that is, the low-side switch M2 increases. The increase in the drain voltage of the low side switch M2 is detected by the reverse current detection circuit 25, and the gate drive signal of the low side switch M2 generated from the control drive circuit 24 in the period T _RESET in response to the detection signal of the reverse current detection circuit 25 Is maintained at a low level. As a result, since the switching operation of the switching power supply is forcibly stopped during this period T _RESET , the smoothing capacitor C2 is discharged by the load current, and the output voltage _VOUT decreases linearly.

The voltage level of the feedback signal FB increases contrary to the decrease of the output voltage _{VOUT. When} the feedback signal FB rises above the high level threshold VthH of the hysteresis comparison circuit 13, the output control signal of the voltage comparator COMP1. CNT becomes low level, the constant current value of the second constant current circuit CS2 is set to a small current, and the voltage between the terminals of the resistor R2 is set to the low level threshold VthL. Further, the low level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 is supplied to the gate of the N-channel MOS field effect transistor Q2 for soft start control, and the transistor Q2 is controlled to be in the off state. The 18 soft start terminals SS start to rise from the ground potential GND toward the power supply voltage V _DD . Therefore, the PWM comparator 12 selects a low level signal from the feedback signal FB and the soft start signal SS, and outputs a drive output signal GD that has been pulse width modulated by voltage comparison between the selected low level signal and the triangular wave oscillation signal OSC. To be generated.

Therefore, according to the DC-DC converter according to the third embodiment shown in FIG. 5, when the switching operation is restarted (restarted) from the forced stop period T _RESET of the switching operation at a light load as shown in FIG. In addition, it becomes possible to supply the drive output signal GD, whose pulse width is controlled from narrow to gradually wide, to the gate of the transistor Q1. Therefore, according to the DC-DC converter according to the third embodiment shown in FIG. 5, it is possible to reduce the fluctuation ripple of the output voltage _VOUT . Furthermore, the DC-DC converter according to the third embodiment shown in FIG. 5 can solve the problem that the transistor Q1 is destroyed and the problem that noise is generated. .

<Operation of soft start circuit when power supply voltage is turned on>
In the DC-DC converter according to the third embodiment shown in FIG. 5, immediately after the power supply voltage V _DD is turned on, the voltage level of the feedback signal FB is the resistance R7 connected to the error amplifier 23 and the capacitors C5 and C6. While being maintained at a low level by a constant, the high level threshold VthH is supplied to the non-inverting input terminal + of the voltage comparator COMP1. Therefore, the soft start control N-channel MOS field effect transistor Q2 is maintained in the ON state by the high level output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13, and the soft start terminal SS of the soft start circuit 18 is softened. The start signal SS is maintained at the ground potential GND. Thereafter, the voltage level of the feedback signal FB rises due to the time constants of the resistor R7 and the capacitors C5 and C6 connected to the error amplifier 23, so that the voltage level of the feedback signal FB is the high level threshold VthH in the hysteresis comparison circuit 13. Become a higher level. As a result, the output control signal CNT of the voltage comparator COMP1 of the hysteresis comparison circuit 13 changes from high level to low level, and the soft start control N-channel MOS field effect transistor Q2 changes from on to off. Therefore, the soft start signal SS at the soft start terminal SS of the soft start circuit 18 starts to rise from the ground potential GND toward the power supply voltage V _DD . As a result, it is possible to gradually increase the pulse width of the drive output signal GD immediately after turning on the power supply voltage. Therefore, according to the DC-DC converter according to the third embodiment shown in FIG. 5, the fluctuation ripple of the output voltage _VOUT can be reduced when the power supply voltage is turned on. Furthermore, according to the DC-DC converter according to the third embodiment shown in FIG. 5, it is possible to solve the problem that the transistor Q1 is destroyed when the power supply voltage is turned on and the problem that sound is generated. It will be.

  As mentioned above, the invention made by the present inventor has been specifically described based on various embodiments. However, the present invention is not limited thereto, and various modifications can be made without departing from the scope of the invention. Needless to say.

  For example, the switch element Q1 in FIG. 1 and the high-side switch M1 and low-side switch M2 in FIG. 5 are not limited to N-channel power MOS transistors. For example, these transistors may be constituted by N-channel insulated gate bipolar transistors (IGBTs). As is well known, an insulated gate bipolar transistor (IGBT) has an insulated gate MOS transistor structure with a high input impedance and a bipolar transistor structure with a collector / emitter current path with a low output impedance at the input and output sections, respectively. I have it.

  The semiconductor integrated circuit IC for configuring the switching regulator type DC-DC converter of FIGS. 1, 4 and 5 is a hybrid semiconductor integrated circuit IC configured in a system-in-package (SIP) form. It is not limited only. For example, this semiconductor integrated circuit IC is integrated in one semiconductor chip of a monolithic semiconductor integrated circuit in which the switching element Q1 of FIG. 1 and the high-side switch M1 and low-side switch M2 of FIG. 5 are integrated. Is also possible.

DESCRIPTION OF SYMBOLS 1 ... Semiconductor integrated circuit 2 ... Transformer 3 ... Rectification smoothing and output voltage detection circuit Q1 ... N channel power MOS field effect transistor 11 ... Feedback circuit 12 ... PWM comparator 13 ... Hysteresis comparison circuit 14 ... Triangular wave oscillator 16 ... Control flip-flop 17 ... Drive circuit 18 ... Soft start circuit
V _IN ... input voltage Vcc ... power supply voltage V _OUT ... output voltage R1, R2 ... resistors C1, C2, C3, C4 ... capacitance D1 ... rectifier diode ZD1 ... zener diode PC1 ... photocoupler light emitting part PC2 ... photocoupler light receiving part COMP1 ... Voltage comparator CS1 ... first constant current circuit CS2 ... second constant current circuit SS ... soft start terminal FB ... feedback terminal OSC ... triangular wave oscillation signal VthH ... high threshold voltage VthL ... low threshold voltage

Claims (20)

The semiconductor integrated circuit includes a switch element, a feedback circuit, a PWM comparator, a hysteresis comparison circuit, a triangular wave oscillator, a control drive circuit, a soft start circuit, and a soft start control transistor.
The switch element is connectable to an output circuit, and the output circuit generates an output voltage in response to a switching operation of the switch element,
The feedback circuit generates a feedback signal in response to the output voltage generated from the output circuit, and the feedback signal is supplied to the PWM comparator and the hysteresis comparison circuit;
The soft start control transistor is connected between a soft start terminal of the soft start circuit and a ground potential, and an on state and an off state of the soft start control transistor are an on control output signal and an off control of the hysteresis comparison circuit. Each controlled by output signal,
The soft start circuit includes a charging circuit connected between a power supply voltage and the soft start terminal, and a soft start capacitor is connectable between the soft start terminal and the ground potential. A soft start signal is generated from the soft start terminal by controlling the control transistor to be in an off state,
The triangular wave oscillator generates a triangular wave oscillation signal, and the triangular wave oscillation signal is supplied to the PWM comparator and the control drive circuit,
The hysteresis comparison circuit generates a low threshold voltage and a high threshold voltage that is higher than the low threshold voltage.
The hysteresis comparison circuit sets the soft start control transistor to the on state in response to the feedback signal falling below the low threshold voltage due to reaching the maximum value of the output voltage at light load. Generating the on-control output signal to control,
The PWM comparator and the control drive circuit are responsive to the on-control output signal generated from the hysteresis comparison circuit upon reaching the maximum value of the output voltage at the light load. It stops the switching operation of the element,
The hysteresis comparison circuit turns off the soft start control transistor in response to the feedback signal rising above the high threshold voltage due to a decrease in the output voltage due to the switching operation of the switch element being stopped. Generating the off-control output signal to control the state,
The soft start circuit, the PWM comparator, and the control drive circuit are configured to perform the switching operation of the switch element in response to the OFF control output signal of the hysteresis comparison circuit after the stop of the switching operation of the switch element. When resuming, a drive output signal that drives the switch element to an on state is generated,
The pulse width of the drive output signal is controlled from a narrower width to a wider width when restarting the switching operation. In claim 1,
When the switching operation is resumed, the PWM comparator and the control drive circuit are configured such that the falling edge of the triangular wave oscillation signal generated from the triangular wave oscillator and the off control output signal generated from the hysteresis comparison circuit In response to and driving the switch element to the ON state,
Upon restarting the switching operation, the PWM comparator selects a low level signal from the feedback signal of the feedback circuit and the soft start signal of the soft start circuit, and the selected low level signal In response to the triangular wave oscillation signal becoming a higher level, generating a comparison output signal for controlling the switch element to the off state,
In response to the comparison output signal generated from the PWM comparator, the control drive circuit drives the switch element to the OFF state. In claim 2,
The switch element is connectable to a primary winding of a transformer, a secondary winding of the transformer is connected to the output circuit, and the output circuit includes a light emitting unit of a photocoupler,
The feedback circuit is a semiconductor integrated circuit in which the output circuit includes a light receiving portion of the photocoupler optically coupled to the light emitting portion of the photocoupler. In claim 3,
The output circuit further includes a rectifier diode and a smoothing capacitor,
The anode of the rectifier diode is connected to one end of the secondary winding of the transformer, and one end of the light receiving unit of the photocoupler and one end of the smoothing capacitor are connected to the cathode of the rectifier diode, and the second of the transformer A semiconductor integrated circuit in which the other end of the light receiving portion of the photocoupler and the other end of the smoothing capacitor are connected to the other end of the next winding. In claim 4,
The primary winding and the secondary winding of the transformer are connected in reverse polarity, and the semiconductor integrated circuit, the switch element, the transformer, and the output circuit constitute a flyback converter. . In claim 5,
The switch element is constituted by a semiconductor chip of a transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, and the soft start control transistor are integrated on one semiconductor chip of the semiconductor integrated circuit. ,
The semiconductor integrated circuit in which the semiconductor chip of the transistor constituting the switch element and the one semiconductor chip of the semiconductor integrated circuit are sealed in one package of a system-in-package. In claim 5,
The switch element is constituted by a transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, and the transistor constituting the switch element are monolithic semiconductor integrated circuits. A semiconductor integrated circuit integrated on one chip. In claim 2,
The switch element includes a high side switch and a low side switch,
The high side switch and the low side switch are complementarily driven by the control drive circuit,
The switching node to which the high-side switch and the low-side switch are connected is connectable to one end of an inductor,
The other end of the inductor is connectable to one end of a smoothing capacitor, the other end of the smoothing capacitor is connected to the ground potential, and an output voltage is output from a connection node between the other end of the inductor and the one end of the smoothing capacitor. A semiconductor integrated circuit that can be generated. In claim 8,
The PWM comparator and the control drive circuit are responsive to the on-control output signal generated from the hysteresis comparison circuit upon reaching the maximum value of the output voltage at the light load. A semiconductor integrated circuit that stops the switching operation of the side switch. In claim 9,
The semiconductor integrated circuit further includes a reverse current detection circuit that generates a detection signal by detecting a voltage increase of the switching node due to polarity inversion of the current flowing through the low-side switch,
The control driving circuit is a semiconductor integrated circuit that stops the switching operation of the low-side switch in response to the detection signal of the reverse current detection circuit during a period in which the switching operation of the high-side switch is stopped. In claim 10,
The semiconductor integrated circuit, the high-side switch, the low-side switch, the inductor, and the smoothing capacitor constitute a synchronous rectification switching regulator. In claim 11,
The high side switch and the low side switch are respectively configured by a first semiconductor chip of a first transistor and a second semiconductor chip of a second transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, and the reverse current detection circuit are included in one of the semiconductor integrated circuits. Integrated on a semiconductor chip,
A semiconductor integrated circuit in which the first semiconductor chip, the second semiconductor chip, and the one semiconductor chip of the semiconductor integrated circuit are sealed in one package of a system-in-package. In claim 11,
The high-side switch and the low-side switch are each constituted by a first transistor and a second transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, the reverse current detection circuit, the first transistor, and the second transistor. Is a semiconductor integrated circuit integrated on a single chip of a monolithic semiconductor integrated circuit. A method of operating a semiconductor integrated circuit comprising a switch element, a feedback circuit, a PWM comparator, a hysteresis comparison circuit, a triangular wave oscillator, a control drive circuit, a soft start circuit, and a soft start control transistor,
The switch element is connectable to an output circuit, and the output circuit generates an output voltage in response to a switching operation of the switch element,
The feedback circuit generates a feedback signal in response to the output voltage generated from the output circuit, and the feedback signal is supplied to the PWM comparator and the hysteresis comparison circuit;
The soft start control transistor is connected between a soft start terminal of the soft start circuit and a ground potential, and an on state and an off state of the soft start control transistor are an on control output signal and an off control of the hysteresis comparison circuit. Each controlled by output signal,
The soft start circuit includes a charging circuit connected between a power supply voltage and the soft start terminal, and a soft start capacitor is connectable between the soft start terminal and the ground potential. A soft start signal is generated from the soft start terminal by controlling the control transistor to be in an off state,
The triangular wave oscillator generates a triangular wave oscillation signal, and the triangular wave oscillation signal is supplied to the PWM comparator and the control drive circuit,
The hysteresis comparison circuit generates a low threshold voltage and a high threshold voltage that is higher than the low threshold voltage.
The hysteresis comparison circuit sets the soft start control transistor to the on state in response to the feedback signal falling below the low threshold voltage due to reaching the maximum value of the output voltage at light load. Generating the on-control output signal to control,
The PWM comparator and the control drive circuit are responsive to the on-control output signal generated from the hysteresis comparison circuit upon reaching the maximum value of the output voltage at the light load. It stops the switching operation of the element,
The hysteresis comparison circuit turns off the soft start control transistor in response to the feedback signal rising above the high threshold voltage due to a decrease in the output voltage due to the switching operation of the switch element being stopped. Generating the off-control output signal to control the state,
The soft start circuit, the PWM comparator, and the control drive circuit are configured to perform the switching operation of the switch element in response to the OFF control output signal of the hysteresis comparison circuit after the stop of the switching operation of the switch element. When resuming, a drive output signal that drives the switch element to an on state is generated,
The pulse width of the drive output signal is a method for operating a semiconductor integrated circuit, wherein the pulse width of the drive output signal is controlled from narrow to gradually wider when the switching operation is resumed. In claim 14,
When the switching operation is resumed, the PWM comparator and the control drive circuit are configured such that the falling edge of the triangular wave oscillation signal generated from the triangular wave oscillator and the off control output signal generated from the hysteresis comparison circuit In response to and driving the switch element to the ON state,
Upon restarting the switching operation, the PWM comparator selects a low level signal from the feedback signal of the feedback circuit and the soft start signal of the soft start circuit, and the selected low level signal In response to the triangular wave oscillation signal becoming a higher level, generating a comparison output signal for controlling the switch element to the off state,
In response to the comparison output signal generated from the PWM comparator, the control drive circuit operates the semiconductor integrated circuit to drive the switch element to the OFF state. In claim 15,
The switch element is connectable to a primary winding of a transformer, a secondary winding of the transformer is connected to the output circuit, and the output circuit includes a light emitting unit of a photocoupler,
The feedback circuit is a method of operating a semiconductor integrated circuit, wherein the output circuit includes a light receiving portion of the photocoupler optically coupled to the light emitting portion of the photocoupler. In claim 16,
The output circuit further includes a rectifier diode and a smoothing capacitor,
The anode of the rectifier diode is connected to one end of the secondary winding of the transformer, and one end of the light receiving unit of the photocoupler and one end of the smoothing capacitor are connected to the cathode of the rectifier diode, and the second of the transformer A method for operating a semiconductor integrated circuit, wherein the other end of the photocoupler and the other end of the smoothing capacitor are connected to the other end of the next winding. In claim 17,
The primary winding and the secondary winding of the transformer are connected in reverse polarity, and the semiconductor integrated circuit, the switch element, the transformer, and the output circuit constitute a flyback converter. How it works. In claim 18,
The switch element is constituted by a semiconductor chip of a transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, and the soft start control transistor are integrated on one semiconductor chip of the semiconductor integrated circuit. ,
A method of operating a semiconductor integrated circuit, wherein the semiconductor chip of the transistor constituting the switch element and the one semiconductor chip of the semiconductor integrated circuit are sealed in one package of a system-in-package. In claim 18,
The switch element is constituted by a transistor,
The feedback circuit, the PWM comparator, the hysteresis comparison circuit, the triangular wave oscillator, the control drive circuit, the soft start circuit, the soft start control transistor, and the transistor constituting the switch element are monolithic semiconductor integrated circuits. A method of operating a semiconductor integrated circuit integrated on one chip.

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