JP2012154640A - Radar device - Google Patents

Abstract

Even if there are a plurality of targets within the distance resolution and the Doppler frequency of the signal of the reflected wave from each target is approximately the same, the accuracy of direction estimation of each target is improved.
A radar transmitter converts a pulse compression code having a predetermined code length into a high-frequency transmission signal and transmits it from a transmission antenna. The radar receiving unit has an antenna system processing unit for each of a plurality of receiving antennas. The antenna system processing unit converts the received signal into digital data at a sampling rate twice the pulse transmission rate in the pulse compression code, and a first correlation between the data converted according to the odd sample timing and the pulse compression code A value is calculated, and a second correlation value between the data converted according to the even sample timing and the pulse compression code is calculated. The radar receiving unit includes a high resolution processing unit that separates and detects each target when a plurality of targets are present within the distance resolution of the high-frequency transmission signal based on the first correlation value and the second correlation value.
[Selection] Figure 3

Description

  The present invention relates to a radar apparatus that receives a reflected wave signal reflected from a target by a receiving antenna and detects the target.

  The radar device radiates radio waves from a measurement point into space, receives a reflected wave signal reflected by the target, and measures at least one of the distance and direction between the measurement point and the target. Particularly in recent years, development of radar devices that can be detected as targets including not only automobiles but also pedestrians by measuring with high resolution using short-wave radio waves including microwaves or millimeter waves has been underway.

  In addition, the radar apparatus may receive a signal in which reflected waves from a target existing at a short distance and a target existing at a long distance are mixed. In particular, a range side lobe is generated by a reflected wave signal from a target existing at a short distance, and therefore a range side lobe and a main lobe of a reflected wave signal from a target present at a long distance may be mixed. Regarding the detection of a target existing at a long distance in a radar apparatus, the detection accuracy may be deteriorated.

  Therefore, a pulse compression code having an autocorrelation characteristic (hereinafter referred to as a “low range sidelobe characteristic”) having a low range sidelobe level is used for a radar apparatus that requires high resolution measurement for a plurality of targets. It is required to transmit a pulse wave or pulse modulated wave.

  In addition, when a car and a pedestrian are present at the same distance from the measurement point, the radar apparatus mixed signals of reflected waves from a car and a pedestrian having different radar cross sections (RCS). A signal may be received. In general, the radar reflection cross section of a pedestrian is lower than the radar reflection cross section of an automobile.

  For this reason, the radar apparatus is required to appropriately receive a reflected wave signal from not only the automobile but also the pedestrian even if the automobile and the pedestrian exist at the same distance from the measurement point. Depending on the distance or type of the target, the reflected wave signal at the reception signal level changes. The radar device is required to have a reception dynamic range that is large enough to receive signals of reflected waves having various reception signal levels.

  As an example of a conventional radar apparatus, a pulse wave or a pulse modulated wave is transmitted by mechanically scanning an antenna or electronically scanning a narrow-angle directional beam, and a reflected wave reflected from a target is detected. There is something to receive.

  In an example radar apparatus, since radio waves are transmitted and received by a single antenna, it takes time to scan the antenna for target detection. For this reason, in order to detect a target moving at high speed, a lot of scanning is required as high-resolution measurement is required, and it is difficult to detect following the movement of the target.

  On the other hand, as another example of a conventional radar device, the reflected wave signal reflected from the target is received by a plurality of antennas, and the phase of each received signal is measured, so that the directivity of the beam for each antenna is exceeded. Some estimate the angle of arrival using high resolution.

  According to another example of the radar apparatus, it is possible to estimate the arrival angle by thinning the scanning interval and processing the signal as compared with the radar apparatus using a single antenna, thereby improving the target detection accuracy. it can. Even if the target moves at high speed, the angle of arrival can be estimated following the movement of the target.

  Here, it is known that the distance resolution ΔR in the pulse compression radar having the pulse width Tp is generally expressed by Expression (1). Parameter C is the speed of light.

  FIG. 22 is an explanatory diagram illustrating an example of a direction estimation result in which a plurality of targets having different angular directions from the pulse compression radar exist within the distance resolution ΔR. FIG. 4A is an explanatory diagram showing the positional relationship between a pulse compression radar using a plurality of antennas (array antennas) and a plurality of targets. FIG. 6B is an explanatory diagram showing the received power estimation value for the arrival direction θ where only the target A exists. FIG. 4C is an explanatory diagram showing the received power estimation value for the arrival direction θ where the target A and the target B exist.

  As shown in FIG. 22, when a pulse compression radar using a plurality of antennas measures a plurality of targets existing within the distance resolution ΔR of the pulse compression radar, the range finding performance and angle measurement performance of the target are as follows. There are challenges as shown.

  In ranging performance, multiple targets are in the same range bin from the pulse compression radar. The pulse compression radar outputs a signal in which reflected wave signals from the targets are mixed in the pulse compression processing. This makes it difficult to detect each target separately.

  In the angle measurement performance, since there are a plurality of targets in directions close to each other within the beam width of the plurality of antennas, the correlation of the signals of the reflected waves from the plurality of targets becomes high. Therefore, a direction component obtained by vector synthesis of the signals of the reflected waves is output. This makes it difficult to detect each target separately. In addition, the direction estimation value for each of the plurality of targets may indicate an intermediate arrival direction of the plurality of targets. In this case, the direction estimation accuracy may be greatly deteriorated.

  For example, Patent Document 1 is known as a technique for separating signals of reflected waves from a plurality of targets having high correlation and performing angle calculation on the signal arrival direction from each signal. The super-resolution antenna of Patent Document 1 performs Fourier transform on a received signal by a plurality of element antennas, and weights signal components for each filter bank according to the Doppler frequency generated by Fourier transform. Further, the super-resolution antenna extracts only a desired filter bank component from the weighted signal component, and estimates the signal arrival direction of the target from the extracted signal component of the element antenna.

  As a result, if there are multiple targets within the distance resolution, the correlation of each target is high, and the Doppler frequency of each target is sufficiently different, the received signal from each target is separated, and the arrival direction of each target is estimated can do.

JP 2003-194919 A

  However, in Patent Document 1, it is necessary that the Doppler frequencies of signals of reflected waves from a plurality of targets are different so that a weighted signal component can be extracted by a filter bank. Accordingly, when there are signals of reflected waves from a plurality of targets having the same Doppler frequency within the distance resolution ΔR, the super resolution antenna shown in Patent Document 1 separates and detects the plurality of targets. Difficult to do.

  The present invention has been made in view of the above-described conventional circumstances, and there are a plurality of targets within the distance resolution, and even if the Doppler frequency of each reflected wave signal from each target is the same, the direction and distance of each target. An object of the present invention is to provide a radar apparatus that separates and detects each target.

  The present invention is the above-described radar apparatus, which converts a pulse compression code into a high-frequency transmission signal, transmits the high-frequency transmission signal from a transmission antenna, and a reflection that is the high-frequency transmission signal reflected by the target. A radar receiver that estimates an arrival time and direction to a target using a wave signal using a sampling rate N (integer) times a pulse transmission rate in the pulse compression code, The receiving unit is configured to output a correlation value between the received signal and the pulse compression code, and based on the outputs of the plurality of antenna system processing units, the arrival time and the arrival direction of the target individually. And the N arrival time direction estimation units, and the N arrival time direction estimation units, A plurality of targets existing within the resolution, and a high-resolution processing unit that detects each target separately, and the plurality of antenna system processing units each receive a reflected wave signal; A reception RF unit that converts the received signal into a baseband reception signal, and an A / D conversion unit that converts the converted reception signal into digital data using the N (integer) multiple sampling rate And N correlation value calculation units for calculating correlation values between the digital data of the received signal converted at every 1 / N sample timing and the transmission pulse compression code.

  According to the present invention, even when there are a plurality of targets within the distance resolution and the Doppler frequency of each reflected wave signal from each target is approximately the same, in order to improve the estimation accuracy of the direction and distance of each target, The target can be detected separately.

Explanatory drawing explaining the property of a complementary code, (a) Explanatory drawing showing an autocorrelation calculation result of one complementary code sequence, (b) Explanatory drawing showing an autocorrelation calculation result of the other complementary code sequence, (c) 2 Explanatory drawing which shows the addition value of the autocorrelation calculation result of two complementary code sequences Explanatory drawing explaining the autocorrelation characteristics based on oversampling at the time of reception, (a) Sampling rate twice that of received signal y (t) with respect to pulse width Tp of a pulse compression code using a complementary code to be transmitted (B) An explanatory diagram for explaining the autocorrelation characteristics for an odd number of discrete sample values among the results of oversampling twice during reception, and (c) for reception. Explanatory drawing explaining an autocorrelation characteristic about the even number of discrete sample values among the results of oversampling twice. The block diagram which shows simply the internal structure of the radar apparatus of 1st Embodiment The block diagram which shows the internal structure of the radar apparatus of 1st Embodiment in detail Explanatory drawing which shows the relationship between the transmission interval and transmission period of a high frequency transmission signal The block diagram which shows the other internal structure of a transmission signal generation part in detail Explanatory drawing which shows the relationship between the transmission section of a high frequency transmission signal, a transmission period, and a measurement range Block diagram showing in detail the internal configuration of the high-resolution processor Explanatory diagram showing the relationship between estimation results and measurement time Explanatory drawing which shows the relationship between a coincidence determination range and an estimation result An explanatory diagram showing the configuration of each block divided into three of the match determination range, (a) an explanatory diagram showing a configuration in which the number of estimation results included in the match determination range is six, and (b) a match determination range An explanatory diagram showing an example of a configuration in which the number of estimation results included in 8 is eight, and (c) an explanatory diagram showing another example of a configuration in which the number of estimation results included in the match determination range is eight Explanatory drawing explaining the operation | movement outline | summary of a high-resolution process part, (a) Explanatory drawing which made the relative delay in arrival of the signal of the reflected wave from the target A and the signal of the reflected wave from the target B the pulse width Tp, (b) ) An explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is a pulse width Tp / 2, (c) the reflected wave signal from the target A and the target B Explanatory drawing which made the relative delay in arrival with the signal of the reflected wave of the pulse width Tp / 3 Flow chart explaining operation of high resolution processing unit Flow chart explaining operation of high resolution processing unit Explanatory diagram schematically showing the estimation result of the high resolution processing unit, (a) The relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is set to a pulse width of 1.5 Tp. Explanatory drawing, (b) Explanatory drawing which made the relative delay in arrival of the reflected wave signal from target A and the reflected wave signal from target B the pulse width Tp, (c) Reflected wave signal from target A An explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target B is a pulse width Tp / 2, (d) relative in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B Explanatory diagram with delay as pulse width Tp / 3 Explanatory drawing which showed typically the estimation result of the high-resolution process part of the radar apparatus of the modification 1 of 1st Embodiment. The block diagram which shows in detail the internal structure of the radar apparatus of the modification 2 of 1st Embodiment. Timing chart regarding operation of radar device of modification 2 of first embodiment The block diagram which shows in detail the internal structure of the 1st antenna system | strain processing part in the radar receiver of the radar apparatus of the modification 3 of 1st Embodiment. Timing chart about operation | movement of each A / D conversion part in the 1st antenna system | strain process part in the radar receiver of the radar apparatus of the modification 3 of 1st Embodiment. The block diagram which shows in detail the internal structure of the radar receiver of the radar apparatus of the modification 4 of 1st Embodiment. An explanatory view showing an example of a direction estimation result in which a plurality of targets having different angular directions from the pulse compression radar exist within the distance resolution ΔR, (a) a pulse compression radar using a plurality of antennas (array antennas), a plurality of targets, and (B) An explanatory diagram showing a received power estimation value for the arrival direction θ where only the target A exists, (c) A received power estimation value for the arrival direction θ where the target A and the target B exist Illustration showing

  Before describing each embodiment of the radar apparatus according to the present invention, as the technical content that is the premise of each embodiment described below, each of the autocorrelation characteristics based on pulse compression processing, complementary codes, and oversampling at the time of reception, respectively. Briefly described.

(Pulse compression processing)
First, the pulse compression process will be described. For example, pulse compression that transmits a high-frequency radar transmission signal using a pulse compression code including at least one of a Barker code, an M-sequence code, and a complementary code as the pulse wave or pulse modulated wave having the low-range sidelobe characteristics described above Radar is known.

  With pulse compression, a radar device performs pulse modulation or phase modulation on a pulse signal and transmits it using a signal with a wide pulse width (pulse compression code), and demodulates the received signal in signal processing after reception of the reflected wave. The correlation value is calculated after being converted (compressed) into a narrow signal. According to this pulse compression, the received power can be increased equivalently, the target detection distance can be increased, and the distance estimation accuracy with respect to the detection distance can be improved.

(Complementary code)
Next, complementary codes will be described. The complementary code is a code configured using a plurality of, for example, two complementary code sequences (a _n , b _n ). Complementary code, by adding at each autocorrelation calculation result of one of the complementary code sequences a _n and the other of the complementary code sequence b _n, to match the delay time tau [sec] of each autocorrelation calculation result, the range side It has the property that the lobe becomes zero. The parameter n is n = 1, 2,. The parameter L indicates the code sequence length or simply the code length.

A method for generating a complementary code is disclosed, for example, in Reference Non-Patent Document 1 below.
(Reference Non-Patent Document 1) BUDISIN, S. Z, “NEW COMPLEMENTARY PAIRS OF SEQUENCES”, Electron. Lett. , 26, (13), pp. 881-883 (1990)

The nature of such complementary codes will be described with reference to FIG. FIG. 1 is an explanatory diagram for explaining the properties of complementary codes. FIG (a) is an explanatory diagram showing an auto-correlation value calculation result of one of the complementary code sequences a _n. FIG (b) is an explanatory diagram showing an auto-correlation value calculation result of the other complementary code sequence b _n. FIG. 6C is an explanatory diagram showing the added value of the autocorrelation value calculation results of two complementary code sequences (a _n , b _n ). The code length L of the complementary code used in FIG.

Complementary code sequences (a _{n, b n)} one of the complementary code sequences a _n autocorrelation value calculation result of the is calculated according to equation (2). Autocorrelation value calculation result of the other complementary code sequence b _n is calculated according to equation (3). The parameter R indicates the autocorrelation value calculation result. However, when n> L or n <1, the complementary code sequences a _n and b _n are set to zero (that is, when n> L or n <1, a _n = 0 and b _n = 0). The asterisk * indicates a complex conjugate operator.

Equation autocorrelation value calculation result of the computed complementary code sequence a _n R _{aa (τ)} in accordance with (2), as shown in FIG. 1 (a), the delay time (or shift time) when tau is zero peak When the delay time τ is not zero, a range side lobe exists. Similarly, the autocorrelation value calculation result R _bb (τ) of the complementary code sequence b _n calculated according to Equation (3) has a peak when the delay time τ is zero, as shown in FIG. However, when the delay time τ is not zero, there is a range side lobe.

The sum of these autocorrelation value calculation results (R _aa (τ), R _bb (τ)) has a peak when the delay time τ is zero, as shown in FIG. If τ is other than zero, the range side lobe does not exist and becomes zero. Hereinafter, a peak that occurs when the delay time τ is zero is referred to as a “main lobe”. This relationship is shown in Equation (4). 1A to 1C, the horizontal axis indicates the delay time (τ) in the autocorrelation value calculation, and the vertical axis indicates the calculated autocorrelation value calculation result.

  In order to receive a signal that is a mixture of reflected waves from a target at a short distance and a target at a long distance, in general, the longer the code length of a pulse-compressed code, the higher the required reception dynamic range. Is known to grow.

  In the complementary code, the peak sidelobe level can be reduced with a shorter code length from the above-described autocorrelation characteristics. For this reason, the complementary code using a short code length can reduce the reception dynamic range even when a signal in which reflected waves from a target existing at a short distance and a target existing at a long distance are mixed is received. .

Further, by using a Barker code and an M-sequence code having a code length L, the peak side lobe ratio is given by 20 log ₁₀ (1 / L) [dB], and by increasing the code length L, an excellent low-range side A lobe characteristic is obtained.

(Autocorrelation characteristics based on oversampling during reception)
Next, autocorrelation characteristics based on oversampling (sampling at a predetermined sampling rate) at the time of reception will be described with reference to FIG. FIG. 2 is an explanatory diagram illustrating autocorrelation characteristics based on oversampling at the time of reception. In the following description, the parameter Tp [seconds] indicates a pulse (transmission time) width at the time of transmission of a pulse compression code using a complementary code, simply a pulse width.

  In FIG. 6A, a sampling rate (= 2 / Tp) twice that of the received signal y (t) is used for the pulse width Tp of the pulse compression code (transmission code) using the complementary code to be transmitted. It is explanatory drawing explaining the autocorrelation characteristic oversampled.

The autocorrelation calculation is performed by using the received signal y ((kTp / 2) + t ₀ ) sampled at a sampling rate twice the pulse width and the transmitted complementary code sequence of the code length L, as shown in Equation (5). pulse width a _n to a correlation calculation between the sample signal aa _s at twice the sampling rate. The parameter S = 1, ..., a _{2L, aa 2n-1 = a} n, a _aa 2n = _{a n.} The time t ₀ is the timing at which the target reflected wave arrives.

  FIG. 6B is an explanatory diagram for explaining the autocorrelation characteristics for odd-numbered discrete sample values among the results of oversampling twice during reception.

The autocorrelation calculation is performed by using an odd number of discrete sample values of the received signal y (kTp / 2 + t ₀ ) sampled at a sampling rate twice the pulse width and the transmitted code length, as shown in Equation (6). L and signal a _n sampled at 1 times the sample rate relative to the pulse width of complementary code sequence a _n of the correlation calculation. The time t ₀ is the timing at which the target reflected wave arrives.

  FIG. 7C is an explanatory diagram for explaining the autocorrelation characteristics for even-numbered discrete sample values among the results of double oversampling at the time of reception.

The autocorrelation calculation, as shown in Equation (7), includes an even number of discrete sample values of the received signal y (kTp / 2 + t ₀ ) sampled at a sampling rate twice the pulse width and the transmitted code length. L and signal a _n sampled at 1 times the sample rate relative to the pulse width of complementary code sequence a _n of the correlation calculation. The time t ₀ is the timing at which the target reflected wave arrives.

  In FIG. 2, the arrival timing of the reflected wave signal from the target coincides with the sample timing at the time of reception. The adjacent discrete time, that is, the sample timing becomes Tp / 2, and three sample timings at which the autocorrelation value does not become zero appear as shown in FIG. Hereinafter, as shown in FIG. 2A, a range in which the autocorrelation value does not become zero at adjacent sample timings is defined as a “correlation spread range”.

  Therefore, as shown in FIG. 2A, the autocorrelation value is zero before and after the peak of the autocorrelation value due to oversampling at a predetermined time (for example, a sampling rate twice as high as the pulse width Tp) at the time of reception. There is a sample timing that does not become. That is, in FIG. 2A, it can be considered that a range side lobe has occurred around the peak of the autocorrelation value.

  From the time when the peak of the autocorrelation value is obtained, it is affected by the discrete sample value at the time of reception of the delay time ± Tp / 2. When there are a plurality of targets at ΔR / 2, which is a distance within the distance resolution ΔR of the radar device, the reflected wave signal reflected by each target is received with some delay, and the self of each target is received by the above range side lobe. Correlation values are added, and the angle measurement estimation performance for a plurality of targets deteriorates.

  On the other hand, FIG. 2B shows an autocorrelation characteristic in which the arrival timing of the reflected wave signal from the target coincides with the sample timing at the time of reception. Unlike the autocorrelation calculation result shown in FIG. 2A, when the adjacent discrete time, that is, the sample timing is Tp, the sample timing at which the autocorrelation value does not become zero is 1 as shown in FIG. There is only one.

  Therefore, in the autocorrelation calculation result of FIG. 2B, even if oversampling is actually performed using a double sampling rate, it is affected by the discrete sample value at the time of reception of delay time ± Tp / 2. do not become. Even if there are a plurality of targets within the distance resolution ΔR of the radar apparatus, the reflected wave signal reflected by each target is received with some delay, but the angle measurement estimation performance for the plurality of targets deteriorates. Don't be.

  FIG. 2C shows an autocorrelation characteristic in which the arrival timing of the reflected wave signal from the target does not coincide with the sample timing at the time of reception. The sample timing shown in FIG. 2C and the sample timing shown in FIG. 2B are shifted by a discrete time Tp / 2.

  Here, it is assumed that the sample timing of the odd-numbered discrete sample values shown in FIG. 2B matches the arrival timing of the reflected wave signal reflected by the target. As shown in FIG. 2C, unlike the autocorrelation calculation result shown in FIG. 2B, there are two sample timings in which the autocorrelation value does not become zero at the sample timings separated by the adjacent discrete time Tp.

  Therefore, the autocorrelation calculation result in FIG. 2C is affected by the sample value at the time of reception of the delay time ± Tp / 2. Since there are a plurality of targets at ΔR / 2, which is a distance within the distance resolution ΔR of the radar apparatus, the autocorrelation value of each target is added by the above range side lobe, and a reflected wave signal reflected by each target is obtained. The angle estimation performance with respect to a plurality of targets is deteriorated due to a partial delay.

  In the above description, the sample timing at the time of reception shown in FIG. 2B matches the arrival timing of the signal of the reflected wave from the target, and the sample timing and the reflected wave at the time of reception shown in FIG. It is assumed that the signal arrival timing does not match.

  On the other hand, the sample timing at the time of reception shown in FIG. 2C matches the arrival timing of the reflected wave signal, and the sample timing at the time of reception and the arrival timing of the reflected wave signal shown in FIG. Are not matched.

  In this case, the autocorrelation calculation result shown in FIG. 2B is affected by the sample value at the time of reception of the delay time ± Tp / 2, and the autocorrelation calculation result shown in FIG. 2 is not affected by the sample value at the time of reception.

(Embodiments of the present invention)
Next, embodiments of the present invention will be described with reference to the drawings. As shown in FIG. 2B and FIG. 2C, the radar apparatus according to the present invention, for example, at the time of reception with respect to the pulse width Tp of the pulse compression code (transmission code) using the transmitted complementary code, A double oversampled correlation value characteristic (correlation value calculation result) is used. Furthermore, the radar apparatus according to the present invention separates the reflected wave signal from each target based on the correlation value calculation result even if there are a plurality of targets within the distance resolution of the radar apparatus. To detect.

  In the following description, the radar apparatus according to the present invention has a plurality of receiving antennas for receiving a reflected wave signal from a target. For example, a configuration having four receiving antennas (array antennas) is shown as an example, but the present invention is not limited to this.

[First Embodiment]
The configuration and operation of the radar apparatus 1 according to the first embodiment will be described with reference to FIGS. FIG. 3 is a block diagram schematically showing the internal configuration of the radar apparatus according to the first embodiment. FIG. 4 is a block diagram showing in detail the internal configuration of the radar apparatus 1 according to the first embodiment. FIG. 5 is an explanatory diagram showing the relationship between the transmission interval Tw of the high-frequency transmission signal and the transmission cycle Tr. FIG. 6 is a block diagram showing in detail another internal configuration of the transmission signal generation unit 4r.

FIG. 7 is an explanatory diagram showing the relationship among the transmission interval Tw, the transmission cycle Tr, and the measurement range of the high-frequency transmission signal. FIG. 8 is a block diagram showing in detail the internal configuration of the high resolution processing unit 26. FIG. 9 is an explanatory diagram showing the relationship between the estimation result DOUT _p (q, m) and the discrete time (p, q).

FIG. 10 is an explanatory diagram showing the relationship between the match determination range and the estimation result DOUT _p (q, m). FIG. 11 is an explanatory diagram showing the configuration of each block divided into three match determination ranges. FIG. 6A is an explanatory diagram showing a configuration in which the number of estimation results included in the coincidence determination range is six. FIG. 6B is an explanatory diagram showing an example of a configuration in which the number of estimation results included in the coincidence determination range is eight. FIG. 7C is an explanatory diagram showing another example of the configuration in which the number of estimation results included in the coincidence determination range is eight.

  FIG. 12 is an explanatory diagram for explaining an outline of the operation of the high resolution processing unit 26. FIG. 7A is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is defined as a pulse width Tp. FIG. 4B is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is set to the pulse width Tp / 2. FIG. 6C is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is set to a pulse width Tp / 3.

  FIG. 13 and FIG. 14 are flowcharts for explaining the operation of the high resolution processing unit 26.

  The radar apparatus 1 according to the first embodiment transmits (emits) the high-frequency transmission signal generated by the radar transmission unit 2 from the transmission antenna AN1. The radar apparatus 1 receives a reflected wave signal obtained by reflecting a transmitted high-frequency transmission signal by a target, for example, by four receiving antennas AN2a to AN2d (not shown, the same applies hereinafter) as shown in FIG. The radar apparatus 1 detects the presence or absence of a target by performing signal processing on signals received by the receiving antennas AN2a to AN2d.

  The target is an object to be detected by the radar apparatus 1 and includes, for example, a car or a person, and the same applies to each of the following embodiments.

  First, the configuration of each part of the radar apparatus 1 will be briefly described.

  The radar apparatus 1 includes a radar transmitter 2 and a radar receiver 3 as shown in FIG. The radar transmission unit 2 includes a transmission signal generation unit 4 and a transmission RF unit 5 connected to the transmission antenna AN1. The radar transmitter 2 and the radar receiver 3 are both connected to the reference signal oscillator Lo, and a signal is supplied from the reference signal oscillator Lo, so that the processing of the radar transmitter 2 and the radar receiver 3 is synchronized. ing.

  The radar receiver 3 includes D antenna system processors 11-1 to 11 -D, N arrival time direction estimators 24-1 to 24 -N, and a high resolution processor 26. Parameter D and parameter N are both integers of 2 or more. Each antenna system processing unit has the same configuration, and in the following description, the antenna system processing unit 11-1 will be described as an example.

  The antenna system processing unit 11-1 includes at least a reception RF unit 12, an A / D conversion unit 17, and N correlation value calculation units 20-1 to 20-N connected to the reception antenna AN2.

(Radar transmitter)
Next, the configuration of each part of the radar transmitter 2 will be described in detail. Hereinafter, the configuration of each part of the radar transmitter 2 will be described with reference to FIG.

  As shown in FIG. 4, the radar transmission unit 2 includes a transmission signal generation unit 4 and a transmission RF unit 5 to which a transmission antenna AN1 is connected.

  As shown in FIG. 4, the transmission signal generation unit 4 includes a code generation unit 6, a modulation unit 7, and an LPF (Low Pass Filter) 8. In FIG. 4, the transmission signal generation unit 4 is configured to include the LPF 8, but the LPF 8 may be configured in the radar transmission unit 2 independently of the transmission signal generation unit 4.

  The transmission RF unit 5 includes a frequency conversion unit 9 and an amplifier 10 as shown in FIG.

  Next, the operation of each part of the radar transmitter 2 will be described in detail.

  The transmission signal generation unit 4 generates a signal obtained by multiplying the reference signal by a predetermined factor based on the reference signal generated by the reference signal oscillator Lo. Each unit of the transmission signal generation unit 4 operates based on the generated signal.

Transmission signal generating unit 4 modulates the code sequence a _n of the code length L, the transmission signal r baseband shown in equation (8) (k, M) ( also referred to as pulse compression code) to periodically generate. Here, the parameter n = 1, · · ·, L, and the parameter L represents the code length of the code sequence _{a n.} The parameter j is an imaginary unit that satisfies j ² = −1. The parameter k is a discrete time satisfying k = 1 to (Nr + Nu). The range of the discrete time k is the same in each embodiment described later.

  The baseband transmission signal r (k, M) shown in Expression (8) indicates a transmission signal at a discrete time k in the Mth transmission cycle Tr, and an in-phase signal component Ir (k, M), This is indicated by the addition result with the orthogonal signal component Qr (k, M) multiplied by the imaginary unit j.

Further, the transmission signal generated by the transmission signal generation unit 4 is not a continuous signal. As shown in FIG. 5, for example, the transmission interval Tw [sec] of each transmission period Tr, to the code sequence a _n of the code length L, samples No per one pulse code [pieces] are present. Therefore, Nr (= No × L) samples are included in the transmission interval Tw. Further, in the non-transmission section (Tr−Tw) [seconds] of each transmission cycle Tr, there are Nu [number] samples as baseband transmission signals.

Code generator 6, for each transmission period Tr, to generate a transmission code for pulse compression code sequence a _n of the code length L. This transmission code is preferably a code including at least one of a code sequence, a Barker code sequence, or an M-sequence code constituting the above-described complementary code pair.

Code generator 6 outputs a transmission code of the generated code sequence a _n to the modulator 7. Hereinafter, the transmission code of the code sequence a _n, are described as conveniently transmitted symbols a _n.

Furthermore, in _order to generate a pair of complementary codes as a transmission code an in a certain transmission cycle Tr, the code generation unit 6 uses two transmission cycles and uses a pair of codes Pn alternately in each transmission cycle. , Qn are generated respectively. That is, in the M-th transmission period (Tr), transmits the code Pn as a pulse compression code _a n (M), as followed by the (M + 1) th transmission period (Tr) in the pulse compression code _a n (M + 1) , Qn is transmitted. Thereafter, in the (M + 2) th and subsequent transmission cycles, the repetition codes Pn and Qn are generated in the same manner with the Mth transmission cycle and the (M + 1) th two transmission cycles as one unit.

Modulation unit 7 inputs the transmission code a _n output by the code generator 6. Modulation unit 7, by pulse-modulating a transmission code a _n input, generates a transmission signal r baseband shown in equation (8) (k, M) . The pulse modulation is amplitude modulation, ASK (Amplitude Shift Keying) or phase modulation (PSK (Phase Shift Keying)), and the modulation unit 7 generates a transmission signal r (k) generated via the LPF 8. , M), a transmission signal r (k, M) that is equal to or smaller than a preset limit band is output to the transmission RF unit 5.

  The transmission RF unit 5 generates a signal obtained by multiplying the reference signal by a predetermined multiple based on the reference signal generated by the reference signal oscillator Lo. The transmission RF unit 5 operates based on the generated signal.

  The frequency conversion unit 9 receives the transmission signal r (k, M) generated by the transmission signal generation unit 4, up-converts the input baseband transmission signal r (k, M), and sets the carrier frequency band. A high frequency transmission signal is generated. The frequency conversion unit 9 outputs the generated high-frequency transmission signal to the amplifier 10.

  The amplifier 10 receives the output high-frequency transmission signal, amplifies the level of the input high-frequency transmission signal to a predetermined level, and outputs it to the transmission antenna AN1. The amplified high-frequency transmission signal is transmitted so as to be radiated to the space via the transmission antenna AN1.

  The transmission antenna AN1 transmits the high-frequency transmission signal output from the transmission RF unit 5 so as to radiate it into space. As shown in FIG. 5, the high-frequency transmission signal is transmitted during the transmission interval Tw in the transmission cycle Tr, and is not transmitted during the non-transmission interval (Tr−Tw).

  The transmission RF unit 5 and the reception RF units 12a to 12d (not shown, the same applies hereinafter) of the antenna system processing units 11a to 11d are multiplied by a predetermined number of reference signals generated by the reference signal oscillator Lo. The signal is supplied in common. Thereby, the synchronization between each transmission RF part 5 and reception RF part 12a-12d can be established.

Note that without providing the code generating unit 6 described above to the transmission signal generator 4, as shown in FIG. 6, provided with a transmission code storage unit CM for previously storing a transmission code a _n generated by the transmission signal generator 4 Also good. Furthermore, the complementary code is generated by the transmission signal generator 4, the complementary code pair, for example, it is preferable that the transmission code a _n and b _n are stored together in the transmission code storage unit CM.

  Note that the transmission code storage unit CM illustrated in FIG. 6 is not limited to the first embodiment, and can be similarly applied to each embodiment described later. As shown in FIG. 6, the transmission signal generation unit 4r includes a transmission code storage unit CM, a transmission code control unit CT3, a modulation unit 7r, and an LPF 8r.

6, Send code controller CT3 based reference signal output by the reference signal oscillator Lo with the multiplied signal to the predetermined number of times, each transmission period Tr, which constitutes the transmission code a _{n (or} complementary code The code a _n and the transmission code b _n ) are read cyclically from the transmission code storage unit CM and output to the modulation unit 7r. Since the operation after the output to the modulation unit 7r is the same as that of the modulation unit 7 and the LPF 8, the description of the operation is omitted.

(Radar receiver)
Next, the configuration of each part of the radar receiver 3 will be described. The radar receiver 3 includes a plurality of, for example, four array antennas. Hereinafter, the configuration of each part of the radar receiver 3 will be described with reference to FIG. In the radar receiver 3 shown in FIG. 4, the number of antenna system processors is 4, and the number of correlation value calculators and arrival time direction estimators is 2.

  As shown in FIG. 4, the radar receiver 3 includes four antenna system processors 11a to 11d, a first arrival time direction estimator 24, a second antenna system processor 11a to 11d provided corresponding to the number of receiving antennas constituting the array antenna. An arrival time direction estimation unit 25 and a high resolution processing unit 26 are provided. In the following description, since the configuration and operation of each part of the four antenna system processing units 11a to 11d are the same, the antenna system processing unit 11a will be described as an example, and the same applies to each embodiment described later.

  The antenna system processing unit 11a includes a reception RF unit 12a to which a reception antenna AN2a is connected, and a signal processing unit 13a. The reception RF unit 12a includes an amplifier 14a, a frequency conversion unit 15a, and a quadrature detection unit 16a. The signal processor 13a includes A / D converters 17a and 18a, an S / P converter 19a, a first correlation value calculator 20a, a second correlation value calculator 21a, a first addition processor 22a, and a second addition processor. 23a. The radar receiver 3 periodically calculates each transmission cycle Tr as a signal processing section in signal processors 13a to 13d (not shown, the same applies hereinafter).

  As shown in FIG. 8, the high resolution processing unit 26 includes a reception signal level determination unit 27, a match determination range extraction unit 28, and a match determination unit 29.

  Next, the operation of each part of the radar receiver 3 will be described.

  The reception antenna AN2a receives a reflected wave signal obtained by reflecting the high-frequency transmission signal transmitted from the radar transmitter 2 by the target. The reception signal received by the reception antenna AN2 is output to the reception RF unit 12a.

  Similarly to the transmission RF unit 5, the reception RF unit 12a generates a signal obtained by multiplying the reference signal by a predetermined number based on the reference signal generated by the reference signal oscillator Lo. The reception RF unit 12a operates based on the generated signal.

  The amplifier 14a receives a high-frequency band reception signal received by the reception antenna AN2a, amplifies the level of the input reception signal, and outputs the amplified signal to the frequency converter 15a.

  The frequency conversion unit 15a receives the high frequency band reception signal output from the amplifier 14a, down-converts the input high frequency band reception signal to baseband, and outputs the down-converted reception signal to the quadrature detection unit 16a. To do.

  The quadrature detection unit 16a performs quadrature detection on the received baseband signal output from the frequency conversion unit 15a, thereby forming a baseband using an in-phase signal and a quadrate signal. The received signal is generated. The quadrature detection unit 16a outputs an in-phase signal among the generated reception signals to the A / D conversion unit 17a, and outputs a quadrature signal among the generated reception signals to the A / D conversion unit 18a.

  The A / D converter 17a samples the baseband in-phase signal output from the quadrature detector 16a at the discrete time k, and converts the in-phase signal of analog data into digital data. The A / D conversion unit 17a outputs the in-phase signal component of the converted digital data to the S / P conversion unit 19a.

  The A / D conversion unit 17a samples the transmission signal r (k, M) generated by the radar transmission unit 2 by Ns [number] per one pulse width (pulse time) Tp (= Tw / L). . The sampling rate of the A / D conversion unit 17a is Ns.

  Similarly, the A / D conversion unit 18a samples the baseband quadrature signal output from the quadrature detection unit 16a at the discrete time k, and converts the quadrature signal of analog data into digital data. The A / D conversion unit 18a outputs the orthogonal signal component of the converted digital data to the S / P conversion unit 19a.

  The A / D conversion unit 18a samples the transmission signal r (k, M) generated by the radar transmission unit 2 by Ns [number] per one pulse width (pulse time) Tp (= Tw / L). . The sampling rate of the A / D conversion unit 18a is Ns. FIG. 4 shows the configuration of the radar receiver 3 in which each sampling rate Ns of the A / D converter 17a and the A / D converter 18a is 2.

  The received signal at the discrete time k in the Mth transmission cycle Tr converted by the A / D converters 17a and 18a is the in-phase signal component I (k, M) of the received signal and the quadrature signal component of the received signal. Using Q (k, M), it is expressed as a complex signal x (k, M) in equation (9). The same applies to the following embodiments.

  Here, the discrete time k = 1 indicates the start time of each transmission cycle Tr. Further, the discrete time k = Nr × (Ns / No) indicates the end point of the transmission section Tw in each transmission cycle Tr. Further, the discrete time k = (Nr + Nu) × (Ns / No) indicates the time immediately before the end of each transmission cycle Tr.

  The S / P converter 19a receives the in-phase signal component and the quadrature signal component of the digital data output by the A / D converter 17a and the A / D converter 18a, respectively. The S / P conversion unit 19a outputs the output of the complex number x (k, M) composed of the input in-phase signal component and quadrature signal component according to the sampling rate Ns of the A / D conversion units 17a and 18a at discrete time. Switch sequentially every time. In each embodiment of the present invention, since the parameter Ns = 2 is described as an example, two correlation value calculation units are provided, but the parameter Ns is not limited to two.

  Specifically, the S / P conversion unit 19a outputs the complex number x (k, M) at the discrete time k to the {mod (k-1, Ns) +1} -th correlation value calculation unit. Thus, the complex number (hereinafter referred to as discrete sample value) x (k, M) input to the first to Nsth correlation value calculation units has a sampling rate of 1 per pulse, and the sample timing is Tp / Ns = (Tw / L) / Ns shifted. Note that mod (k−1, Ns) is an operator that outputs a remainder obtained by dividing (k−1) using Ns.

In the following description, the discrete sample value output from the S / P conversion unit 19a to the p-th correlation value calculation unit is represented as y _p (q, M). The parameter p is p = 1,..., Ns. The parameter q is q = 1,... (Nr + Nu) / No.

The first correlation value calculation unit 20a receives the discrete sample value y _p (q, M) output by the S / P conversion unit 19a. The first correlation value calculation unit 20a establishes synchronization with the operation of the transmission signal generation unit 4, and similarly to the transmission signal generation unit 4, based on the reference signal generated in the reference signal oscillator Lo, A signal multiplied by a predetermined factor is generated. In FIG. 4, the input of the reference signal to the first correlation value calculation unit 20a is omitted.

First correlation value calculating section 20a, based on a signal multiplying the reference signal to a predetermined multiple, discrete time according to q, the pulse compression code a _n of the code length L to be transmitted in the M-th transmission period Tr ( M) is generated periodically. Here, parameters n = 1,..., L.

The first correlation computing unit 20a computes the inputted discrete sample values _y p (q, M) and the pulse compression code _a n (M) and the first correlation value _AC p (q, M) of the . The parameter AC _p (q, M) indicates the first correlation value at the discrete time q. In the first correlation value AC _p (q, M) calculated by the first correlation value calculation unit 20a, the parameter p is p = 1.

Specifically, the first correlation value calculation unit 20a performs the first correlation value AC _p (in accordance with Expression (10) in each transmission cycle Tr shown in FIG. 7, that is, at discrete times q = 1 to (Nr + Nu) / No. q, M). The first correlation value calculation unit 20a outputs the first correlation value AC _p (q, M) calculated according to Equation (10) to the addition processing unit 22a. The parameter u indicates the delay time.

  The first stage in FIG. 7 shows the transmission timing of the high-frequency transmission signal, and the second stage shows the reception timing of the reflected wave. The reflected wave is a wave in which the high-frequency transmission signal transmitted during the transmission section Tw is reflected by the target.

The second correlation value calculation unit 21a receives the discrete sample value y _p (q, M) output from the S / P conversion unit 19a. The second correlation value calculation unit 21a establishes synchronization with the operation of the transmission signal generation unit 4 so that, similarly to the transmission signal generation unit 4, the reference signal is generated based on the reference signal generated in the reference signal oscillator Lo. A signal multiplied by a predetermined factor is generated. In FIG. 4, the input of the reference signal to the second correlation value calculation unit 21a is omitted.

Second correlation value calculating unit 21a, based on a signal multiplying the reference signal to a predetermined multiple, discrete time according to q, the pulse compression code a _n of the code length L to be transmitted in the M-th transmission period Tr ( M) is generated periodically. Here, n = 1,..., L.

The second correlation computing unit 21a computes the input discrete sample values _y p (q, M), the pulse compression code _a n (M) and the second correlation value _AC p (q, M) of the . The parameter AC _p (q, M) indicates the second correlation value at the discrete time q. In the second correlation value AC _p (q, M) calculated by the second correlation value calculation unit 21a, the parameter p is p = 2.

Specifically, the second correlation value calculating unit 21a performs the second correlation value AC _p (in accordance with Expression (10) in each transmission cycle Tr shown in FIG. 7, that is, at discrete times q = 1 to (Nr + Nu) / No. q, M). The second correlation value calculation unit 21a outputs the second correlation value AC _p (q, M) calculated according to Equation (10) to the addition processing unit 23a.

  Each calculation of the first correlation value calculation unit 20a and the second correlation value calculation unit 21a is executed at discrete times q = 1 to (Nr + Nu) / No. Note that the measurement range (the range of q) is further narrowed to be, for example, q = (L + 1) to {((Nr + Nu) / No) −L} depending on the presence range of the target to be measured by the radar apparatus 1. You may do.

  As shown in the second stage of FIG. 7, the discrete time q = (L + 1) indicates the discrete time next to the end time of the transmission interval Tw. Further, the discrete time q = (L + 1) is a reception start time at which the reflected wave signal starts after a delay time τ1 from the discrete time q = 0. This delay time τ1 is expressed by Equation (11).

  Further, as shown in the second stage of FIG. 7, the discrete time q = {((Nr + Nu) / No) −L} is the time before the transmission interval Tw (= Tp × L) from the end time of the transmission cycle Tr. Equivalent to. Further, the discrete time q = {((Nr + Nu) / No) −L} is the reception start time when the reflected wave signal is started with a delay time τ2 from the discrete time q = 0. This delay time τ2 is expressed by Equation (12).

  Accordingly, the first correlation value calculation unit 20a and the second correlation value calculation unit 21a are represented by the above-described formula (10) at least in the range of the discrete time q = (L + 1) to {((Nr + Nu) / No) −L}. The first correlation value and the second correlation value may be calculated. Thereby, the radar apparatus 1 can reduce the calculation amount of the 1st correlation value calculating part 20a and the 2nd correlation value calculating part 21a. That is, the radar apparatus 1 can reduce the power consumption based on the reduction of the calculation amount by the signal processing unit 13a. The same applies to the other antenna system processing units 11b to 11d.

  Furthermore, the radar apparatus 1 does not measure in the transmission section Tw of the high-frequency transmission signal, and even if the high-frequency transmission signal directly wraps around the radar receiver 3, the radar apparatus 1 can perform measurement while eliminating the influence of the wraparound. By limiting the measurement range (the range of the discrete time q), a first addition processing unit 22a, a second addition processing unit 23a, a first arrival time direction estimation unit 24, a second arrival time direction estimation unit 25, which will be described later, The operation of the high resolution processing unit 26 is also limited to a similar measurement range.

The first addition processing unit 22a receives the first correlation value AC _p (q, M) output from the first correlation value calculation unit 20a. The first addition processing unit 22a adds and averages a plurality of R times using the first correlation value AC _p (q, M) multiplied in the M-th transmission cycle Tr as a unit. The parameter R is a natural number.

That is, the first addition processing unit 22a, the first correlation value in the m-th transmission period Tr _AC p (q, m) the (m + R-1) -th first correlation value _AC p (q in the transmission cycle Tr of , M + R−1) as a unit, the average first correlation value aveAC _p (q, M) is calculated with the timings of the discrete times q aligned. In the average first correlation value aveAC _p (q, M), the parameter p is p = 1. The first addition processing unit 22 a outputs the calculated average correlation value aveAC _p (q, M) to the first arrival time direction estimation unit 24. The parameter m is a multiple of the natural number R including zero.

The second addition processing unit 23a receives the second correlation value AC _p (q, M) output from the second correlation value calculation unit 21a. The second addition processing unit 23a performs addition averaging a plurality of R times using the second correlation value AC _p (q, M) multiplied in the Mth transmission cycle Tr as a unit.

That is, the second addition processing unit 23a, the second correlation value in the m-th transmission period Tr _AC p (q, m) the (m + R-1) -th second correlation value _AC p (q in the transmission cycle Tr of , M + R−1) as a unit, the average second correlation value aveAC _p (q, M) is calculated with the timings of the discrete times q aligned. In the average second correlation value aveAC _p (q, M), the parameter p is p = 2. The second addition processing unit 23 a outputs the calculated average correlation value aveAC _p (q, M) to the second arrival time direction estimation unit 25.

  By the operations of the first addition processing unit 22a and the second addition processing unit 23a, the radar apparatus 1 can further improve the SNR (Signal Noise Ratio) by further suppressing the noise component. Furthermore, the radar apparatus 1 can improve the measurement performance regarding the estimation of the arrival distance of the target.

The first arrival time direction estimation unit 24 outputs each average correlation value aveAC _p (q, M) output by each first addition processing unit 22a to 22d (not shown, the same applies hereinafter) of each antenna system processing unit 11a to 11d. Enter. The first arrival time direction estimation unit 24 calculates the received power of the reflected wave from the target based on each average correlation value aveAC _p (q, M) calculated by each antenna system processing unit using four reception antennas. Estimate and measure the arrival time based on the estimation result. Further, for each discrete time q, the arrival direction from the radar apparatus 1 to the target is estimated and calculated based on the reception phase difference of the radar measurement target by the plurality of reception antennas.

The estimation calculation result (estimation result) includes the arrival direction estimation result DOA _p (q, m) in the maximum received power direction and the received power estimation result POW _p (q, m). The first arrival time direction estimation unit 24 outputs the estimation result [POW _p (q, m), DOA _p (q, m)] to the high resolution processing unit 26.

  In addition, the estimation calculation of the arrival direction to the target by the 1st arrival time direction estimation part 24 is an already well-known technique, for example, can be implement | achieved by referring the following nonpatent literature 2.

  (Reference Non-Patent Document 2) JAMES A. Cadzow, “Direction of Arrival Estimating Signal Subspace Modeling”, IEEE, Vol. 64-79 (1992)

The second arrival time direction estimation unit 25 outputs each average correlation value aveAC _p (q, M) output by each of the second addition processing units 23a to 23d (not shown, the same applies hereinafter) of each of the antenna system processing units 11a to 11d. Enter. The second arrival time direction estimation unit 25 calculates the received power of the reflected wave from the target based on each average correlation value aveAC _p (q, M) calculated by each antenna system processing unit using four reception antennas. Estimate and measure the arrival time based on the estimation result. Further, the arrival direction from the radar apparatus 1 to the target is estimated and calculated for each discrete time q.

Similarly, the estimation calculation result (estimation result) includes the arrival direction estimation result DOA _p (q, m) in the maximum reception power direction and the reception power estimation result POW _p (q, m). The second arrival time direction estimation unit 25 outputs the estimation result [POW _p (q, m), DOA _p (q, m)] to the high resolution processing unit 26.

  In addition, the estimation calculation of the arrival direction to the target by the 2nd arrival time direction estimation part 25 is an already well-known technique, For example, it is realizable by referring the above-mentioned reference nonpatent literature 2. FIG.

The high resolution processing unit 26 inputs each estimation result [POW _p (q, m), DOA _p (q, m)] output by the first arrival time direction estimation unit 24a and the second arrival time direction estimation unit 25a. To do. The operation of the high resolution processing unit 26 will be described with reference to FIGS. 13 and 14.

The reception signal level determination unit 27 includes a reception power estimation result POW _p (q, m) as a predetermined reception signal among the input estimation results [POW _p (q, m), DOA _p (q, m)]. It is determined whether the level is equal to or higher than level Lc (S11). The predetermined reception signal level Lc is preferably set to a reception SNR that exceeds the predetermined target detection rate and is lower than the target false detection rate. That is, of the discrete times (p, q) determined by the parameters p and q, the estimation result DOUT _r (s, m), which is the discrete time (r, s) satisfying Equation (13), is sent to the coincidence determination range extraction unit 28. Output (S11). Here, the discrete time determined by the parameters p and q may be called measurement time or sample timing.

The coincidence determination range extraction unit 28 receives the estimation result DOUT _r (s, m) output from the reception signal level determination unit 27. The coincidence determination range extraction unit 28 is a predetermined time range corresponding to the sampling rate Ns corresponding to the measurement times before and after each discrete time (r, s), specifically twice the correlation spread range (2Ns−1). Among the estimation results DOA _p (q, m) corresponding to the number of s, the set extracted as DOUT _r (s, m) is output to the coincidence determination unit 29 as a unit of the coincidence determination range (S12).

  A set of estimation results corresponding to the unit of the matching determination range output by the matching determination range extraction unit 28 is extracted in the order of the estimation results shown in FIG.

  In addition, the match determination range extraction unit 28 shifts the set of estimation results for the number of units of the match determination range by ceil [(2Ns−1) / 2] according to the correlation spread range (2Ns−1). Things are further extracted (S13). Hereinafter, each estimation result of the match determination range extracted by the match determination range extraction unit 28 is denoted as DOUT_SEL (w).

Further, the coincidence determination range extraction unit 28 repeats the extraction until the estimation result DOUT _Ns ((Nr + Nu) / No, m) for the discrete sample value until the end of the discrete time in the transmission cycle Tr (S13). Similarly, each of these extracted match determination ranges is output to the match determination unit 29.

  Note that ceil [(2Ns-1) / 2] is an operator for rounding up the decimal part of (2Ns-1) / 2 to an integer value. For example, when Ns = 2, ceil [(2Ns−1) / 2] = ceil [1.5] is obtained, and each estimation result is shifted by two when the coincidence determination range extraction unit 28 extracts the coincidence determination range. It will be.

  A set of estimation results corresponding to the units of the extracted match determination ranges is shown in FIG. FIG. 10 shows the w-th, (w + 1) -th, and (w + 2) -th match determination ranges where Ns = 2. The parameter w is a natural number.

  Further, the match determination range extraction unit 28 divides each set of estimation results corresponding to the unit of the match determination range extracted in step S13 into, for example, three blocks (B1, B2, B3) (S14). An example in which each estimation result is divided is shown in FIG. For example, since it is difficult to equally divide into three, it is preferable to divide so that many estimation results are included in the central block B2, as shown in FIGS. 11 (b) and 11 (c). .

  The operations after step S15 are operations of the coincidence determination unit 29, and are operations performed on DOUT_SEL (w) extracted in steps S12 and S13.

  The coincidence determination unit 29 uses the set of estimation results included in the w-th coincidence determination range DOUT_SEL (w) output from the coincidence determination range extraction unit 28 to estimate the arrival direction estimation result from the radar apparatus 1 to the target. Determine the consistency of. Furthermore, the coincidence determination unit 29 repeats the operation of step S15 for all the coincidence determination ranges in a certain transmission cycle Tr extracted in steps S12 and S13. Details of the operation in step S15 will be described below.

The coincidence determination unit 29 includes the maximum received power estimation result (POW _{p_peak} (q_peak, m)) among the estimation results that are included in the w-th coincidence determination range DOUT_SEL (w) and satisfy Expression (10). Is selected (S15-1). In the coincidence determination unit 29, the discrete time (p_peak, q_peak) when obtaining the maximum received power estimation result of the selected estimation results is the discrete time in each estimation result of the central block B2 divided in step S14. Is determined (S15-2).

  Assume that it is determined that it is not a discrete time in each estimation result of the central block B2 (S15-2, NO). The coincidence determination unit 29 ends the operation for the wth coincidence determination range DOUT_SEL (w), and performs the operation of step S15 for the next (w + 1) th coincidence determination range DOUT_SEL (w + 1) (S15−). 4).

Assume that it is determined that it is a discrete time in each estimation result of the central block B2 (S15-2, YES). The coincidence determination unit 29 calculates the absolute value of the difference between the arrival direction estimation result DOA _{p_peak} (q_peak, m) of the measurement time (p_peak, q_peak) and the arrival direction estimation result with respect to the discrete sample values at discrete times before and after the time. Is less than a predetermined value (S15-3).

  This predetermined value is a threshold for determining whether or not there are a plurality of targets within the distance resolution ΔR of the radar apparatus 1 (see Equation (1)).

  Assume that it is determined in step S15-3 that the absolute value of the difference is less than a predetermined value (S15-3, YES). The coincidence determination unit 29 determines that a plurality of targets do not exist within the distance resolution ΔR of the radar device 1 (S15-5). Furthermore, the coincidence determination unit 29 outputs the maximum received power estimation result and the arrival direction estimation result in the estimation result including the maximum received power estimation result as a high resolution arrival direction estimation result (see FIG. 8).

  Assume that it is determined in step S15-3 that the absolute value of the difference is equal to or greater than a predetermined value (S15-3, NO). The coincidence determination unit 29 determines that there are a plurality of targets within the distance resolution ΔR of the radar device 1 (S15-6). Furthermore, the coincidence determination unit 29 has the coincidence of the arrival direction estimation results among the estimation results in the respective correlation spread ranges (2Ns−1) around the time centered on the maximum received power POWp_peak (q_peak, m). It is determined whether there is a high one (S15-7).

  The high coincidence of the arrival direction estimation results means that the reliability of the arrival direction estimation results in the radar apparatus 1 for each target is high after it is determined that there are a plurality of targets. For example, it is assumed that there are a target A and a target B as a plurality of targets. When the arrival direction estimation results are highly consistent, the relative delay in the arrival of the reflected wave signals reflected by the target A and the target B is equal to or greater than the pulse width Tp. The correlation spread range of the arrival direction estimation result for the target A and the arrival direction estimation result for the target B do not overlap.

  On the other hand, if the arrival direction estimation results are not highly consistent, the relative delay in arrival of the reflected wave signals reflected by the target A and the target B is less than the pulse width Tp. The correlation spread range of the arrival direction estimation result for the target A and the arrival direction estimation result for the target B overlap.

  Assume that it is determined in step S15-7 that there is a high match of arrival direction estimation results (S15-7, YES). The coincidence determination unit 29 obtains a high-resolution arrival direction estimation result (see FIG. 8) that has a high coincidence of the arrival direction estimation results among the estimation results in the correlation spread ranges before and after the time centered on the maximum received power. ) Is output (S15-8).

  Suppose that it is determined in step S15-7 that there is no high match in arrival direction estimation results (S15-7, NO). The coincidence determination unit 29 has a discrete time that is farthest in time among the discrete times satisfying Equation (10) from the estimation results in the correlation spread ranges before and after the time centered on the maximum received power estimation result. The direction-of-arrival estimation result is output (S15-9).

  Here, since there are a plurality of targets in the range below the distance resolution of the radar apparatus 1, as shown in FIG. 12, the signal of each reflected wave from each target has a relative delay time of 1.0 pulse width Tp or less. Suppose that it has come within.

  For example, as shown in FIG. 12A, it is assumed that the relative delay in the arrival of each reflected wave signal from each target is 1.0 pulse width Tp. The radar apparatus 1 selects arrival direction estimation results (reference numerals C1 and C2 in the figure) using correlation values at discrete times when the signals of the reflected waves are not synthesized. Thereby, the radar apparatus 1 can isolate | separate and detect the several target which exists within the distance resolution of the radar apparatus 1, and can improve the estimation precision of the arrival direction of a target. Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

  For example, as shown in FIG. 12B and FIG. 12C, the relative delay in arrival of each reflected wave signal from each target is 0.5 pulse width (Tp / 2) or 0.3... It is assumed that the width is (Tp / 3). The radar apparatus 1 determines the arrival direction estimation results using the correlation values at discrete times when the influence of the synthesis of the signals of the reflected waves is small (codes C3 and C4 in FIG. 12B, codes C5 and C6 in FIG. 12C). ) Is selected. Thereby, the radar apparatus 1 can isolate | separate and detect the several target which exists within the distance resolution of the radar apparatus 1, and can improve the estimation precision of the arrival direction of a target. Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

  The effect of the radar apparatus 1 of the first embodiment described above will be described with reference to FIG. FIG. 15 is an explanatory diagram schematically showing an estimation result of the high resolution processing unit 26. FIG. 6A is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is set to a pulse width of 1.5 Tp. FIG. 4B is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is defined as a pulse width Tp. FIG. 4C is an explanatory diagram in which the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is set to a pulse width Tp / 2. FIG. 4D is an explanatory diagram in which the relative delay in the arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is a pulse width Tp / 3.

In FIG. 14, it is assumed that the received signal level of the reflected wave signal from target A and the received signal level of the reflected wave signal from target B are substantially equal. That is, it is assumed that the target A and the target B exist in the range of the distance resolution of the radar apparatus 1. The coordinates of the center point of the circle (◯) in FIG. 14 indicate the arrival direction estimation result DOA _p (q, m), and the size of the circle indicates the size of the received power estimation result POW _p (q, m). .

  It is assumed that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is 1.5 Tp, which is a time longer than the pulse width Tp (see FIG. 15A). The correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B do not intersect. That is, the correlation spread range of the arrival direction estimation result for the target A matches the arrival direction estimation result in the correlation spread range of the arrival direction estimation result for the target B, and the radar apparatus 1 uses the arrival direction estimation results as targets A and B. It can be estimated as the true value of.

  Assume that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is equal to a pulse width of 1.0 Tp (see FIG. 15B). A discrete time is generated when the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B intersect. At this discrete time, the arrival direction estimation result for the target A and the arrival direction estimation result for the target B are combined in a vector manner. For this reason, the reception power estimation result at discrete times may be high, but the reception power estimation result and arrival direction estimation result are likely to be incorrect.

  In the conventional radar apparatus, the arrival direction estimation value at the discrete time when the reception power estimation result is high is selected, or the arrival direction estimation result at the discrete time before and after the discrete time is synthesized as the final estimation result. . For this reason, in the conventional radar apparatus, when there are a plurality of targets within the distance resolution of the radar apparatus, an estimation error with respect to the arrival direction estimation result for each target becomes large.

  In the radar apparatus 1 according to the present invention, the high resolution processing unit 26 performs the operation of step S15-8 described above. Therefore, the radar apparatus 1 uses the arrival direction estimation result at a discrete time when the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B do not intersect as the received power estimation value. Can be selected and output. Thereby, according to the radar apparatus 1, when there are a plurality of targets within the distance resolution, the direction estimation accuracy of each target can be improved even if the Doppler frequency of the signal of each reflected wave from each target is approximately the same. . Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

  Assume that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is equal to the pulse width Tp / 2 (see FIG. 15C). Similarly, a discrete time occurs when the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B intersect. At this discrete time, the arrival direction estimation result for the target A and the arrival direction estimation result for the target B are combined in a vector manner. For this reason, the reception power estimation result at discrete times may be high, but the reception power estimation result and arrival direction estimation result are likely to be incorrect.

  In the conventional radar apparatus, the arrival direction estimation value at the discrete time when the reception power estimation result is high is selected, or the arrival direction estimation result at the discrete time before and after the discrete time is synthesized as the final estimation result. . For this reason, in the conventional radar apparatus, when there are a plurality of targets within the distance resolution of the radar apparatus, an estimation error with respect to the arrival direction estimation result for each target becomes large.

  In the radar apparatus 1 according to the present invention, the high resolution processing unit 26 performs the operation of step S15-9 described above. Therefore, the radar apparatus 1 selects the arrival direction estimation result at the discrete time that is the most discrete in time where the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B do not intersect. Output. Thereby, according to the radar apparatus 1, when there are a plurality of targets within the distance resolution, the direction estimation accuracy of each target can be improved even if the Doppler frequency of the signal of each reflected wave from each target is approximately the same. . Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

  It is assumed that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is equal to the pulse width Tp / 3 (see FIG. 15D). Similarly, a discrete time occurs when the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B intersect. At this discrete time, the arrival direction estimation result for the target A and the arrival direction estimation result for the target B are combined in a vector manner. For this reason, the reception power estimation result at discrete times may be high, but the reception power estimation result and arrival direction estimation result are likely to be incorrect.

  In the conventional radar apparatus, the arrival direction estimation value at the discrete time when the reception power estimation result is high is selected, or the arrival direction estimation result at the discrete time before and after the discrete time is synthesized as the final estimation result. . For this reason, in the conventional radar apparatus, when there are a plurality of targets within the distance resolution of the radar apparatus, an estimation error with respect to the arrival direction estimation result for each target becomes large.

  In the radar apparatus 1 according to the present invention, the high resolution processing unit 26 performs the operation of step S15-9 described above. Therefore, the radar apparatus 1 selects the arrival direction estimation result at the discrete time that is the most discrete in time where the correlation spread range of the arrival direction estimation result for the target A and the correlation spread range of the arrival direction estimation result for the target B do not intersect. Output. Thereby, according to the radar apparatus 1, when there are a plurality of targets within the distance resolution, the direction estimation accuracy of each target can be improved even if the Doppler frequency of the signal of each reflected wave from each target is approximately the same. . Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

  Furthermore, since the radar apparatus 1 according to the first embodiment does not use the filter bank disclosed in Patent Document 1 described above, an increase in the circuit scale of the radar apparatus 1 can be suppressed.

  In the first embodiment, the correlation spread range has been described as (2Ns-1). However, the correlation spread range may be set to 2Ns in consideration that the reception timing (arrival timing) of the reflected wave signals from the plurality of targets in the radar receiver 3 does not coincide with the sampling rate of the radar receiver 3.

[Variation 1 of the first embodiment]
In the first modification of the first embodiment, the operation of step S15-9 of the coincidence determination unit 29 of the high resolution processing unit 26 of the radar device 1 of the first embodiment is replaced with the operation described later. Since the other operations of the radar apparatus according to the first modification of the first embodiment excluding the replaced operation are the same as those of the radar apparatus 1 according to the first embodiment, only the contents of the replacement operation will be described below. Description of other operations other than the described operations is omitted.

  In the radar device according to the first modification of the first embodiment, the coincidence determination unit corresponds to step S15-9 of the coincidence determination unit 29 of the high resolution processing unit 26 of the radar device 1 according to the first embodiment. The operation like this is performed.

Specifically, the coincidence determination unit focuses on the maximum received power estimation result, and the DOA estimation result DOA at the discrete time farthest from the correlation spread range on the front side among the discrete times satisfying Equation (13). Select _p1 (q1, m). Further, the coincidence determination unit centering on the maximum received power estimation result, the arrival direction estimation result DOA _p2 (at the discrete time farthest from the correlation spread range on the rear side among the discrete times satisfying Equation (13). q2, m).

Further, the coincidence determination unit uses the arrival direction estimation result DOA _{p_peak} (q_peak, m) at the discrete time (p_peak, q_peak) when obtaining the maximum received power estimation result, to select the arrival direction estimation result DOA _p1 ( q1, m) and DOA _p2 (q2, m) are extrapolated.

This extrapolation correction is performed as shown in Equation (14) and Equation (15). The coincidence determination unit outputs the corrected arrival direction estimation results cDOA _p1 (q1, m) and cDOA _p2 (q2, m) as high-resolution arrival direction estimation results (see FIG. 8). For example, linear interpolation is used for extrapolation. The parameter α is a predetermined coefficient.

The parameter α may be a fixed value, but the received power estimation results POW _p1 (q1, m), POW _p2 (q2, m) at discrete times (p1, q1), (p2, q2), (p_peak, q_peak). ) And POW _{p_peak} (q_peak, m). Further, in Equation (14) and Equation (15), ΔT [(p1, q1) − (p_peak, q_peak)] represents a time difference between the discrete time (p1, q1) and the discrete time (p_peak, q_peak).

FIG. 16 is an explanatory diagram schematically illustrating an estimation result of the high resolution processing unit of the radar apparatus according to the first modification of the first embodiment. In FIG. 16, arrival direction estimation results cDOA _p1 (q1, m) and cDOA _p2 (q2, m) calculated by the extrapolation correction described above are represented by black circles (●). However, the parameter α = 1.

  As described above, the radar apparatus according to the first modification of the first embodiment can improve the direction estimation accuracy of each target more than the radar apparatus 1 according to the first embodiment by the extrapolation correction described above. Can do. In particular, it is remarkable that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is equal to the pulse width Tp / 3.

  If the relative delay in the arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is equal to the pulse width Tp / 2 or Tp / 3, step S15- in FIG. 9 and the operation of extrapolation correction may be used in combination.

Specifically, it is assumed that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is about a pulse width Tp / 3. The coincidence determination unit focuses on the maximum received power estimation result, and the arrival direction estimation result DOA _p1 (q1, m at the discrete time farthest from the correlation spread range in the temporal direction among the discrete times satisfying Expression (13). ) Is selected. Further, the coincidence determination unit centering on the maximum received power estimation result, the arrival direction estimation result DOA _p2 (at the discrete time most distant from the correlation spread range in the time among the discrete times satisfying Expression (10). q2, m).

Further, the coincidence determination unit uses the arrival direction estimation result DOA _{p_peak} (q_peak, m) at the discrete time (p_peak, q_peak) when obtaining the maximum received power estimation result, to select the arrival direction estimation result DOA _p1 ( q1, m) and DOA _p2 (q2, m) are extrapolated.

  On the other hand, it is assumed that the relative delay in arrival of the reflected wave signal from the target A and the reflected wave signal from the target B is about the pulse width Tp / 2. The coincidence determination unit has the arrival direction at the most discrete time among the discrete times satisfying Expression (10) from the estimation results in the respective correlation spread ranges before and after the time centered on the maximum received power estimation result. Output the estimation result.

  Thereby, the direction estimation accuracy of each target can be further improved as compared with the radar apparatus according to the first modification of the first embodiment. Further, distance measurement can be performed from the arrival times of the separated targets, and the distance estimation accuracy is improved.

[Modification 2 of the first embodiment]
In Modification 2 of the first embodiment, instead of Ns times oversampling in the radar receiver 3 of the radar apparatus 1 of the first embodiment, the output timing of the transmission code in the radar transmitter 2 is set to a predetermined sampling rate. Is periodically shifted every transmission cycle Tr.

  FIG. 17 is a block diagram illustrating in detail the internal configuration of the radar apparatus 1x according to the second modification of the first embodiment. FIG. 18 is a timing chart regarding the operation of the radar apparatus 1x according to the second modification of the first embodiment.

  Components having the same configuration and operation in each part of the radar device 1x and the radar device 1 of the first embodiment are denoted by the same reference numerals. Hereinafter, in the description of the configuration and operation of the radar apparatus 1x, the contents of the same configuration and operation will be omitted, and contents different from the configuration and operation of the radar apparatus 1 will be described. In the following description, the sampling rate Ns is assumed to be Ns = 2, but is not limited to Ns = 2.

  In FIG. 17, a radar apparatus 1x includes a transmission timing control unit 33, a radar transmission unit 2x connected to a transmission antenna AN1, and antenna system processing units 11ax to 11dx (not shown) connected to the reception antennas AN2a to AN2d. , The same shall apply hereinafter). The radar transmitter 2x and the radar receiver 3x are both connected to the reference signal oscillator Lo, and a signal is supplied from the reference signal oscillator Lo, so that the processing of the radar transmitter 2x and the radar receiver 3x is synchronized. ing.

(Radar transmitter)
Next, the configuration of each part of the radar transmitter 2x will be described.

  As shown in FIG. 17, the radar transmission unit 2 x includes a transmission signal generation unit 4 x and a transmission RF unit 5. The transmission signal generation unit 4x includes a code generation unit 6, a modulation unit 7, an LPF 8, a delay unit 31, and a transmission timing switching unit 32.

  Next, the operation of each part of the radar transmitter 2x will be described.

  The transmission signal generation unit 4x generates a signal obtained by multiplying the reference signal by a predetermined multiple based on the reference signal generated by the reference signal oscillator Lo. Each unit of the transmission signal generation unit 4x operates based on the generated signal.

  The baseband transmission signal r (k, M) generated by the modulation unit 7 is output to the delay unit 31 and the transmission timing switching unit 32 via the LPF 8.

  The delay unit 31 gives a delay of a predetermined delay time ΔD to the transmission signal r (k, M) output via the LPF 8, and receives the transmission signal r (k, M) given the delay time ΔD. The data is output to the transmission timing switching unit 32. For this predetermined delay time ΔD, Equation (16) is established.

  The transmission timing switching unit 32 receives the transmission signal r (k, M) output through the LPF 8 and the transmission signal r (k, M) output from the delay unit 31. The transmission timing switching unit 32 transmits the transmission signal r (k, M) output via the LPF 8 or the transmission signal r output by the delay unit 31 according to the transmission timing control signal output by the transmission timing control unit 33. Switches the output of (k, M). The transmission timing switching unit 32 outputs the transmission signal r (k, M) selected by the switching to the transmission RF unit 5.

  The transmission timing control unit 33 operates based on a signal obtained by multiplying the reference signal generated by the reference signal oscillator Lo by a predetermined factor. The transmission timing control unit 33 transmits a transmission timing control signal for switching the output of the transmission timing switching unit 32 for each transmission cycle Tr or every integral multiple of the transmission cycle Tr based on the signal, and for the transmission timing switching unit 32 and the correlation. The value is output to the value calculation input switching unit 34a. The transmission timing control signal includes a signal for switching the output of the correlation value calculation input switching unit 34a.

(Radar receiver)
Next, the configuration of each part of the radar receiver 3x will be described. A plurality of, for example, four array antennas are configured in the radar receiver 3x, as in the radar device 1 of the first embodiment. Further, similarly to the radar apparatus 1 of the first embodiment, each receiving antenna constituting the array antenna is provided for each antenna system processing unit.

  As shown in FIG. 17, the radar receiver 3x includes four antenna system processors 11ax to 11dx, a first arrival time direction estimator 24, a second antenna provided in correspondence with the number of receiving antennas constituting the array antenna. An arrival time direction estimation unit 25 and a high resolution processing unit 26 are provided. In the following description, since the configuration and operation of each of the four antenna system processing units 11ax to 11dx are the same, the antenna system processing unit 11ax will be described as an example.

  The antenna system processing unit 11ax includes a reception RF unit 12a to which the reception antenna AN2a is connected, and a signal processing unit 13ax. The reception RF unit 12a includes an amplifier 14a, a frequency conversion unit 15a, and a quadrature detection unit 16a. The signal processing unit 13ax includes A / D conversion units 17a and 18a, a correlation value calculation input switching unit 34a, a first correlation value calculation unit 20a, a second correlation value calculation unit 21a, a first addition processing unit 22a, and a second addition process. A portion 23a is provided. The radar receiver 3x periodically calculates each transmission cycle Tr as a signal processing section in signal processors 13ax to 13dx (not shown, the same applies hereinafter).

  Next, the operation of each part of the radar receiver 3x will be described.

  The in-phase signal component of the digital data converted by the A / D conversion unit 17a and the quadrature signal component of the digital data converted by the A / D conversion unit 18a are output to the correlation value calculation input switching unit 34a.

  The correlation value calculation input switching unit 34a inputs the in-phase signal component and the quadrature signal component of the digital data output from the A / D conversion unit 17a and the A / D conversion unit 18, respectively. The correlation value calculation input switching unit 34a transmits the output of the complex number x (k, M) composed of the input in-phase signal component and quadrature signal component according to the sampling rate Ns of the A / D conversion units 17a, 18a. In accordance with the transmission timing control signal output by the timing control unit 33, switching is performed sequentially at discrete times. The correlation value calculation input switching unit 34a outputs the complex number x (k, M) selected by switching to the first correlation value calculation unit 20a or the second correlation value calculation unit 21a.

  FIG. 18 shows a transmission timing control signal, a high-frequency transmission signal, a reception signal with a delay time τ1, sample timings of the A / D conversion units 17a and 18a, and an output destination of the correlation value calculation input switching unit 34a. .

  As shown in FIG. 18, when the transmission timing control signal does not select the output from the delay unit 31 (delay output OFF), the high-frequency transmission signal is transmitted over the code transmission period Tw in synchronization with the start timing of the transmission cycle Tr. Is done. On the other hand, when the transmission timing control signal selects the output from the delay unit 31 (delay output ON), the high-frequency transmission signal is transmitted over the code transmission section Tw with a delay of the delay time ΔD described above from the start timing of the transmission cycle Tr. Is done.

  Therefore, when the delay device output is OFF, the radar device 1x starts receiving the high-frequency transmission signal at the timing of the delay time τ1 from the start timing of the transmission cycle Tr. On the other hand, when the delay device output is ON, the radar apparatus 1x starts reception of the high-frequency transmission signal at the delay time (τ1 + ΔD) from the start timing of the transmission cycle Tr.

  On the other hand, since sampling of an integral multiple is performed in synchronization with the transmission cycle Tr, the sample timing of the A / D converters 17a and 18b is the delay time ΔD (formula) when the delayer output is OFF and the delayer output is ON. (Refer to (13)). By setting the delay time ΔD to a value smaller than the pulse width Tp, a discrete sample value using the sampling rate Ns of the first embodiment can be equivalently obtained.

  The correlation value calculation input switching unit 34a outputs the discrete sample value x (k, M) with the delay unit output OFF to the first correlation value calculation unit 20a, and the discrete sample value x (k, M) with the delay unit output ON. Is output to the Ns (= 2) th correlation calculation unit (second correlation value calculation unit 21a). Subsequent operations are the same as those in the first embodiment.

  As described above, the radar device 1x according to the second modification of the first embodiment receives the reflected wave from the target using a unit smaller than the sample timing of the A / D conversion units 17a and 18a of the radar receiving unit 3x. be able to. Thereby, according to the radar apparatus 1x, the sampling rate (oversample number) at the time of the reception of the reflected wave from the target can be equivalently increased.

  As a result, the radar device 1x uses a sampling rate similar to the sampling rate of the A / D conversion obtained in the radar receiver 3 of the radar device 1 of the first embodiment, and a slower A / D converter. Can also be achieved.

  In general, if the A / D converter has a slower sampling rate, it is easier to increase the number of quantization bits. For this reason, the reception dynamic range in the radar receiver 3x can be increased, and the detection accuracy of targets including vehicles / pedestrians can be increased. Furthermore, the power consumption in the radar apparatus 1x can be reduced by using an A / D converter having a low sampling rate.

[Modification 3 of the first embodiment]
In the third modification of the first embodiment, instead of the S / P converter 19a of the radar receiver 3 of the radar apparatus 1 of the first embodiment, two more A / D converters having a low sampling rate are provided. Use one.

  FIG. 19 is a block diagram illustrating in detail the internal configuration of the first antenna system processing unit 11ay in the radar receiving unit 3y of the radar apparatus 1y according to the third modification of the first embodiment. FIG. 20 is a timing chart regarding the operation of each A / D conversion unit in the first antenna system processing unit 11ay in the radar receiving unit 3y of the radar device 1y according to the third modification of the first embodiment.

  Components having the same configuration and operation in each part of the radar apparatus 1y and the radar apparatus 1 of the first embodiment are denoted by the same reference numerals. The configuration of the radar transmission unit of the radar device 1y is the same as that of the radar transmission unit 2 of the radar device 1 of the first embodiment shown in FIG. 4, and FIG. 19 shows the configuration of the radar transmission unit of the radar device 1y. Omitted.

  Hereinafter, in the description of the configuration and operation of the radar apparatus 1y, the contents of the same configuration and operation will be omitted, and contents different from the configuration and operation of the radar apparatus 1 will be described. In the following description, the sampling rate Ns is assumed to be Ns = 2, but is not limited to Ns = 2.

  In the radar receiver 3y of the radar device 1y and the radar receiver 3 of the radar device 1 of the first embodiment, the first antenna system processor 11ay is different from the first antenna system processor 11a. The other configurations and operations of the first arrival time direction estimation unit 24, the second arrival time direction estimation unit 25, and the high resolution processing unit 26 are the same, and thus the description of the configuration and operation is omitted.

(Antenna system processing part)
Next, the configuration of each part of the antenna system processing unit 11ay will be described. Note that a plurality of, for example, four array antennas are configured in the radar receiver 3y of the radar apparatus 1y, as in the radar apparatus 1 of the first embodiment. Further, similarly to the radar apparatus 1 of the first embodiment, each receiving antenna constituting the array antenna is provided for each antenna system processing unit.

  As shown in FIG. 19, the antenna system processing unit 11ay includes a reception RF unit 12a to which a reception antenna AN2a is connected, and a signal processing unit 13ay. The reception RF unit 12a includes an amplifier 14a, a frequency conversion unit 15a, and a quadrature detection unit 16a. The signal processor 13ay includes a delay unit 35a, four A / D converters 17a1, 17a2, 18a1, and 18a2, a first correlation value calculator 20a, a second correlation value calculator 21a, a first addition processor 22a, and a second adder. An addition processing unit 23a is provided. The antenna system processing unit 11ay periodically calculates each transmission cycle Tr as a signal processing section in the signal processing units 13ay to 13dy (not shown, the same applies hereinafter).

  Next, the operation of each part of the antenna system processing unit 11ay will be described.

  The delay unit 35a gives a delay of a predetermined delay time ΔTD to a signal obtained by multiplying the reference signal output from the reference signal oscillator Lo by a predetermined multiple, and the signal provided with the delay time ΔTD is supplied to the A / D converter. Input to 18a1 and 18a2, respectively. With respect to the predetermined delay time ΔTD, Expression (17) is established.

  The A / D converter 17a1 samples 2 Tp / Ns at a timing synchronized with a signal obtained by multiplying the reference signal output from the reference signal oscillator Lo by a predetermined value with respect to the in-phase signal output from the quadrature detector 16a. Sampling using rate. The A / D converter 17a1 converts the in-phase signal of analog data into digital data. The A / D converter 17a1 outputs the in-phase signal component of the converted digital data to the first correlation value calculator 20a.

  The A / D converter 17a2 has a sampling rate of 2 Tp / Ns at a timing synchronized with a signal obtained by multiplying the quadrature signal output by the quadrature detector 16a by a predetermined multiple of the reference signal output by the reference signal oscillator Lo. To sample. Thereby, the A / D conversion unit 17a2 converts the orthogonal signal of the analog data into digital data. The A / D converter 17a2 outputs the orthogonal signal component of the converted digital data to the first correlation value calculator 20a.

  The A / D conversion unit 18a1 is a timing output from the delay unit 35a with respect to a signal obtained by multiplying the reference signal output by the reference signal oscillator Lo by a predetermined factor with respect to the in-phase signal output by the quadrature detection unit 16a. , Sampling is performed using a sampling rate of 2 Tp / Ns. Thereby, the A / D conversion unit 18a1 converts the in-phase signal of the analog data into digital data. The A / D converter 18a1 outputs the in-phase signal component of the converted digital data to the second correlation value calculator 21a.

  The A / D conversion unit 18a2 is configured to output a signal obtained by multiplying the quadrature signal output from the quadrature detection unit 16a by a predetermined multiple to the reference signal output from the reference signal oscillator Lo at the timing output from the delay unit 35a. Sampling is performed using a sampling rate of 2 Tp / Ns. Thereby, the A / D conversion unit 18a2 converts the orthogonal signal of the analog data into digital data. The A / D converter 18a2 outputs the orthogonal signal component of the converted digital data to the second correlation value calculator 21a.

  As shown in FIG. 20, the sample timings of the A / D converters 17a1 and 17a2 and the A / D converters 18a1 and 18a2 have a relative delay of the delay time ΔTD shown by the equation (17). Further, the A / D converters 17a1 and 17a2 and the A / D converters 18a1 and 18a2 respectively sample using the sampling rate 2Tp / Ns, and the sampled discrete sample values are compared with the first correlation value calculator 20a and the second correlation value calculator 20a. It outputs to the correlation value calculating part 21a.

  Therefore, the delay unit 35a and the A / D converters 17a1, 17a2, 18a1, and 18a2 described above, the A / D converters 17a and 18a and the S / P converter 19a with the sampling rate Tp / Ns in the first embodiment It is possible to perform the same operation.

  Further, the radar apparatus 1y according to the third modification of the first embodiment requires a plurality of A / D conversion units as compared with the radar apparatus 1 according to the first embodiment. However, in the radar apparatus 1y, the sampling rate of each A / D conversion unit can be further reduced.

  Thereby, according to the radar apparatus 1y, it becomes easy to increase the quantization accuracy at the time of A / D conversion in each A / D converter, that is, to increase the number of quantization bits, and the reception dynamic range in the radar apparatus 1y. Can be increased.

  In the antenna system processing unit 11ay shown in FIG. 19, the A / D conversion units 17a1, 17a2, 18a1, and 18a2 are provided. However, more A / D conversion units may be provided, and similar effects are provided. Can be obtained. For this reason, the reception dynamic range in the radar receiver 3y can be increased, and the detection accuracy of targets including vehicles / pedestrians can be increased. Furthermore, the power consumption in the radar apparatus 1y can be reduced by using an A / D converter having a low sampling rate.

[Modification 4 of the first embodiment]
In the fourth modification of the first embodiment, an arrival time direction selection unit 36 is further provided in the radar receiver 3 of the radar apparatus 1 of the first embodiment. By providing the arrival time direction selection unit 36, the processing amount and processing time of the first arrival time direction selection unit 24z, the second arrival time direction selection unit 25z, and the high resolution processing unit 26 can be reduced.

  FIG. 21 is a block diagram illustrating in detail the internal configuration of the radar receiver 3z of the radar apparatus 1z according to the fourth modification of the first embodiment. Components having the same configuration and operation in each part of the radar device 1z and the radar device 1 of the first embodiment are denoted by the same reference numerals. The configuration of the radar transmission unit of the radar device 1z is the same as that of the radar transmission unit 2 of the radar device 1 of the first embodiment shown in FIG. 4, and FIG. 21 shows the configuration of the radar transmission unit of the radar device 1z. Omitted.

  Hereinafter, in the description of the configuration and operation of the radar apparatus 1z, the contents of the same configuration and operation will be omitted, and contents different from the configuration and operation of the radar apparatus 1 will be described. In the following description, the sampling rate Ns is assumed to be Ns = 2, but is not limited to Ns = 2.

  Compared with the radar receiver 3 of the radar device 1 of the first embodiment, the radar receiver 1z of the radar device 1z is further provided with an arrival time direction selector 36, and the first arrival time direction estimator 24z and the second The arrival time direction selection unit 25z is different from the first arrival time direction estimation unit 24 and the second arrival time direction estimation unit 25 in operation. The other antenna system processing units 11a to 11d are the same as the antenna system processing units 11a to 11d of the radar receiving unit 3 of the radar apparatus 1 according to the first embodiment, and thus the description thereof is omitted.

(Radar receiver)
Next, the configuration of each part of the radar receiver 3z will be described. A plurality of, for example, four array antennas are configured in the radar receiver 3z.

  As shown in FIG. 21, the radar receiver 3z includes four antenna system processors 11a to 11d, a first arrival time direction estimator 24z, a second antenna provided in correspondence with the number of receiving antennas constituting the array antenna. An arrival time direction estimation unit 25z, an arrival time direction selection unit 36, and a high resolution processing unit 26 are provided. In the following description, the antenna system processing unit 11a among the four antenna system processing units 11a to 11d will be exemplified and described as necessary.

In the radar receiver 3z of the radar apparatus 1z, the average first correlation value aveAC ₁ (q, M) output by the first addition processor 22a is sent to the first arrival time direction estimation unit 24z and the arrival time direction selection unit 36. Is output. The average second correlation value aveAC ₂ (q, M) output by the second addition processing unit 23a is output to the second arrival time direction estimation unit 25z.

  As shown in FIG. 21, the average first correlation values output from the first addition processing units of the other antenna system processing units 11b to 11d are respectively the first arrival time direction estimation unit 24z and the arrival time direction selection. Is output to the unit 36. The average second correlation values output by the second addition processing units of the other antenna system processing units are output to the second arrival time direction estimation unit 25z, respectively.

  Based on the average first correlation values output from the first addition processing units of the antenna system processing units 11a to 11d, the arrival time direction selection unit 36 determines that the reflected wave signal from the target has a predetermined received signal level. Choose a higher discrete time. The arrival time direction selection unit 36 obtains selected discrete time information including the selected discrete time and discrete times before and after the discrete time as the first arrival time direction estimation unit 24z and the second arrival time direction estimation unit 25z. Respectively.

  Hereinafter, the operation of the arrival time direction selection unit 36 will be described in detail.

  The arrival time direction selection unit 36 receives the average first correlation values output by the first addition processing units of the antenna system processing units 11a to 11d. The arrival time direction selection unit 36 calculates the average received power of the array antenna at each discrete time (z, q) based on each input average first correlation value according to Equation (18). The parameter q is q = 1 to (Nr + Nu) / No.

The arrival time direction selection unit 36 selects a discrete time (z, q _s ) in which the average received power calculated according to Equation (18) is greater than the predetermined received power Lp. The parameter q _s is a discrete time satisfying Expression (19) among q _s = 1 to (Nr + Nu) / No.

The arrival time direction selecting unit 36 selects the discrete time (z, q _s ), the discrete time (z, q _s −1) and (z, q _s ) that are temporally adjacent to the discrete time (z, q _s ). Discrete time information including _s + 1) is output to the first arrival time direction estimation unit 24z and the second arrival time direction estimation unit 25z as selected discrete time information. Below, the information of said discrete time is described as (q, m). (Q, m) includes a discrete time (z, q _s ) selected on the condition that the mathematical formula (19) is satisfied, and a discrete time (z, q _s ) that is before and after the discrete time (z, q _s ). q _s -1) and (z, q _s +1).

  The first arrival time direction estimation unit 24z is included in the selected discrete time information output by the arrival time direction selection unit 36 among the average first correlation values output by the first addition processing unit of each antenna system processing unit. The average first correlation value at the discrete time (q, m) is extracted. The first arrival time direction estimation unit 24z estimates the reception power of the reflected wave from the target based on the extracted average first correlation value, and measures the arrival time based on the estimation. Further, the arrival direction from the radar apparatus 1z to the target is estimated and calculated for each selected discrete time.

The estimation calculation result (estimation result) includes an arrival direction estimation result DOA _p (maximum received power direction at each discrete time (q, m) included in the selected discrete time information output by the arrival time direction selection unit 36 (DOA _p ( q, m) and the received power estimation result POW _p (q, m). The first arrival time direction estimation unit 24z outputs the estimation result [POW _p (q, m), DOA _p (q, m)] to the high resolution processing unit 26. Here, p = 1.

  Similarly, the second arrival time direction estimation unit 25z selects selected discrete time information output by the arrival time direction selection unit 36 from the average second correlation values output by the second addition processing unit of each antenna system processing unit. The average second correlation value at each discrete time (q, m) included in is extracted. The second arrival time direction estimation unit 25z estimates the reception power of the reflected wave from the target based on the extracted average second correlation value, and measures the arrival time based on the estimation. Further, the arrival direction from the radar apparatus 1z to the target is estimated and calculated for each selected discrete time.

The estimation calculation result (estimation result) includes an arrival direction estimation result DOA _p (maximum received power direction at each discrete time (q, m) included in the selected discrete time information output by the arrival time direction selection unit 36 (DOA _p ( q, m) and the received power estimation result POW _p (q, m). The second arrival time direction estimation unit 25z outputs the estimation result [POW _p (q, m), DOA _p (q, m)] to the high resolution processing unit 26. Here, p = 2.

The high resolution processing unit 26 estimates the arrival direction of the maximum received power direction at each discrete time (q, m) of the selected discrete time information output by the first arrival time direction estimation unit 24z and the second arrival time direction estimation unit 25z. Each estimation result [POW _p (q, m), DOA _p (q, m)] which is the result DOA _p (q, m) and the received power estimation result POW _p (q, m) is input. Here, p = 1,2. The following description of the operation of the high-resolution processing unit 26 is the same as that of the high-resolution processing unit 26 of the radar apparatus 1 of the first embodiment described above, and is therefore omitted.

  As described above, each arrival time direction estimation unit arrives at the target only for a part of discrete times selected based on the average received power satisfying Expression (19) from all the discrete times q = 1 to (Nr + Nu) / No. Estimate direction. Thereby, according to the radar apparatus 1z, it is possible to reduce the estimated processing amount and the estimated processing time while preventing the target detection rate from deteriorating.

  While various embodiments have been described above with reference to the drawings, it goes without saying that the radar apparatus of the present invention is not limited to such an example. It will be apparent to those skilled in the art that various changes and modifications can be made within the scope of the claims, and these are naturally within the technical scope of the present invention. Understood.

In the fourth modification of the first embodiment, the arrival time direction selection unit 36 performs the following process instead of selecting the discrete time (z, q) that satisfies the above-described equation (16). Also good. Specifically, the arrival time direction selection unit 36 obtains each N_Lp average received power in order from the largest average received power from the average received power ave_P (q, m) calculated according to Equation (18). Select each discrete time. Each selected discrete time is represented as (z, q _{s ′} )). The parameter q _{s ′} is the upper N_Lp discrete times among the discrete times q = 1 to (Nr + Nu) / No sorted in descending order of the average received power ave_P (q, m).

  Thereby, since the upper limit of the discrete time in the arrival direction estimation process of each arrival time direction estimation unit is fixed, it is possible to suppress the estimation processing amount and the estimation processing time required to the maximum in the radar apparatus 1z. .

  Since the distance RT to the target can be converted as RT = C × TOA / 2 based on the arrival time estimation result TOA, the arrival time estimation in this embodiment is replaced with the arrival distance estimation. Also good. Here, the parameter C indicates the speed of light.

  The present invention is useful as a radar device that improves the direction estimation accuracy of each target even if the Doppler frequency of each reflected wave signal from each target is the same when a plurality of targets exist within the distance resolution.

1, 1x, 1y, 1z Radar device 2, 2x Radar transmitter 3, 3x, 3z Radar receiver 4, 4r, 4x Transmit signal generator 5 Transmit RF unit 6 Code generator 7, 7r Modulator 8, 8r LPF
9, 15a Frequency conversion unit 10, 14a Amplifier 11a, 11b, 11c, 11d, 11ax, 11ay, 11bx, 11cx, 11dx Antenna system processing unit 12a Reception RF unit 13a, 13ax, 13ay Signal processing unit 16a Quadrature detection unit 17a, 17a1 , 17a2, 18a, 18a1, 18a2 A / D converter 19a S / P converter 20a First correlation value calculator 21a Second correlation value calculator 22a First addition processor 23a Second addition processor 24, 24z First Arrival time direction estimation unit 25, 25z Second arrival time direction estimation unit 26 High resolution processing unit 27 Reception signal level determination unit 28 Match determination range extraction unit 29 Match determination unit 31, 35a Delay unit 32 Transmission timing switching unit 33 Transmission timing control Unit 34a correlation value calculation input switching unit 36 arrival time direction selection unit AN1 transmission antenna AN2a receiving antenna CM transmission code storage unit CT3 transmission code controller Lo reference signal oscillator Tp pulse width Tr transmission period Tw transmission interval .tau.1, .tau.2 delay time

Claims (18)

A radar transmitter that converts a pulse compression code into a high-frequency transmission signal and transmits the high-frequency transmission signal from a transmission antenna;
The arrival time and the arrival direction to the target using the reflected wave signal that is the high-frequency transmission signal reflected by the target, using a sampling rate N (integer) times the pulse transmission rate in the pulse compression code, A radar receiver for estimation, and
The radar receiver
A plurality of antenna system processing units for outputting a correlation value between a received signal and the pulse compression code;
N arrival time direction estimation units that individually estimate the arrival time and arrival direction of the target based on the outputs of the plurality of antenna system processing units;
A plurality of targets existing within the range resolution of the high-frequency transmission signal based on the respective estimation results in the N arrival time direction estimation units; Including
Each of the plurality of antenna system processing units is
A receiving antenna for receiving the reflected wave signal;
A receiving RF unit for converting the received signal into a baseband received signal;
An A / D converter that converts the converted received signal into digital data using a sampling rate of N (integer) times;
A radar apparatus comprising: N correlation value calculation units for calculating a correlation value between the digital data of the received signal converted at every 1 / N sample timing and the pulse compression code. The radar apparatus according to claim 1,
Each estimation result includes a reception power estimation result at each sample timing,
The high-resolution processor is
A radar apparatus having a reception signal level determination unit that determines whether or not the reception power estimation result is equal to or higher than a predetermined reception signal level. The radar apparatus according to claim 2,
The high-resolution processor is
A radar apparatus, further comprising: a coincidence determination range extraction unit that extracts each estimation result in a predetermined time range before and after a sample timing at which the reception power estimation result is equal to or higher than the predetermined reception signal level. The radar apparatus according to claim 3,
The coincidence determination range extraction unit further includes:
A radar apparatus that extracts the number of estimation results after being shifted in time according to the sampling rate in units of the number of estimation results in each of the predetermined time ranges. The radar device according to claim 3 or 4,
The coincidence determination range extraction unit further includes:
A radar apparatus that divides each estimation result of the extracted number into a predetermined number of blocks. The radar apparatus according to claim 5,
The high-resolution processor is
A radar apparatus further comprising: a coincidence determination unit that selects an estimation result including a maximum received power estimation result from each of the extracted estimation results in each predetermined time range. The radar apparatus according to claim 6, wherein
The match determination unit further includes:
A radar apparatus that determines whether a sample timing when obtaining the selected estimation result is a sample timing in each estimation result of a central block among the divided blocks. The radar apparatus according to claim 7, wherein
Each of the estimation results further includes a direction of arrival estimation result at each of the sample timings,
The match determination unit further includes:
By determining that the sampling timing when obtaining the selected estimation result is the sampling timing in each estimation result of the central block, the arrival direction estimation result of the selected estimation result is temporally continuous. A radar apparatus that determines whether or not the difference between the estimation result of each estimation result at the preceding and following sample timings is less than a predetermined value. The radar apparatus according to claim 8, wherein
The match determination unit further includes:
The distance resolution of the high-frequency transmission signal is determined by determining that the difference between the direction estimation result of the selected estimation result and the direction estimation result of each estimation result at the sample timing before and after time continuation is less than a predetermined value. Radar device that determines that there are no multiple targets within. The radar apparatus according to claim 8 or 9, wherein
The match determination unit further includes:
The distance resolution of the high-frequency transmission signal is determined by determining that the difference between the direction estimation result of the selected estimation result and the direction estimation result of each estimation result at the sample timing before and after time continuation is greater than a predetermined value. Radar device that determines that there are multiple targets within. The radar apparatus according to claim 10, wherein
The match determination unit further includes:
A radar apparatus that determines whether or not there is an arrival direction estimation result with high coincidence of arrival direction estimation results from each estimation result in a predetermined time range before and after the maximum reception power estimation result. The radar apparatus according to claim 11, wherein
The match determination unit further includes:
Based on the determination that there is an arrival direction estimation result with high consistency of the arrival direction estimation result, an arrival direction estimation result with high consistency of the arrival direction estimation result is output from each estimation result in the predetermined time range before and after Radar device. The radar apparatus according to claim 11 or 12,
The match determination unit further includes:
Each of the received power estimation results equal to or higher than the predetermined reception signal level is determined from the estimation results in the predetermined time range before and after the determination that there is no arrival direction estimation result with high consistency of the arrival direction estimation results. A radar apparatus that outputs each arrival direction estimation result at a sample timing that is most discrete in time within a predetermined time range before and after the sample timing of the estimation result. The radar apparatus according to claim 11 or 12,
The match determination unit further includes:
Each of the received power estimation results equal to or higher than the predetermined reception signal level is determined from the estimation results in the predetermined time range before and after the determination that there is no arrival direction estimation result with high consistency of the arrival direction estimation results. A radar apparatus that extrapolates each arrival direction estimation result at a sample timing that is most temporally discrete in a predetermined time range before and after the sample timing of the estimation result. The radar apparatus according to any one of claims 1 to 14,
The antenna system processing unit is
A radar apparatus further comprising: an S / P converter that sequentially switches the output of the converted digital data of the received signal to the N correlation value calculators according to the sampling rate. The radar apparatus according to any one of claims 1 to 14,
The radar transmitter
A first delay unit that gives a delay of a predetermined time to the output of the transmission code of the predetermined code length;
A transmission timing switching unit that switches the output of the transmission code or the transmission code given a delay of the predetermined time, and
The antenna system processing unit is
A correlation value calculation input switching unit that switches output of the digital data of the converted reception signal to the N correlation value calculation units;
The radar device is
A radar apparatus comprising: a transmission timing control unit that outputs a control signal for switching an output of the transmission timing switching unit and an output of the correlation value calculation input switching unit for each transmission cycle of the high-frequency transmission signal. The radar apparatus according to any one of claims 1 to 14,
The antenna system processing unit is
A second delay unit that gives a delay of a predetermined time to the reception signal converted by the reception RF unit;
A radar apparatus comprising: a second A / D converter that converts the received signal given the delay of the predetermined time into digital data using the predetermined multiple sampling rate. The radar apparatus according to any one of claims 1 to 14,
A radar apparatus, further comprising: an arrival time direction selection unit that selects a sample timing at which a level of a reflected wave reflected by the target is equal to or higher than a predetermined reception signal level according to outputs of the N correlation calculation units.

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