JP2012095528A - Controlling device for rotary machine - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve the problem that controllability of torque drops when switching from the control by a torque feedback control part to a current feedback control part.SOLUTION: A current feedback control part 20 performs maximum torque control. A voltage estimator 48 estimates a norm of a voltage vector for achieving an estimated torque Te by the maximum torque control with the estimated torque Te and electric angular speed ω as input. A switch control part 46 switches to the control by the current feedback control part 20 when the norm is below a threshold.

Description

The present invention relates to a rotating machine that controls an actual torque of the rotating machine to a required torque by operating a power conversion circuit including a switching element that electrically connects each of a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine. about the control equipment of.

  This type of control device calculates a command value (command voltage) of a voltage to be applied to each phase in order to feedback control the current flowing in each phase of the three-phase motor to a command value, A device that performs PWM control for operating the switching element of the inverter based on the size of the carrier has been put into practical use. Thereby, the voltage applied to each phase of the three-phase motor can be used as a command voltage, and the current flowing through each phase can be controlled as desired.

  However, in the high rotation speed region of the three-phase motor, the command voltage rises and the amplitude becomes “½” or more of the input voltage of the inverter, so that the actual output voltage of the inverter becomes the command voltage. Can not be. Here, in the high rotation speed region of the three-phase motor, so-called rectangular wave control is also put into practical use in which the ON / OFF cycle of the switching element of the inverter substantially matches the rotation cycle of the electrical angle of the three-phase motor. Yes. However, the voltage utilization factor of the rectangular wave control is discontinuously large compared to the voltage utilization factor at the time when the amplitude of the command voltage in the PWM control becomes a value of “½” of the input voltage of the inverter. It has become.

  Therefore, conventionally, for example, as seen in Patent Document 1 below, when the amplitude of the command voltage of the three-phase motor is “½” or more of the input voltage of the inverter, the current on the dq axis for current feedback control It has also been proposed to operate the inverter based on the phase calculated based on the command voltage and the pulse pattern stored in the ROM. Specifically, when the vector norm of the command voltage is greater than or equal to a predetermined value, the inverter is operated with the pulse pattern stored in the ROM. Thereby, a voltage utilization factor can be raised to the voltage utilization factor of rectangular wave control.

JP-A-9-47100

  By the way, when the required torque is suddenly reduced under the condition that the inverter is operated based on the pulse pattern stored in the ROM, the vector norm of the command voltage is suddenly reduced and switched to PWM control. However, in this case, the voltage required for realizing the torque generated by the electric motor may become a voltage that cannot be realized by PWM control. In this case, torque controllability is reduced.

The present invention has been made to solve the above problems, upon which control the actual torque of the rotating machine to the required torque, the rotating machine capable of maintaining a high controllability of times turning point to provide a control equipment.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

  According to the first aspect of the present invention, the actual torque of the rotating machine is changed to the required torque by operating a power conversion circuit including a switching element that electrically connects each of the positive electrode and the negative electrode of the DC power source to the terminal of the rotating machine. In the control device for the rotating machine to be controlled, the first control means for operating the power conversion circuit by PWM processing to control the voltage applied to the rotating machine to a command voltage, and the electric power according to the electrical angle of the rotating machine At the time of control by either one of the second control means for operating the power conversion circuit based on the operation signal waveform of the power conversion circuit, and the first control means and the second control means, information related to the control is input, Switching means for switching to one of the other controls, and the operation signal waveform includes a voltage utilization factor smaller than that of the rectangular wave control.

1 is a system configuration diagram according to a first embodiment. FIG. The block diagram which shows the process regarding the production | generation of the operation signal of the inverter concerning the embodiment. The figure which shows the detail of a process of the command electric current setting part concerning the embodiment. The figure which shows the setting method of the voltage vector norm concerning the embodiment. The figure which shows the detail of a process of the norm calculation part concerning the embodiment. The figure which shows the detail of a process of the voltage estimator concerning the embodiment. The flowchart which shows the procedure of the switching process of the current feedback control and torque feedback control concerning the embodiment. The flowchart which shows the procedure of the setting process of the initial value at the time of the said switching. The flowchart which shows the procedure of the switching process of the current feedback control and torque feedback control concerning 2nd Embodiment. The flowchart which shows the procedure of the switching process of the current feedback control concerning 3rd Embodiment, and torque feedback control. The flowchart which shows the procedure of the switching process of the current feedback control concerning 4th Embodiment, and torque feedback control. The flowchart which shows the procedure of the switching process of the current feedback control concerning 5th Embodiment, and torque feedback control. The time chart which points out the problem of 5th Embodiment. The flowchart which shows the procedure of the switching process of the current feedback control concerning 6th Embodiment, and torque feedback control. The flowchart which shows the procedure of the switching process of the current feedback control and torque feedback control concerning 7th Embodiment. The flowchart which shows the procedure of the switching process of the electric current feedback control and torque feedback control concerning 8th Embodiment. The system block diagram concerning 9th Embodiment. The figure which shows the detail of a process of the norm calculation part concerning 10th Embodiment. The system block diagram concerning 11th Embodiment.

(First embodiment)
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a hybrid vehicle will be described with reference to the drawings.

  FIG. 1 shows the overall configuration of a motor generator control system according to this embodiment. The motor generator 10 is a three-phase permanent magnet synchronous motor. The motor generator 10 is a rotating machine (saliency pole machine) having saliency. Specifically, the motor generator 10 is an embedded magnet synchronous motor (IPMSM).

  The motor generator 10 is connected to a high voltage battery 12 via an inverter IV and a boost converter CV. Here, the boost converter CV boosts the voltage (for example, “288V”) of the high-voltage battery 12 up to a predetermined voltage (for example, “666V”). On the other hand, the inverter IV includes a series connection body of the switching elements Sup and Sun, a series connection body of the switching elements Svp and Svn, and a series connection body of the switching elements Swp and Swn. The points are connected to the U, V, and W phases of the motor generator 10, respectively. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements Sup, Sun, Svp, Svn, Swp, and Swn. In addition, diodes Dup, Dun, Dvp, Dvn, Dwp, and Dwn are connected in antiparallel to these.

  In this embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter IV. First, a rotation angle sensor 15 that detects a rotation angle θ (electrical angle) of the motor generator 10 is provided. Further, current sensors 16, 17, and 18 that detect currents iu, iv, and iw flowing through the phases of the motor generator 10 are provided. Furthermore, a voltage sensor 19 for detecting an input voltage (power supply voltage VDC) of the inverter IV is provided.

  The detection values of the various sensors are taken into the control device 14 constituting the low pressure system via the interface 13. The control device 14 generates and outputs an operation signal for operating the inverter IV and the converter CV based on the detection values of these various sensors. Here, the signals for operating the switching elements Sup, Sun, Svp, Svn, Swp, Swn of the inverter IV are the operation signals gup, gun, gvp, gvn, gwp, gwn. The signals for operating the two switching elements of the boost converter CV are the operation signals gup and gcn.

  FIG. 2 shows a block diagram of processing relating to generation of the operation signal of the inverter IV.

As illustrated, the present embodiment includes a current feedback control unit 20 and a torque feedback control unit 30. Hereinafter, “processing of current feedback control unit 20”, “processing of torque feedback control unit 30”, “design of current feedback control unit 20”, “design of torque feedback control unit 30”, “processing of current feedback control unit 20” The process will be described in the order of “switching process between process and process of torque feedback control unit 30”.
<Processing of Current Feedback Control Unit 20>
The currents iu, iv, iw flowing through the phases of the motor generator 10 are converted into an actual current id on the d axis and an actual current iq on the q axis, which are actual currents in the rotating two-phase coordinate system, in the two-phase conversion unit 40. Converted. Specifically, the high-frequency component of the q-axis component of the output of the two-phase conversion unit 40 is cut by the low-pass filter 23, and the high-frequency component of the d-axis component output by the two-phase conversion unit 40 is converted by the low-pass filter 24. Cut. On the other hand, the command current setting unit 22 sets a command current idc on the d axis and a command current iqc on the q axis, which are current command values of the rotating two-phase coordinate system, based on the required torque Td. The feedback control unit 25 calculates a command voltage vdc on the d axis as an operation amount for performing feedback control of the actual current id on the d axis to the command current idc. On the other hand, the feedback control unit 26 calculates a command voltage vqc on the q axis as an operation amount for performing feedback control of the actual current iq on the q axis to the command current iqc. Specifically, the feedback controllers 25 and 26 perform the above calculation using proportional integral control.

The three-phase conversion unit 28 converts the command voltages vdc and vqc in the rotating two-phase coordinate system into three-phase command voltages vuc, vvc and vwc. The PWM signal generation unit 29 generates operation signals gup, gun, gvp, gvn, gwp, and gwn by PWM processing based on the three-phase command voltages vuc, vvc, and vwc and the power supply voltage VDC. In the present embodiment, in particular, an operation signal is generated based on a magnitude comparison between a signal obtained by two-phase modulation of the three-phase command voltages vuc, vvc, and vwc and normalized by the power supply voltage VDC and a triangular wave carrier.
<Processing of Torque Feedback Control Unit 30>
The torque estimator 42 calculates an estimated torque Te that is an estimated value of the torque of the motor generator 10 based on the actual currents id and iq of the rotating two-phase coordinate system. On the other hand, the deviation calculating unit 32 calculates a difference between the required torque Td and the estimated torque Te. The phase setting unit 34 sets the phase δ in the rotating two-phase coordinate system of the output voltage of the inverter IV based on the proportional integral calculation of the output of the deviation calculating unit 32. Here, when the estimated torque Te is insufficient with respect to the required torque Td, the phase δ is advanced, and when the estimated torque Te is excessive with respect to the required torque Td, the phase δ is retarded. To do.

  The norm setting unit 36 sets the norm Vn of the output voltage vector of the inverter IV in the rotating two-phase coordinate system based on the electrical angular velocity ω of the motor generator 10 and the required torque Td. Here, the norm of the vector is defined by the square root of the sum of the squares of the components of the vector. Specifically, the norm setting unit 36 includes a norm calculation unit 36a that calculates the norm Vn1 based on the required torque Td and the electrical angular velocity ω. Further, the norm setting unit 36 includes a PI control unit 36b. The PI control unit 36b calculates the norm Vn2 by proportional-integral calculation of the difference between the norm Vn2 that is the output and the norm Vn1 that is output from the norm calculation unit 36a. In the selector 36c, when the difference between the norm Vn2 output from the PI control unit 36b and the norm Vn1 output from the norm calculation unit 36a is larger than a specified value, the output of the PI control unit 36b is set as the norm Vn set by the norm setting unit 36. adopt. On the other hand, when the difference between the norm Vn2 output from the PI control unit 36b and the norm Vn1 output from the norm calculation unit 36a is equal to or less than a specified value, the output of the norm calculation unit 36a is adopted as the norm Vn set by the norm setting unit 36. To do.

  On the other hand, in the operation signal generation unit 38, the operation signals gup, gun, and the like are based on the phase δ set by the phase setting unit 34, the norm Vn set by the norm setting unit 36, the power supply voltage VDC, and the rotation angle θ. gvp, gvn, gwp, and gwn are generated. Specifically, the operation signal generation unit 38 stores an operation signal waveform for one rotation period of the electrical angle as map data for each voltage utilization rate. Here, as shown in FIG. 2, the stored operation signal waveforms are all periods in which the high potential side switching elements Sup, Svp, Swp are turned on and the low potential side switching elements Sun, The waveform is such that the period during which Svn and Swn are turned on is halved. This is a setting for making the output voltage of the inverter IV balanced in one rotation cycle of the electrical angle. Furthermore, each of the operation signal waveforms has symmetry with respect to the center (180 °) of one rotation period of the electrical angle, as exemplified in one of the waveforms in FIG. Specifically, the logical values of a pair of timings that are equidistant with respect to the center are reversed. Here, the logic “H” corresponds to the ON state of the switching elements Sup, Svp, Swp on the high potential side, and the logic “L” corresponds to the ON state of the switching elements Sun, Svn, Swn on the low potential side. This is a setting for approximating the output voltage of the inverter IV to a sinusoidal voltage as much as possible.

  The operation signal generation unit 38 calculates a voltage utilization rate based on the power supply voltage VDC and the norm Vn, and selects a corresponding operation signal waveform accordingly. Here, the upper limit of the voltage utilization rate is set to “0.78”, which is the voltage utilization rate during rectangular wave control. For this reason, when the voltage utilization rate reaches the maximum value “0.78”, the operation signal waveform is the switching element Sup, Svp, A waveform (one pulse waveform) is selected in which the period in which Swp is turned on and the period in which the low potential side switching elements Sun, Svn, Swn are turned on are generated once. On the other hand, the lower limit of the voltage utilization rate is an upper limit value that can realize three line voltages corresponding to the command voltages vuc, vvc, and vwc set by the current feedback control unit 20 by the input voltage of the inverter IV. 0.71 ". That is, it is set to an upper limit value that can realize a two-phase modulated signal by the input voltage of the inverter IV.

When the operation signal waveform is selected in this way, the operation signal generation unit 38 generates an operation signal by setting the output timing of this waveform based on the phase δ set by the phase setting unit 34.
<Design of Current Feedback Control Unit 20>
FIG. 3 shows details of the command current setting unit 22. In the present embodiment, the command currents idc and iqc are set so as to realize the maximum torque control. Here, the maximum torque control is a control that realizes the maximum torque with the minimum current, and the maximum torque here is a positive maximum torque during power running control, and a negative absolute value during regenerative control. It means the maximum torque. This will be described in detail below.

  The torque T of the motor generator 10 is expressed by the following equation (c1) using the armature winding interlinkage magnetic flux number Φ, the q-axis inductance Lq, the d-axis inductance Ld, the resistance R, and the number P of pole pairs. .

ここで、図示されるように、指令電流ｉｄｃ，ｉｑｃを、（−Ｉｓｉｎβ、Ｉｃｏｓβ）とすると、以下の式（ｃ２）を得る。

Here, as shown in the figure, when the command currents idc and iqc are (−Isin β, Icos β), the following equation (c2) is obtained.

上記の式（ｃ２）において、最大トルク制御の条件、すなわちトルクＴのβによる偏微分がゼロであるとの条件を課すことで、βは、以下の式（ｃ３）となる。

In the above equation (c2), by imposing the condition of maximum torque control, that is, the condition that the partial differentiation of the torque T by β is zero, β becomes the following equation (c3).

上記の式（ｃ３）は、位相βが電流Ｉの関数であることを意味する。したがって、上記の式（ｃ２）により、トルクＴを電流Ｉのみの関数ｓ（Ｉ）にて表現することが可能である。一方、上記の式（ｃ３）は、電流Ｉが位相βによって表現できることをも意味する。このため、上記の式（ｃ２）により、トルクＴを位相βのみの関数ｕ(β)によって表現することができる。そして、これら関数ｓ（Ｉ）及び関数ｕ（Ｉ）の逆関数の値を算出することで、要求トルクＴｄを入力として、最大トルク制御を実現するための指令電流ｉｄｃ，ｉｑｃを出力させることができる。
＜トルクフィードバック制御部３０の設計＞
先の図２に示したように、トルクフィードバック制御部３０では、要求トルクＴｄ及び電気角速度ωに基づき、ノルムＶｎを設定した。これにより、要求トルクＴｄが与えられた場合に、比較的自由にノルムＶｎを設定することができる。このため、例えばノルムＶｎを極力小さくすることで、電圧利用率を抑制することができる。そしてこの場合には、操作信号生成部３８の生成する操作信号波形として、よりパルス数の多い波形を選択することなどができ、ひいてはインバータＩＶの出力電圧を正弦波形状の電圧により近づけることができる。このため、インバータＩＶの出力電圧の高調波歪を低減することができ、ひいては高調波電流を抑制することが可能となる。

The above equation (c3) means that the phase β is a function of the current I. Therefore, the torque T can be expressed by the function s (I) of only the current I by the above formula (c2). On the other hand, the above formula (c3) also means that the current I can be expressed by the phase β. For this reason, the torque T can be expressed by the function u (β) of only the phase β by the above equation (c2). Then, by calculating values of inverse functions of the functions s (I) and u (I), the command currents idc and iqc for realizing the maximum torque control can be output with the required torque Td as an input. it can.
<Design of torque feedback control unit 30>
As shown in FIG. 2, the torque feedback control unit 30 sets the norm Vn based on the required torque Td and the electrical angular velocity ω. Thereby, when the required torque Td is given, the norm Vn can be set relatively freely. For this reason, for example, voltage utilization can be suppressed by making norm Vn as small as possible. In this case, a waveform having a larger number of pulses can be selected as the operation signal waveform generated by the operation signal generation unit 38. As a result, the output voltage of the inverter IV can be made closer to a sinusoidal voltage. . For this reason, the harmonic distortion of the output voltage of the inverter IV can be reduced, and consequently the harmonic current can be suppressed.

  Hereinafter, the setting of the norm Vn0 (norm before the guard process is performed on the norm Vn1, which is the final output) by the norm calculation unit 36a according to the present embodiment will be described. FIG. 4 shows basic restrictions imposed on the norm Vn0 during powering of the motor generator 10 in this embodiment. As shown in the figure, in the present embodiment, there is a restriction that the norm Vn0 is within the region surrounded by the boundary lines BL1 to BL4. Here, the boundary line BL4 indicates that the voltage utilization factor is “0.78”. This corresponds to the fact that the voltage utilization factor of the rectangular wave control is the maximum voltage utilization factor that can be realized. In the following, prior to derivation of the conditions corresponding to each of the boundary lines BL1 to BL3, an expression expressing the torque T and current vector (id, iq) of the motor generator 10 by the norm Vn0, the phase δ, and the electrical angular velocity ω. Is derived.

  Here, the voltage equation becomes the following equation (c4).

上記の式（ｃ４）から、下記の式（ｃ５）を得る。

From the above formula (c4), the following formula (c5) is obtained.

上記の式（ｃ５）を上記の式（ｃ１）に代入することで、下記の式（ｃ６）を得る。

By substituting the above equation (c5) into the above equation (c1), the following equation (c6) is obtained.

ここで、図４に示す境界線ＢＬ１に対応する条件である「トルクＴの位相δによる偏微分が正となるとの条件」は、上記の式（ｃ６）に基づき、下記の式（ｃ７ａ）及び（ｃ７ｂ）によって表現される。この条件は、要求トルクＴｄに対して推定トルクＴｅが不足する場合に、位相δを進角させることによってその不足を低減させて且つ、要求トルクＴｄに対して推定トルクＴｅが過剰である場合に、位相δを遅角させることによってその過剰分を低減させることを可能とするための条件である。

Here, the condition corresponding to the boundary line BL1 shown in FIG. 4 that is “the condition that the partial differentiation of the torque T by the phase δ is positive” is based on the above formula (c6) and the following formula (c7a) and (C7b). This condition is obtained when the estimated torque Te is insufficient with respect to the required torque Td, the shortage is reduced by advancing the phase δ, and the estimated torque Te is excessive with respect to the required torque Td. This is a condition for making it possible to reduce the excess by retarding the phase δ.

なお、本実施形態では、位相δは、力行時には、「０≦δ＜π／２」、回生時には、「π／２＜δ≦３π／２」となるように制御設計をするとの前提を設けているため、位相δが「３π／２」以上となる条件を削除した。また、境界線ＢＬ２に対応する条件である「ｄ軸電流がゼロ以下であるとの条件」は、下記の式（ｃ８）等よって表現される。

In the present embodiment, it is assumed that the control design is such that the phase δ is “0 ≦ δ <π / 2” during power running and “π / 2 <δ ≦ 3π / 2” during regeneration. Therefore, the condition that the phase δ is “3π / 2” or more is deleted. Further, the “condition that the d-axis current is zero or less”, which is a condition corresponding to the boundary line BL2, is expressed by the following equation (c8) or the like.

また、境界線ＢＬ３に対応する条件である「ｑ軸電流が力行時においてはゼロ以上である旨の条件」は、下記の式（ｃ９）によって表現される。

Further, “a condition that the q-axis current is zero or more during powering”, which is a condition corresponding to the boundary line BL3, is expressed by the following equation (c9).

なお、回生時においては、ｑ軸電流がゼロ以下であるとの条件を課す。

Note that a condition that the q-axis current is zero or less is imposed during regeneration.

  In the present embodiment, the norm Vn0 is set within the allowable region as exemplified in FIG. The norm Vn0 is not uniquely determined even if the phase δ and the electrical angular velocity ω are set. For this reason, since there is a certain degree of freedom in setting the norm Vn0, the norm Vn0 can be designed freely. Here, from the viewpoint of suppressing harmonic distortion of the output voltage of the inverter IV, it is desirable to reduce the norm Vn0 as much as possible. In order to minimize the norm Vn0, it is required to impose a condition that the partial differential coefficient of the torque T by the norm Vn0 is zero. However, in this case, when using the above model, the inventors have found that it is impossible to have a one-to-one correspondence between the phase δ and the norm Vn0.

  Therefore, in the present embodiment, the norm Vn0 is set so that the maximum torque control for realizing the maximum torque with the minimum current can be performed. This also makes it possible to reduce the norm Vn0 as much as possible in realizing the required torque Td.

  FIG. 5 shows details of the norm calculation unit 36a according to the present embodiment.

  As shown in the figure, in the present embodiment, a condition expressed by the following equation (c10) is imposed to execute the maximum torque control.

この式は、例えば「埋込磁石同期モータの設計と制御：武田洋次ら オーム社」の２３ページに記載されている。上記の式（ｃ１０）から、上記の式（ｃ５）によって電流ベクトル（ｉｄ，ｉｑ）を消去することで、ノルムＶｎ０を、電気角速度ωと位相δとの関数とすることができる。特に、トルクフィードバック制御を行う領域は電気角速度ωが大きい領域のため、抵抗Ｒを無視することで、以下の式（ｃ１１）とすることができる。

This equation is described, for example, on page 23 of “Design and Control of Embedded Magnet Synchronous Motor: Yoji Takeda et al. Ohm Company”. By eliminating the current vector (id, iq) from the above equation (c10) by the above equation (c5), the norm Vn0 can be made a function of the electrical angular velocity ω and the phase δ. In particular, since the region where torque feedback control is performed is a region where the electrical angular velocity ω is large, the following equation (c11) can be obtained by ignoring the resistance R.

上記の式（ｃ１１）では、ノルムＶｎ０が、位相δ及び電気角速度ωの関数とされている。以下では、これに基づき、ノルムＶｎ０を、要求トルクＴｄと電気角速度ωとによって表現することを考える。上記の式（ｃ６）において、抵抗Ｒが小さいとする近似を行うことで、下記の式（ｃ１２）を得る。

In the above equation (c11), the norm Vn0 is a function of the phase δ and the electrical angular velocity ω. In the following, based on this, it is considered that the norm Vn0 is expressed by the required torque Td and the electrical angular velocity ω. In the above equation (c6), the following equation (c12) is obtained by approximating that the resistance R is small.

上記の式（ｃ１２）における関数ｆは、位相δを独立変数として、電気角速度ωによって規格化されたノルムＶｎ０（速度規格化ノルム）を従属変数とするものである。ここで、ノルムＶｎ０が、電気角速度ωに依存しない関数ｆと電気角速度ωとの積として定義できるのは、上記の式（ｃ１１）を根拠としている。すなわち、抵抗Ｒが無視できるとの近似を前提としている。上記の式（ｃ１２）によれば、位相δを独立変数として且つトルクＴを従属変数とする関数ｇを定義することができる。このため、関数ｇの逆関数を用いることで、上記関数ｆの独立変数を位相δからトルクＴに変換することができる。これにより、トルクＴを独立変数として且つ速度規格化ノルム（Ｖｎ０／ω）を従属変数とする関数ｈを定義することができる。

The function f in the above equation (c12) has a phase δ as an independent variable and a norm Vn0 (speed normalized norm) normalized by the electrical angular velocity ω as a dependent variable. Here, the reason why the norm Vn0 can be defined as the product of the function f and the electrical angular velocity ω that does not depend on the electrical angular velocity ω is based on the above equation (c11). That is, it is assumed that the resistance R can be ignored. According to the above equation (c12), the function g having the phase δ as an independent variable and the torque T as a dependent variable can be defined. For this reason, the independent variable of the function f can be converted from the phase δ to the torque T by using the inverse function of the function g. As a result, the function h having the torque T as an independent variable and the speed normalization norm (Vn0 / ω) as a dependent variable can be defined.

  FIG. 5 shows details of the process of the norm calculation unit 36a shown in FIG. As shown in the figure, the norm calculating unit 36a multiplies the map 50 having the required torque Td as an independent variable and the speed normalized norm (Vn0 / ω) as a dependent variable, and the output of the map 50 by the electrical angular velocity ω. And a voltage limiter 54 for performing a guard process on the norm Vn0 output from the multiplier 52. Here, when it is determined by the norm Vn0 that the voltage usage rate is “0.71” or less, the voltage limiter 54 performs guard processing so that the voltage usage rate is “0.71”. On the other hand, when the voltage usage rate becomes “0.78” or more due to the norm Vn0, guard processing is performed so that the voltage usage rate becomes “0.78”. The output of the voltage limiter 54 thus subjected to the guard process is the norm Vn1 output from the norm calculation unit 36a.

  The map 50 is calculated by calculating the function f indicating the relationship between the phase δ and the speed normalized norm and the function g indicating the relationship between the phase δ and the required torque Td by numerical calculation. As a result, the relationship between the required torque Td and the norm Vn0 can be mapped in advance so that the norm Vn0 is within the region shown in FIG. As a result, when the required torque Td and the electrical angular velocity ω are given, the motor generator 10 can be driven with the minimum current.

In this specification, a “map” is a mapping in which one output value is defined for each discrete input value.
<Switching process between process of current feedback control unit 20 and process of torque feedback control unit 30>
In the present embodiment, current feedback control is performed in a region where the voltage utilization factor is small, and torque feedback control is performed in a region where the voltage utilization factor is large. This is in view of the fact that controllers having high actual torque responsiveness to the required torque Td differ depending on the voltage utilization rate. However, in this case, if the responsiveness of the current feedback control is reduced due to an increase in the voltage utilization factor, it is necessary to switch to torque feedback control. Further, when the responsiveness of the torque feedback control is lowered due to the decrease in the voltage utilization rate, it is necessary to switch to the current feedback control. Here, when each of the current feedback control unit 20 and the torque feedback control unit 30 sets a threshold value for switching to the other based on its own voltage information, the following problem occurs.

  That is, when the required torque Td changes suddenly or when a sudden load change occurs, the voltage information changes suddenly and exceeds the switching threshold. However, in this case, depending on the control by the controller after switching, a situation may occur in which the required voltage vector cannot be realized. Here, in the torque feedback control unit 30, the actual torque adjustable range by the operation of the phase δ is large, but the operation amount of the current feedback control unit 20 is only the command voltages vdc and vqc, which is a problem. That is, when switching to the control by the current feedback control unit 20 when the required torque Td is suddenly decreased, there is a possibility that the control by the current feedback control cannot be appropriately realized immediately after the switching due to the limitation of the power supply voltage VDC.

  Therefore, in this embodiment, a common reference is provided to both controllers without using the output of each controller when switching. As described above, this utilizes the fact that both the current feedback control unit 20 and the torque feedback control unit 30 design the controller so as to perform maximum torque control, as described above. That is, in this case, it is possible to determine which controller is appropriate for control based on voltage information (information on the voltage utilization rate) for realizing the required torque Td by maximum torque control.

  Specifically, as shown in FIG. 2, the present embodiment includes a voltage estimator 48. As shown in FIG. 6, the voltage estimator 48 includes a map 56 and a multiplier 58 having the same configuration as the map 50 and the multiplier 52 of the norm calculation unit 36a shown in FIG. In other words, the norm calculation unit 36a is configured to exclude the voltage limiter 54. However, actual torque (estimated torque Te) is used as an input parameter instead of the required torque Td.

  Based on the norm (estimated norm Vne) of the two-dimensional voltage vector output from the voltage estimator 48, the switching control unit 46 shown in FIG. 2 performs control by the current feedback control unit 20 or by the torque feedback control unit 30. Switch whether to control. FIG. 7 shows a procedure of processing performed by the switching control unit 46. This process is repeatedly executed at a predetermined cycle, for example.

  In this series of processes, first, in step S10, the norm of the voltage vector (the estimated norm Vne) when the estimated torque Te is realized by the maximum torque control is estimated. This process is the process of the voltage estimator 48. In a succeeding step S12, it is determined whether or not the estimated norm Vne is larger than the threshold value Vth. This process is for determining which of the control by the current feedback control unit 20 and the control by the torque feedback control unit 30 is used. Here, the threshold value Vth is set to a norm that is a voltage utilization factor (“0.71”) corresponding to an upper limit value that can realize a signal obtained by two-phase modulation of the command voltages vuc, vvc, and vwc by the inverter IV. Yes.

  If it is determined that the value is larger than the threshold value Vth, in step S14, the torque feedback control mode is set to perform control by the torque feedback control unit 30. On the other hand, if it is determined that the threshold value Vth is equal to or less than the threshold value Vth, the current feedback control mode is set in step S16 in order to perform control by the current feedback control unit 20.

  FIG. 8 shows a procedure of initial value setting processing immediately after the switching. This process is repeatedly executed at a predetermined cycle, for example. In this series of processes, when it is time to switch to the process of the torque feedback control unit (step S20: YES), initial values of the phase δ and norm Vn2 by the torque feedback control unit 30 are given in step S22. Here, the initial value δ0 of the phase δ set by the phase setting unit 34 is based on the command voltages vdc and vqc of the rotating two-phase coordinate system of the current feedback control unit 20 immediately before switching (current) “δ0 = arctan (− vdc / vqc) ". On the other hand, the initial value of the norm Vn2 calculated by the PI control unit 36b is the square root of the sum of the squares of the command voltages vdc and vqc. Specifically, this initial value is the initial value of the integral element of the PI control unit 36b. Thereby, in the norm setting unit 36 shown in FIG. 2, the given initial value is set to the norm Vn to be initially set, and then the norm Vn to be set is gradually changed to the norm Vn1 calculated by the norm calculation unit 36a. Change it. Therefore, even when there is a deviation between the norm of the command voltages vdc and vqc set by the current feedback control unit 20 and the norm calculated by the norm calculation unit 36a at the time of switching, Sudden changes can be avoided.

  On the other hand, when it is time to switch to control by the current feedback control unit 20 (step S24: YES), initial values of the command voltages vdc and vqc are given in step S26. Here, the initial values of the command voltages vdc and vqc of the current feedback control unit 20 are set to be the output voltage vector of the inverter IV by the torque feedback control unit 30. In other words, the initial values of the command voltages vdc and vqc are set to vectors (−Vn · sin δ, Vn · cos δ). Specifically, the initial values of the integral terms of the feedback control units 25 and 26 are assumed to be vectors (−Vn · sin δ, Vn · cos δ).

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) When control by the torque feedback control unit 30 is being performed, if the output voltage of the inverter IV required for realizing the actual torque (estimated torque Te) by the maximum torque control is equal to or less than a predetermined value, current feedback Switched to control by control. Thereby, the controllability of the motor generator 10 can be maintained at the time of switching.

  (2) A torque estimator 42 that estimates the actual torque by using the current (actual current id, iq) flowing through the motor generator 10 as an input is provided. Thereby, since a torque sensor becomes unnecessary, the increase in the hardware means required for control can be suppressed.

  (3) The estimated norm Vne is calculated by providing a map 56 that receives the estimated torque Te and outputs the speed normalized norm Vne / ω. Thereby, the calculation load of the control apparatus 14 can be reduced.

  (4) When switching to the torque feedback control unit 30, the initial value of the norm Vn2 is set based on the command voltages vdc and vqc calculated by the current feedback control unit 20. Thereby, the continuity of the norm can be maintained before and after switching.

  (5) By including the PI control unit 36b and the selector 36c, the norm Vn set by the norm setting unit 36 is calculated by the norm calculation unit 36a from the initial values based on the command voltages vdc and vqc of the current feedback control unit 20. It was gradually changed to. Thus, a sudden change in norm caused by the switching can be suitably avoided.

  (6) When switching from the control by the torque feedback control unit 30 to the control by the current feedback control unit 20, the initial values of the command voltages vdc and vqc by the current feedback control unit 20 are set to the norm Vn and the phase set by the norm setting unit 36. It was set based on the phase δ set by the setting unit 34. Thereby, the continuity of the output voltage of the inverter IV can be maintained even at the time of switching.

  (7) Both the torque feedback control unit 30 and the current feedback control unit 20 are designed according to the same request (request to perform maximum torque control). Thereby, the torque fluctuation | variation at the time of switching between both control can be suppressed suitably. Furthermore, a common reference (estimated norm Vne output from the voltage estimator 48) can be used as a reference for determining the validity of both controls.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 9 shows the procedure of the switching process according to this embodiment. This process is repeatedly executed at a predetermined cycle, for example.

  In this series of processing, in step S30, the estimated norm Vne is calculated by the same processing as in step S10. In a succeeding step S32, it is determined whether or not the torque feedback control mode is set. If it is determined that the torque feedback control mode is selected, it is determined in step S34 whether or not the estimated norm Vne is greater than a threshold value Vth. This process is to determine whether or not to switch to current feedback control. If it is determined in step S34 that it is equal to or lower than the threshold value Vth, the current feedback control mode is switched in step S36.

  On the other hand, if a negative determination is made in step S32, it is determined in step S38 whether or not the estimated norm Vne is smaller than a value obtained by multiplying the threshold value Vth by the coefficient α. This process is for determining whether or not to switch to torque feedback control. Here, the coefficient α is a positive number smaller than “1”. This is for providing hysteresis in the switching condition to the current feedback control and the switching condition to the torque feedback control. If a negative determination is made in step S38, the torque feedback control mode is switched in step S40.

  In addition, when the process of step S36, S40 is completed, or when affirmation determination is made in step S34, S38, this series of processes is once complete | finished.

  According to the present embodiment described above, the following effects can be obtained in addition to the above-described effects of the first embodiment.

  (8) By providing hysteresis in the switching condition to the current feedback control and the switching condition to the torque feedback control, frequent switching can be avoided.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 10 shows the procedure of the switching process according to this embodiment. This process is repeatedly executed at a predetermined cycle, for example. In FIG. 10, processes corresponding to the processes shown in FIG. 7 are given the same step numbers for convenience.

  As shown in the figure, in this embodiment, even when the d-axis current flowing through the motor generator 10 (the output of the low-pass filter 24) is positive (step S50: NO), switching to the current feedback control is performed. This is because loss is known to increase when the d-axis current becomes positive.

  In the case where the control by the torque feedback control unit 30 is performed and the d-axis current is positive, it is considered that the controllability of the current feedback control unit 20 is not in a low region. This is because, in the region where the control by the torque feedback control unit 30 is performed, if it is difficult to apply a voltage required to realize the required torque Td, the applied voltage is obtained by performing field-weakening control with the d-axis current being negative. This is because it is considered that the d-axis current is never positive. Even when the required torque Td is suddenly reduced, the phase δ is calculated using an integrator, so that the d-axis current temporarily becomes positive in synchronization with the sudden decrease in the required torque Td. It seems that there is nothing.

  In the present embodiment described above, the following effects can be obtained in addition to the effects of the first embodiment.

  (9) When the d-axis component of the current flowing through the motor generator 10 is positive, the control is switched to the control by the current feedback control unit 20. Thereby, it can avoid suitably that the control which becomes large in loss is continued.

(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the third embodiment.

  FIG. 11 shows the procedure of the switching process according to this embodiment. This process is repeatedly executed at a predetermined cycle, for example. In FIG. 11, processes corresponding to the processes shown in FIG. 7 are given the same step numbers for convenience.

  As illustrated, in the present embodiment, the current feedback control is also switched when the absolute value of the voltage phase δ is less than the threshold value δth (step S52: NO). This is a process for performing current feedback control when the d-axis current is positive. Here, the reason why the d-axis current is not directly used is that it is difficult to simultaneously determine whether the switching condition is satisfied with high accuracy and to make the determination quickly. That is, since noise is mixed in the current flowing through the motor generator 10, when using the value on the upstream side of the low-pass filter 24, the accuracy of determining whether or not the d-axis current is positive decreases. On the other hand, when the value on the downstream side of the low-pass filter 24 is used, there is a possibility that it is delayed to determine that when the condition is satisfied due to the delay by the filter.

  Therefore, in the present embodiment, it is determined whether or not the d-axis current is equal to or less than zero using the phase δ. This is possible for the following reasons. That is, in the above formula (c4), when the voltage vector (vd, vq) is expressed as (−Vsinδth, Vcosδth) using the threshold δth and the d-axis current is zero, “V / ω = Φ / cosδth” Get. By substituting this into the above equation (c12), “tan δth = TLq / PΦΦ” is obtained. Accordingly, it can be seen that the d-axis current becomes zero or less when the absolute value of the phase δ becomes equal to or larger than the absolute value of the threshold δth. Here, as shown in the lower part of FIG. 11, the region where the threshold value δth is equal to or smaller than the absolute value is a region where the torque T is positive when the torque T is positive and the torque T is negative. Is a region that is equal to or less than the threshold δth.

  In the present embodiment described above, in addition to the effects of the first embodiment and the effect (9) of the third embodiment, the following effects can be obtained.

  (10) Based on the phase of the output voltage of the inverter IV, it is determined whether or not the d-axis component is positive. As a result, the polarity of the d-axis current can be determined quickly and accurately.

(Fifth embodiment)
Hereinafter, the fifth embodiment will be described with reference to the drawings with a focus on differences from the third embodiment.

  FIG. 12 shows the procedure of the switching process according to this embodiment. This process is repeatedly executed at a predetermined cycle, for example.

  In this series of processes, in step S60, the same process as in step S10 is performed, and then in step S62, it is determined whether or not the torque feedback control mode is set. If the determination in step S62 is affirmative, it is determined in step S64 whether or not the estimated norm Vne is larger than the threshold value Vth. If the determination is negative, in step S66, the mode is switched to the current feedback control mode. On the other hand, if it is determined that the torque feedback control mode is not selected, it is determined in step S68 whether or not the norm of the command voltage vector (vdc, vqc) by the current feedback control unit 20 is smaller than the threshold value Vth. This process determines whether or not to switch to control by the torque feedback control unit 30. If a negative determination is made in step S68, the torque feedback control mode is switched in step S70.

  As described above, in this embodiment, when switching to the control by the current feedback control unit 20, the estimated norm Vne output from the voltage estimator 48 is used, but when switching to the control by the torque feedback control unit 30. The command value of the current feedback control unit 20 is used. This is a setting in view of the concern that a situation may occur in which the estimated norm Vne does not reach the threshold value Vth even though the norm of the command voltage vector (vdc, vqc) has reached the threshold value Vth. Such a situation may occur when there is an estimation error of the voltage estimator 48 or when the required torque Td increases near the voltage saturation region of the motor generator 10. In the latter case, the increase in the estimated torque Te is delayed due to the low control responsiveness of the current feedback control unit 20, and therefore the increase in the estimated norm Vne is delayed.

  In the present embodiment described above, the following effects can be obtained in addition to the effects of the first embodiment.

  (11) When the control by the current feedback control unit 20 is being performed, the control is switched to the control by the torque feedback control unit 30 on condition that the norm of the command voltage vector is equal to or greater than a predetermined value. Thereby, the situation where the controllability of current feedback control falls can be judged suitably.

(12) The threshold value to be compared with the norm of the command voltage vector is a value when the fluctuation range of the two-phase modulated voltage is equal to or greater than the peak value of the carrier. Thereby, the current feedback control can be performed only in a region where the controllability to the required torque by the current feedback control can be maintained high.

(Sixth embodiment)
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  When switching to torque feedback control based on the norm of the command voltage vector as in the fifth embodiment, the norm sets the threshold value Vth due to the overshoot of the norm of the command voltage vector resulting from the rapid increase in the required torque Td. There is a risk of exceeding. FIG. 13 shows an example of this.

  FIG. 13A shows the change of torque, and FIG. 13B shows the change of norm of the command voltage vector. As shown in the figure, when the required torque Td increases rapidly, the command current setting unit 22 increases the command currents idc and iqc (particularly, the q-axis command current iqc). As a result, the difference between the command currents idc and iqc and the actual currents id and iq is increased. For this reason, the feedback control units 25 and 26 increase the command voltages vdc and vqc. However, until the difference between the command currents idc and iqc and the actual currents id and iq once becomes zero, the integrators of the feedback control units 25 and 26 continue to change to increase the command voltages vdc and vqc. . For this reason, even after the estimated torque Te reaches the required torque Td, the command voltages vdc and vqc are operated to increase, resulting in overshoot. In this case, when the estimated torque Te follows the required torque Td due to overshooting, the norm of the command voltage vector converges to a value smaller than the peak value. In the example shown in FIG. 13, the convergence value of the norm of the command voltage vector itself is less than the threshold value Vth, but an example is shown in which the threshold value Vth is temporarily exceeded due to overshoot. In this case, when switching to torque feedback control due to overshoot, switching to current feedback control is performed again, and a switching hunting phenomenon occurs.

  In order to avoid such a situation, in the present embodiment, the processing shown in FIG. 14 is performed. FIG. 14 shows the procedure of the switching process according to the present embodiment. This process is repeatedly executed at a predetermined cycle, for example. In FIG. 14, processes corresponding to the processes shown in FIG. 12 are given the same step numbers for convenience.

  In this series of processes, when current feedback control is performed (step S62: NO), the process proceeds to step S68a. In step S68a, (a) the condition that the norm of the command voltage vector is smaller than the threshold value Vth, and (b) the vector norm Vn0 required for realizing the required torque Td by the maximum torque control is a coefficient for the threshold value Vth. It is determined whether or not a logical sum condition with a condition that the value is smaller than a value multiplied by γ is satisfied. Here, the condition (A) is for determining whether or not the required torque Td can be realized by the current feedback control while maintaining the controllability of the torque. Here, the coefficient γ is a positive value smaller than “1”. Thereby, the value compared with norm Vn0 becomes smaller than the threshold value Vth regarding the norm of the command voltage vector. This takes into account the calculation error of the norm Vn0.

  If a negative determination is made in step S68a, the process proceeds to torque feedback control in step S70.

  In the present embodiment described above, the following effects can be obtained in addition to the effects of the fifth embodiment.

  (13) The torque feedback control is switched on condition that the output voltage of the inverter IV required for realizing the required torque Td by the current feedback control is equal to or higher than a threshold value. Thereby, it is possible to avoid unnecessary switching to torque feedback control due to overshooting of the command voltages vdc and vqc.

(Seventh embodiment)
Hereinafter, the seventh embodiment will be described with reference to the drawings with a focus on differences from the sixth embodiment.

  FIG. 15 shows the procedure of the switching process according to the present embodiment. This process is repeatedly executed at a predetermined cycle, for example. In FIG. 15, processes corresponding to the processes shown in FIG. 14 are given the same step numbers for convenience.

  As illustrated, in the present embodiment, the absolute value of the phase δ is less than the absolute value of the threshold δth in addition to the switching process of the sixth embodiment, in view of the reason described in the fourth embodiment. If there is (step S52: NO), a process of switching to the current feedback control mode is added.

(Eighth embodiment)
Hereinafter, the eighth embodiment will be described with reference to the drawings, centering on differences from the previous sixth embodiment.

  FIG. 16 shows the procedure of the switching process according to the present embodiment. This process is repeatedly executed at a predetermined cycle, for example. In FIG. 16, processes corresponding to the processes shown in FIG. 14 are given the same step numbers for convenience.

  As illustrated, in the present embodiment, in addition to the switching process of the sixth embodiment, in addition to the reason described in the third embodiment, when the detected value of the d-axis current is positive (step S50: NO), a process of switching to the current feedback control mode is added.

(Ninth embodiment)
Hereinafter, the ninth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  When the command voltage vuc, vvc, vwc is set by the current feedback control unit 20 and the operation signals gup, gun, gvp, gvn, gwp, gwn are generated based on the command voltages vuc, vvc, vwc, the upper arm and the lower arm are short-circuited. A dead time is generated to avoid it. For this reason, the operation signal is a signal having an error corresponding to the dead time with respect to the pulse signal directly determined by the magnitude relationship between the signal in which the command voltages vuc, vvc, and vwc are two-phase modulated and the carrier. Here, when the polarity of the current flowing through the motor generator 10 is positive, the current flows through the diodes Dun, Dvn, Dwn of the lower arm during the dead time period, so that the positive applied voltage period of the inverter IV is insufficient. On the other hand, when the polarity of the current flowing through motor generator 10 is negative, the current flows through upper arm diodes Dup, Dvp, Dwp during the dead time period, so that the negative voltage application period of inverter IV is insufficient. For this reason, the output voltage of the inverter IV is insufficient with respect to the command voltages vuc, vvc, vwc. Specifically, the voltage error ΔV is “ΔV = VDC · td · fc” using the power supply voltage VDC, the dead time period td, and the carrier frequency fc.

  Here, when the current feedback control is performed, the voltage error ΔV due to the dead time is compensated by the feedback control. However, when switching to torque feedback control based on the comparison between the norm of the command voltage vector and the threshold value Vth as in the fifth embodiment, the switching is performed because the actual norm is “Vth−ΔV”. Will be made.

  Therefore, in the present embodiment, a dead time compensator 60 is provided as shown in FIG. 17 in order to remove the error caused by the voltage error ΔV from the determination of whether or not the command voltage vector is equal to or higher than the threshold value Vth. Here, the dead time compensator 60 calculates a correction amount for setting the output voltage of the inverter IV to the command voltages vdc and vqc output from the feedback control units 25 and 26. This can be realized by calculating the phase δ from the command currents idc and iqc and converting the voltage error ΔV into a correction value for the voltage on the dq axis at this phase δ.

  Thus, the command voltages vdc and vqc are feedforward corrected by the output of the dead time compensator 60, so that the output voltage of the inverter IV becomes the command voltages vdc and vqc output from the feedback control units 25 and 26. Moreover, the switching control unit 46 compares the norm of the vectors of the command voltages vdc and vqc output from the feedback control units 25 and 26 with the threshold value Vth, so that the norm of the actual output voltage vector of the inverter IV and the threshold value Vth are compared. Can be compared.

  In the present embodiment described above, the following effects can be obtained in addition to the effects of the fifth embodiment.

  (14) A dead time compensator 60 that corrects the command voltages vdc and vqc output from the feedback control units 25 and 26, and a norm and a threshold value Vth of the command voltages vdc and vqc before correction by the dead time compensator 60. Based on these comparisons, switching to torque feedback control was performed. Thereby, it can be determined with high accuracy whether or not the norm of the output voltage vector of the inverter IV is equal to or greater than the threshold value Vth.

(Tenth embodiment)
Hereinafter, the tenth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  FIG. 18 shows the processing of the part of the norm calculation unit 36a according to the present embodiment excluding the voltage limiter 54. As shown in the figure, in this embodiment, a norm V2 and a phase δ2 of a voltage vector obtained by deleting a portion canceled by the induced voltage of the motor generator 10 from the output voltage of the inverter IV are defined, and the torque T is defined as the norm. It is expressed by V2 and phase δ2. Then, the norm Vn0 is set so that the partial differential coefficient of the torque T due to the phase δ2 becomes zero.

  Here, the torque T can be expressed by the following equation (c13) by the norm V2 and the phase δ2.

上記の式（ｃ１３）において、位相δ２によるトルクＴの偏微分係数がゼロとなるとの条件の下、ノルムＶ２及び位相δ２を、位相δ及びノルムＶｎ０にて消去することで、ノルムＶｎ０を位相δ及び電気角速度ωにて表現することができる。特に、抵抗Ｒをゼロとする近似では、下記の式（ｃ１４）が成立する。

In the above equation (c13), the norm Vn and the phase δ2 are eliminated by the phase δ and the norm Vn0 under the condition that the partial differential coefficient of the torque T by the phase δ2 becomes zero, so that the norm Vn0 is changed to the phase δ. And the electrical angular velocity ω. In particular, in the approximation in which the resistance R is zero, the following equation (c14) is established.

上記の式（ｃ１４）においても、速度規格化ノルム「Ｖｎ０／ω」は、位相δの関数ｆとして表現できる。これにより、先の第１の実施形態の要領で、速度規格化ノルムをトルクＴの関数ｈとして定義することができる。本実施形態でも、関数ｈを数値計算によって算出することで、マップ化しておく。図１８には、マップ化された関数ｈを示している。

Also in the above equation (c14), the speed normalized norm “Vn0 / ω” can be expressed as a function f of the phase δ. Thereby, the speed normalization norm can be defined as a function h of the torque T in the manner of the first embodiment. Also in this embodiment, the function h is mapped by calculating by numerical calculation. FIG. 18 shows the mapped function h.

  According to this embodiment described above, in addition to the effects (1) to (6) of the previous first embodiment and the effects of the fifth embodiment, the following effects are further obtained. Be able to.

  (15) The norm Vn0 is set so that the maximum torque is generated by the current generated according to the portion of the output voltage of the inverter IV excluding the voltage component canceled by the induced voltage of the motor generator 10. Thereby, the norm Vn0 that minimizes the voltage contributing to the torque can be selected. For this reason, it is considered that the current vectors id and iq can be reduced as much as possible in view of the voltage equation shown in the equation (c4).

(Eleventh embodiment)
Hereinafter, the eleventh embodiment will be described with reference to the drawings, centering on differences from the first embodiment.

  FIG. 19 shows a block diagram of processing relating to generation of an operation signal of the inverter IV according to the present embodiment. In FIG. 19, processes corresponding to the processes shown in FIG. 2 are given the same reference numerals for convenience.

  As illustrated, the norm calculation unit 36a according to the present embodiment sets the norm Vn1 with the phase δ set by the phase setting unit 34 and the electrical angular velocity ω as inputs. This can be realized by changing the norm calculation unit 36a so as to perform the process of setting the norm Vn0 based on the above formula (c9).

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the second embodiment, the threshold value for switching to the torque feedback control mode is made smaller than the threshold value for switching to the current feedback control mode, but it may be reversed.

  -You may change the said 6th-8th embodiment by the change of 9th Embodiment with respect to 5th Embodiment.

  The sixth to ninth and eleventh embodiments may be changed according to changes of the tenth embodiment with respect to the fifth embodiment.

  -You may change the said 2nd-10th embodiment by the change of 11th Embodiment with respect to 1st Embodiment.

  The means for removing the error due to the dead time when determining whether or not the condition that the estimated norm Vne is equal to or greater than a predetermined value (≧ Vth) is satisfied is not limited to that illustrated in the ninth embodiment. For example, the threshold value to be compared with the estimated norm Vne may be a value obtained by adding the error ΔV to the threshold value Vth.

  The command current setting unit 22 is not limited to setting a command current for maximum torque control. For example, it is good also as control designed based on the request | requirement similar to the torque feedback control part 30 in previous 10th Embodiment. In this case, the voltage estimator 48 may receive the estimated torque Te instead of the required torque Td in the process shown in FIG. For example, the command current iq may be set under the condition of “id = 0”. The command current setting unit 22 is preferably configured to set the command currents idc and iqc on the condition of “id ≦ 0”.

  The current feedback control unit 20 is not limited to a signal that is subjected to two-phase modulation processing on the three-phase command voltages vuc, vvc, and vwc as a signal to be compared with the carrier. For example, the two-phase modulation process may not be performed. However, in this case, the threshold value Vth is preferably set to about “0.68 VDC”. In other words, it may be about the value when the average frequency of the operation signal is equal to or lower than the carrier frequency fc.

  The configurations of the switching control unit 46 and the voltage estimator 48 are not limited to those illustrated in the above embodiments. Here, in each of the above embodiments, the estimated torque Te is calculated based on the actual currents id and iq, and the estimated norm Vne is calculated using this and the electrical angular velocity ω as input. This means that a three-dimensional map having the actual currents id, iq and the electrical angular velocity ω as inputs and the estimated norm Vne as an output can be configured. Therefore, the voltage estimator 48 may be configured to include this three-dimensional map.

  The configuration of the voltage estimator 48 is not limited to means for calculating the estimated norm Vne of the voltage vector of the two-phase coordinate system (two-dimensional coordinate system). For example, the peak value of the phase voltage may be estimated as a parameter related to the voltage required for realizing the estimated torque Te by the current feedback control unit 20.

  The norm calculation unit 36a is not limited to the one that performs the control exemplified in the above embodiments. For example, the norm Vn1 for performing the control with “id = 0” may be calculated.

  As the changing means for gradually changing the norm set by the norm setting unit 36 from the initial value based on the command voltages vdc and vqc of the current feedback control unit 20 to the norm calculated by the norm calculation unit 36a, each of the above embodiments It is not restricted to what was illustrated in. For example, a value based on the command voltages vdc and vqc of the current feedback control unit 20 may be used as an initial value, and a value obtained by performing a weighted average process on this value and the output of the norm calculation unit 36a may be used as the output of the norm setting unit 36.

  The operation signal generation unit 38 is not limited to storing data defining an operation signal waveform whose voltage utilization rate is equal to or higher than the threshold value Vth. For example, a pattern having a voltage utilization factor lower than the threshold value Vth may be stored.

  The operation signal generation unit 38 is not limited to one that stores in advance data that defines an operation signal waveform for each voltage usage rate, and uses the corresponding data according to the voltage usage rate each time. For example, the operation signal may be calculated each time by PWM processing or the like based on the norm of the voltage vector.

  If the operation signal generation unit 38 can assume that the input voltage of the inverter IV is constant, only the norm output from the norm setting unit 36 may be used as a parameter for searching for the operation signal waveform, instead of the voltage utilization rate.

  In each of the above embodiments, the operation signal waveform generated by the operation signal generation unit 38 is a signal having symmetry with respect to the center of one period of the electrical angle, but is not limited thereto. If the operation signal waveform is set based on the phase δ and the norm Vn, it is possible to freely design an operation signal waveform that is difficult depending on the setting by the processing of the current feedback control unit 20 or the like.

  As a method for setting the phase δ as an operation amount of feedback control of the actual torque (estimated torque Te) to the required torque Td by the phase setting unit 34, the phase δ is set by proportional integral control of these differences. Not exclusively. For example, the phase δ may be set by integral control or proportional integral derivative control of the difference between the required torque Td and the estimated torque Te.

  In each of the above embodiments, in the current feedback control unit 20, the outputs of the feedback control units 25 and 26 are directly commanded voltages vdc and vqc. However, the present invention is not limited to this. For example, the command voltages vdc and vqc may be calculated by adding a non-interference term as a feedforward term to the outputs of the feedback control units 25 and 26.

  The feedback control units 25 and 26 are not limited to those that perform proportional integral control, but may be those that perform integral control, proportional integral differential control, or the like, for example.

  The high voltage control means for controlling the required torque in the region where the voltage utilization rate is higher than that of the current feedback control unit 20 is not limited to those exemplified in the above embodiments. For example, what was described in the said patent document 1 may be used.

  -The salient pole machine is not limited to IPMSM. For example, a synchronous reluctance motor (SynRM) may be used.

  -The rotating machine is not limited to a salient pole machine, and may be a non-salient pole machine. At this time, for a model expressing torque by the phase δ and the norm Vn0, if the norm Vn0 that satisfies the condition that the partial differential coefficient of the torque T by the norm Vn0 is zero and the phase δ have a one-to-one relationship. The operation signal may be generated based on the norm Vn0 that satisfies this condition.

  -A rotary machine is not restricted to what is mounted in a hybrid vehicle, For example, you may mount in an electric vehicle. Furthermore, the rotating machine is not limited to one constituting a vehicle drive system.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 12 ... High voltage battery, 14 ... Control apparatus (one Embodiment of the control apparatus of a power converter circuit), 20 ... Current feedback control part, 30 ... Torque feedback control part, 34 ... Phase setting part, 36 ... Norm Setting unit, 38 ... operation signal generation unit, 48 ... voltage estimator, 46 ... switching control unit, IV ... inverter, CV ... converter.

Claims (10)

In a control device for a rotating machine that controls an actual torque of the rotating machine to a required torque by operating a power conversion circuit including a switching element that electrically connects each of a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine ,
First control means for operating the power conversion circuit by PWM processing to control the voltage applied to the rotating machine to a command voltage;
Second control means for operating the power conversion circuit based on an operation signal waveform of the power conversion circuit according to an electrical angle of the rotating machine;
Switching means for switching to one of the other controls by using information related to the control as an input during control by either one of the first control means and the second control means;
The operation signal waveform includes a control device for a rotating machine that includes a voltage utilization factor smaller than that of rectangular wave control.   The control device for a rotating machine according to claim 1, wherein the switching means performs the switching by inputting information on a rotation speed as information on the control. When operating the switching element of the power conversion circuit, a dead time is set to avoid a short-circuit state of the positive and negative electrodes of the DC power supply,
The switching means performs the switching by inputting information on the output voltage vector of the power conversion circuit as information on the control, and removes an error caused by the dead time from the output voltage vector as the input The rotating machine control device according to claim 1, further comprising:   4. The switching unit according to claim 1, wherein the switching unit performs the switching by using, as input, information related to an output voltage vector of the power conversion circuit required by the control by either one of the information regarding the control. The control device for a rotating machine according to Item 1.   5. The control of a rotating machine according to claim 4, wherein the switching unit performs the switching based on a comparison between a norm of an output voltage vector of the power conversion circuit required by the control by either one and a threshold value. apparatus. The first control means operates the switching element based on a magnitude relationship between a signal corresponding to the command voltage and a carrier wave, which is a pair of signals standardized based on the voltage of the DC power supply,
The switching means switches from the first control means to the second control means based on a fluctuation width of a signal corresponding to the command voltage to be compared with the carrier wave being equal to or greater than a peak value of the carrier wave. The control device for a rotating machine according to any one of claims 3 to 5, wherein:   The control device for a rotating machine according to any one of claims 1 to 6, wherein the switching means performs the switching by inputting information on a current flowing through the rotating machine as information on the control.   8. The rotating machine control device according to claim 7, wherein the switching unit performs the switching by inputting information on a d-axis current flowing through the rotating machine as information on the control.   9. The control device for a rotating machine according to claim 8, wherein the switching unit performs the switching by inputting information on a polarity of a d-axis current flowing through the rotating machine as information on the control.   When switching from either one of the controls to the other control, the initial value of the other command voltage is set based on the command voltage in either one, and the other one is independent of the initial value. The control device for a rotating machine according to claim 1, further comprising changing means for gradually changing to a command voltage calculated by the formula (1).

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