JP2011244660A - Switching power supply circuit and recording device - Google Patents

Abstract

To provide a switching power supply capable of almost uniformly controlling an overpower and overcurrent protection operating point with a simple configuration and not depending on an input voltage from a commercial power supply, and a recording apparatus using the same. is there.
There is provided a transformer that inputs and rectifies an AC commercial power supply having a different voltage, inputs the AC voltage rectified on the primary side, converts the AC voltage to a DC voltage, and outputs the DC voltage from the secondary side. The switching power supply circuit has the following configuration. A switching element connected in series to the primary side of the transformer and controlling the input of AC voltage to the primary side, and a detection circuit connected in series to the switching element and detecting the voltage on the primary side of the transformer Prepare. Further, the gain changes according to the detected voltage, the detected voltage is amplified according to the gain, and the on / off of the switching element is controlled according to the amplified output voltage.
[Selection] Figure 3

Description

  The present invention relates to a switching power supply circuit and a recording apparatus using the power supply, and more particularly to a switching power supply circuit that includes an overpower protection circuit and can correct a difference in overpower protection operating point due to a difference in input voltage. The present invention relates to a recording apparatus using the.

  A switching power supply circuit for generating a stable DC voltage from a commercial AC power supply is used in many home appliances and office machine products today. Such switching power supply circuits are generally designed using commercially available control ICs, and many of them incorporate an overpower protection function or an overcurrent protection function. However, these features have many technical compromises. For example, there is a drawback that a difference in input voltage, for example, when the input voltage is AC100V and AC200V (generally AC230V in Europe, Australia, etc.) causes a large difference in the protection operation point, which affects safety.

  In general, the protection operating point when the input voltage is 200 VAC tends to be excessive compared to that when the input voltage is 100 VAC. The reasons are typically due to (1) the time delay from the overcurrent detection time by the control IC to the actual protection operation, and (2) difficulty in realizing an accurate power simulation in the continuous conduction mode. . Hereinafter, these reasons will be described in detail.

  First, in the case of a switching power supply circuit generally having a power consumption of about 60 to 80 W, a flyback method is often employed from the viewpoint of cost advantage.

FIG. 8 is a diagram showing a basic configuration of a flyback power supply circuit. In this configuration, power is stored in the primary winding of the transformer when the transistor Tr ₁ is on, and current flows through the secondary winding of the transformer when Tr ₁ is off. Then, the energy stored in the transformer is released to the output side (Vo). Stable DC power is supplied by repeating this cycle.

FIG. 9 is a diagram showing the current flowing through the primary winding of the transformer, and hence the current flowing through the transistor Tr ₁ operating as a switching element, in the flyback power supply circuit shown in FIG.

  In FIG. 9, (a) shows the case where the switching power supply circuit and the recording apparatus using the power supply have a light load, and (b) shows the case of a heavy load. In the flyback method, operation up to a specific power load is performed in a mode called a discontinuous conduction mode as shown in FIG. However, when the electric power is exceeded, it operates in a continuous conduction mode as shown in FIG. 9 (b). In the discontinuous conduction mode shown in FIG. 9A, the current flowing through the switching element gradually increases from the zero point during the on period (Ton). Then, after reaching the peak point, it is turned off, and becomes zero level during the off period. Note that the waveform of this current (I) is expressed by equation (1) and is proportional to the input voltage.

I = V _DC / Lp · t (1)
Here, _VDC is the DC voltage after rectifying and smoothing the input AC voltage, Lp is the primary inductance of the transformer, and t is the elapsed time after the switching element is turned on. V _DC has approximately the following relationship with the input voltage V _in , V _DC = 1.4 V _in . Therefore, Formula (1) is expressed as follows.

I = 1.4V _in / Lp · t (2)
In other words, the gradient of the current I becomes a result that is proportional to the input voltage V _in, the slope of current increase when AC200V is twice the time of AC100V.

  On the other hand, in the continuous conduction mode, as shown in FIG. 9B, the initial current in the ON period starts to increase with a certain offset, not from zero. Next, the discontinuous conduction mode and the continuous conduction mode will be described in relation to the BH curve shown in FIG.

<Description of BH curve>
The BH curve shown in FIG. 10 shows the relationship between H (magnetic field) and B (magnetic flux density) as is well known. As the magnetic field is applied from the initial state where the core is not magnetized (point O), the magnetic flux density B increases and finally reaches the saturation point P. The magnetic flux density Bs at this time is called a saturation magnetic flux density. Thereafter, as the magnetic field decreases, the magnetic flux density also decreases. However, even when the magnetic field becomes zero, the magnetic flux density does not become zero, and the residual magnetic flux density Br ₀ remains. When a magnetic field is further applied in the opposite direction, the magnetic flux density eventually becomes zero. The strength (Hc) of the magnetic field at this time is called holding force. Thereafter, when the magnetic field is further increased, the magnetic flux density is saturated at the Q point. The magnetic flux density at this time is -Bs. When the magnetic field is weakened again, the residual magnetic flux density -Br remains when the magnetic field is zero, and then the magnetic flux density becomes zero when the holding force Hc is applied by applying a positive magnetic field again. When the magnetic field is further increased, the saturation point P is reached again.

  Thus, the BH curve has a hysteresis characteristic, and the area surrounded by the hysteresis is a heat loss due to the core of the transformer.

  By the way, unlike the push-pull method in which a magnetic field is applied in both directions, the flyback method applies only a magnetic field in one direction. Therefore, it can be said that the actual operation of the flyback method uses only the first quadrant of the BH curve shown in FIG.

  Next, the correlation between the discontinuous conduction mode and the continuous conduction mode described above with reference to FIG. 9 and the BH curve shown in FIG. 10 will be described.

The discontinuous conduction mode shown in FIG. 9A operates in the X0 region of the BH curve of FIG. 10, and the magnetic flux density changes between the residual magnetic flux density Br ₀ and the maximum magnetic flux density Bm _0. . In FIG. 10, this change in magnetic flux density is indicated by ΔB ₀ . That is, in the discontinuous conduction mode, the magnetic field H is reset to zero and the excitation is repeated. In other words, in the discontinuous conduction mode, the energy accumulated in the primary winding of the transformer during the turn-on period of the switching element is completely released by the secondary winding during the turn-off period, and then the next cycle is resumed. .

On the other hand, the continuous conduction mode shown in FIG. 9B operates in the X1 region of FIG. 10, and the magnetic flux density changes between the residual magnetic flux density Br ₁ and the maximum magnetic flux density Bm ₁ . In FIG. 10, this change in magnetic flux density is indicated by ΔB ₁ . That is, in the continuous conduction mode, the next excitation is repeated in a state where the magnetic field H is not reset to zero. In other words, in the discontinuous conduction mode, the energy accumulated in the primary winding of the transformer during the turn-on period of the switching element is not completely released by the secondary winding during the turn-off period, and the next cycle resumes. Is done.

  Next, energy generated by the transformer in each case of the discontinuous conduction mode and the continuous conduction mode will be described. The generated energy in the discontinuous conduction mode is expressed by Equation (3).

E = (1/2) · f · Lp · (Ip ² ) (3)
Here, f is the switching frequency, Lp is the inductance of the primary winding of the transformer, and Ip is the peak value of the current flowing through the primary winding, which are shown as Ip1 and Ip2 in FIG.

  On the other hand, the generated energy in the case of the continuous conduction mode is expressed by Expression (4).

E = (1/2) · f · Lp · (Ip ² −Iq ² ) (4)
Here, Iq is shown as Iq1 and Iq2 in FIG. 9 and indicates an offset current at the time of turn-on. If a discontinuous conduction mode is to be realized even under heavy loads, the size of the transformer (ie, core size) must be increased, which is not practical due to cost and space requirements. turn into.

  Next, the reason why the continuous conduction mode represented by the formula (4) can generate large energy corresponding to a heavy load will be briefly described.

  As shown in FIG. 9B, the offset current Iq at turn-on is considerably smaller than the peak current Ip, and Iq is designed to be about half of Ip even at a point where the overcurrent protection function is normally applied. . Now, since the equation (4) uses the square difference between Ip and Iq as a factor, the effect of decrement by Iq is actually only about 25%. As a result, even if the decrement by Iq is subtracted, Ip If the value is increased, a large energy can be generated. This is why the continuous conduction mode can generate a large amount of energy using a moderately sized transformer.

  By the way, since the overcurrent protection operation is designed to occur at a heavy load, that is, in the continuous conduction mode, the expression (4) is realized in the control IC in order to accurately realize the overcurrent protection operation. It is necessary to provide a circuit. However, it is actually very difficult to realize the equation (4). In practice, overcurrent detection is generally performed while ignoring the offset component Iq. That is, only the peak current Ip is detected to detect whether or not it is an overcurrent. In this case, as described above, a factor of about 25% due to the offset component Iq is ignored, so that a large error occurs from a desired result.

  In order to reduce such an error, a device has been proposed so far in which an external circuit is provided separately from the control IC and correction corresponding to the offset current Iq is performed. However, such contrivances also include, for example, (1) a decrease in dynamic range that can lead to erroneous detection of an overcurrent detection voltage, or (2) an increase in power consumption due to correction (ie, a deterioration in energy saving level). It can cause secondary harm.

  Another cause that makes accurate overcurrent detection difficult is a time delay from the overcurrent detection time by the control IC to the actual protection operation. This point will be described with reference to FIG.

In FIG. 11, the upper current change is the current flowing in the transformer primary winding when the input is AC 100V, and the lower current change is the current flowing in the switching element (Tr _{1 in} FIG. 8) when the input is AC 200V. Show. As described above, the slope of the current is 200 times AC 200 times that of AC 100 V. Referring to the case of AC 100 V, the control IC detects an overcurrent when it reaches Ip _1A in FIG. 11, and tries to turn off the switching element. However, in practice, there is a time delay Δt from the moment of detection to the time when the drive signal of the switching element is actually turned off. Therefore, the current when the switching element is actually turned off is Ip _1B . That is, the current at turn-off becomes larger than the desired peak current value by the difference ΔI ₁ between Ip _1B and Ip _1A , and the overcurrent increases accordingly.

  On the other hand, such a delay occurs similarly in the case of AC 200 V, and the delay time Δt is the same as AC 100 V. For this reason, since the current gradient is as large as twice that of AC 100 V, the peak current increment ΔI 2 is also twice as large as ΔI 1. That is, it becomes as shown in Formula (5).

ΔI ₂ = 2ΔI ₁ (5)
As described in relation to the equation (4), the peak current Ip contributes to the generated energy by its square, and therefore, the overcurrent protection operating point at 200 VAC is considerably larger than that at 100 VAC. It becomes.

  As described above, in a switching power supply using a commercially available control IC, it is caused by the time delay from the overcurrent detection time by the control IC to the actual protection operation or the difficulty of realizing an accurate power simulation in the continuous conduction mode. Thus, the overcurrent protection operating point is affected by the input voltage. In order to solve such drawbacks, techniques disclosed in, for example, Patent Document 1, Patent Document 2, and Patent Document 3 have been proposed so far.

JP-A-8-340672 JP 2002-191172 A JP-A-8-149821 Japanese Patent Laid-Open No. 11-178332

  However, the prior art as described above still has the following drawbacks.

  In Patent Document 1, the detection resistance value is variably controlled by linearly conducting conduction control of a power MOS transistor connected in parallel to a detection resistor that detects a current flowing through a switching element. However, in the technique disclosed in Patent Document 1, the power MOS transistor needs to have the same specifications as the main switching element, and the circuit for linearly controlling the power MOS transistor is considerably complicated. There are drawbacks.

  Patent Document 2 has a drawback that the circuit becomes complicated, such as generation of a negative power source by using an auxiliary winding of a transformer.

  Patent Document 3 proposes an overcurrent protection circuit for an active filter. However, there is basically no technical relationship with the flyback method of the present invention, and what is the problem of the flyback method that is the subject of the present invention? No solution has been proposed.

  Patent Document 4 realizes a desired overcurrent detection by increasing the switching frequency as the input voltage increases. Similar to Patent Document 3, the problem with the flyback method that is the subject of the present invention is disclosed. No solution is proposed.

  The present invention has been made in view of the above-described conventional example, and is a switching power supply circuit that can control overpower and overcurrent protection operating points almost uniformly with a simple configuration and without depending on an input voltage from a commercial power supply. And a recording apparatus using the same.

  In order to achieve the above object, the recording head of the present invention has the following configuration.

  That is, an AC commercial power supply having a different voltage is input to be rectified, and the rectified AC voltage is input to the primary side to be converted into a DC voltage, and the DC voltage is output from the secondary side. A switching power supply circuit, which is connected in series to the primary side of the transformer and controls the input of the AC voltage to the primary side, and is connected to the switching element and connected to the primary side of the transformer A detection circuit that detects a voltage; a gain that varies according to the voltage detected by the detection circuit; an amplifier that amplifies the voltage detected by the detection circuit according to the gain; and a voltage output from the amplifier according to the voltage. And a control circuit for controlling on / off of the switching element.

  In another aspect of the present invention, a recording apparatus using the switching power supply circuit having the above configuration is provided.

  Therefore, according to the present invention, it is possible to control the on / off state of the switching element that controls the input of the AC voltage in accordance with the output voltage from the amplifier whose gain changes substantially in accordance with the input voltage of the commercial power supply. Thereby, even if the input voltage changes, the on / off operation point of the switching element, that is, the overpower or overcurrent protection operation point can be maintained substantially constant, and the safety of the device can be improved.

1 is a schematic perspective view illustrating a configuration of an ink jet recording apparatus that is a typical embodiment of the present invention. FIG. 2 is a block diagram illustrating a control configuration of the recording apparatus illustrated in FIG. 1. It is a figure which shows the power circuit of the flyback system according to Example 1 of this invention. It is a figure which shows the comparison operation performed with control IC. It is a figure explaining overcurrent protection operation. It is a figure explaining the difference in the overcurrent detection point according to the difference in input commercial voltage. It is a figure which shows the flyback system power supply circuit according to Example 2 of this invention. It is a figure which shows the basic composition of the power supply circuit of the conventional flyback system. FIG. 10 is a diagram showing a current flowing in the primary winding of the transformer at light load and heavy load in the power supply circuit shown in FIG. 9. It is a figure which shows the relationship between H (magnetic field) and B (magnetic flux density) of a transformer. It is a figure which shows the electric current which flows into a transformer primary winding.

  Hereinafter, preferred embodiments of the present invention will be described more specifically and in detail with reference to the accompanying drawings. However, the relative arrangement and the like of the constituent elements described in this embodiment are not intended to limit the scope of the present invention only to those unless otherwise specified.

  In this specification, “recording” (sometimes referred to as “printing”) is not limited to the case of forming significant information such as characters and graphics, but may be significant. Furthermore, it also represents a case where an image, a pattern, a pattern, or the like is widely formed on a recording medium or a medium is processed regardless of whether or not it is manifested so that a human can perceive it visually.

  “Recording medium” refers not only to paper used in general recording apparatuses but also widely to cloth, plastic film, metal plate, glass, ceramics, wood, leather, and the like that can accept ink. Shall.

  Further, “ink” (sometimes referred to as “liquid”) should be interpreted widely as in the definition of “recording (printing)”. Therefore, by being applied on the recording medium, it is used for formation of images, patterns, patterns, etc., processing of the recording medium, or ink processing (for example, solidification or insolubilization of the colorant in the ink applied to the recording medium). It shall represent a liquid that can be made.

  Furthermore, unless otherwise specified, the “recording element” collectively refers to an ejection port or a liquid path communicating with the ejection port and an element that generates energy used for ink ejection.

<Description of Inkjet Recording Apparatus (FIG. 1)>
FIG. 1 is an external perspective view showing an outline of the configuration of an ink jet recording apparatus 1 which is a typical embodiment of the present invention.

  As shown in FIG. 1, an ink jet recording apparatus (hereinafter referred to as a recording apparatus) has an ink jet recording head (hereinafter referred to as a recording head) 3 that performs recording by discharging ink in accordance with an ink jet system. Recording is performed by reciprocating in the direction. A recording medium P such as recording paper is fed through the paper feeding mechanism 5 and conveyed to a recording position, and recording is performed by discharging ink from the recording head 3 to the recording medium P at the recording position.

  In addition to mounting the recording head 3 on the carriage 2 of the recording apparatus 1, an ink cartridge 6 for storing ink to be supplied to the recording head 3 is mounted. The ink cartridge 6 is detachable from the carriage 2.

  The recording apparatus 1 shown in FIG. 1 is capable of color recording. For this reason, the carriage 2 contains four inks containing magenta (M), cyan (C), yellow (Y), and black (K) inks, respectively. An ink cartridge is installed. These four ink cartridges are detachable independently.

  The recording head 3 of this embodiment employs an ink jet system that ejects ink using thermal energy. For this reason, an electrothermal converter is provided. The electrothermal transducer is provided corresponding to each of the ejection ports, and ink is ejected from the corresponding ejection port by applying a pulse voltage to the corresponding electrothermal transducer in accordance with the recording signal.

  In the example shown in FIG. 1, the recording head 3 and the ink cartridge 6 are separated, but a head cartridge in which the recording head and the ink cartridge are integrated may be used.

<Control Configuration of Inkjet Recording Apparatus (FIG. 2)>
FIG. 2 is a block diagram showing a control configuration of the recording apparatus shown in FIG.

  As shown in FIG. 2, the controller 600 includes an MPU 601, a ROM 602, a special purpose integrated circuit (ASIC) 603, a RAM 604, a system bus 605, an A / D converter 606, and the like. Here, the ROM 602 stores a program corresponding to a control sequence to be described later, a required table, and other fixed data. The ASIC 603 generates control signals for controlling the carriage motor M1, the transport motor M2, and the recording head 3. The RAM 604 is used as a development area for image data, a work area for program execution, and the like. A system bus 605 connects the MPU 601, the ASIC 603, and the RAM 604 to each other to exchange data. The A / D converter 606 inputs analog signals from the sensor group described below, performs A / D conversion, and supplies a digital signal to the MPU 601.

  In FIG. 2, reference numeral 610 denotes a computer (or a reader for image reading, a digital camera, etc.) serving as a supply source of image data, and is collectively referred to as a host device. Image data, commands, status signals, and the like are transmitted and received between the host apparatus 610 and the recording apparatus 1 via an interface (I / F) 611. This image data is input in a raster format, for example.

  Reference numeral 620 denotes a switch group, which includes a power switch 621, a print switch 622, a recovery switch 623, and the like.

  Reference numeral 630 denotes a sensor group for detecting the apparatus state, and includes a position sensor 631, a temperature sensor 632, and the like.

  Further, 640 is a carriage motor driver that drives a carriage motor M1 for reciprocating scanning of the carriage 2 in the direction of arrow A, and 642 is a conveyance motor driver that drives a conveyance motor M2 for conveying the recording medium P. Reference numeral 644 denotes a head driver that drives the recording head based on recording data and control signals transferred from the controller 600.

  The ASIC 603 transfers data for driving a recording element (ejection heater) to the recording head while directly accessing the storage area of the RAM 604 during recording scanning by the recording head 3.

  The switching power supply circuit 650 is a switching power supply circuit that inputs an AC voltage having an AC commercial power supply frequency of 50 Hz to 60 Hz and a voltage of 100 V to 240 V and supplies a direct current (DC) voltage to each unit of the recording apparatus.

  Next, an embodiment of a power supply circuit used by the recording apparatus having the above configuration will be described.

  FIG. 3 is a diagram showing a configuration of a flyback power supply circuit according to the first embodiment of the present invention. This power supply circuit corresponds to a universal power supply, and the input voltage range is AC100 to 240V. The input voltage is rectified by the bridge diode BD1, and then smoothed by the electrolytic capacitor C1 to be a direct current (DC). The smoothed DC voltage is input to the primary side of the transformer T1 and switched through the transformer T1 and the switching element Q1 (here, MOSFET), and a desired DC voltage Vo is generated on the secondary side of the transformer T1. Further, Vo is controlled to be maintained at a constant voltage through a feedback circuit 300 surrounded by a broken line.

  The feedback circuit 300 includes an error detection / amplification circuit 301 and a photocoupler IC 302. The output signal FB of the photocoupler IC 302 is compared with the reference signal REF in the control IC 400.

  FIG. 4 is a diagram showing a state of this comparison. As shown in FIG. 4, the reference signal REF is a triangular wave (or a sawtooth wave), and its frequency is equal to the on / off switching frequency f of the switching element Q1. On the other hand, the output signal FB of the photocoupler IC 302 increases when the secondary side load current increases, and conversely decreases when the load current decreases. Then, a comparison result between the two signals is output as a drive signal DRV for the switching element Q1. The drive signal DRV is applied to the switching element Q1, here the gate of the MOSFET, via the resistor R2. The diode D1 connected in parallel to the resistor R2 serves to promote the elimination of minority carriers when the switching element Q1 is turned off, and to shorten the off time.

  As is clear from FIG. 4, in the cycle T1 of the drive signal DRV, the level of the signal FB is low, that is, the load current is small, so that the ON period (DRV high level period) of the drive signal DRV is short. On the other hand, in the period T2, the level of the signal FB is high. Therefore, the load current is increased, so that the ON period of the drive signal DRV is long. As described above, the level of the signal FB changes due to the change in the load current on the secondary side, and thereby, the ON period of the drive signal DRV of the switching element Q1 is controlled, and as a result, the output voltage Vo is controlled to be constant. Is done.

  Returning to FIG. 3 and continuing the description, the circuit 100 surrounded by a broken line in the figure is a detection circuit that detects the level of the current flowing through the switching element Q1. The circuit 200 is a thyristor circuit triggered by receiving the output of the circuit 100. When this thyristor circuit is triggered, the gate signal of the switching element Q1 is forcibly pulled to a low level through the diode D2, and the switching element Q1 is permanently turned off. In other words, this thyristor circuit serves as a control circuit for controlling the operation of the switching element Q1.

  First, the circuit 100 will be described. The current flowing through the switching element Q1 is detected by the resistor R1. The voltage generated in the resistor R1 is input to the (+) input which is one input of the amplifier (IC) 101 via the resistor R3, and then amplified according to the gain A of the amplifier 101. Gain A is determined by several resistors connected to the other input of the amplifier, the inverting (-) input, in this embodiment resistors R6, R7, R8 are associated therewith. Therefore, the amplifier (IC) 101 operates as a differential amplifier in this embodiment. Furthermore, since the resistor R6 is bypassed by the transistor Q2 when the input voltage of the commercial power supply is 200 VAC, it does not contribute to the gain A at that time, and consequently contributes to the gain A only in the case of 100 VAC. Become.

  This point will be described in detail. That is, when the input voltage of the commercial power supply is AC100V, the positive side voltage (node 150 in the figure) of the electrolytic capacitor C1 is about DC135V. Further, when the Zener diode ZD1 selects the Zener voltage of, for example, 200V, the Zener diode ZD1 is cut off when the voltage of the node 150 is DC135V. Therefore, no current flows through the resistors R4 and R5 connected in series with the Zener diode ZD1, so that the transistor Q2 is turned off. Therefore, when the input voltage of the commercial power supply is AC 100 V, the transistor Q2 is turned off, and the gain A of the amplifier (IC) 101 is as shown in Expression (6).

A = R8 / (R6 + R7) (6)
On the other hand, when the input voltage of the commercial power supply is AC200V, the voltage at the node 150 is about DC270V, which is higher than the Zener voltage 200V of the Zener diode ZD1. For this reason, a current flows through the resistors R4 and R5 through the Zener diode ZD1, and the transistor Q2 is turned on. As a result, the resistor R6 is bypassed by the transistor Q2. Therefore, the gain A of the amplifier (IC) 101 when the input voltage of the commercial power supply is 200 VAC is as shown in Expression (7).

A = R8 / R7 (7)
As can be seen from a comparison between Equation (6) and Equation (7), in this embodiment, it can be seen that the gain A when the input voltage of the commercial power supply is 200 VAC is larger than that when 100 VAC. This means that even when the current flowing through the resistor R1 is the same, the voltage appearing at the output of the amplifier (IC) 101 (node 160 in the figure) is larger in the case of AC200V than in the case of AC100V. Means.

Note that the voltage of the node 160 is divided by the resistors R9 and R10 and transmitted to the latch circuit in the subsequent stage (the circuit 200 surrounded by a broken line in the drawing).
As described above, the latch circuit 200 is basically a thyristor circuit, and includes two transistors Q3 and Q4. When the base voltage of the transistor Q3 shown as the node 170 in the figure exceeds about 0.6V, the transistor Q3 is turned on. At that moment, the PNP transistor Q4 is also turned on, and then both the transistors Q3 and Q4 are permanently turned on. To maintain. In this embodiment, when the thyristor circuit 200 is turned on, the gate voltage of the switching element Q1 is forcibly pulled to a low level via the diode D2, and thereafter, the driving of the switching element Q1 is stopped.

  The operations described above are summarized as shown in FIG. As shown in FIG. 5, the current flowing through the switching element Q1 gradually increases, and the peak output voltage of the amplifier (IC) 101 shown as the node 160 also increases accordingly. As a result, the base voltage of transistor Q3 shown as node 170 gradually increases. Then, at the time point P when the trigger level of the transistor Q3, that is, about 0.6V is reached, the thyristor circuit 200 composed of the transistors Q3 and Q4 transitions from the off state to the on state, and then the driving of the switching element Q1 stops. . Here, the trigger sensitivity of the transistor Q3 is determined by a time constant circuit including two resistors R9 and R10 and one capacitor C3. Usually, in order to avoid malfunction due to noise, the trigger sensitivity is several tens of milliseconds to several It has a time constant of 100 milliseconds. Therefore, as can be seen from FIG. 5, unless the peak value of the current pulse flowing through the switching element Q1 continues to take a large value for a certain period, the transistor Q3 is not triggered and the overcurrent protection operation does not operate.

  Next, with reference to FIG. 6, the difference of the overcurrent detection point according to the difference of the input voltage of a commercial power supply is demonstrated.

Fig.6 (a) is a figure which shows the current waveform at the time of AC100V. A waveform 91 shows the output of the amplifier (IC) 101, and a waveform 92 shows the input voltage of the transistor Q3, that is, the voltage of the node 170. As is apparent from FIG. 6A, the capacitor C3 is charged through the resistors R9 and R10 during the on period Ton, and the capacitor C3 is discharged during the off period Toff. However, as the peak voltage Ip ₁ of the IC 302 increases, the level of the waveform 92 gradually increases. When the trigger level of the transistor Q3 is reached (point P1), the thyristor circuit 200 is triggered as described above. Here, as shown in FIG. 6A, the slope of the output waveform of the IC 101 at AC 100 V is α.

Next, in the case of AC200V, assuming that the gain of the amplifier (IC) 101 is given by equation (6) as in the case of AC100V, the output current of the amplifier (IC) 101 is the solid line in FIG. 6 (b). The waveform 93 and the waveform 95 shown in FIG. Here, the slope of the output waveform of the IC 101 is twice that of AC 100 V, and is therefore 2α. Theoretically, the timing at which the overcurrent protection operation should be activated is approximately the timing at which the peak voltage Ip ₂ becomes equal to Ip ₁ . This is because the generated energy is given by equation (4), the magnitude of which is largely determined by Ip, and the proportion contributed by Iq is considerably smaller than Ip.

Incidentally, as apparent from FIG. 6 (b), even though the output voltage of IC101 reached Ip _2, the input of the transistors Q3, i.e., the node 170 to not reach the trigger level, remains point P2 Yes. The reason for this is that, as is clear from FIG. 6B, in the case of AC 200V, the on period Ton is approximately one half of that in the case of AC 100V, but the off period Toff is longer. That is, while the charging time is shortened by about half, the discharging time becomes longer, and as a result, the potential of the capacitor C3 does not rise as in the case of AC100V. For these reasons, if the gain of the amplifier at AC100V is equal to that at AC200V, the overcurrent protection operating point at AC200V will be considerably larger than that at AC100V, which may cause a safety problem. Result.

  In this embodiment, in order to solve such a problem, there is provided means for setting the gain of the amplifier (IC) 101 at the time of AC 200 V to a value larger than that at the time of AC 100 V. Specifically, as described above, the gain is variably controlled by bypassing a specific one of the plurality of resistors related to the gain determination according to the situation. As a result, waveforms 94 and 96 indicated by broken lines in FIG. 6B are obtained. Waveform 94 is obtained at node 160 based on the amplifier gain obtained in equation (7), and waveform 96 shows the corresponding node 170 waveform, ie, the input voltage of transistor Q3.

  According to the embodiment described above, the value of the bypass resistor R6 is adjusted to set the overcurrent protection operating point at 200V AC to a value almost equal to that at 100V AC (P3 in the figure). It becomes possible. As a result, even if the input voltage of the commercial power supply used is high or low, the overcurrent protection operating point can be maintained almost constant, and the circuit, and finally the recording apparatus itself can be protected.

  Thus, by controlling the gain of the amplifier for detecting the current flowing through the switching element, it is possible to maintain an almost constant overpower protection operation point or overcurrent protection operation point according to the input voltage of the commercial power supply. Can do.

  Note that the gain A of the amplifier is also affected by factors such as overcurrent response sensitivity associated with the above-described time constant circuit, and therefore it is desirable that the amplifier gain A be appropriately determined through separate simulations and experiments.

  FIG. 7 is a diagram showing a configuration of a power supply circuit according to the second embodiment of the present invention. In FIG. 7, the same components as those already described with reference to FIG. 3 are denoted by the same reference numerals, and description thereof is omitted. Here, instead of controlling the gain of the amplifier (IC) 101 as in the case of the first embodiment, the gain remains constant, the division ratio of the output is controlled, and the result is transmitted to the thyristor circuit 200 at the subsequent stage. It is characterized by doing.

  Hereinafter, the characteristic configuration and operation of this embodiment will be described in detail.

As in the case of the first embodiment, when the AC voltage is 200 V, the transistor Q2 is turned on. As a result, the PNP transistor Q5 is also turned on, and the resistor R9 is bypassed. Therefore, the voltage level V _nd3 appearing at the node 170 is expressed by Expression (8).

V _nd3 = {R10 / (R10 + R16)} · V _nd2 (8)
Here, V _nd2 indicates the output of the amplifier (IC) 101, that is, the voltage appearing at the node 160.

On the other hand, since transistor Q2 is off at AC _100V , transistor Q5 is also off, and voltage level _Vnd3 appearing at node 170 is expressed by equation (9).

V _nd3 = {R10 / (R9 + R10 + R16)} · V _nd2 (9)
From equations (8) and (9), it can be seen that the voltage appearing at node 170 when AC is 200 V is larger than that when AC is 100 V because the divisor does not include the factor of resistor R9.

  Therefore, according to the embodiment described above, controlling the output voltage division ratio of the amplifier can achieve the same effect as in the first embodiment.

  In this way, by controlling the division ratio of the output voltage of the amplifier for detecting the current flowing through the switching element, an almost constant overpower protection operation point or overcurrent protection operation point according to the input voltage of the commercial power supply. Can be maintained.

Claims (7)

Switching power supply comprising a transformer for inputting and rectifying an AC commercial power supply having a different voltage, inputting the rectified AC voltage to a primary side, converting the DC voltage into a DC voltage, and outputting the DC voltage from a secondary side A circuit,
A switching element connected in series to the primary side of the transformer and controlling the input of the AC voltage to the primary side;
A detection circuit connected to the switching element and detecting a voltage on a primary side of the transformer;
An amplifier whose gain changes according to the voltage detected by the detection circuit and amplifies the voltage detected by the detection circuit according to the gain;
And a control circuit that controls on / off of the switching element in accordance with a voltage output from the amplifier. The switching element is a MOSFET;
The control circuit is a thyristor circuit;
The output of the thyristor circuit is input to the gate of the MOSFET,
2. The switching power supply circuit according to claim 1, wherein the MOSFET is turned on / off by an output voltage from the thyristor circuit. The detection circuit includes a Zener diode;
The amplifier is a differential amplifier;
3. The switching power supply circuit according to claim 1, wherein a voltage detected by the detection circuit is input to one input of the differential amplifier. A transistor that operates according to an output voltage from the Zener diode;
Two resistors connected in series to the other input of the differential amplifier;
The transistor is connected in parallel to one of the two resistors, and is turned on / off according to an output voltage from the Zener diode, thereby bypassing a current flowing through the one resistor connected in parallel. The switching power supply circuit according to claim 3. The detection circuit includes a Zener diode;
A transistor that operates according to an output voltage from the Zener diode;
Two resistors for dividing the output voltage of the amplifier;
A transistor connected between the two resistors and the output of the amplifier and operated by the transistor;
The switching power supply circuit according to claim 2, wherein the divided voltage is connected as an output voltage to the control circuit.   The gain of the amplifier is larger when the input voltage of the commercial power supply is 200V than when the input voltage of the commercial power supply is 100V. Switching power supply circuit.   A recording apparatus using the switching power supply circuit according to claim 1.

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