JP2011091976A - Motor controller and motor control system - Google Patents

Abstract

The present invention provides a stable control even when a rotational frequency of a permanent magnet motor is in a low speed region and a high load torque is applied.
A motor control system includes a permanent magnet motor, a power converter, and a vector control device that performs vector control of the permanent magnet motor via the power converter with a frequency command value ωr ^* as a target value. The vector control apparatus 200 includes an axis error estimator 5 that estimates an axis error that is a deviation between the rotational phase estimated value θdc obtained by integrating the frequency estimated value ω ^ and the rotational phase value of the permanent magnet motor. , A frequency estimator 6 that estimates the output frequency by proportionally integrating the difference between the estimated value Δθc of the axis error and the command value Δθc ^* of the axis error, and the proportional-integral calculation when the estimated output frequency is equal to or less than a predetermined value. And a gain calculator 7 for limiting the upper limits of the control gains Kp and Ki, and control is performed so that the estimated value of the axis error matches the axis error command value.
[Selection] Figure 1

Description

  The present invention relates to a motor control device and a motor control system for a permanent magnet motor, and more particularly to a motor control technique capable of realizing highly stable control characteristics even in a low speed region.

  The permanent magnet motor control system detects the rotational position of the motor and controls the rotational frequency and the like to approach the target value. In this control system, it is preferable to omit a position sensor for detecting the rotational position of the motor in order to reduce the space of the motor drive unit. As a position sensorless vector control method with this position sensor omitted, the axis error is estimated using the voltage command value, current detection value, and frequency estimation value, which are the vector control outputs, and this estimated value is used as the axis error command. A control technique is disclosed in which proportional and integral calculations are performed so as to match the values to obtain the estimated frequency value (see Patent Document 1).

JP 2001-251889 A

The control technology described in Patent Document 1 realizes high-stability control from a low load to a high load when the rotational frequency of the permanent magnet motor is medium to high, or high stability only for a low load at a low speed. .
However, according to the control technique described in Patent Document 1, when a high load torque is applied in a low speed region (excluding zero speed), the rotational frequency of the permanent magnet motor is lowered, which is related to the motor constant and the load size. As a result, an unstable phenomenon in which the estimated value of the shaft error vibrates, and the permanent magnet motor is out of step.

  The present invention has been made to solve such problems, and a motor that can be stably controlled even when a rotational frequency of a permanent magnet motor is in a low speed region and a high load torque is applied. It is an object to provide a control device and a motor control system.

In order to achieve the above object, a motor control device of the present invention includes a vector control arithmetic unit that performs vector control of a permanent magnet motor (1) through a power converter (2) using a frequency command value (ωr ^* ) as a target value. (50) and an axis error estimating unit (5) for estimating an axis error which is a deviation between the rotational phase estimated value (θdc) obtained by integrating the frequency estimated value (ω ^) and the rotational phase value of the permanent magnet motor. ) And a frequency estimator that estimates a difference between the estimated value of the shaft error and the command value of the shaft error by proportional-integral calculation, and the estimated value of the shaft error matches the shaft error command value In the motor control device that controls to perform, when the estimated output frequency is equal to or lower than a predetermined value, the motor control device includes a gain calculation unit (7) that limits or resets the control gain of the proportional-integral calculation. And Here, the control gain is determined using a known motor constant, current value, or estimated frequency value. The numbers and symbols in parentheses are examples.

  According to the present invention, the rotational frequency of the permanent magnet motor can be controlled stably even when a high load torque is applied in a low speed region.

It is a block diagram of the motor control apparatus of the permanent magnet motor which shows 1st Embodiment of this invention. Control response frequency omega _c is set to the frequency estimator is a diagram showing an operating characteristic of the lower case _{(ω c = 60rad / s)} . Control response frequency omega _c is set to the frequency estimator is a diagram showing operating characteristics of the high case _{(ω c = 240rad / s)} . It is a flowchart of the gain calculation part which is the characteristic structure of this invention. It is a figure which shows the driving | running characteristic at the time of using 1st Embodiment of this invention ((omega) _c = 240rad / s). It is a block diagram of the vector control apparatus of the permanent magnet motor which shows 2nd Embodiment of this invention. It is a flowchart of the gain calculating part 7a which is the characteristic structure of 2nd Embodiment of this invention. It is a figure which shows the control characteristic at the time of using the gain calculating part 7 when the electric current detection value Idc of d-axis is "positive". It is a figure which shows the control characteristic at the time of using the gain calculating part 7a when the electric current detection value Idc of d-axis is "positive". It is a block diagram of the motor control apparatus of the permanent magnet motor which shows 3rd Embodiment of this invention. It is a flowchart of the gain calculating part 7b which is the characteristic structure of 3rd Embodiment of this invention. It is another flowchart of the gain calculating part 7b which is the characteristic structure of 3rd Embodiment of this invention. It is a block diagram of the motor control apparatus of the permanent magnet motor which shows 4th Embodiment of this invention. It is an external view of a general-purpose inverter device as a motor control device of the present invention.

<First Embodiment>
FIG. 1 is a configuration diagram of a motor control system for a permanent magnet motor according to a first embodiment of the present invention.
The motor control system 100 (100a) includes a motor control device 200 (200a), a DC power source 21, a power converter 2 that converts DC power supplied from the DC power source 21 into three-phase power, and a power converter 2 supplied. The permanent magnet motor 1 driven by the three-phase electric power and the current detector 3 for detecting the three-phase current flowing through the permanent magnet motor 1 are used, and the frequency command value ωr ^* input to the motor control device 200 is obtained. The rotational frequency ωr of the permanent magnet motor 1 is vector-controlled to the target value. In the present embodiment, the frequency command value ωr ^* is set to a predetermined value (for example, 1/10 of the rated rotational frequency) or more.

  In the permanent magnet motor 1, a rotor (not shown) is rotatably provided inside a stator (not shown), and a three-phase rectangular wave voltage is applied to the stator winding, so that the rotating shaft of the rotor has a rotational frequency. Rotating at ωr, a load torque T is generated by synthesizing the torque component pΦ · iq caused by the magnetic flux of the permanent magnet and the reluctance torque component p (Ld−Lq) · id · iq caused by the difference in the inductance direction of the armature winding. At this time, the mechanical output P is P = ωr · T, and p is the number of pole pairs.

The power converter 2 converts the DC power supplied from the DC power source 21 into three-phase AC power and drives the permanent magnet motor 1. The power converter 2 compares a three-phase AC voltage command value vu ^* , vv ^* , vw ^* with a triangular wave, and converts a PWM (Pulse Width Modulation) modulated three-phase rectangular wave voltage into a permanent magnet motor 1. Applied to the stator winding.

  The current detector 3 is composed of a Hall element or the like, detects three-phase alternating currents iu, iv, and iw flowing through the permanent magnet motor 1, and outputs the detected values Iuc, Ivc, and Iwc to the motor control device 200.

  The motor control device 200 includes a CPU (Central Processing Unit), a ROM (Read Only Memory), a RAM (Random Access Memory), and the like, and the CPU executes a program stored in the ROM, thereby , Axis error estimator 5, frequency estimator 6, gain calculator 7 (7 a), phase estimator 8, speed control calculator 9, d-axis current command setting unit 10, q-axis current control calculation The functions of the unit 11, the d-axis current control calculation unit 12, the vector control calculation unit 13, the coordinate conversion unit 14, and the adders 15a, 15b, 15c, and 15d are realized. The motor control device 200 estimates the rotational position of the permanent magnet motor 1 using the signals of the current detection values Iuc, Ivc, and Iwc, and uses the estimation result to set the frequency command value ωr * as the target value and the permanent magnet motor. 1 is vector controlled.

  The coordinate conversion unit 4 performs coordinate conversion on the detected values of the three-phase alternating currents iu, iv, iw and the rotational phase estimated value θdc to convert the d-axis current detection value Idc and the q-axis current detection. The value Iqc is calculated and output to the power converter 2.

The coordinate conversion unit 14 performs coordinate conversion of the voltage command values Vdc ^* and Vqc ^** and the rotation phase estimation value ωdc to convert the voltage command values Vu ^* , Vv ^* and Vw ^* of three-phase AC into adders 15b and 15c, And output to the axis error estimator 5.

The adder 15 a calculates the deviation between the frequency command value ωr ^* and the frequency estimated value ω ^, and outputs the calculation result to the speed control calculation unit 9.
The speed control calculation unit 9 calculates the q-axis current command value Iq ^* from the deviation between the frequency command value ωr ^* and the estimated frequency value ω ^, and outputs the calculation result to the adder 15b.
The d-axis current command setting unit 10 outputs a first d-axis current command value Id ^* that is zero. In the field weakening control, the d-axis current command setting unit 10 outputs a predetermined value that is not zero.

The adder 15 b calculates a deviation between the first q-axis current command value Iq ^* and the q-axis current detection value Iqc, and outputs the calculation result to the q-axis current control calculation unit 11. The adder 15 c calculates a deviation between the first d-axis current command value Id ^* and the d-axis current detection value Idc, and outputs the calculation result to the d-axis current control calculation unit 12.
The q-axis current control calculation unit 11 outputs a second q-axis current command value Iq ^** from the deviation between the first q-axis current command value Iq ^* and the q-axis current detection value Iqc. The d-axis current control calculation unit 12 outputs the second d-axis current command value Id ^** from the deviation between the first d-axis current command value Id ^* and the d-axis current detection value Idc.

The vector control calculation unit 13 is based on the electric constants (R, Ld, Lq, Ke) of the permanent magnet motor 1, the second current command values Id ^** , Iq ^** , and the frequency estimated value ω ^ (1) The d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* are calculated according to the equation and output to the coordinate conversion unit 14.

ここに、
Ｒ ：抵抗値
Ｌｄ ：ｄ軸インダクタンス値、 Ｌｑ ：ｑ軸インダクタンス値
Ｋｅ ：誘起電圧係数 ＊ ：設定値
である。

here,
R: resistance value Ld: d-axis inductance value, Lq: q-axis inductance value Ke: induced voltage coefficient *: set value

  First, assuming that the control gain set in the frequency estimator 6 is a fixed value, the basic operation of vector control voltage control and phase control will be described.

The axis error estimation unit 5 uses the voltage command values Vdc ^* , Vqc ^* , the current detection values Idc, Iqc, the frequency estimation value ω ^, and the set values of the electrical constants (R, Ld, Lq) of the permanent magnet motor 1. Then, an axis error estimated value Δθc (= θdc−θd), which is a deviation between the rotational phase estimated value θdc and the rotational phase value θd, is estimated and calculated by the equation (2).

The adder 15d subtracts the axis error estimated value Δθc from the axis error command value Δθc ^* which is “zero”. The frequency estimation unit 6 performs a proportional / integral calculation on the deviation between the axis error command value Δθc ^* and the axis error estimated value Δθc, and outputs the calculation result as a frequency estimated value ω ^.

ここに、
Ｋｐ：比例ゲイン、Ｋｉ：積分ゲイン
であり、ゲイン演算部７が（４）式に示すように、Ｋｐ，Ｋｉを設定している。

ここに、
ωｃ：周波数推定部６に設定する制御応答周波数［ｒａｄ／ｓ］
Ｎ ：比例・積分の折れ点比（通常は、自然数１から２０までの範囲で設定される。
である。 Further, the frequency estimation unit 6 performs PLL control so that the estimated phase error value Δθc becomes “zero”, calculates the estimated speed value ω ^ using the equation (3), and adds the adder 15 a and the phase estimation unit 8. , Output to the axis error estimation unit 5 and the gain calculation unit 7.

here,
Kp: proportional gain, Ki: integral gain, and Kp and Ki are set by the gain calculation unit 7 as shown in equation (4).

here,
ωc: Control response frequency [rad / s] set in the frequency estimation unit 6
N: Proportional / integral breakpoint ratio (usually set in the range of natural numbers 1 to 20.
It is.

ここに、
Ｓ：ラプラス演算子
以上が、電圧制御と位相制御との基本動作の説明である。 The phase estimation unit 8 outputs a signal obtained by integrating the frequency estimation value ω ^ to the coordinate conversion units 4 and 14 as the rotation phase estimation value θdc by the calculation shown in the equation (5).

here,
S: Laplace operator The above is the description of the basic operation of voltage control and phase control.

  The gain calculation unit 7 receives the frequency estimation value ω ^, calculates the proportional gain Kp and the integral gain Ki set in the frequency estimation unit 6, and outputs the calculation result to the frequency estimation unit 6.

From here, the effect of the “gain calculation part 7” which is the characteristic configuration of the present embodiment will be described. First, control characteristics when the gain calculation unit 7 is not used as a conventional method will be described.
FIG. 2 is a diagram illustrating “control characteristics when the control response frequency ωc is low (ωc = 60 rad / s)” set in the frequency estimation unit 6. The upper vertical axis in FIG. 2 indicates the load torque T, the middle vertical axis indicates the motor rotation frequency ωr, the lower vertical axis indicates the axis error estimated value Δθc, and the horizontal axis indicates time [s]. The frequency estimation unit 6 sets (resets) the proportional gain Kp and the integral gain Ki using the equation (4), and the frequency command value ωr ^*, which is the command value of the rotation frequency of the permanent magnet motor 1, is the rated rotation. It is set to 1/10 of the frequency.

  In FIG. 2, a load is applied to the permanent magnet motor 1 so as to increase the load torque T linearly (in a ramp state) from point a (0 Nm) to point b (100 Nm) (upper stage). The permanent magnet motor 1 is controlled stably (middle stage) while the rotational frequency ωr is decreased during the change of the load torque T. However, the axis error estimated value Δθc is generated about 20 ° at the point a (lower stage), the primary current of the permanent magnet motor 1 is increased, and the efficiency is deteriorated.

FIG. 3 is a diagram showing “control characteristics when the control response frequency ωc is increased (ωc = 240 rad / s)”. 2, the upper vertical axis in FIG. 3 represents the load torque T, the middle vertical axis represents the motor rotation frequency ωr, the lower vertical axis represents the estimated axis error Δθc, and the horizontal axis represents time. [S] is shown.
Near the point a where the linear increase of the load torque T starts, the estimated shaft error Δθc is almost zero, indicating that highly efficient control is being performed. However, the estimated shaft error Δθc diverges from the point c, and the motor rotational frequency ωr rapidly decreases at the point d. That is, the motor control system 100 has a problem that the permanent magnet motor 1 is stepped out due to the divergence of the axis error estimated value Δθc.

ここで、ｄ軸電圧値Ｖｄ、及びｑ軸電圧値Ｖｑは、（７）式で示すことができる。

（７）式のｄ軸電圧値Ｖｄは、ｑ軸電圧指令値Ｖｑｃ^＊の情報が含まれている。このｄ軸電圧値Ｖｄの変動により、ｄ軸電流Ｉｄ⇒ｑ軸電圧Ｖｑ⇒ｑ軸電流Ｉｑが変動する。軸誤差推定部５は、この電流検出値Ｉｄｃ，Ｉｑｃを用いた演算を行っている。このため、モータ制御装置２００は、軸誤差の推定値Δθｃに関係する一巡の不安定ループが発生することになる。
ここで、「軸誤差の指令値Δθｃ^＊」から「軸誤差の推定値Δθｃ」までの、不安定ループに関係する伝達関数ゲインをＧθとすると、

ここに、バーＩｑｃ：ｑ軸の電流検出の大きさ［平均値］、バーＩｄｃ：ｄ軸電流検出［平均値］、バーω＾：周波数推定値の平均値
である。
（８）式より、軸誤差の推定値Δθｃの安定条件を考えると、伝達関数ゲインＧθがすべての周波数域において、「１」以下であれば閉ループのシステムは安定であるという技術常識から、この場合の安定条件は（９）式となる。

（９）式から、周波数推定部６の比例ゲインＫｐ、周波数推定値の平均値、ｄ軸電流検出の平均値、及びｑ軸電流検出の平均値、が安定性に関係していることが分かる。
（９）式より、ゲイン演算部が設定し、あるいは制限すべき比例ゲインＫｐ（＝ωｃ）は、（１０）式となる。

通常のベクトル制御装置は、ｑ軸電流検出値ＩｄｃがＩｄｃ＝０になるように制御するため、比例ゲインＫｐを（１１）式により決定する。

つまり、安定に運転するためには、ゲイン演算部は（１１）式の関係で、制御応答周波数ωｃを設定し、あるいは制限する必要がある。 Hereinafter, “the cause of the divergence of the axis error estimated value Δθc” that occurs when the control response frequency ωc is set to a high value of 240 rad / s will be described.
When there is an axis error Δθ (= θdc−θd) that is a deviation between the rotational phase estimated value θdc and the rotational phase value θd, the conversion matrix from the control side (dc−qc) to the motor shaft (dq) is (6)

Here, the d-axis voltage value Vd and the q-axis voltage value Vq can be expressed by Equation (7).

The d-axis voltage value Vd in the equation (7) includes information on the q-axis voltage command value Vqc ^* . Due to the fluctuation of the d-axis voltage value Vd, the d-axis current Id → q-axis voltage Vq → q-axis current Iq varies. The axis error estimation unit 5 performs a calculation using the detected current values Idc and Iqc. For this reason, in the motor control device 200, one round of unstable loop related to the estimated value Δθc of the axis error occurs.
Here, if the transfer function gain related to the unstable loop from the “axis error command value Δθc ^* ” to the “axis error estimated value Δθc” is Gθ,

Here, bar Iqc: q-axis current detection magnitude [average value], bar Idc: d-axis current detection [average value], bar ω ^: average frequency estimated value.
From the technical common sense that the closed loop system is stable if the transfer function gain Gθ is “1” or less in all frequency ranges, considering the stability condition of the estimated value Δθc of the axis error from the equation (8). In this case, the stable condition is expressed by equation (9).

From equation (9), it can be seen that the proportional gain Kp of the frequency estimation unit 6, the average value of the frequency estimation value, the average value of the d-axis current detection, and the average value of the q-axis current detection are related to the stability. .
From the equation (9), the proportional gain Kp (= ωc) to be set or restricted by the gain calculation unit is the equation (10).

Since the normal vector control device controls the q-axis current detection value Idc to be Idc = 0, the proportional gain Kp is determined by the equation (11).

That is, in order to operate stably, the gain calculation unit needs to set or limit the control response frequency ωc according to the relationship of the expression (11).

ここに、
Ｉｑｃ＿ｍａｘ：最大トルクに比例するｑ軸電流検出値の大きさ
である。
ステップＳ７２では、ゲイン演算部７は、（１２）式で演算したωｃ＿ｌｍｔ０と任意の初期値で設定してあるωｃ＿ｉｎｉの比較を行い、比較結果に応じて、ステップＳ７３，Ｓ７４で（１３）式の判断処理を行う。

ステップＳ７５では、ゲイン演算部７は、（１４）式を用いて、周波数推定部６に設定する比例ゲインＫｐ、及び積分ゲインＫｉを演算し、処理を終了する。

Therefore, the processing of the “gain calculation unit 7”, which is a feature of the present embodiment, will be described with reference to FIG. In step S71, the gain calculation unit 7 calculates the control response frequency ωc_lmt0 that can be stably operated up to the maximum torque, using the equation (12).

here,
Iqc_max: The magnitude of the q-axis current detection value proportional to the maximum torque.
In step S72, the gain calculation unit 7 compares ωc_lmt0 calculated in equation (12) with ωc_ini set as an arbitrary initial value, and in steps S73 and S74, the equation (13) is compared according to the comparison result. Judgment processing is performed.

In step S75, the gain calculator 7 calculates the proportional gain Kp and the integral gain Ki set in the frequency estimator 6 using equation (14), and ends the process.

Here, control characteristics when the gain calculation unit 7 of the present embodiment is used will be described.
FIG. 5 is a diagram showing control characteristics when the control gain of the frequency estimation unit 6 is limited using the equations (12) to (14) (ωc = 240 rad / s used in FIG. 2 is an initial value). Set as). The upper vertical axis in FIG. 5 represents the load torque T, the middle vertical axis represents the motor rotation frequency ωr, the lower vertical axis represents the axis error estimated value Δθc, and the horizontal axis represents time [s].

As shown in FIG. 3, in the conventional motor control device, since the estimated shaft error Δθc is likely to diverge, the permanent magnet motor 1 has stepped out. However, the motor control device 200 of the present embodiment has a control gain. By limiting the motor rotational frequency ωr slightly (FIG. 5B), the estimated shaft error Δθc operates stably (FIG. 5C), and a highly stable control characteristic is realized. .
Further, the motor control device 200 of the present embodiment does not need to lower the control response frequency ωc more than necessary, and the axial error estimated value Δθc generated at point a is smaller than in the case of FIG. Control operation can be realized.

The motor control device 200 performs an operation of setting the frequency command value ωr ^* to a constant value and increasing or decreasing the load torque in a ramp shape. At this time, in the direction in which the load torque increases, the motor control device 200 performs control so as to decrease the control gain used for the speed estimation calculation, and the fluctuation of the axis error estimated value Δωc increases. As a result, the motor efficiency is deteriorated and more current than necessary is generated.
On the other hand, in the direction in which the load torque decreases, the motor control device 200 performs control so that the control gain used for the speed estimation calculation increases, and the fluctuation of the axis error estimated value Δωc becomes small. For this reason, the motor efficiency does not deteriorate, and a current corresponding to the load torque is generated.

Second Embodiment
Since the permanent magnet motor 1 (see FIG. 1) has a small induced voltage at the time of start-up and low-speed rotation, sensorless control tends to become unstable. Therefore, at the time of start-up, the d-axis current detection value Idc is gradually increased to fix the rotor at a predetermined rotational position and perform synchronous operation before shifting to sensorless control. In this case, the motor control device needs to gently change the d-axis current detection value Idc in order to suppress the current fluctuation at the time of switching, and the frequency estimation value ω ^ is calculated using the d-axis current detection value Idc. It is preferable to calculate.

FIG. 6 is a configuration diagram of the motor control system 100 (100b) according to the second embodiment of the present invention.
The motor control device 200 (200a) of the first embodiment sets or restricts the control response frequency ωc using the frequency estimation value ω ^ as the d-axis current detection value Idc = 0, but in the second embodiment, The motor control device 200b is different in that the value of the control response frequency ωc is obtained using both the estimated frequency value ω ^ and the detected d-axis current value Idc.

ステップＳ７ａ２では、ゲイン演算部７ａは、（１５）式で演算したωｃ＿ｌｍｔ１と任意の初期値で設定してあるωｃ＿ｉｎｉの比較を行い、この判断結果に応じて、ステップＳ４ａ３，Ｓ７ａ４で、（１６）式の判断処理を行う。

ステップＳ７ａ５では、ゲイン演算部７ａは、（１７）式を用いて、周波数推定部６に設定する比例ゲインＫｐ、及び積分ゲインＫｉを演算し、演算結果を周波数推定部６に対して設定（再設定）する。

In FIG. 6, reference numerals 1 to 6, 8, 9, 11, 14 and 21 are the same as those shown in the block diagram of FIG.
The gain calculation unit 7a receives the frequency estimation value ω ^ and the d-axis current detection value Idc, and outputs a proportional gain Kp and an integral gain Ki set in the frequency estimation unit 6.
The d-axis current command setting unit 10 outputs Id0 as a predetermined value as a d-axis current command value Id ^* .
Therefore, the operation of the “gain calculation unit 7a”, which is a characteristic configuration of this embodiment, will be described with reference to the flowchart of FIG.
In step S7a1, the gain calculation unit 7a calculates the control response frequency ωc_lmt1 that can be stably operated up to the maximum torque using the equation (15).

In step S7a2, the gain calculation unit 7a compares ωc_lmt1 calculated in equation (15) with ωc_ini set with an arbitrary initial value, and in steps S4a3 and S7a4, (16) Perform formula decision processing.

In step S7a5, the gain calculation unit 7a calculates the proportional gain Kp and the integral gain Ki set in the frequency estimation unit 6 using the equation (17), and sets the calculation result in the frequency estimation unit 6 (re-set). Setting).

Here, control characteristics when the gain calculation unit 7a of the present embodiment is used will be described.
FIG. 8 sets a predetermined value of “positive” for the d-axis current command value Id ^* , and sets the control gain of the frequency estimator 6 using the equations (12) to (14) described in the first embodiment. It is a figure which shows the control characteristic at the time of limiting. The upper vertical axis in FIG. 8 indicates the load torque T, the lower vertical axis indicates the axis error estimated value Δθc, and the horizontal axis indicates time [s].
It can be seen that the axis error estimated value Δωc diverges from the vicinity of the point e in FIG. This is because the term related to the detected d-axis current value Idc included in the equation (10) is omitted.
Therefore, in the present embodiment, the control gain of the frequency estimation unit 6 is set using Equations (15) to (17) in consideration of the d-axis current detection value Idc.

FIG. 9 is a diagram illustrating control characteristics when the configuration of the present embodiment is applied.
Since the magnitude of the control gain is smaller than when using the formulas (12) to (14), the estimated value Δωc of the axis error near the point a (upper stage) is slightly larger (lower stage), but the d axis Even when a predetermined value Idc0 of “positive” is set in the current command setting unit 10, highly stable control characteristics can be realized.

  If the predetermined value Idc0 is a “negative” predetermined value, the stable range is widened, so that there is no problem that the axis error estimated value Δωc diverges as in the first embodiment. That is, when Id0, which is a “negative” predetermined value, is set in the d-axis current command setting unit 10, the magnitude of the control gain may be determined using equations (12) to (14).

<Third Embodiment>
FIG. 10 is a configuration diagram of a motor control system according to the third embodiment of the present invention.
In the first embodiment and the second embodiment, the upper limit value of the control gain of the frequency estimator 6 is set or the control gain is limited during actual operation. However, in the third embodiment, before the operation (at least 1 time), the control gain initial value of the frequency estimation unit 6 is automatically set. The d-axis current command value Id ^* 10 is set to Id0 as a predetermined value, as in the first embodiment.

10, reference numerals 1 to 6, 8 to 14, and 21 denote the same elements as those in FIG.
The gain calculation unit 7b calculates the proportional gain Kp and the integral gain Ki at the sensorless lowest frequency ω0 at least once in advance, and outputs them as the control gain initial value of the frequency estimation unit 6.
Therefore, with reference to FIG. 11, the “gain calculation unit 7b” which is a characteristic configuration of the present embodiment will be described.
In step S7b1, the gain calculation unit 7b calculates a control response frequency ωc_ini0 that can be stably operated up to the maximum torque even at the sensorless minimum rotation frequency ω0 using the equation (18).

本実施形態では、センサレス最低回転周波数ω０における制御ゲインを設定するため、第１実施形態の場合と比べて、軸誤差推定値Δωｃは多少大きくなるが、運転中における演算を省略することができる。
また、ｄ軸電流指令設定部１０に「正」の所定値Ｉｄｃ０に設定する場合は、図１２を用いて説明する。
ステップＳ７ｂ３では、（１８）の代わりに（２０）を演算すればよい。

さらに、ステップＳ７ｂ４において、ステップＳ７ｂ３は、（２１）を用いて、周波数推定部６に設定あるいは制限する比例ゲインＫｐ、積分ゲインＫｉを演算する。

このような構成を取れば、制御演算が軽くなる利点がある。 Further, in step S7b2, the gain calculation unit 7b calculates the proportional gain Kp and the integral gain Ki that are set or restricted in the frequency estimation unit 6 using the equation (19), and the calculation result is used as the frequency estimation unit 6 Set to (reset).

In this embodiment, since the control gain at the sensorless minimum rotation frequency ω0 is set, the shaft error estimated value Δωc is somewhat larger than in the case of the first embodiment, but the calculation during operation can be omitted.
Further, the case where the “positive” predetermined value Idc0 is set in the d-axis current command setting unit 10 will be described with reference to FIG.
In step S7b3, (20) may be calculated instead of (18).

Further, in step S7b4, step S7b3 uses (21) to calculate the proportional gain Kp and integral gain Ki that are set or restricted in the frequency estimation unit 6.

With such a configuration, there is an advantage that the control calculation becomes light.

Until this embodiment, the second current command value (Id ^** , Iq ^** ) is created from the first current command value (Id ^* , Iq ^* ) and the current detection value (Idc, Iqc). Although the vector control calculation was performed using the second current command value, the deviation between the first current command value (Id ^* , Iq ^* ) and the current detection value (Idc, Iqc) was directly calculated by proportional / integral calculation. The present invention can also be applied to vector control calculations that output voltage command values (Vd ^* , Vq ^* ) for the d-axis and q-axis.

<Fourth embodiment>
FIG. 13 is a configuration diagram of a motor control system according to the fourth embodiment of the present invention.
The motor control system 100 (100d) has a configuration in which the permanent magnet motor 1 is connected to the inverter board 18b. The inverter board 18b includes the motor control device 200d, the power converter 2, and the DC power source 21, and the motor control device. 200d includes a vector control calculation unit 50d, an axis error estimation unit 5, a frequency estimation unit 6, a gain calculation unit 7, and an adder 15d.
Here, the vector control calculation unit 50d is different from the first embodiment in that it has a simple configuration in which the d-axis current control calculation unit and the q-axis current control calculation unit are omitted. The motor control device 200d is configured by one or more 32-bit class microcomputers 18a, and functions are realized by executing programs.

In FIG. 13, components 1 to 8, 10, 14, and 21 are the same as those in FIG.
The low-pass filter 15 outputs a first-order lag value Iqctd of the q-axis current detection value Iqc, and this output signal is fed back to the vector control calculation unit 13a.

座標変換部１４は、電圧指令値Ｖｄｃ^＊＊、Ｖｄｃ^＊＊と回転位相推定値θｄｃから３相交流の電圧指令値ｖｕ^＊、ｖｖ^＊、ｖｗ^＊を出力する。
このようなｄ軸電流制御演算部、及びｑ軸電流制御演算部を省略したベクトル制御装置においても、第１実施形態と同様な効果を得ることができる。
また、本実施形態では、図１のゲイン演算部７を用いているが、図６、１０のゲイン演算部７ａ，７ｂを用いても同様の効果が得られる。 The vector control calculation unit 13a is based on the electric constant of the permanent magnet motor 1 and the d-axis current command value Id ^* which is “zero”, the primary delay value Iqctd of the q-axis current detection value Iqc, and the frequency command value ωr ^* . According to the equation (22), the d-axis voltage command value Vdc ^** and the q-axis voltage command value Vqc ^** are output.

The coordinate conversion unit 14 outputs three-phase AC voltage command values vu ^* , vv ^* , vw ^* from the voltage command values Vdc ^** , Vdc ^**, and the rotational phase estimation value θdc.
Even in such a vector control apparatus in which the d-axis current control calculation unit and the q-axis current control calculation unit are omitted, the same effect as that of the first embodiment can be obtained.
Further, in the present embodiment, the gain calculation unit 7 of FIG. 1 is used, but the same effect can be obtained by using the gain calculation units 7a and 7b of FIGS.

<Fifth Embodiment>
FIG. 14 is an external view of a general-purpose inverter device using the motor control device of the fifth embodiment of the present invention.
The general-purpose inverter device 18 incorporates the inverter board 18b (FIG. 13) of the fourth embodiment, and the general-purpose permanent magnet motor 1a is connected to the inverter board 18b.
The general-purpose inverter device 18 receives a frequency command value ωr ^* from an operation panel 18c mounted on the front surface.
The general-purpose inverter device 18 can realize highly stable control characteristics even with the general-purpose permanent magnet motor 1a. In the direction in which the load torque increases, the general-purpose inverter device 18 performs control so as to decrease the control gain used for the speed estimation calculation, and the fluctuation of the axis error estimated value Δωc increases. As a result, the motor efficiency is deteriorated and more current than necessary is generated.
On the other hand, the general-purpose inverter device 18 performs control so that the control gain used for the speed estimation calculation increases in the direction in which the load torque decreases, and the fluctuation of the axis error estimated value Δωc becomes small. For this reason, the motor efficiency does not deteriorate, and a current corresponding to the load torque is generated.

  In the first embodiment to the fifth embodiment, the three-phase alternating currents iu to iw detected by the expensive current detector 3 are detected, but are attached for detecting the overcurrent of the power converter 2. The three-phase motor currents iu ^, iv ^, iw ^ can be reproduced from the direct current flowing through the one-shunt resistor, and the "low cost system" using this reproduced current value can be dealt with.

DESCRIPTION OF SYMBOLS 1 Permanent magnet motor 1a General purpose permanent magnet motor 2 Power converter 3 Current detector 4 Coordinate conversion part 5 Axis error estimation part 6 Frequency estimation part 7, 7a, 7b Gain calculation part 8 Phase estimation part 9 Speed control calculation part 10 d axis Current command setting unit 11 q-axis current control calculation unit 12 d-axis current control calculation unit 13, 13a Vector control calculation unit 14 Coordinate conversion unit 15 Low-pass filters 15a, 15b, 15c, 15d Adder 18 General-purpose inverter device 18b Inverter board 18c Operation Panel 21 DC power supply 50, 50a, 50b, 50c, 50d Vector control operation unit 100, 100a100b, 100c, 100d Motor control system 200, 200a, 200b, 200c, 200d Motor controller Id d-axis current Iq q-axis current Idc d-axis Current detection value Iqc q-axis current detection value Vd d-axis voltage V q-axis voltage Id ^* No. 1d-axis current command value ^{Id **.} 2d-axis current command value Iq ^* first 1q axis current command value ^{Iq **} the 2q-axis current command value ωc_lmt0
ωc_lmt1 Control response frequency calculated by gain calculation unit 7 ωc_ini Control response frequency Kp set to initial value of gain calculation unit 7 Proportional gain Ki Integral gain Iqctd Primary delay signal ωr Rotational frequency ω ^ Frequency estimated value ωr ^* Frequency command value θdc Rotation Phase value θdc Estimated rotational phase value Δθc Estimated axis error value Δθ Axis error value T Load torque P Mechanical output

Claims (11)

A vector control calculation unit that vector-controls the permanent magnet motor through the power converter with the frequency command value as a target value,
An axis error estimator for estimating an axis error which is a deviation between the rotational phase estimated value obtained by integrating the frequency estimated value and the rotational phase value of the permanent magnet motor;
A frequency estimator for estimating an output frequency by performing a proportional-integral operation on a difference between the estimated value of the axis error and the command value of the axis error, so that the estimated value of the axis error matches the command value of the axis error A motor control device for controlling,
The motor control device further comprising: a gain calculation unit that limits an upper limit of a control gain of the proportional integration calculation when the estimated output frequency is equal to or less than a predetermined value. A vector control calculation unit that vector-controls the permanent magnet motor through the power converter with the frequency command value as a target value,
An axis error estimator for estimating an axis error which is a deviation between the rotational phase estimated value obtained by integrating the frequency estimated value and the rotational phase value of the permanent magnet motor;
A frequency estimator for estimating an output frequency by performing a proportional-integral operation on a difference between the estimated value of the axis error and the command value of the axis error, so that the estimated value of the axis error matches the command value of the axis error In the motor control device to control,
A motor control device comprising: a gain calculation unit that resets a control gain of the proportional integration calculation when the estimated output frequency is equal to or less than a predetermined value.   The control gain is calculated using a motor constant of the permanent magnet motor, an estimated value or command value of a rotation frequency, and at least one of a d-axis current value and a q-axis current value. The motor control device according to claim 1 or 2. The control gain includes a proportional gain Kp and an integral gain Ki.
The gain calculation unit sets a value when the following control response frequency ωc_lmt0 is equal to or higher than an initial set value to the proportional gain Kp, and sets a value of Kp ² / N to the integral gain Ki. The motor control device according to claim 1, wherein the motor control device is characterized.

here,
Ke: induced voltage coefficient (V / (rad / s)), Iqc_max: maximum value of q-axis current (A)
ω ^: estimated frequency value (rad / s), N: ratio of break points (arbitrary natural number)   The gain calculation unit sets the value of the control gain operable up to a maximum torque as an initial set value of the proportional-integral calculation even at the minimum rotation frequency of the permanent magnet motor. Motor control device. The control gain includes a proportional gain Kp and an integral gain Ki.
The gain calculation unit sets a value when the following control response frequency ωc_lmt0 is equal to or higher than an initial set value to the proportional gain Kp, and sets a value of Kp ² / N to the integral gain Ki. The motor control device according to any one of claims 1 to 5, wherein the motor control device is characterized in that:

here,
Ld: d-axis inductance value (H), Lq: q-axis inductance value (H)
Ke: induced voltage coefficient (V / (rad / s)), Iqc_max: maximum value of q-axis current (A)
ω ^: estimated frequency value (rad / s), N: ratio of break points (arbitrary natural number) The control gain includes a proportional gain Kp and an integral gain Ki.
The gain calculation unit sets a value of the control response frequency ωc_ini0 when the sensorless minimum rotation frequency ω0 or more is set to the proportional gain Kp, and sets a value of Kp ² / N to the integral gain Ki. The motor control device according to claim 3, wherein the motor control device is characterized by the following.

here,
ω0: Sensorless minimum rotation frequency (rad / s)
Ke: induced voltage coefficient (V / (rad / s)), Iqc_max: maximum value of q-axis current (A)
ω ^: Estimated frequency (rad / s), N: Breakpoint ratio (arbitrary natural number)   The motor control device according to any one of claims 1 to 7, wherein the shaft error command value is zero.   The motor control device according to any one of claims 1 to 8, wherein the predetermined output frequency is a rotation frequency equal to or less than 10% of a rated rotation frequency of the permanent magnet motor. A permanent magnet motor;
A power converter for driving the permanent magnet motor;
A motor control system comprising a vector controller for vector-controlling a permanent magnet motor through the power converter with a frequency command value as a target value,
The vector controller is
An axis error estimator for estimating an axis error which is a deviation between the rotational phase estimated value obtained by integrating the frequency estimated value and the rotational phase value of the permanent magnet motor;
A frequency estimator for estimating an output frequency by performing a proportional-integral operation on a difference between the estimated value of the axial error and the command value of the axial error;
When the estimated output frequency is equal to or lower than a predetermined value, a gain calculation unit that limits the upper limit of the control gain of the proportional integral calculation,
The motor control system is characterized in that control is performed so that the estimated value of the axis error coincides with an axis error command value. A permanent magnet motor;
A motor control system comprising an inverter board for speed-controlling the permanent magnet motor, using a frequency command value as a target value,
The inverter board is
A power converter for driving the permanent magnet motor;
An axis error estimator for estimating an axis error which is a deviation between the rotational phase estimated value obtained by integrating the frequency estimated value and the rotational phase value of the permanent magnet motor;
A frequency estimator for estimating an output frequency by performing a proportional-integral operation on a difference between the estimated value of the axial error and the command value of the axial error;
When the estimated output frequency is equal to or lower than a predetermined value, a gain calculation unit that limits the upper limit of the control gain of the proportional integral calculation,
A motor control device that vector-controls the permanent magnet motor via the power converter so that the estimated value of the axis error matches an axis error command value;
A motor control system comprising:

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