JP2010004704A - Dc-dc converter - Google Patents

Abstract

A step-up ratio is obtained without increasing the number of windings of a winding.
A first switch Tr1 connected to both ends of a DC power source Vi via a primary winding 1a and a first reactor Lr1, and a first switch 2 connected via a first winding 2a and a second reactor Lr2. Two switches Tr2, a series circuit of a winding 1b, a diode D1, and a smoothing capacitor Co connected to both ends of a series circuit of the first reactor and the first switch Tr1, and a connection point of the first reactor and the first switch Tr1 A diode D2 connected to one end of the smoothing capacitor Co, a series circuit of a winding 2b, a diode D3, and a smoothing capacitor Co connected to both ends of a series circuit of the second reactor Lr2 and the second switch Tr2, a second reactor Diode D4, secondary winding 1c and secondary connected to the connection point between Lr2 and second switch Tr2 and one end of smoothing capacitor Co The third reactor L1 is connected in series circuit across the line 2c, having a control circuit 10.
[Selection] Figure 1

Description

  The present invention relates to a DC-DC converter including a step-up chopper circuit, and more particularly to a DC-DC converter applied to an electric vehicle.

  FIG. 5 is a circuit configuration diagram of a conventional DC-DC converter. The step-up DC-DC converter includes a DC power supply Vdc1, transformers T3 and T4, a reactor L3, switches Q1 and Q2, diodes D3 and D4, a smoothing capacitor C1, and a control circuit 100.

  The transformer T3 is electromagnetically coupled to the primary winding 5a (number of turns np), the winding 5b (number of turns np1) connected in series to the primary winding 5a, the primary winding 5a, and the winding 5b. Secondary winding 5c (number of turns ns). The transformer T4 is configured in the same manner as the transformer T3, and includes a primary winding 6a (number of turns np), a winding winding 6b (number of turns np1) connected in series to the primary winding 6a, a primary winding 6a, and A secondary winding 6c (number of turns ns) that is electromagnetically coupled to the winding 6b is provided.

  Both ends of the DC power supply Vdc1 are connected between the drain and source of the switch Q1 made of a MOSFET or the like via the primary winding 5a of the transformer T3. Both ends of the DC power supply Vdc1 are connected between the drain and source of the switch Q2 made of a MOSFET or the like via the primary winding 6a of the transformer T4. A connection point between the primary winding 5a of the transformer T3 and the drain of the switch Q1 and the source of the switch Q1 are connected to a first series circuit including a winding 5b of the transformer T3, a diode D3, and a smoothing capacitor C1. ing. A connection point between the primary winding 6a of the transformer T4 and the drain of the switch Q2 and the source of the switch Q2 are connected to a second series circuit including a winding winding 6b of the transformer T4, a diode D4, and a smoothing capacitor C1. ing.

  A reactor L3 is connected to both ends of the series circuit of the secondary winding 5c of the transformer T3 and the secondary winding 6c of the transformer T4. The control circuit 100 turns on / off the switches Q1 and Q2 with a phase difference of 180 ° based on the output voltage Vo of the smoothing capacitor C1.

  According to the conventional DC-DC converter configured as described above, when the switch Q1 is turned on by the Q1 control signal Q1g from the control circuit 100, the current flows through a path of Vdc1 plus → 5a → Q1 → Vdc1 minus. For this reason, the current Q1i of the switch Q1 increases linearly. At the same time, a voltage is also generated in the secondary winding 5c of the transformer T3, and a current L3i flows through the reactor L3 through a path of 5c → L3 → 6c → 5c.

  The current L3i flows according to the law of equiamp turn of the transformer, accumulates energy in the reactor L3, and the same current also flows in the secondary winding 6c of the transformer T4. For this reason, a voltage corresponding to the number of turns is induced in the primary winding 6a and the winding winding 6b of the transformer T4.

  When the winding ratio of the transformer T4 is A = (np + np1) / np, the diode D4 has a current 1 / A of the current Q1i of the switch Q1 Vdc1 plus → 6a → 6b → D4 → C1 → Vdc1 minus. It flows in the route. The current D4i of the diode D4 flows until the time when the switch Q2 is turned on. The output voltage Vo of the smoothing capacitor C1 is the sum of the voltage (input voltage) of the DC power supply Vdc1, the voltage generated in the primary winding 6a of the transformer T4, and the voltage generated in the winding winding 6b of the transformer T4.

  The voltage generated in the transformer T4 is A · Vdc1 · D, where D is the on-duty (D = Ton / T) of the switch Q1. Ton is the ON time of the switch Q1. T is a period for switching the switch Q1. The output voltage Vo of the smoothing capacitor C1 is Vo = Vdc1 (1 + A · D), and the output voltage Vo can be controlled by varying the on-duty D.

  Next, the switch Q1 is turned off by the Q1 control signal Q1g from the control circuit 100. At this time, the current D3i flows through a path of Vdc1 plus → 5a → 5b → D3 → C1 → Vdc1 minus.

  Next, the switch Q2 is turned on by the Q2 control signal Q2g from the control circuit 100. At this time, the current flows along a path of Vdc1 plus → 6a → Q2 → Vdc1 minus. For this reason, the current Q2i of the switch Q2 increases linearly. At the same time, a voltage is also generated in the secondary winding 6c of the transformer T4, and a current L3i flows through the reactor L3 while increasing in a path of 6c → 5c → L3 → 6c.

  The current L3i flows according to the law of equi-ampere turn of the transformer, accumulates energy in the reactor L3, and the same current also flows in the secondary winding 5c of the transformer T3. For this reason, a voltage corresponding to the number of turns is induced in the primary winding 5a and the winding winding 5b of the transformer T3.

  In addition, when the winding ratio of the transformer T3 is A = (np + np1) / np, the diode D3 has a current 1 / A of the current Q2i of the switch Q2 Vdc1 plus 5a → 5b → D3 → C1 → Vdc1 minus. It flows in the route. The current D3i of the diode D3 flows until the time when the switch Q1 is turned on. The output voltage Vo of the smoothing capacitor C1 is the sum of the voltage (input voltage) of the DC power supply Vdc1, the voltage generated in the primary winding 5a of the transformer T3, and the voltage generated in the winding winding 5b of the transformer T3.

As described above, in the multiphase translink type boost chopper circuit shown in FIG. 5, two independent phases are coupled by a transformer. In this way, it is possible to boost the operation of two cores that are necessary with only one core.
JP 2006-262601 A

  However, in the DC-DC converter shown in FIG. 5, recovery losses of the diodes D3 and D4 occur. In addition, a switching loss occurs when the switches Q1 and Q2 are turned on.

  Therefore, by connecting a reactor La (not shown) between the diode D3 and the winding winding 5b and connecting a reactor Lb (not shown) between the diode D4 and the winding winding 6b, the diode D3 is connected. , D4 recovery loss can be suppressed.

  However, even if the recovery loss of the diodes D3 and D4 can be suppressed, when the step-up ratio of the output voltage is modulated by the transformer turns ratio, the characteristics relating to the modulation of the step-up ratio are not so good, and the output current may vary due to load fluctuations, etc. If it changes, there is a problem that the characteristics of the step-up ratio change greatly.

  Further, since the reactors La and Lb are connected in series to the winding windings 5b and 6b, the reactors La and Lb hinder the voltage applied to the winding windings 5b and 6b when a voltage is output. Since the step-up ratio decreases, it is necessary to increase the number of turns of the windings 5b and 6b in order to obtain a necessary step-up ratio.

  Furthermore, if various conditions such as the inductance of the reactors La and Lb and the value of the output current are changed, the period during which the reactors La and Lb block the voltage applied to the windings 5b and 6b also changes. End up. For this reason, it is difficult to perform an optimal design with a predetermined duty ratio.

  An object of the present invention is to provide a DC-DC converter capable of suppressing a recovery loss of a diode and a switching loss when the switch is turned on. It is another object of the present invention to provide a DC-DC converter that can obtain a necessary step-up ratio without increasing the number of winding windings and that can be easily designed optimally with a predetermined duty ratio.

  In order to solve the above-mentioned problems, the invention of claim 1 is a DC-DC converter that boosts the voltage of a DC power supply, and includes a primary winding of a first transformer and a first reactor at both ends of the DC power supply. A first switch connected via the first power source, a second switch connected to both ends of the DC power source via a primary winding of a second transformer and a second reactor, the first reactor and the first switch, A first series circuit comprising a winding winding of the first transformer, a first diode and a smoothing capacitor connected to both ends of the series circuit of the first transformer and connected in series to the primary winding of the first transformer; A second diode connected to a connection point between the first reactor and the first switch and one end of the smoothing capacitor; and connected to both ends of a series circuit of the second reactor and the second switch; A second series circuit comprising a winding winding of the second transformer, a third diode and the smoothing capacitor connected in series to the primary winding of the lance, and a connection between the second reactor and the second switch A fourth diode connected to a point and one end of the smoothing capacitor; a second winding of the first transformer; and a second winding of the second transformer are connected to both ends of a series circuit connected in series. The third reactor, the first switch, and the second switch are alternately turned on every ½ cycle, the first switch is turned off during the ON period of the second switch, and the second switch is turned on And a control circuit for turning off during the ON period of the first switch.

  According to the present invention, since the first reactor is connected in series to the first switch and the second reactor is connected in series to the second switch, the recovery loss of the first, second, third and fourth diodes and the second Switching loss when the first and second switches are turned on can be suppressed. In addition, a necessary step-up ratio can be obtained without increasing the number of windings of the winding, and an optimum design can be easily performed with a predetermined duty ratio.

  Hereinafter, embodiments of a DC-DC converter of the present invention will be described in detail with reference to the drawings.

  FIG. 1 is a circuit configuration diagram illustrating a DC-DC converter according to a first embodiment. The DC-DC converter shown in FIG. 1 includes a multi-phase translink type boost chopper circuit.

  The DC-DC converter includes a DC power source Vi, a transformer T1 (first transformer), a transformer T2 (second transformer), a reactor Lr1 (first reactor), a reactor Lr2 (second reactor), a reactor L1 (third reactor), A switch Tr1 (first switch), a switch Tr2 (second switch), diodes D1, D2, D3, D4, a smoothing capacitor Co, and a control circuit 10 are provided.

  The transformer T1 includes a primary winding 1a (number of turns n1), a winding winding 1b (number of turns n3) connected in series to the primary winding 1a, and a secondary winding 1c electromagnetically coupled to the primary winding 1a. (The number of turns n2). The transformer T2 has the same configuration as the transformer T1, and includes a primary winding 2a (number of turns n4), a winding winding 2b (number of turns n6) connected in series to the primary winding 2a, and a primary winding 2a. And a secondary winding 2c (the number of turns n5) to be electromagnetically coupled.

  The collector and emitter of a switch Tr1 made of an IGBT (insulated gate bipolar transistor) are connected to both ends of the DC power source Vi via a primary winding 1a of a transformer T1 and a reactor Lr1. The collector and emitter of the switch Tr2 made of IGBT are connected to both ends of the DC power source Vi through the primary winding 2a of the transformer T2 and the reactor Lr2.

  A first series circuit including a winding 1b of the transformer T1, a diode D1, and a smoothing capacitor Co is connected to both ends of the series circuit of the reactor Lr1 and the switch Tr1. A second series circuit comprising a winding 2b of the transformer T2, a diode D3 and a smoothing capacitor Co is connected to both ends of the series circuit of the reactor Lr2 and the switch Tr2.

  A diode D2 is connected between a connection point between the reactor Lr1 and the switch Tr1 and one end of the smoothing capacitor Co. A diode D4 is connected between the connection point between the reactor Lr2 and the switch Tr2 and one end of the smoothing capacitor Co.

  A reactor L1 is connected to both ends of a series circuit in which the secondary winding 1c of the transformer T1 and the secondary winding 2c of the transformer T2 are connected in series. Based on the output voltage Vo of the smoothing capacitor Co, the control circuit 10 controls the switch Tr2 to be turned on before the switch Tr1 is turned off after the switch Tr1 is turned on, and the switch Tr1 is turned on before the switch Tr2 is turned off. . That is, there is an overlap period in which the switch Tr1 and the switch Tr2 are turned on simultaneously every half cycle.

  The transformer T1, the reactor Lr1, the diode D1, the diode D2, and the switch Tr1 constitute a first converter, and the transformer T2, the reactor Lr2, the diode D3, the diode D4, and the switch Tr2 constitute a second converter.

  Next, the operation of the DC-DC converter of the first embodiment configured as described above will be described with reference to the timing chart shown in FIG.

  In FIG. 2, time t0 to t3 is a ½ cycle, and time t0 to t1 and time t3 to t4 are overlapping periods in which the switch Tr1 and the switch Tr2 are turned on simultaneously.

  First, at time t0, the switch Tr1 is turned on by the gate signal Tr1g of the switch Tr1 from the control circuit 10. At this time, the current flows along a path of Vi plus → 1a → Lr1 → Tr1 → Vi minus. For this reason, the current i1 flowing through the primary winding 1a of the transformer T1 increases. At the same time, a voltage is also generated in the secondary winding 1c of the transformer T1, and a current L1i flows through the reactor L1 through a path of 1c → 2c → L1 → 1c.

  This current L1i flows according to the law of equiamp turn of the transformer, energy is accumulated in the reactor L1, and the same current also flows in the secondary winding 2c of the transformer T2. For this reason, a voltage corresponding to the number of turns is induced in the primary winding 2a and the winding winding 2b of the transformer T2.

  Further, when the winding ratio of the transformer T2 is A = (n4 + n6) / n4, the diode D3 has a current 1 / A of the current of the switch Tr1 as follows: Vi plus → 2a → 2b → D3 → Co → Vi minus It flows in the route. The current D3i of the diode D3 flows from time t1 to time t3 when the switch Tr2 is turned on. The output voltage Vo of the smoothing capacitor Co is the sum of the voltage (input voltage) of the DC power source Vi, the voltage generated in the primary winding 2a of the transformer T2, and the voltage generated in the winding 2b of the transformer T2.

  The voltage generated in the transformer T2 is A · Vi · D when the on-duty (D = Ton / T) of the switch Tr1 is D. Ton is the on time of the switch Tr1. T is a period for switching the switch Tr1. The output voltage Vo of the smoothing capacitor Co is Vo = Vi (1 + A · D). For this reason, the output voltage Vo can be controlled by varying the on-duty D.

  Next, at time t1, the switch Tr2 is turned off by the gate signal Tr2g from the control circuit 10, and the collector-emitter voltage Tr2v of the switch Tr2 rises. Then, first, a current flows through a path of Vi plus → 2a → Lr2 → D4 → Co → Vi minus. For this reason, the current D4i flows through the diode D4.

  However, the current of the reactor Lr2 is commutated to the diode D3 by the voltage applied to the winding 2b of the transformer T2. For this reason, the current D3i flowing through the diode D3 increases. Along with this, the current D4i of the diode D4 gradually decreases. When the current in the primary winding 2a and the winding 2b of the transformer T2 has finished commutating to the diode D3, the diode D4 is turned off at time t2. Since the current gradually decreases and the diode D4 is turned off, occurrence of recovery loss in the diode D4 is suppressed.

  From time t2 to time t3, the transformer current is completely commutated to the diode D3, and the current is output only through the diode D3.

  At time t3, when the switch Tr2 is turned on by the gate signal Tr2g of the switch Tr2 from the control circuit 10, the current in the primary winding 2a and the winding winding 2b of the transformer T2 is commutated from the diode D3 to the switch Tr2. start.

  At this time, the increase in the current of the switch Tr2 is moderated by the reactor Lr2, and a zero current turn-on operation can be realized. Along with this, the current decrease of the diode D3 also becomes moderate, and the occurrence of recovery loss at turn-off is suppressed.

 The current flows through a path of Vi plus → 2a → Lr 2 → Tr 2 → Vi minus. For this reason, the current i2 flowing through the primary winding 2a of the transformer T2 increases. At the same time, a voltage is also generated in the secondary winding 2c of the transformer T2, and a current L1i flows through the reactor L1 through the path 2c → L1 → 1c → 2c.

  This current L1i flows according to the law of equi-ampere turn of the transformer, energy is accumulated in the reactor L1, and the same current also flows in the secondary winding 1c of the transformer T1. For this reason, a voltage corresponding to the number of turns is induced in the primary winding 1a and the winding winding 1b of the transformer T1.

  Further, when the winding ratio of the transformer T1 is A = (n1 + n3) / n1, the diode D1 has a current 1 / A of the current of the switch Tr2 as follows: Vi plus → 1a → 1b → D1 → Co → Vi minus It flows in the route. The current D1i of the diode D1 flows from time t4 to time t6 when the switch Tr1 is turned on. The output voltage Vo of the smoothing capacitor Co is the sum of the voltage (input voltage) of the DC power source Vi, the voltage generated in the primary winding 1a of the transformer T1, and the voltage generated in the winding winding 1b of the transformer T1.

  The voltage generated in the transformer T1 is A · Vi · D when the on-duty (D = Ton / T) of the switch Tr2 is D. Ton is the ON time of the switch Tr2. T is a period for switching the switch Tr2. The output voltage Vo of the smoothing capacitor Co is Vo = Vi (1 + A · D). For this reason, the output voltage Vo can be controlled by varying the on-duty D.

  Next, at time t4, the switch Tr1 is turned off by the gate signal Tr1g from the control circuit 10, and the collector-emitter voltage Tr1v of the switch Tr1 rises. Then, first, a current flows through a path of Vi plus → 1a → Lr1 → D2 → Co → Vi minus. For this reason, the current D2i flows through the diode D2.

  However, the current of the reactor Lr1 is commutated to the diode D1 by the voltage applied to the winding 1b of the transformer T1. For this reason, the current D1i flowing through the diode D1 increases. Along with this, the current D2i of the diode D2 gradually decreases. When the current of the primary winding 1a and the winding winding 1b of the transformer T1 finishes commutating to the diode D1, the diode D2 is turned off at time t5. Since the current gradually decreases and the diode D2 is turned off, occurrence of recovery loss in the diode D2 is suppressed.

  From time t5 to time t6, the transformer current is completely commutated to the diode D1, and the current is output only through the diode D1.

  When the switch Tr1 is turned on at time t6 (same as time t0), the current in the primary winding 1a and the winding winding 1b of the transformer T1 starts to commutate from the diode D1 to the switch Tr1.

  At this time, the increase in the current of the switch Tr1 is moderated by the reactor Lr1, and a zero current turn-on operation can be realized. Along with this, the decrease in the current of the diode D1 also becomes moderate, and the occurrence of recovery loss at the time of turn-off is suppressed.

  In the first embodiment, since the reactors Lr1 and Lr2 are arranged on the switches Tr1 and Tr2 side, the voltage applied to the windings 1b and 2b of the transformers T1 and T2 is not hindered by the reactors Lr1 and Lr2. Circuit (a circuit in which a reactor La (not shown) is connected between the diode D3 and the winding 5b in FIG. 5 and a reactor Lb (not shown) is connected between the diode D4 and the winding 6b) The problem no longer occurs.

  That is, in the circuit of the first embodiment, a necessary boost ratio can be obtained without increasing the number of windings 1b and 2b of the transformers T1 and T2, and the characteristics of the boost ratio with respect to modulation should be analyzed relatively easily. Therefore, an optimum design can be easily performed with a predetermined duty ratio.

  Further, the inductance values of the reactors Lr1 and Lr2 can be small. Further, the smoothing capacitor Co can be reduced in size.

  FIG. 3 shows waveforms of voltages and currents at various parts of the conventional DC-DC converter. In FIG. 3, a reactor La (not shown) is connected between the diode D3 and the winding winding 5b in FIG. 5, and a reactor Lb (not shown) is connected between the diode D4 and the winding winding 6b. A circuit was used.

  FIG. 4 shows waveforms of voltages and currents at various parts of the DC-DC converter according to the first embodiment. 3 and 4, the reactor current is a current flowing through the reactors La and Lb and the reactors Lr1 and Lr2. The diode current is a current flowing through the diodes D2 and D4. The capacitor ripple voltage is the ripple voltage of the smoothing capacitor Co.

  In contrast to FIG. 3, in FIG. 4, it can be seen that the ripple of the reactor current and the ripple of the capacitor ripple voltage are greatly reduced.

1 is a circuit configuration diagram showing a DC-DC converter of Example 1. FIG. 3 is a timing chart of each part of the DC-DC converter of Example 1. FIG. It is a figure which shows the waveform of the voltage of each part of the conventional DC-DC converter, and an electric current. It is a figure which shows the waveform of the voltage of each part of the DC-DC converter of Example 1, and an electric current. It is a circuit block diagram of the conventional DC-DC converter.

Explanation of symbols

Vi DC power supply Co Smoothing capacitor T1, T2 Transformer Tr1, Tr2 Switch D1-D4 Diode Ro Load resistance L1, Lr1, Lr2 Reactor 1a, 2a Primary winding 1b, 2b Winding winding 1c, 2c Secondary winding 10 Control circuit

Claims (1)

A DC-DC converter that boosts the voltage of a DC power source,
A first switch connected to both ends of the DC power source via a primary winding of a first transformer and a first reactor;
A second switch connected to both ends of the DC power source via a primary winding of a second transformer and a second reactor;
A winding winding, a first diode, and a smoothing capacitor of the first transformer connected to both ends of a series circuit of the first reactor and the first switch and connected in series to the primary winding of the first transformer. A first series circuit comprising:
A second diode connected to a connection point between the first reactor and the first switch and one end of the smoothing capacitor;
A winding winding of the second transformer, a third diode, and the smoothing connected to both ends of a series circuit of the second reactor and the second switch, and connected in series to the primary winding of the second transformer. A second series circuit comprising a capacitor;
A fourth diode connected to a connection point between the second reactor and the second switch and one end of the smoothing capacitor;
A third reactor connected to both ends of a series circuit in which the secondary winding of the first transformer and the secondary winding of the second transformer are connected in series;
The first switch and the second switch are alternately turned on every half cycle, the first switch is turned off during the on period of the second switch, and the second switch is turned on. A control circuit to turn off,
The DC-DC converter characterized by having.

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