JP2009284598A - Controller for alternating-current motors - Google Patents

Abstract

To provide a control device for an AC motor capable of current control with a high response speed even in a pulse mode.
A control device 10 for controlling an inverter 3 to control driving of an AC motor 4, and based on a magnetic flux reference Φ * and a torque reference Trq *, a current reference iq *, id on a dq axis coordinate. * Is calculated, the output current of the inverter 3 is detected, and based on the deviation between the detected torque current iq and the calculated torque current reference iq *, the torque current reference iq * is corrected and the flux weakening control is performed. .
[Selection] Figure 1

Description

  The present invention relates to a control device for an AC motor.

  Generally, in various fields such as electric power, industry, and traffic, a current control method is used for a control device of a voltage source inverter for driving an AC motor.

  Next, an example of the configuration of the speed control device for an induction motor based on the dq axis current proportional integral control method will be described as a current control method for the voltage source inverter.

  First, the three-phase alternating current output from the inverter is detected and converted into a direct current amount on the dq-axis coordinates based on the magnetic flux position of the alternating current motor. Next, the deviation between the dq axis current reference and each detected current is amplified by a proportional integral type current control circuit to obtain a voltage command on the dq axis. Again, the voltage command on the dq axis is converted into the voltage command on the stator stationary coordinate based on the magnetic flux position of the AC motor. The voltage command on the stator coordinates is given to a PWM control circuit such as a triangular wave comparison PWM (Pulse Width Modulation), and the inverter is controlled based on the PWM signal output from the PWM control circuit. By this control, the inverter outputs a voltage proportional to the voltage command for each phase. As a result, the three-phase AC voltage output from the inverter is applied to the electric motor, and a desired current can flow through the electric motor.

  The dq axis current reference is obtained by vector calculation of the magnetic flux reference and the torque reference.

  The magnetic flux reference is obtained by a magnetic flux weakening function generator. The flux weakening function generator is given the speed detected by the strong flux reference and a rotation sensor that detects the position of the rotor of the motor. The magnetic flux weakening function generator outputs the strong magnetic flux reference as it is as the magnetic flux reference at a predetermined speed or less, and outputs the weak magnetic flux reference as a magnetic flux reference that is inversely proportional to the speed at a predetermined speed or higher.

  The torque reference is obtained by amplifying a speed deviation between the speed of the rotor of the electric motor and the speed reference, and adjusting the speed so as to follow the speed reference.

  Since this method performs proportional integral control after converting the current into a DC amount on the dq axis coordinates, it is characterized in that it can be controlled without a steady deviation even with a high frequency AC current of several hundred hertz.

  On the other hand, as a current control method for an electric motor, a method for directly generating a PWM signal (hereinafter referred to as “current tracking PWM method”) so that current detection follows a current reference has been proposed (for example, a patent). Reference 1 and Patent Document 2).

  Next, a control device using a current tracking type PWM method will be described.

  First, the dq-axis current reference is converted to a stator stationary coordinate three-phase current reference. Moreover, the three-phase alternating current output from the inverter is detected. Next, the difference between the three-phase current reference and the three-phase alternating current is applied to a current tracking PWM control circuit. This control circuit generates a PWM signal such that the detected three-phase alternating current follows the three-phase current reference. The control circuit performs on / off control of the switching elements constituting the inverter with the PWM signal.

This method is characterized in that the current response is extremely fast because a PWM signal that directly follows the command value is generated without using a carrier wave.
Japanese Patent No. 3267524 JP 2003-235270 A

  However, the above-described AC motor control device has the following problems.

  There are two problems with the control device based on the proportional-integral control method. The first problem is that the current control response speed is limited by the modulation frequency of the PWM control provided in the subsequent stage of the current control circuit. The second problem is that current control becomes impossible when the voltage command output from the current control circuit exceeds the upper limit of the voltage level that can actually be output.

  The first problem is that the performance of an industrial large-sized drive using a large-current switching element, a train main machine drive, or the like is limited. Since a large current element has a large switching loss, the modulation frequency cannot be increased. For this reason, only a very slow current response can be obtained compared to a small drive. For this reason, the modulation frequency is selected so that both performance and efficiency are satisfied to some extent in consideration of the current response required for the device and the switching loss allowed from the cooling device or the like.

  As a countermeasure against the second problem, since the operation of the device should not be hindered, the magnetic flux is weakened early so that the current can be controlled within the voltage range that the inverter can output in the high-speed region. Is the current situation. However, for this reason, the capacity of the electric motor cannot be output, resulting in a decrease in operating efficiency. That is, in order to control in this way, the output capacity of the electric motor is limited and the efficiency is also deteriorated.

  In the case where the operating frequency range of the motor is wide despite the low modulation frequency, such as a train drive, complicated control of switching the PWM control method according to the operating frequency is also required. Specifically, (1) In a range where the operating frequency of the motor is low, asynchronous PWM control for generating a PWM signal by comparing a triangular carrier wave having a constant modulation frequency and a voltage reference sine wave is performed. (2) When the operating frequency increases and the frequency of the voltage reference sine wave approaches the frequency of the triangular carrier wave, the fluctuation of the fundamental wave component included in the PWM signal increases in the triangular wave with a constant modulation frequency. For this reason, synchronous PWM control is performed in which the voltage reference sine wave and the triangular carrier wave are synchronized, and the frequency of the triangular wave is changed according to the frequency of the voltage reference sine wave to eliminate the voltage fluctuation. (3) Further, if the operating frequency is increased and the number of pulses of the PWM signal included in one cycle of the operating frequency is a region where the number of pulses is extremely small such as 3 pulses or 5 pulses, the efficiency of the motor is affected. PWM control is performed based on a pulse pattern that intensively removes low-order harmonics such as large fifth and seventh orders. Thus, the control device is switched according to the operating frequency.

  If the current control is fast, the fluctuation of the current caused by the low-order harmonic voltage caused by the PWM control and the voltage fluctuation caused by the frequency difference between the carrier wave and the voltage reference can be suppressed by the control. However, it is difficult to suppress by current control at a response speed obtained by combined control of dq-axis current control and PWM control as in this proportional integral control method. For this reason, the PWM control is switched so that a PWM signal that causes undesirable harmonic voltage and voltage fluctuation is not output from the beginning. However, switching of the PWM control itself causes another problem. The pulse pattern changes by switching the PWM control. It is inevitable that the output voltage varies depending on the pulse pattern. For this reason, depending on the voltage change at the time of control switching, an abrupt change in current, a torque fluctuation, an excessive current protection operation, or the like may occur. In order to avoid these, it is necessary to perform transitional switching control such as selecting and switching a phase that does not cause a sudden current change, and narrowing the current reference at the time of switching. This switching control complicates the adjustment.

  On the other hand, the current tracking type PWM method can obtain a very fast current response even at a low modulation frequency. Further, the PWM waveform is automatically and continuously switched depending on the operation frequency. For this reason, intentional PWM control switching like the above-described proportional-integral control is not required. In addition, it does not fall out of control at high speeds.

  However, there is a steady error in principle in the current tracking type PWM method. In the current tracking type PWM, the current reference on the stator stationary coordinate is compared with the current detection, and the PWM signal is generated only by the magnitude relationship, so the proportional gain is infinite. For this reason, since the switching frequency becomes too high when operated as it is, it is necessary to limit the switching frequency by providing a dead zone due to hysteresis and a delay time due to a timer. A stationary error occurs due to the dead zone and the delay time. The lower the switching frequency, the larger the steady-state error and the greater the influence on the motor control performance.

  Next, the 1 pulse mode will be described.

  The 1-pulse mode is an inverter control mode that outputs a square wave voltage by turning on the positive side element of the inverter for 180 degrees electrical angle and turning on the negative side element for the remaining 180 degrees without PWM control at all. It is. This one-pulse mode is a control mode that is performed when it is desired to increase the output voltage of the PWM inverter as much as possible because it does not have to be a sine wave.

  By increasing the output voltage of the inverter, the constant magnetic flux region can be expanded to a high rotational speed. For this reason, the output capacity of the electric motor can be improved even in a combination of the same electric motor and inverter. Further, the number of ways of weakening the magnetic flux in the magnetic flux weakening region can be reduced. Since the generated torque of the electric motor is proportional to the product of the torque current and the magnetic flux component current, if the magnetic flux is weakened, the ratio of the current flowing through the electric motor to the torque decreases. Less weakening means that less current is required to generate the same torque, which means that efficiency can be improved. For this reason, the control is performed in the train drive.

  However, in the configuration based on the above-described proportional-integral control method, only sine wave PWM control can be performed, and thus control in one pulse mode cannot be performed.

  For example, in a train drive, when control in the 1-pulse mode is performed, the current is indirectly controlled by switching to another control method such as phase control instead of current control. However, the PWM waveform and the one-pulse waveform cannot be switched simply because there is a great leap in the fundamental wave and low-order harmonics of the voltage. For this reason, complicated switching control is performed. In addition, the phase control cannot control the current as fast as the current control.

  Further, in the above-described current tracking type PWM method, there are the following problems when the control in the 1 pulse mode is performed.

  In this current follow-up type PWM system, if attention is paid only to the aspect of PWM control, control does not become impossible even in a high-speed region, so that it is possible to shift to one-pulse operation continuously. However, the current deviation is within the allowable error in the sine wave PWM possible area, but the deviation increases when the sine wave PWM possible area is exceeded. As the deviation increases, the voltage waveform approaches one pulse.

  In a high-speed region where sine wave PWM is not possible, the steady-state deviation increases as current control is difficult. Since current control is not impossible as in the above-described proportional-integral control method, if the current reference is changed, the current also changes accordingly. Therefore, the current control is established with a very large steady-state deviation. Since an error is generated between the magnetic flux reference and the magnetic flux and the torque reference and the torque due to the steady deviation, if the steady deviation is too large, the motor control is not established.

  Therefore, an object of the present invention is to provide a control device for an AC motor that can perform current control with a fast response speed even in the 1-pulse mode.

  An AC motor control device according to an aspect of the present invention is an AC motor control device that controls an inverter by controlling an inverter, the magnetic flux reference for controlling the magnetic flux of the AC motor, and the Based on the torque reference for controlling the torque of the AC motor, the torque current reference for controlling the torque on the dq axis coordinate of the current flowing to the AC motor and the dq axis coordinate of the current flowing to the AC motor Vector calculation means for calculating a magnetic flux component current reference for controlling the magnetic flux component of the current, three-phase alternating current detection means for detecting the three-phase alternating current output from the inverter, and detection by the three-phase alternating current detection means Detection current coordinate conversion means for converting the three-phase AC current thus converted into a torque component current that is a torque component on the dq axis coordinate and a magnetic flux component current that is a magnetic flux component on the dq axis coordinate; Torque component current deviation calculating means for calculating a torque component current deviation which is a deviation between the torque component current coordinate-converted by the detected current coordinate converter and the torque component current reference calculated by the vector calculator; Based on the torque component current deviation calculated by the torque component current deviation calculator, a torque component current reference for calculating a torque component current reference correction amount that is a correction amount of the torque component current reference calculated by the vector calculator Torque component current reference correction amount for limiting the torque component current reference correction amount calculated by the correction amount calculation means and the torque component current reference correction amount calculation unit to a level value or less that starts the weakening of the magnetic flux of the AC motor. Based on the torque component current reference correction amount restricted by the limiting device and the torque component current reference correction amount restriction device, Torque component current reference correction means for correcting the torque component current reference calculated by the torque calculation means, and correction for weakening the magnetic flux reference based on the torque component current deviation calculated by the torque component current deviation calculation means. Magnetic flux reference weakening correction means, and each of the three-phase alternating currents output from the inverter, the magnetic flux current reference calculated by the vector calculation means and the torque current reference corrected by the torque current reference correction means. A current reference coordinate conversion means for converting coordinates to a phase current reference, a current reference for each phase coordinate-converted by the current reference coordinate conversion means, and a three-phase alternating current detected by the three-phase alternating current detection means Based on the three-phase AC current deviation calculation means for calculating the deviation for each phase, and the deviation calculated for each phase by the three-phase AC current deviation calculation means And an inverter control means for controlling the inverter.

  ADVANTAGE OF THE INVENTION According to this invention, the control apparatus of the alternating current motor which can perform electric current control with a quick response speed also in 1 pulse mode can be provided.

  Embodiments of the present invention will be described below with reference to the drawings.

(First embodiment)
FIG. 1 is a configuration diagram showing a configuration of an electric motor drive system 1 according to a first embodiment of the present invention. In addition, the same code | symbol is attached | subjected to the same part in drawing, the detailed description is abbreviate | omitted, and a different part is mainly described. In the following embodiments, the same description is omitted.

  The electric motor drive system 1 includes a DC power source PW, a smoothing capacitor CN connected in parallel with the DC power source PW, an electric motor 4, an inverter 3 that supplies electric power for driving the electric motor 4, and a control that controls the inverter 3. The device 10 includes halls CT5U, 5V, 5W provided in each phase for detecting a three-phase alternating current output from the inverter 3, and a rotation sensor 6 provided in the electric motor 4.

  The DC power source PW is a power source that supplies driving power to the electric motor 4.

  Smoothing capacitor CN smoothes the output voltage of DC power supply PW.

  The inverter 3 converts the DC power supplied from the DC power source PW into three-phase AC power and supplies it to the electric motor 4. The inverter 3 includes three inverter circuits including two switching elements SW. Switching of the switching element SW is controlled by a control signal from the control device 10. By this control, the three-phase AC power output from the inverter 3 is controlled.

  The electric motor 4 is an induction motor that is driven by three-phase AC power supplied from the inverter 3.

  Hall CT (current transformer) 5U, 5V, 5W is a current detector which detects an electric current using a Hall element. The halls CT5U, 5V, 5W are provided for measuring the motor current by the control device 10. Halls CT5U, 5V, 5W are provided in each phase of the three-phase alternating current output from the inverter 3. Hall CT5U, 5V, 5W detects the current which flows into the phase corresponding to each of U phase of the three-phase alternating current output from inverter 3, V phase, and W phase, and outputs an output signal to control device 10.

  The rotation sensor 6 is a sensor that detects the position of the rotor of the electric motor 4. The rotation sensor 6 outputs an output signal to the control device 10 according to the position of the rotor of the electric motor 4.

  The control device 10 controls the drive of the electric motor 4 by controlling the inverter 3 based on the output signals output from the hall CT 5U, 5V, 5W and the rotation sensor 6. Therefore, the control device 10 is also a control device for the inverter 3.

  The control device 10 includes a rotation detection circuit 7, a speed control circuit 9, a vector calculation circuit 11, a current detection circuit 13, coordinate conversion circuits 14 and 19, a current tracking type PWM control circuit 21, and d-axis side correction. Control circuit 22d, q-axis side correction control circuit 22q, magnetic flux weakening control circuit 26, limiters 23 and 27, absolute value circuit 29, adders 12, 24d and 24q, and subtractors 8, 15d, 15q, 25, 28, 20U, 20V, 20W.

  The rotation detection circuit 7 receives an output signal from the rotation sensor 6. The rotation detection circuit 7 calculates a rotor position signal (electrical angle signal) θr and a speed detection signal ωr corresponding to the position of the rotor of the electric motor 4 based on the output signal input from the rotation sensor 6. The rotation detection circuit 7 outputs the calculated rotor position signal θr to the adder 12. The rotation detection circuit 7 outputs the calculated speed detection signal ωr to the subtracter 8.

  The subtracter 8 calculates a speed deviation which is a difference between the speed detection signal ωr output from the rotation detection circuit 7 and the speed reference ωr *. The subtracter 8 outputs the calculated speed deviation as a signal to the speed control circuit 9.

  The speed control circuit 9 amplifies the speed deviation signal input from the subtracter 8 and adjusts the torque reference Trq * so that the speed detection signal ωr follows the speed reference ωr *. The speed control circuit 9 outputs the adjusted torque reference Trq * to the vector calculation circuit 11.

  The vector calculation circuit 11 is based on a magnetic flux reference Φ * input from a subtractor 28, which will be described later, and a torque reference Trq * input from the speed control circuit 9, and a magnetic flux current reference id *, a torque shared current reference iq *, And the slip angle θs is calculated. The vector calculation circuit 11 outputs the calculated magnetic flux component current reference id * to the subtracter 15d and the adder 24d. The vector calculation circuit 11 outputs the calculated torque current reference iq * to the subtracter 15q and the adder 24q. The vector calculation circuit 11 outputs the calculated slip angle θs to the adder 12.

  The adder 12 adds the rotor position signal θr input from the rotation detection circuit 7 and the slip angle θs input from the vector calculation circuit 11 to calculate the magnetic flux position signal θ0. The adder 12 outputs the calculated magnetic flux position signal θ0 to the coordinate conversion circuits 14 and 19, respectively.

  The current detection circuit 13 calculates the current detection signals iu, iv, iw by matching the output signals input from the hall CTs 5U, 5V, 5W with the scaling in the control circuit. The current detection signals iu, iv, and iw are signals corresponding to currents flowing in the U phase, the V phase, and the W phase, respectively. The current detection circuit 13 outputs the calculated current detection signals iu, iv, iw to the coordinate conversion circuit 14. The current detection circuit 13 outputs the current detection signals iu, iv, and iw to the corresponding phase adders 20U, 20V, and 20W, respectively.

  The coordinate conversion circuit 14 uses the magnetic flux position signal θ0 input from the adder 12 to convert the current detection signals iu, iv, iw input from the current detection circuit 13 on the dq axis coordinate in synchronization with the magnetic flux of the motor 4. Is converted into a magnetic flux component current detection signal id and a torque component current detection signal iq. The coordinate conversion circuit 14 outputs the converted magnetic flux component current detection signal id to the subtractor 15d. The coordinate conversion circuit 14 outputs the converted torque component current detection signal iq to the subtractor 15q.

  The subtractor 15d calculates the magnetic flux component current deviation by taking the difference between the magnetic flux component current reference id * input from the vector calculation circuit 11 and the magnetic flux component current detection signal id input from the coordinate conversion circuit 14. The subtractor 15d outputs the calculated magnetic flux component current deviation to the d-axis side correction control circuit 22d.

  The subtractor 15q calculates the torque current deviation by taking the difference between the torque current reference iq * input from the vector calculation circuit 11 and the torque current detection signal iq input from the coordinate conversion circuit 14. The subtractor 15q outputs the calculated torque current deviation to the q-axis side correction control circuit 22q.

  The d-axis side correction control circuit 22d amplifies the magnetic flux current deviation input from the subtractor 15d, and calculates a correction signal idc * for correcting the steady deviation. The d-axis side correction control circuit 22d outputs the calculated correction signal idc * to the adder 24d.

  The q-axis side correction control circuit 22q amplifies the torque current deviation input from the subtractor 15q and calculates a correction signal iqc * for correcting the steady deviation. The q-axis side correction control circuit 22q outputs the calculated correction signal iqc * to the limiter 23 and the absolute value circuit 29.

  The limiter 23 is an upper / lower limiter that limits the upper and lower limits of the correction signal iqc * output from the q-axis side correction control circuit 22q. The limiter 23 sets the upper limit so that the correction signal iqc * is weakened and becomes equal to or less than the control start level iqcLIM *. Therefore, the upper limit value is set equal to or higher than the weakening control start level iqcLIM *. The limiter 23 outputs the limited correction signal iqc * to the adder 24q.

  The adder 24d adds the magnetic flux component current reference id * input from the vector calculation circuit 11 and the correction signal idc * input from the d-axis side correction control circuit 22d to calculate the magnetic flux correction current reference id **. To do. The adder 24 d outputs the calculated magnetic flux component correction current reference id ** to the coordinate conversion circuit 19.

  The adder 24q adds the torque component current reference iq * input from the vector calculation circuit 11 and the correction signal iqc * input from the limiter 23 to calculate the torque correction current reference iq **. The adder 24q outputs the calculated torque correction current reference iq ** to the coordinate conversion circuit 19.

  The coordinate conversion circuit 19 uses the magnetic flux position signal θ0 input from the adder 12, and uses the magnetic flux correction current reference id ** input from the adder 24d and the torque correction current reference iq input from the adder 24q. ** is converted into a three-phase current reference iu *, iv *, iw * in the stator stationary coordinate system. The coordinate conversion circuit 19 outputs the converted three-phase current references iu *, iv *, iw * to the subtracters 20U, 20V, 20W corresponding to the respective phases.

  The subtracter 20 </ b> U calculates a deviation between the U-phase current reference iu * input from the coordinate conversion circuit 19 and the current detection signal iu input from the current detection circuit 13. The subtractor 20 </ b> U outputs the calculated U-phase deviation to the current tracking type PWM control circuit 21. Similarly, the subtracters 20V and 20W calculate the deviations of the V phase and the W phase, respectively, and output them to the current tracking type PWM control circuit 21.

  The current tracking type PWM control circuit 21 generates a PWM signal such that the current detection signals iu, iv, iw track the three-phase current references iu *, iv *, iw *. The current follow-up type PWM control circuit 21 outputs the generated PWM signal and controls the switching element SW of the inverter 3 on and off. A desired current is caused to flow from the inverter 3 to the motor 4 by the control by the current tracking type PWM control circuit 21.

  The absolute value circuit 29 calculates the absolute value of the correction signal iqc * output from the q-axis side correction control circuit 22q. The absolute value circuit 29 outputs the calculated absolute value of the correction signal iqc to the subtracter 25.

  The subtractor 25 calculates the difference between the absolute value of the correction signal iqc * output from the absolute value circuit 29 and the start level value iqcLIM *. The subtractor 25 outputs the difference between the calculation results to the magnetic flux weakening control circuit 26.

  The magnetic flux weakening control circuit 26 amplifies the signal input from the subtractor 25. The magnetic flux weakening control circuit 26 outputs the amplified signal to the subtracter 28 via the limiter 27 as a signal for weakening the magnetic flux.

  The limiter 27 limits the lower limit of the signal output from the magnetic flux weakening control circuit 26 to zero.

  The subtractor 28 is input with a magnetic flux command Φ ** that strengthens the magnetic flux. The subtracter 28 subtracts the signal output from the magnetic flux weakening control circuit 26 via the limiter 27 from the input strong magnetic flux command Φ **. The subtracter 28 outputs the value obtained by the subtraction to the vector calculation circuit 11 as the magnetic flux reference Φ *. Thereby, the magnetic flux reference Φ * becomes a magnetic flux command according to the state of the electric motor 4.

  Next, the operation of the electric motor drive system 1 will be described.

  The current detection circuit 13 detects the three-phase alternating current output from the inverter 3 as current detection values iu, iv, iw.

  The coordinate conversion circuit 14 converts the current detection values iu, iv, iw detected by the current detection circuit 13 into current detection values id, iq in quantities on the dq axis coordinates.

  The subtractors 15 d and 15 q obtain deviations between the current detection values id and iq converted by the coordinate conversion circuit 14 and the current reference id * and iq * on the dq axis coordinates output from the vector calculation circuit 11.

  The correction control circuits 22d and 22q amplify the deviations of the dq axes obtained by the subtracters 15d and 15q.

  The adder 24d adds the correction signal idc *, which is the output of the d-axis side correction control circuit 22d, and the magnetic flux component current reference id * to obtain the magnetic flux component correction current reference id **.

  An excessive portion of the correction signal iqc *, which is the output of the q-axis side positive control circuit 22q, is cut by the limiter 23.

  The adder 24q adds the correction signal iqc * limited by the limiter 23 and the torque component current reference iq * to obtain a torque component correction current reference iq **.

  The coordinate conversion circuit 19 converts the correction current references id ** and iq ** into the three-phase current references iu *, iv * and iw * on the stator stationary coordinates.

  Differences between the three-phase current references iu *, iv *, iw * converted by the coordinate conversion circuit 19 and the current detection values iu, iv, iw detected by the current detection circuit 13 are input to the current tracking type PWM control circuit 21. Is done.

  The current tracking type PWM control circuit 21 controls the current output from the inverter 3 based on the inputted difference.

  Next, the effect of the electric motor drive system 1 will be described.

  If the current detection id, iq is smaller than the current reference id *, iq *, the correction control circuits 22d, 2q increase the values of the correction signals idc *, iqc * to be output. As a result, the correction current references id ** and iq ** input to the current tracking type PWM control circuit 21 increase. For this reason, the current follow-up type PWM control circuit 21 increases the motor currents id and iq flowing in the motor 4, so that the difference from the original current reference id * and iq * becomes small.

  By providing the correction control circuits 22d and 22q with integral elements, even if the deviation output by the subtracters 15d and 15q is very small, it is integrated to correct the correction current references id ** and iq **. Any steady deviation of the d-axis and q-axis can be made zero.

  In this manner, current control without steady deviation can be performed in the middle / low speed region of the motor 4.

  In the middle / low speed region, the correction signal iqc * output from the q-axis side correction control circuit 22q is very small. For this reason, the output signal of the flux weakening control circuit 26 is negative. This output signal is limited to the lower limit value 0 by the limiter 27. Therefore, as the magnetic flux command Φ * given to the vector calculation circuit 11, the strong magnetic flux command Φ ** is given as it is.

  When the rotation speed of the electric motor 4 increases and the current cannot flow, the correction signal iqc of the q-axis side correction control circuit 22q increases rapidly. When the correction signal iqc of the q-axis side correction control circuit 22q exceeds the weakening start level iqcLIM *, the output of the subtracter 25 turns positive. As a result, the output of the flux weakening control circuit 26 begins to increase. The magnetic flux reference Φ * input to the vector arithmetic circuit 11 is obtained by subtracting the output signal of the magnetic flux weakening control circuit 26 from the strong magnetic flux command Φ **. Therefore, when the output of the flux weakening control circuit 26 increases, the magnetic flux reference id * output from the vector calculation circuit 11 decreases. Therefore, the magnetic flux Φ of the electric motor 4 is reduced. This limits the increase in induced voltage. The correction signal iqc * of the q-axis side correction control circuit 22q is controlled to the weakening start level iqcLIM *.

  Here, the role of the limiter 23 will be described.

  The correction signal iqc * of the q-axis side correction control circuit 22q is added to the current reference iq * via the limiter 23. If this limiter 23 is not provided, the magnetic flux weakening may not be in time and control may be lost.

  For example, when a large-amplitude power running torque command (torque reference Trq *) is given, the correction signal iqc * becomes a very large signal transiently. As a result, the flux weakening control operates, and the d-axis side current reference id * is narrowed down. For this reason, the d-axis side current reference id * is smaller than the magnetic flux component current detection signal id. Thereby, the correction signal idc * is swung to the negative side. On the other hand, the correction signal iqc * swings to the positive side. Since this is a step change, the q-axis side current deviation is large. Further, the increase rate of the correction signal iqc * is fast.

  In the 1 pulse region, the q-axis current cannot flow unless the magnetic flux is weakened. For this reason, the correction signal iqc * further increases to the positive side. On the other hand, since the magnetic flux current reference id * is narrowed, the correction signal idc * increases to the negative side.

  Based on the current references id * and iq * corrected by these correction signals, the current follow-up type PWM control circuit 21 tries to control the current on both the d-axis side and the q-axis side without steady deviation. The current follow-up type PWM control circuit 21 tries to perform such control, and on the contrary, the d-axis side and the q-axis side cannot flow currents as standard.

  When the limiter 23 is inserted, the current follow-up type PWM control circuit 21 controls the d-axis side current so as to eliminate a steady deviation. The q-axis side limited by the limit 23 is controlled with a steady deviation. Accordingly, since the d-axis side current is controlled with priority, the d-axis side current easily follows the current reference id *.

  Since the correction signal iqc * input to the magnetic flux weakening control circuit 26 is not limited by the limiter 23, while the correction signal iqc * exceeds the weakening control start level iqcLIM *, the control device 10 Keep weakening. Eventually, the magnetic flux is sufficiently weakened, and the correction signal iqc * is controlled to be constant at the weakening control start level iqcLIM *.

  By setting the upper limit of the limiter 23 to be equal to or higher than the weakening control start level iqcLIM *, when the correction signal iqc * is constantly controlled to the weakening control start level iqcLIM *, the correction signal iqc * is The limiter 23 is not applied. This indicates that the torque component current iq follows the torque component current reference iq *.

  In this way, the control device 10 performs the minimum weakening control that can control the torque component current iq equal to the torque component current reference iq *. The control apparatus 10 maintains the torque component current iq so as to be controllable by performing control to further weaken the magnetic flux as the rotational speed increases.

  According to the present embodiment, the deviation (id **-id), (iq **-) between the corrected current reference id **, iq ** and the current detection signal id, iq input to the current tracking type PWM control circuit 21. By holding the steady portion of iq) by the integration elements of the correction control circuits 22d and 22q, a current equal to the current reference id * and iq * output from the vector calculation circuit 11 can be passed. Thereby, highly accurate current control can be performed using current tracking type PWM control excellent in current response. Therefore, the control device 10 can perform high-performance vector control excellent in both accuracy and response.

  Further, by providing the magnetic flux weakening control circuit 26, the magnetic flux of the electric motor 4 can be weakened to a minimum level that allows the q-axis current to flow. For example, when a combination method of PI control type (proportional integration type) dq-axis current control and triangular wave comparison PWM control is adopted, the q-axis current output (voltage reference) exceeds the q-axis voltage that the inverter 3 can output. The magnetic flux must be weakened quickly so that nothing happens. In the control according to the present embodiment, an increase in the steady deviation is detected by current control to weaken the magnetic flux. That is, the magnetic flux is weakened by detecting that the voltage for current control is insufficient. Thereby, the inverter 3 can output a complete one-pulse voltage. That is, the control device 10 can control the inverter 3 in the 1 pulse mode.

  Here, with reference to FIG. 2 and FIG. 3, the performance of the control in the one-pulse mode will be described.

  FIG. 2 is a graph showing a correlation between the inverter output voltage and the rotation speed in the control in the one-pulse mode of the control device 10 according to the present embodiment. FIG. 3 is a graph showing the correlation between the magnetic flux and the rotation speed in the control in the one-pulse mode of the control device 10 according to the present embodiment.

  2 and 3 show a comparison between the control in the 1-pulse mode and the sine wave PWM control for the sake of explanation. The sine wave PWM control here is control for performing sine wave PWM control based on, for example, a voltage command based on proportional-integral current control.

  The rotational speeds r1, r2, and r3 shown in FIG. 2 and FIG. 3 are magnetic fluxes at the theoretical limit at the time of control in the 1 pulse mode, at the sine wave PWM control, and at the practical limit at the sine wave PWM control, respectively. The rotation speed at which the weakening control is started is shown.

The maximum value (instantaneous value) of the line voltage that can be output by the voltage source inverter by sine wave PWM control is ± Edc when the DC voltage of the inverter is Edc. Therefore, as a sine wave voltage,
± Edc · sinθ
Is the maximum.

  However, if the voltage command by proportional-integral current control exceeds the outputtable voltage, control becomes impossible. In order to avoid such a situation, in the sine wave PWM control, as shown in a graph attached with “practical limit at the time of sine wave PWM” shown in FIG. Current controller saturation level). The practical limit at the time of sine wave PWM is used at 95% of the output possible voltage Edc by sine wave PWM control, for example. At this time, the magnetic flux command is a graph with “sine wave PWM practical limit” shown in FIG.

On the other hand, the magnitude of the fundamental wave component of the output line voltage of the inverter controlled in 1 pulse mode is

It becomes. The magnitude of the amplitude at this time (a graph labeled “during one pulse operation” shown in FIG. 2) is about 1.103 · Edc. This amplitude is about 10% larger than the theoretical limit by the sine wave PWM control (the graph labeled “theoretical limit during sine wave PWM” shown in FIG. 2).

  Therefore, by performing the control in the 1-pulse mode, the voltage can be increased by about 15% compared to the case where the sine wave PWM control is used at 95% of the output possible voltage Edc. As a result, as shown in FIG. 3, in the control in the 1-pulse mode, the constant magnetic flux region is expanded to about 15% higher than the sine wave PWM control (that is, the start of the flux weakening control is delayed). be able to. That is, by performing the control in the 1-pulse mode, the motor output capacity can be improved by about 15% with the completely same motor 4 and inverter 3 as compared with the sine wave PWM control. Further, by controlling in the 1-pulse mode, the way of weakening the magnetic flux in the magnetic flux weakening region can be reduced.

  Furthermore, since the control device 10 performs control by the current tracking type PWM control method, the current response can be speeded up regardless of the switching frequency. Thus, if the control device 10 is applied to an industrial large-sized drive or a train main machine drive that uses a large-current switching element, the performance can be dramatically improved.

  Further, the control device 10 can shift the control to the one-pulse mode without switching the PWM control or switching the current control to the phase control. This eliminates the need for complicated control switching, thereby simplifying the control. Therefore, the control apparatus 10 can be easily adjusted and can improve reliability. Further, the control device 10 can compensate for an increase in the steady-state deviation even in the high-speed region, and can pass a current according to the reference to the motor 4. Therefore, since current control is performed even in the one-pulse mode region, current control can be performed at a speed higher than that of phase control, for example.

(Second Embodiment)
FIG. 4 is a configuration diagram showing the configuration of an electric motor drive system 1A according to the second embodiment of the present invention.

  In the electric motor drive system 1 according to the first embodiment shown in FIG. 1, the electric motor drive system 1A replaces the control device 10 with a control device 10A. The control device 10A is the same as the control device 10 except that the multiplication circuits 31d and 31t, the subtracter 32, and the square root circuit 33 are added, and the adder 24q, the subtracter 25, and the limiter 23 are replaced with the adder 24qA, the subtracter 25A, and the limiter 23A, respectively. In this configuration, the absolute value circuit 29 is removed. Other points are the same as those of the electric motor drive system 1.

  The multiplication circuit 31d receives two identical correction signals idc * from the d-axis side correction control circuit 22d. The multiplier circuit 31d multiplies the two input correction signals idc *. That is, the multiplication circuit 31d calculates the square of the correction signal idc *. The multiplication circuit 31d outputs the calculation result to the adder 32.

  The multiplier circuit 31t receives two same limit values iqC *. The multiplication circuit 31q multiplies the two input correction signals iqC *. That is, the multiplication circuit 31t calculates the square of the correction signal iqC *. The multiplier circuit 31q outputs the calculation result to the adder 32.

  The subtracter 32 subtracts the calculation result input from the multiplication circuit 31d from the calculation result input from the multiplication circuit 31t. The subtractor 32 outputs the calculation result to the square root circuit 33.

  The square root circuit 33 calculates the square root of the calculation result input from the subtracter 32. The square root circuit 33 outputs the calculation result to the subtracter 25A and the limiter 23A.

  The subtractor 25A calculates the difference between the calculation result output from the square root circuit 33 and the start level value iqcLIM *. The subtractor 25 outputs the difference between the calculation results to the magnetic flux weakening control circuit 26.

  The magnetic flux weakening control circuit 26 performs magnetic flux weakening control based on the calculation result input from the subtractor 25A.

  The limiter 23 </ b> A is an upper / lower limiter that limits the upper limit and the lower limit of the calculation result output from the square root circuit 33. The upper limit value of the limiter 23A is given from the square root circuit 33. The limiter 23A limits the output signal from the square root circuit 33 with a limit value, and outputs the limit signal to the adder 24qA as a correction signal.

  Here, how to obtain the limit value of the limiter 23A will be described.

The steady deviation correction must be applied with priority on the d-axis side. Therefore, when the limit value is iqc * max, the limit value is obtained by the following equation.

  Here, the set value iC * is a set value of the limit of the vector sum of the q-axis side correction signal iqc * and the d-axis side correction signal idc *.

  In this way, the limit value iqc * max of the q-axis side correction signal is obtained from the set value iC * and the d-axis side correction signal idc *.

  The adder 24qA adds the torque component current reference iq * input from the vector calculation circuit 11 and the correction signal input from the limiter 23A, and calculates the torque correction current reference iq **. The adder 24qA outputs the calculated torque correction current reference iq ** to the coordinate conversion circuit 19.

  As described above, the control device 10A calculates the limit value of the q-axis side correction signal by the configuration of the multiplication circuits 31d and 31q, the adder 32, and the square root circuit 33.

  The control device 10A performs weakening control similar to that in the first embodiment based on the calculated vector length for correction. Further, the control device 10A corrects the torque current reference iq * based on the calculated correction vector length.

  According to the present embodiment, in addition to the operational effects of the first embodiment, the following operational effects can be obtained.

  The control device 10A feeds back the vector length of the correction composed of the torque correction signal iqc * and the magnetic flux correction signal idc * to the magnetic flux weakening control circuit 26. For this reason, when the q-axis current command (torque component current reference iq *) increases stepwise, the magnetic flux command (magnetic flux reference Φ *) is more rapidly narrowed down. Therefore, it can be expected that the rise of the q-axis current in one pulse region is accelerated.

  The control device 10A also limits the correction given to the current tracking type PWM control circuit 21 by the vector lengths of idc * and iqc *. If the limit value of the torque correction signal iqc * is determined without considering the value of the magnetic flux correction signal idc * as in the first embodiment, a margin on the d-axis side must be taken into account. On the other hand, by limiting as a vector, it is not necessary to see a margin on the d-axis side. Therefore, the upper limit value for limiting the q-axis side correction signal iqc * can be made larger than in the first embodiment. As a result, the rise of the q-axis current in one pulse region can be accelerated.

(Third embodiment)
FIG. 5 is a configuration diagram showing a configuration of an electric motor drive system 1B according to the third embodiment of the present invention.

  In the electric motor drive system 1 according to the first embodiment shown in FIG. 1, the electric motor drive system 1B replaces the control device 10 with a control device 10B. The control device 10B has a configuration in which, in the control device 10, the subtracter 28 is replaced with a subtractor 28B, and a first-order lag circuit 41 is added. Other points are the same as those of the electric motor drive system 1.

  The subtractor 28B receives a strong d-axis current command idA ** that strengthens the magnetic flux. The subtractor 28 </ b> B receives an output signal from the magnetic flux weakening control circuit 26 via the limiter 27. The subtractor 28B subtracts the output signal from the magnetic flux weakening control circuit 26 from the strong d-axis current command idA ** to obtain a d-axis side current reference id *. The subtractor 28B outputs the calculated d-axis side current reference id * to the primary delay circuit 41, the subtractor 15d, and the adder 24d.

  The primary delay circuit 41 is a primary delay circuit having a motor constant of the AC motor 4. When the d-axis side current reference id * passes through the primary delay circuit 41, the magnetic flux reference Φ * is obtained. The primary delay circuit 41 outputs the calculated magnetic flux reference Φ * to the vector calculation circuit 11.

  According to the present embodiment, the d-axis side current reference id * can be obtained by subtracting the output signal of the magnetic flux weakening control circuit 26 from the stronger d-axis current command idA **. Further, by passing the d-axis side current reference id * through the first-order lag circuit 41, the stronger magnetic flux reference Φ * can be obtained.

  Therefore, by having the strong d-axis current command idA ** without having the strong magnetic flux command Φ **, it is possible to obtain the same effects as those of the first embodiment.

  In the first embodiment and the third embodiment, the absolute value circuit 29 is provided. In the control method of the present invention, when the induced voltage is used in the increase stage of the current deviation, the positional relationship between the steady deviation and the induced voltage is fixed, and the q-axis current (current detection value iq) flowing through the AC motor 4 is It is always smaller than the command value (q-axis side current reference iq *). For this reason, the output of the q-axis side correction control circuit 22q comes only in the positive direction. Therefore, since the output of the q-axis side correction control circuit 22q does not become negative, there is no adverse effect on the control by inserting the absolute value circuit 29. On the other hand, if the correction signal iqc * is output with a negative sign for some unexpected reason, the magnetic flux weakening control is not performed and the current deviation increases without the absolute value circuit 29. In such a case, by providing the absolute value circuit 29, such a situation can be avoided.

  In the first embodiment and the third embodiment, the d-axis correction control circuit 24d may be omitted. Most of the steady deviation appears on the q-axis side. If the correction control circuit 24d is omitted, a slight steady error appears. Therefore, when the switching frequency is high and the steady deviation is originally small, the control devices 10 and 10A may have a configuration in which the d-axis side correction control circuit 24d is omitted.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying the constituent elements without departing from the scope of the invention in the implementation stage. Moreover, various inventions can be formed by appropriately combining a plurality of constituent elements disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, constituent elements over different embodiments may be appropriately combined.

1 is a configuration diagram showing a configuration of an electric motor drive system according to a first embodiment of the present invention. The graph which shows the correlation of the inverter output voltage and rotation speed in control by 1 pulse mode of the control apparatus which concerns on 1st Embodiment. The graph which shows the correlation of the magnetic flux and rotation speed in control by 1 pulse mode of the control apparatus which concerns on 1st Embodiment. The block diagram which shows the structure of the electric motor drive system which concerns on the 2nd Embodiment of this invention. The block diagram which shows the structure of the electric motor drive system which concerns on the 3rd Embodiment of this invention.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Electric motor drive system, 3 ... Inverter, 4 ... Electric motor, 5U, 5V, 5W ... Hall CT, 6 ... Rotation sensor, 7 ... Rotation detection circuit, 8, 15d, 15q, 25, 28, 20U, 20V, 20W ... Subtractor, 9 ... speed control circuit, 11 ... vector operation circuit, 12, 24d, 24q ... adder, 13 ... current detection circuit, 14, 19 ... coordinate conversion circuit, 21 ... current tracking PWM control circuit, 22d ... d Axis side correction control circuit, 22q... Q axis side correction control circuit, 23, 27 ... limiter, 26 ... magnetic flux weakening control circuit, 29 ... absolute value circuit, PW ... DC power supply, CN ... smoothing capacitor.

Claims (10)

An AC motor control device that controls the inverter and controls the drive of the AC motor,
Based on a magnetic flux reference for controlling the magnetic flux of the AC motor and a torque reference for controlling the torque of the AC motor, a torque component for controlling a torque component on the dq axis coordinate of the current flowing to the AC motor. Vector calculation means for calculating a magnetic flux component current reference for controlling a magnetic flux component on a dq axis coordinate of a current flowing through the AC motor and the current reference;
Three-phase alternating current detection means for detecting the three-phase alternating current output from the inverter;
Detection current coordinate conversion for converting the three-phase alternating current detected by the three-phase alternating current detection means into a torque component current that is a torque component on the dq axis coordinate and a magnetic flux component current that is a magnetic flux component on the dq axis coordinate Means,
Torque component current deviation calculating means for calculating a torque component current deviation that is a deviation between the torque component current coordinate-converted by the detected current coordinate converter and the torque component current reference calculated by the vector calculator;
Based on the torque component current deviation calculated by the torque component current deviation calculator, a torque component current for calculating a torque component current reference correction amount that is a correction amount of the torque component current reference calculated by the vector calculator A reference correction amount calculating means;
Torque component current reference correction amount limiting means for limiting the torque component current reference correction amount calculated by the torque component current reference correction amount calculation unit to a level value or less for starting the weakening of the magnetic flux of the AC motor;
A torque component current reference correction unit that corrects the torque component current reference calculated by the vector calculation unit based on the torque component current reference correction amount limited by the torque component current reference correction amount limiting unit;
Magnetic flux reference weakening correction means for performing correction to weaken the magnetic flux reference based on the torque-component current deviation calculated by the torque-component current deviation calculation means;
The coordinate conversion of the magnetic flux component current reference calculated by the vector calculation unit and the torque component current reference corrected by the torque component current reference correction unit into the current reference of each phase of the three-phase alternating current output from the inverter. Current reference coordinate conversion means for
Three-phase alternating current deviation calculating means for calculating, for each phase, a deviation between the current reference of each phase coordinate-converted by the current reference coordinate converting means and the three-phase alternating current detected by the three-phase alternating current detecting means; ,
An AC motor control apparatus comprising: inverter control means for controlling the inverter based on the deviation calculated for each phase by the three-phase AC current deviation calculation means. Magnetic flux component current deviation calculating means for calculating a magnetic flux component current deviation that is a deviation between the magnetic flux divided current coordinate-converted by the detected current coordinate converting means and the magnetic flux component current reference calculated by the vector calculating means;
Based on the magnetic flux component current deviation calculated by the magnetic flux component current deviation calculating means, a magnetic flux current that calculates a magnetic flux current reference correction amount that is a correction amount of the magnetic flux current reference calculated by the vector calculating means. A reference correction amount calculating means;
A magnetic flux component current reference correction unit that corrects the magnetic flux component current reference calculated by the vector calculation unit based on the magnetic flux component current reference correction amount calculated by the magnetic flux component current reference correction amount calculation unit;
The current reference coordinate conversion means outputs a three-phase output from the inverter of the magnetic flux current reference corrected by the magnetic flux current reference correction means and the torque current reference corrected by the torque current reference correction means. 2. The control apparatus for an AC motor according to claim 1, wherein coordinate conversion is performed to a current reference for each phase of the AC current. The magnetic flux reference weakening correction means is
Absolute value calculating means for calculating the absolute value of the torque current reference correction amount calculated by the torque current reference correction amount calculating means;
Difference calculating means for calculating a difference between the absolute value calculated by the absolute value calculating means and a level value at which weakening of the magnetic flux of the AC motor is started;
Amplifying means for amplifying the difference calculated by the difference calculating means;
Limiting means for limiting the difference amplified by the amplifying means not to be negative;
The control apparatus for an AC motor according to claim 1, further comprising a correction unit that corrects the magnetic flux reference based on the difference limited by the limitation unit. The magnetic flux reference weakening correction means is
A vector length calculation means for calculating a vector length obtained from the magnetic flux current deviation calculated by the magnetic flux current deviation calculation means and the torque current deviation calculated by the torque current deviation calculation means;
Difference calculating means for calculating a difference between the vector length calculated by the vector length calculating means and a level value at which weakening of the magnetic flux of the AC motor is started;
Amplifying means for amplifying the difference calculated by the difference calculating means;
Limiting means for limiting the difference amplified by the amplifying means not to be negative;
The control apparatus for an AC motor according to claim 2, further comprising a correcting unit that corrects the magnetic flux reference based on the difference limited by the limiting unit. An AC motor control device that controls the inverter and controls the drive of the AC motor,
Based on a magnetic flux reference for controlling the magnetic flux of the AC motor and a torque reference for controlling the torque of the AC motor, a torque component for controlling a torque component on the dq axis coordinate of the current flowing to the AC motor. Vector computing means for computing a current reference;
Three-phase alternating current detection means for detecting the three-phase alternating current output from the inverter;
Detection current coordinate conversion for converting the three-phase alternating current detected by the three-phase alternating current detection means into a torque component current that is a torque component on the dq axis coordinate and a magnetic flux component current that is a magnetic flux component on the dq axis coordinate Means,
A torque component current deviation calculation unit that calculates a torque component current deviation that is a deviation between the torque component current coordinate-converted by the detected current coordinate conversion unit and the torque component current reference calculated by the vector calculation unit;
Based on the torque component current deviation calculated by the torque component current deviation calculator, a torque component current for calculating a torque component current reference correction amount that is a correction amount of the torque component current reference calculated by the vector calculator A reference correction amount calculating means;
Torque component current reference correction amount limiting means for limiting the torque component current reference correction amount calculated by the torque component current reference correction amount calculation unit to a level value or less for starting the weakening of the magnetic flux of the AC motor;
A torque component current reference correction unit that corrects the torque component current reference calculated by the vector calculation unit based on the torque component current reference correction amount limited by the torque component current reference correction amount limiting unit;
Magnetic flux component current reference weakening correction means for performing correction to weaken the magnetic flux component current reference for controlling the magnetic flux component on the dq axis coordinate based on the torque component current deviation calculated by the torque component current deviation calculation means; Magnetic flux reference calculating means for calculating the magnetic flux reference based on the magnetic flux divided current reference corrected by the magnetic flux divided current reference weakening correcting means;
The magnetic flux component current reference corrected by the magnetic flux component current reference weakening correction unit and the torque component current reference corrected by the torque component current reference correction unit for each phase of the three-phase alternating current output from the inverter. Current reference coordinate conversion means for converting coordinates to a current reference;
Three-phase alternating current deviation calculating means for calculating, for each phase, a deviation between the current reference of each phase coordinate-converted by the current reference coordinate converting means and the three-phase alternating current detected by the three-phase alternating current detecting means; ,
An AC motor control apparatus comprising: inverter control means for controlling the inverter based on the deviation calculated for each phase by the three-phase AC current deviation calculation means. A magnetic flux component current deviation which is a deviation between the magnetic flux component current coordinate-converted by the detected current coordinate conversion means and the magnetic flux component current reference for controlling the magnetic flux component on the dq axis coordinate of the current flowing to the AC motor is calculated. Magnetic flux component current deviation calculating means,
Based on the magnetic flux component current deviation calculated by the magnetic flux component current deviation calculating means, the magnetic flux component which is a correction amount of the magnetic flux component current reference for controlling the magnetic flux component on the dq axis coordinate of the current flowing to the AC motor. Magnetic flux component current reference correction amount calculating means for calculating the current reference correction amount;
A magnetic flux component current reference correction unit for correcting the magnetic flux component current reference based on the magnetic flux component current reference correction amount calculated by the magnetic flux component current reference correction amount calculation unit;
The magnetic flux component current reference weakening correction means corrects to weaken the magnetic flux component current reference before correction by the magnetic flux component current reference correction means,
The current reference coordinate conversion means outputs the magnetic flux current reference corrected by the magnetic flux current reference correction means and the torque current reference corrected by the torque current reference correction means from the inverter. 6. The control apparatus for an AC motor according to claim 5, wherein coordinate conversion is performed to a current reference of each phase of the current. The magnetic flux component current reference weakening correction means,
Absolute value calculating means for calculating the absolute value of the torque current reference correction amount calculated by the torque current reference correction amount calculating means;
Difference calculating means for calculating a difference between the absolute value calculated by the absolute value calculating means and a level value at which weakening of the magnetic flux of the AC motor is started;
Amplifying means for amplifying the difference calculated by the difference calculating means;
Limiting means for limiting the difference amplified by the amplifying means not to be negative;
7. The control apparatus for an AC motor according to claim 5, further comprising a correction unit that corrects the magnetic flux component current reference based on the difference limited by the limitation unit. The magnetic flux component current reference weakening correction means,
A vector length calculation means for calculating a vector length obtained from the magnetic flux current deviation calculated by the magnetic flux current deviation calculation means and the torque current deviation calculated by the torque current deviation calculation means;
Difference calculating means for calculating a difference between the vector length calculated by the vector length calculating means and a level value at which weakening of the magnetic flux of the AC motor is started;
Amplifying means for amplifying the difference calculated by the difference calculating means;
Limiting means for limiting the difference amplified by the amplifying means not to be negative;
7. The control apparatus for an AC motor according to claim 6, further comprising correction means for correcting the magnetic flux component current reference based on the difference restricted by the restriction means. The torque current reference correction amount calculation means is
A vector length obtained from the magnetic flux component current deviation calculated by the magnetic flux component current deviation calculation means and the torque component current deviation calculated by the torque component current deviation calculation means is used as a limit value of the torque component current reference correction amount. The control apparatus for an AC motor according to any one of claims 2 to 4, or 6 to 8. A control device for an AC electric motor according to any one of claims 1 to 9,
The AC motor;
An electric motor drive system comprising the inverter.

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