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JP2009162521A - Radar device - Google Patents

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Publication number
JP2009162521A
JP2009162521A JP2007339712A JP2007339712A JP2009162521A JP 2009162521 A JP2009162521 A JP 2009162521A JP 2007339712 A JP2007339712 A JP 2007339712A JP 2007339712 A JP2007339712 A JP 2007339712A JP 2009162521 A JP2009162521 A JP 2009162521A
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Prior art keywords
signal
frequency
transmission
radar
switch
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JP2007339712A
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Japanese (ja)
Inventor
Yoshihide Agari
良英 上里
Tomohiro Shinomiya
知宏 篠宮
Katsuyuki Oguchi
勝之 大口
Kunihiro Tanaka
邦博 田中
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Denso Ten Ltd
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Denso Ten Ltd
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Priority to JP2007339712A priority Critical patent/JP2009162521A/en
Priority to PCT/JP2008/003692 priority patent/WO2009084159A1/en
Publication of JP2009162521A publication Critical patent/JP2009162521A/en
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S3/00—Direction-finders for determining the direction from which infrasonic, sonic, ultrasonic, or electromagnetic waves, or particle emission, not having a directional significance, are being received
    • G01S3/02—Direction-finders for determining the direction from which infrasonic, sonic, ultrasonic, or electromagnetic waves, or particle emission, not having a directional significance, are being received using radio waves
    • G01S3/14—Systems for determining direction or deviation from predetermined direction
    • G01S3/46—Systems for determining direction or deviation from predetermined direction using antennas spaced apart and measuring phase or time difference between signals therefrom, i.e. path-difference systems
    • G01S3/48—Systems for determining direction or deviation from predetermined direction using antennas spaced apart and measuring phase or time difference between signals therefrom, i.e. path-difference systems the waves arriving at the antennas being continuous or intermittent and the phase difference of signals derived therefrom being measured
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/35—Details of non-pulse systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/32—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated
    • G01S13/34—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal
    • G01S13/345—Systems for measuring distance only using transmission of continuous waves, whether amplitude-, frequency-, or phase-modulated, or unmodulated using transmission of continuous, frequency-modulated waves while heterodyning the received signal, or a signal derived therefrom, with a locally-generated signal related to the contemporaneously transmitted signal using triangular modulation
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G01S2013/9319—Controlling the accelerator
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G01S2013/9321—Velocity regulation, e.g. cruise control
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G01S2013/9325—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles for inter-vehicle distance regulation, e.g. navigating in platoons
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G01S2013/9327—Sensor installation details
    • G01S2013/93271—Sensor installation details in the front of the vehicles
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/93—Radar or analogous systems specially adapted for specific applications for anti-collision purposes
    • G01S13/931—Radar or analogous systems specially adapted for specific applications for anti-collision purposes of land vehicles
    • G01S2013/9327—Sensor installation details
    • G01S2013/93275—Sensor installation details in the bumper area

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Remote Sensing (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Radar Systems Or Details Thereof (AREA)

Abstract

<P>PROBLEM TO BE SOLVED: To provide a radar device capable of preventing decline of detection performance, even in the case of having a constitution equipped with a transmission route pair for transmitting transmission signals to a transmitting antenna pair separated at a prescribed interval or a switch for distributing the transmission signals to the transmission route pair. <P>SOLUTION: This radar device has an oscillator for outputting a radar signal having a lower frequency than a prescribed frequency; the switch for distributing radar signals outputted from the oscillator to the transmission route pair toward the transmitting antenna pair; and a multiplier pair connected respectively to the transmission route pair, for converting the frequency of the radar signal into the prescribed frequency and outputting it to the transmitting antenna pair. Consequently, an insertion loss of the switch or a transmission loss of the transmission route can be suppressed by reducing a frequency of the transmission signal passing the switch or the transmission route, and deterioration of the detection performance caused by intensity decline of the transmission signal can be prevented. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

本発明は、所定周波数のレ−ダ信号を送信する送信アンテナ対と、物標により反射された前記レ−ダ信号を受信信号として受信する受信アンテナ群とを有し、送信アンテナ対がそれぞれ送信を行ったときに得られる受信アンテナ群による受信信号を処理することにより、物標が位置する方位角を検出するレ−ダ装置に関する。     The present invention includes a transmission antenna pair that transmits a radar signal having a predetermined frequency and a reception antenna group that receives the radar signal reflected by a target as a reception signal, and each of the transmission antenna pairs transmits a signal. The present invention relates to a radar apparatus that detects an azimuth angle at which a target is located by processing a received signal from a group of receiving antennas obtained when performing the operation.

車載用のレ−ダ装置のスキャン方式として、電子スキャン式が知られている。電子スキャン式では、レ−ダ装置は、送信アンテナからレ−ダ信号を送信し、所定間隔で配列された複数の受信アンテナにより、物標に反射されたレ−ダ信号を受信信号として受信する。そして、レ−ダ装置は、到来方向に応じて位相と振幅が異なる複数の受信信号を用いて、物標が位置する方位角を検出する。   An electronic scanning method is known as a scanning method for a vehicle-mounted radar device. In the electronic scanning method, the radar apparatus transmits a radar signal from a transmission antenna, and receives the radar signal reflected on the target as a reception signal by a plurality of reception antennas arranged at predetermined intervals. . And a radar apparatus detects the azimuth | direction angle in which a target is located using the several received signal from which a phase and an amplitude differ according to an arrival direction.

図１は、従来の電子スキャン式レ−ダ装置の構成例を示す。レ−ダ装置２では、制御部６が出力する変調信号に従って、電圧制御発振器（ＶＣＯ）８がＦＭ−ＣＷ（Frequency Modulated-Continuous Wave）方式で周波数変調されたレ−ダ信号を出力する。分配器１０は、このレ−ダ信号を送信用のレ−ダ信号（以下、送信信号という）ＳＴと、受信信号の処理に用いるロ−カル信号としてのレーダ信号（以下、単にローカル信号という）ＳＬとに電力分配する。送信信号ＳＴは増幅器１２＿１により増幅され、送信アンテナＡＴ＿１から送信される。この信号が物標により反射されると、一定間隔で配列された受信アンテナＡＲ＿１、ＡＲ＿２、ＡＲ＿３、ＡＲ＿４により受信信号ＳＲ＿１、ＳＲ＿２、ＳＲ＿３、ＳＲ＿４として受信される。受信信号ＳＲ＿ｎ（以下では、既述した複数の構成要素に言及するとき、添え字を「＿ｎ」で表す）はそれぞれ増幅器１４＿１、１４＿２、１４＿３、１４＿４により増幅されて、ミキサ１６＿１、１６＿２、１６＿３、１６＿４に入力される。   FIG. 1 shows an example of the configuration of a conventional electronic scanning radar apparatus. In the radar apparatus 2, the voltage controlled oscillator (VCO) 8 outputs a radar signal frequency-modulated by the FM-CW (Frequency Modulated-Continuous Wave) method in accordance with the modulation signal output from the control unit 6. The distributor 10 uses this radar signal as a transmission radar signal (hereinafter referred to as a transmission signal) ST and a radar signal (hereinafter simply referred to as a local signal) as a local signal used for processing the received signal. Power is distributed to SL. The transmission signal ST is amplified by the amplifier 12_1 and transmitted from the transmission antenna AT_1. When this signal is reflected by the target, it is received as received signals SR_1, SR_2, SR_3, SR_4 by receiving antennas AR_1, AR_2, AR_3, AR_4 arranged at regular intervals. The received signal SR_n (hereinafter, the subscript is represented by “_n” when referring to the plurality of components described above) is amplified by the amplifiers 14_1, 14_2, 14_3, and 14_4, respectively, and mixed by the mixers 16_1, 16_2, 16_3, 16_4.

一方、分配器１０により電力分配されたロ−カル信号ＳＬは、増幅器１２＿２により増幅されてミキサ１６＿ｎに入力される。ミキサ１６＿ｎは、それぞれロ−カル信号ＳＬと受信信号ＳＲ＿ｎとを混合し、ロ−カル信号ＳＬと受信信号ＳＲ＿ｎとの周波数差、つまり送信信号ＳＴと受信信号ＳＲ＿ｎとの周波数差を周波数とする周波数差信号（以下、ビ−ト信号という）ＳＢ＿１、ＳＢ＿２、ＳＢ＿３、ＳＢ＿４を出力する。ビ−ト信号ＳＢ＿ｎは、それぞれＡ／Ｄ変換器１８＿１、１８＿２、１８＿３、１８＿４によりサンプリングされてディジタルデ−タに変換されると、制御部６に取り込まれる。   On the other hand, the local signal SL distributed by the distributor 10 is amplified by the amplifier 12_2 and input to the mixer 16_n. The mixer 16_n mixes the local signal SL and the received signal SR_n, and the frequency difference between the local signal SL and the received signal SR_n, that is, the frequency difference between the transmitted signal ST and the received signal SR_n. Difference signals (hereinafter referred to as beat signals) SB_1, SB_2, SB_3, and SB_4 are output. The beat signal SB_n is sampled by the A / D converters 18_1, 18_2, 18_3, and 18_4 and converted into digital data, and then taken into the control unit 6.

制御部６は、ＣＰＵ、ＲＯＭ、ＲＡＭを備えた周知のマイクロコンピュ−タを中心に構成され、更に、Ａ／Ｄ変換器１８＿ｎを介して取り込んだデ−タについて、高速フ−リエ変換（ＦＦＴ：Fast Fourier Transform）処理を実行するための演算処理装置（例えばＤＳＰ：Digital Signal Processor）を備える。ここでは、制御部６は、ビ−ト信号ＳＢ＿ｎに対しＤＢＦ（Digital Beam Forming）処理を行うことで、物標が位置する方位角を検出する。   The control unit 6 is mainly composed of a well-known microcomputer having a CPU, a ROM, and a RAM. Further, the controller 6 performs high-speed Fourier transform (FFT) on the data taken in via the A / D converter 18_n. : An arithmetic processing unit (for example, DSP: Digital Signal Processor) for executing Fast Fourier Transform (Fast Fourier Transform) processing. Here, the controller 6 detects the azimuth angle at which the target is located by performing DBF (Digital Beam Forming) processing on the beat signal SB_n.

ＤＢＦ処理では、一定間隔で配列された複数の受信アンテナＡＲ＿ｎによる受信信号ＳＲ＿ｎを合成すると、その利得は受信アンテナＡＲ＿ｎの指向性が一致する方位角において最大となることを利用し、かかる方位角を受信信号の到来方向、つまり物標が位置する方位角として逆算する。すなわち、制御部６は、受信信号ＳＲ＿ｎから生成されたビ−ト信号ＳＢ＿ｎの合成振幅を方位角の関数として求め、その合成振幅の最大値に対応する方位角を算出する。   In the DBF processing, when the reception signals SR_n from the plurality of reception antennas AR_n arranged at regular intervals are combined, the gain is maximized at the azimuth angle where the directivities of the reception antennas AR_n match. The calculation is performed backward as the arrival direction of the received signal, that is, the azimuth angle at which the target is located. That is, the control unit 6 obtains the combined amplitude of the beat signal SB_n generated from the received signal SR_n as a function of the azimuth, and calculates the azimuth corresponding to the maximum value of the combined amplitude.

ここにおいて、受信アンテナの数が多いほど、すなわち、合成されるビ−ト信号の数が多いほど、その合成振幅の変化曲線が急峻となるので、合成振幅の最大値に対応する方位角が精度良く求まる。そのためには、多数の受信アンテナを配列することが望ましい。   Here, the greater the number of receiving antennas, that is, the greater the number of beat signals to be combined, the steeper the change curve of the combined amplitude. Therefore, the azimuth corresponding to the maximum value of the combined amplitude is more accurate. I can find it well. For this purpose, it is desirable to arrange a large number of receiving antennas.

しかし一方で、車載用のレ−ダ装置においては、設置スペ−スと製造コストの制約上、小型化が要望される。このため、受信アンテナを多数配列する代わりに、複数の送信アンテナを設けることで、装置規模を拡大させることなく、多数の受信アンテナを配列した構成と等価な構成を実現する方法が提案されている。かかるレ−ダ装置の構成例を図２に示す。   On the other hand, in-vehicle radar devices are required to be downsized due to restrictions on installation space and manufacturing cost. For this reason, there has been proposed a method for realizing a configuration equivalent to a configuration in which a large number of receiving antennas are arranged without increasing the scale of the apparatus by providing a plurality of transmitting antennas instead of arranging a large number of receiving antennas. . A configuration example of such a radar apparatus is shown in FIG.

図２に示すレ−ダ装置２は、分配器１０により分配された送信信号ＳＴをさらに電力分配する分配器２０と、この分配器２０により分配された送信信号ＳＴをそれぞれ増幅する増幅器２２＿１、２２＿２と、増幅された送信信号ＳＴを送信する１対の送信アンテナＡＴ＿１、ＡＴ＿２を有する点において、図１の構成例と異なる。   The radar apparatus 2 shown in FIG. 2 further distributes power of the transmission signal ST distributed by the distributor 10 and amplifiers 22_1 and 22_2 that amplify the transmission signal ST distributed by the distributor 20, respectively. 1 in that it has a pair of transmission antennas AT_1 and AT_2 that transmit the amplified transmission signal ST.

この構成例では、送信アンテナＡＴ＿１、ＡＴ＿２は、受信アンテナＡＲ＿ｎの配列の両端に位置することで一定間隔離間している。よって、その間隔と物標が位置する方位角に応じて、両送信アンテナから物標までの経路長に差が生じる。このため、送信アンテナＡＴ＿１からの送信信号が物標に反射されて受信アンテナＡＲ＿ｎにそれぞれ受信されるときの往復経路長と、送信アンテナＡＴ＿２からの送信信号が物標に反射されて受信アンテナＡＲ＿ｎにそれぞれ受信されるときの往復経路長はいずれも異なる。よって、送信アンテナＡＴ＿１、ＡＴ＿２のそれぞれと、受信アンテナＡＲ＿ｎのそれぞれとの組合せにおいては、いずれも振幅と位相とが異なる複数の受信信号ＳＲ＿ｎが得られる。   In this configuration example, the transmission antennas AT_1 and AT_2 are spaced apart from each other by being positioned at both ends of the array of the reception antennas AR_n. Therefore, a difference occurs in the path length from both transmitting antennas to the target according to the interval and the azimuth angle at which the target is located. Therefore, the round trip path length when the transmission signal from the transmission antenna AT_1 is reflected by the target and received by the reception antenna AR_n, and the transmission signal from the transmission antenna AT_2 is reflected by the target and is received by the reception antenna AR_n. Each of the round trip path lengths when received is different. Therefore, in each combination of the transmission antennas AT_1 and AT_2 and the reception antenna AR_n, a plurality of reception signals SR_n having different amplitudes and phases are obtained.

その結果、図３（Ａ）に示す、送信を行う送信アンテナＡＴ＿１（黒塗りで示す）と受信アンテナＡＲ＿ｎとの位置関係と、図３（Ｂ）に示す、送信を行う送信アンテナＡＴ＿２（黒塗りで示す）と受信アンテナＡＲ＿ｎとの位置関係とをそれぞれ送信を行う送信アンテナの位置を重ねて組み合わせると、図３（Ｃ）に示す、倍の数の受信アンテナを配列した構成と等価な構成が得られる。すなわち、２つの送信アンテナと４つの受信アンテナにより、１つの送信アンテナと８つの受信アンテナからなる構成と等価な構成が実現できる。   As a result, the positional relationship between the transmitting antenna AT_1 (shown in black) and the receiving antenna AR_n shown in FIG. 3A and the transmitting antenna AT_2 (shown in black) shown in FIG. 3) and the positional relationship between the receiving antennas AR_n and combining the positions of the transmitting antennas that transmit each other in combination, a configuration equivalent to the configuration in which a double number of receiving antennas are arranged as shown in FIG. can get. That is, a configuration equivalent to a configuration including one transmission antenna and eight reception antennas can be realized by two transmission antennas and four reception antennas.

ところで、図２のレ−ダ装置２では、分配器２０により電力分配されることで送信信号ＳＴの強度が低下するので、分配された送信信号ＳＴの強度を増幅器２２＿１、２２＿２が所期の強度まで増幅する。しかし、電力分配による強度の低下分を補うためには、有る程度大きい増幅度が必要となる。すると、同時にノイズも増幅され、良好なＳ／Ｎ比の送信信号ＳＴが得られない。そこで、分配器２０の代わりに、送信信号ＳＴを交互に送信アンテナ対ＡＴ＿１、ＡＴ＿２に出力するスイッチを用いることで、かかる事態を回避する方法が、さらに提案されている。   By the way, in the radar apparatus 2 of FIG. 2, since the power of the transmission signal ST is reduced by the power distribution by the distributor 20, the amplifiers 22_1 and 22_2 use the strength of the distributed transmission signal ST as the expected power. Amplify until. However, in order to compensate for the decrease in strength due to power distribution, a certain degree of amplification is required. Then, noise is also amplified at the same time, and a transmission signal ST having a good S / N ratio cannot be obtained. Therefore, a method for avoiding such a situation by using a switch that alternately outputs the transmission signal ST to the transmission antenna pair AT_1 and AT_2 instead of the distributor 20 has been proposed.

図４は、かかる方法の一例におけるレ−ダ装置の構成を示す。このレ−ダ装置２は、送信信号ＳＴを電力分配する分配器２０の代わりに、送信信号ＳＴを送信アンテナＡＴ＿１、ＡＴ＿２に対応する伝送経路Ｐ＿１、Ｐ＿２に交互に出力するスイッチＳＷ＿１を有する点において、図２の構成例と異なる。ここで、スイッチＳＷ＿１は、制御部６により切替制御される。   FIG. 4 shows the configuration of the radar apparatus in an example of such a method. This radar apparatus 2 has a switch SW_1 that alternately outputs the transmission signal ST to the transmission paths P_1 and P_2 corresponding to the transmission antennas AT_1 and AT_2, instead of the distributor 20 that distributes the transmission signal ST. This is different from the configuration example of FIG. Here, the switch SW_1 is switch-controlled by the control unit 6.

図５は、さらに別の例におけるレ−ダ装置の構成を示す。このレ−ダ装置２は、受信アンテナＡＲ＿ｎによる受信信号ＳＲ＿ｎを順次切り替えて１つのミキサ１６に入力するスイッチＳＷ＿２をさらに備え、ミキサ１６が出力するビ−ト信号ＳＢは、１つのＡ／Ｄ変換器１８でディジタルデ−タ化される点において、図４の構成と異なる。ここで、スイッチＳＷ＿２は、制御部６により切替制御される。この構成は、スイッチＳＷ＿２を設けることでミキサとＡ／Ｄ変換器の数を減らし、レ−ダ装置２の回路規模をより小型化している。   FIG. 5 shows a configuration of a radar apparatus in still another example. The radar apparatus 2 further includes a switch SW_2 for sequentially switching the reception signal SR_n from the reception antenna AR_n and inputting the signal to one mixer 16, and the beat signal SB output from the mixer 16 is one A / D conversion. 4 is different from the configuration shown in FIG. Here, the switch SW_2 is switch-controlled by the control unit 6. In this configuration, the number of mixers and A / D converters is reduced by providing the switch SW_2, and the circuit scale of the radar apparatus 2 is further reduced.

また、さらに別の例が、特許文献１に記載されている。特許文献１に記載されたレ−ダ装置は、配列された複数の受信アンテナ群の両端に１対の送信アンテナを設け、それぞれから時分割でレ−ダ信号を送信する構成を有する。
特開２００４−１９８３１２号公報 Still another example is described in Patent Document 1. The radar apparatus described in Patent Document 1 has a configuration in which a pair of transmission antennas are provided at both ends of a plurality of arranged reception antenna groups, and a radar signal is transmitted from each of them in a time division manner.
JP 2004-198312 A

上述の図２、図４、図５、または特許文献１の例では、一定の間隔離間した複数の送信アンテナに対し、分配された送信信号を伝送する伝送経路Ｐ＿１、Ｐ＿２が必須の構成となる。ここで、車載用のレ−ダ装置においては、法規制により使用可能な電波の周波数帯域がミリ波帯に含まれる７６．０〜７７．０ＧＨｚの高周波帯域に限定される。すると、かかるミリ波帯の送信信号が伝送経路Ｐ＿１、Ｐ＿２を伝送されるときに、伝送経路の導体抵抗、つまり、表皮効果による伝送経路の表面抵抗と内部インダクタンスとの和が、送信信号の周波数に応じて大きくなる。さらに、伝送経路Ｐ＿１、Ｐ＿２をマイクロストリップラインのように誘電体上に形成する場合には、誘電体損失も生じる。そのため、送信信号ＳＴが送信アンテナ対まで伝送されるときに、送信信号ＳＴの強度が伝送損失により低下する。   In the above-described FIG. 2, FIG. 4, FIG. 5, or the example of Patent Document 1, the transmission paths P_1 and P_2 for transmitting the distributed transmission signals are indispensable for a plurality of transmission antennas spaced apart by a fixed interval. . Here, in an on-vehicle radar device, the frequency band of radio waves that can be used by law is limited to a high frequency band of 76.0 to 77.0 GHz that is included in the millimeter wave band. Then, when the millimeter-wave band transmission signal is transmitted through the transmission paths P_1 and P_2, the conductor resistance of the transmission path, that is, the sum of the surface resistance of the transmission path due to the skin effect and the internal inductance is the frequency of the transmission signal. It grows according to. Furthermore, when the transmission paths P_1 and P_2 are formed on a dielectric like a microstrip line, dielectric loss also occurs. Therefore, when the transmission signal ST is transmitted to the transmission antenna pair, the strength of the transmission signal ST is reduced due to transmission loss.

また、ＶＣＯ８から出力された送信信号ＳＴを伝送経路Ｐ＿１、Ｐ＿２に分配するスイッチＳＷ＿１が、通過する信号の周波数に応じて挿入損失が大きくなるような場合には、スイッチＳＷ１を通過することによっても送信信号ＳＴの強度が低下する。   Further, when the switch SW_1 that distributes the transmission signal ST output from the VCO 8 to the transmission paths P_1 and P_2 has a large insertion loss according to the frequency of the signal passing through, the switch SW_1 also passes through the switch SW1. The intensity of the transmission signal ST decreases.

そして、このように送信信号ＳＴの強度が低下すると、レ−ダ装置としての探知性能が低下するという問題が生じる。   If the strength of the transmission signal ST is lowered in this way, there arises a problem that the detection performance as the radar apparatus is lowered.

そこで、本発明の目的は、所定間隔離間した送信アンテナ対に送信信号を伝送する伝送経路対や、伝送経路対に送信信号を分配するスイッチを備えた構成であっても、探知性能の低下を防止できるレ−ダ装置を提供することにある。   Therefore, an object of the present invention is to reduce the detection performance even in a configuration including a transmission path pair that transmits a transmission signal to a pair of transmission antennas spaced apart by a predetermined interval and a switch that distributes the transmission signal to the transmission path pair. It is an object of the present invention to provide a radar device that can be prevented.

上記の目的を達成するために、本発明の第１の側面によるレ−ダ装置は、所定周波数のレ−ダ信号を送信する所定間隔離間した送信アンテナ対と、物標により反射された前記レーダ信号を受信信号として受信する受信アンテナ群とを有し、前記送信アンテナ対から交互に前記レーダ信号を送信して得られる受信信号を用いて前記物標が位置する方位角を検出するレ−ダ装置であって、前記所定周波数より低い周波数のレ−ダ信号を出力する発振器と、前記発振器が出力するレ−ダ信号を、第１の前記送信アンテナへの第１の伝送経路と、第２の前記送信アンテナへの第２の伝送経路とに分配するスイッチと、前記第１の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第１の送信アンテナに出力する第１の逓倍器と、前記第２の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第２の送信アンテナに出力する第２の逓倍器とを有することを特徴とする。 In order to achieve the above object, a radar apparatus according to a first aspect of the present invention includes a pair of transmitting antennas that transmit a radar signal having a predetermined frequency, and a radar that is reflected by a target. A radar that detects a azimuth angle at which the target is located by using a reception signal obtained by alternately transmitting the radar signal from the pair of transmission antennas. An oscillator for outputting a radar signal having a frequency lower than the predetermined frequency; a radar signal output by the oscillator; a first transmission path to the first transmitting antenna; A switch that distributes to the second transmission path to the transmission antenna, and the first transmission antenna that is connected to the first transmission path and converts the frequency of the radar signal to the predetermined frequency. Output to the first A multiplier, and a second multiplier connected to the second transmission path and converting the frequency of the radar signal into the predetermined frequency and then outputting the frequency to the second transmission antenna. And

上記側面の好ましい態様では、前記所定の周波数はミリ波帯、前記発振器が出力するレ−ダ信号の周波数はミリ波帯より低いマイクロ波帯である。なお、一般にミリ波帯は３０ＧＨｚ〜３００ＧＨｚの帯域、マイクロ波帯は１ＧＨｚ〜１０００ＧＨｚの帯域をいうが、以下の説明で単にマイクロ波帯というときは、３０ＧＨｚ〜３００ＧＨｚのミリ波帯より低い帯域、つまり１ＧＨｚ〜３０ＧＨｚの帯域を意味するものとする。   In a preferred embodiment of the above aspect, the predetermined frequency is a millimeter wave band, and the frequency of the radar signal output from the oscillator is a microwave band lower than the millimeter wave band. In general, the millimeter wave band is a band of 30 GHz to 300 GHz, and the microwave band is a band of 1 GHz to 1000 GHz. However, in the following description, when simply referred to as a microwave band, a band lower than the millimeter wave band of 30 GHz to 300 GHz, that is, It means a band of 1 GHz to 30 GHz.

上記側面によるレ−ダ装置は、前記所定周波数（ミリ波帯）より低い周波数（マイクロ波帯）のレ−ダ信号を出力する発振器と、第１の前記送信アンテナへの第１の伝送経路と、第２の前記送信アンテナへの第２の伝送経路とに分配するスイッチと、前記第１の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第１の送信アンテナに出力する第１の逓倍器と、前記第２の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第２の送信アンテナに出力する第２の逓倍器とを有する。よって、前記スイッチが、これを通過するレーダ信号の周波数が高いほど挿入損失が大きいスイッチであっても、所定周波数（ミリ波帯）より低い周波数（マイクロ波帯）のレ−ダ信号としてスイッチを通過させることにより、所定周波数のレ−ダ信号のままスイッチを通過させる場合より挿入損失を小さく抑えることができる。さらに、レ−ダ信号が第１、第２の伝送経路を伝送されるときには、その周波数に応じて伝送損失が大きくなるが、送信信号の周波数を低くして第１、第２の伝送経路を伝送させることで、伝送損失を小さく抑えることができる。このように、スイッチと伝送経路を経て送信アンテナまで伝送されるレーダ信号の強度低下を防ぐことができるので、レーダ装置の検出精度の低下を防止できる。   The radar apparatus according to the above aspect includes an oscillator that outputs a radar signal having a frequency (microwave band) lower than the predetermined frequency (millimeter wave band), and a first transmission path to the first transmitting antenna. A switch that distributes to a second transmission path to the second transmission antenna; and a switch that is connected to the first transmission path and converts the frequency of the radar signal to the predetermined frequency and then the first transmission path. A first multiplier that outputs to the second transmission antenna, and a second multiplier that is connected to the second transmission path, converts the frequency of the radar signal to the predetermined frequency, and then outputs the second signal to the second transmission antenna. And a multiplier. Therefore, even if the switch has a higher insertion loss as the frequency of the radar signal passing therethrough is higher, the switch is used as a radar signal having a frequency (microwave band) lower than a predetermined frequency (millimeter wave band). By allowing the signal to pass, the insertion loss can be suppressed to a smaller value than when the switch is passed with the radar signal having a predetermined frequency. Further, when the radar signal is transmitted through the first and second transmission paths, the transmission loss increases according to the frequency. However, the frequency of the transmission signal is lowered to reduce the first and second transmission paths. By transmitting, transmission loss can be suppressed small. As described above, since it is possible to prevent a decrease in the intensity of the radar signal transmitted to the transmission antenna via the switch and the transmission path, it is possible to prevent a decrease in the detection accuracy of the radar apparatus.

以下、図面にしたがって本発明の実施の形態について説明する。但し、本発明の技術的範囲はこれらの実施の形態に限定されず、特許請求の範囲に記載された事項とその均等物まで及ぶものである。   Hereinafter, embodiments of the present invention will be described with reference to the drawings. However, the technical scope of the present invention is not limited to these embodiments, but extends to the matters described in the claims and equivalents thereof.

図６は、本実施形態におけるレ−ダ装置の、第１の構成例を説明する図である。このレ−ダ装置２は、次の点において、図４で示した従来の構成と異なる。   FIG. 6 is a diagram illustrating a first configuration example of the radar apparatus according to this embodiment. This radar device 2 is different from the conventional configuration shown in FIG. 4 in the following points.

第１の構成例では、ＶＣＯ８の発振周波数が、図４に示した従来例の１／Ｎ（Ｎは自然数。ここでは、Ｎ＝４）に設定される。また、伝送経路Ｐ＿１、Ｐ＿２の終端には、スイッチＳＷ＿１により分配された送信信号ＳＴの周波数をＮ倍して送信アンテナＡＴ＿１、ＡＴ＿２に出力する逓倍器３０＿１、３０＿２が接続される。さらに、ロ−カル信号ＳＬの伝送経路Ｐ＿４には、ローカル信号ＳＬの周波数を同じくＮ倍してミキサ１６＿ｎに出力する逓倍器３０＿３が接続される。   In the first configuration example, the oscillation frequency of the VCO 8 is set to 1 / N (N is a natural number; here, N = 4) in the conventional example shown in FIG. Further, multipliers 30_1 and 30_2 that multiply the frequency of the transmission signal ST distributed by the switch SW_1 and output to the transmission antennas AT_1 and AT_2 are connected to the ends of the transmission paths P_1 and P_2. Furthermore, a multiplier 30_3 that similarly multiplies the frequency of the local signal SL by N and outputs it to the mixer 16_n is connected to the transmission path P_4 of the local signal SL.

ここでは、ＶＣＯ８、分配器１０、増幅器１２＿ｎ、スイッチＳＷ＿１、逓倍器３０＿ｎ、増幅器２２＿ｎがモジュール化されて送信用ＭＭＩＣ（Microwave Monolithic IC）により構成され、増幅器１４＿ｎ、ミキサ１６＿ｎ、Ａ／Ｄ変換器１８＿ｎがモジュール化されて受信用ＭＭＩＣにより構成される。そして、各モジュールは、大電力用材料（ＧａＡｓ、ＳｉＧｅなど）と小電力用材料（Ｓｉ）の組合せにより構成される。   Here, the VCO 8, the divider 10, the amplifier 12_n, the switch SW_1, the multiplier 30_n, and the amplifier 22_n are modularized and configured by a transmission MMIC (Microwave Monolithic IC), and the amplifier 14_n, the mixer 16_n, and the A / D converter 18_n. Is modularized and configured by a receiving MMIC. Each module is composed of a combination of a high power material (GaAs, SiGe, etc.) and a low power material (Si).

第１の構成例では、スイッチＳＷ＿１は、ダイオードを用いたスイッチ回路により構成され、制御部６により切替制御される。スイッチＳＷ＿１は１つの入力端子と３つの出力端子を有し、出力端子の１つは伝送経路Ｐ＿１に、他の１つは伝送経路Ｐ＿２に、残りの１つは伝送経路Ｐ＿３に接続される。ここで、伝送経路Ｐ＿３への出力は、伝送経路Ｐ＿１、Ｐ＿２への出力の中間に位置しており、伝送経路Ｐ＿３を抵抗Ｒに接続することにより、伝送経路Ｐ＿１（またはＰ＿２）に出力される送信信号ＳＴの輻射波は伝送経路Ｐ＿３において抵抗Ｒに吸収されるため、輻射波が伝送経路Ｐ＿２（またはＰ＿１）に入り込むことでノイズを生じさせることを防止できる。これにより、スイッチＳＷ＿１のアイソレ−ションを向上させることができる。
また、スイッチＳＷ＿１を通過する信号の周波数に対する挿入損失を図７（Ａ）を用いて示すと、通過する信号の周波数（横軸）が増大するのに対し、通過する信号の挿入損失（縦軸）は増大する。よって、第１の構成例によれば、ミリ波帯の送信信号ＳＴの代わりにマイクロ波帯の送信信号ＳＴがスイッチＳＷ＿１を通過することで、スイッチＳＷ＿１の挿入損失による送信信号ＳＴの強度の低下をミリ波帯のときより小さく抑えることができる。 In the first configuration example, the switch SW_1 is configured by a switch circuit using a diode, and is switched and controlled by the control unit 6. The switch SW_1 has one input terminal and three output terminals. One of the output terminals is connected to the transmission path P_1, the other is connected to the transmission path P_2, and the other is connected to the transmission path P_3. Here, the output to the transmission path P_3 is located in the middle of the outputs to the transmission paths P_1 and P_2, and is output to the transmission path P_1 (or P_2) by connecting the transmission path P_3 to the resistor R. Since the radiated wave of the transmission signal ST is absorbed by the resistor R in the transmission path P_3, it is possible to prevent the generation of noise due to the radiated wave entering the transmission path P_2 (or P_1). As a result, the isolation of the switch SW_1 can be improved.
Further, when the insertion loss with respect to the frequency of the signal passing through the switch SW_1 is shown using FIG. 7A, the frequency of the passing signal (horizontal axis) increases, whereas the insertion loss of the passing signal (vertical axis) ) Will increase. Therefore, according to the first configuration example, the transmission signal ST in the microwave band passes through the switch SW_1 instead of the transmission signal ST in the millimeter wave band, so that the strength of the transmission signal ST is reduced due to the insertion loss of the switch SW_1. Can be kept smaller than in the millimeter wave band.

そして、送信信号ＳＴの周波数が逓倍器３０＿１、３０＿２によりミリ波帯の所定周波数（例えば７６．５ＧＨｚ）まで逓倍されてから、送信信号ＳＴが送信アンテナＡＴ＿１、ＡＴ＿２から送信される。このような構成により、スイッチＳＷ＿１により送信信号ＳＴを択一的に切り替えて交互に出力しても、送信信号ＳＴの強度低下を回避でき、かつ所期の周波数の送信信号ＳＴを送信できる。   Then, after the frequency of the transmission signal ST is multiplied by a predetermined frequency (for example, 76.5 GHz) in the millimeter wave band by the multipliers 30_1 and 30_2, the transmission signal ST is transmitted from the transmission antennas AT_1 and AT_2. With such a configuration, even if the transmission signal ST is alternately switched by the switch SW_1 and alternately output, the strength of the transmission signal ST can be prevented from decreasing and the transmission signal ST having a desired frequency can be transmitted.

また、送信信号ＳＴの周波数が大きくなると、表皮効果における伝送経路Ｐ＿１、Ｐ＿２、またはＰ＿３の表面抵抗は周波数の平方根に比例して大きくなり、内部インダクタンスは周波数に比例して大きくなるので、これらの和である導体抵抗は周波数に応じて大きくなる。さらに、伝送経路Ｐ＿１、Ｐ＿２、またはＰ＿３をマイクロストリップラインのように誘電体上に形成する場合には、誘電体損失も生じる。このことは、図７（Ｂ）において示される。図７（Ｂ）では、伝送される信号の周波数（横軸）の増大に応じて、異なるタイプの伝送経路の伝送損失（縦軸）がいずれも増大することが示される。   When the frequency of the transmission signal ST increases, the surface resistance of the transmission path P_1, P_2 or P_3 in the skin effect increases in proportion to the square root of the frequency, and the internal inductance increases in proportion to the frequency. The conductor resistance, which is the sum, increases with frequency. Further, when the transmission path P_1, P_2, or P_3 is formed on a dielectric like a microstrip line, dielectric loss also occurs. This is shown in FIG. 7 (B). FIG. 7B shows that the transmission loss (vertical axis) of different types of transmission paths increases as the frequency (horizontal axis) of the transmitted signal increases.

この点においても、第１の構成例では、ミリ波帯の送信信号ＳＴの代わりにマイクロ波帯の送信信号ＳＴを伝送させ、送信される直前にこれをミリ波帯まで逓倍して送信アンテナＡＴ＿１、ＡＴ＿２から送信し、あるいは、ミリ波帯まで逓倍してミキサミキサ１６＿ｎに入力するので、伝送経路Ｐ＿１、Ｐ＿２における伝送損失をミリ波帯のときより小さく抑えることができる。よって、送信アンテナＡＴ＿１、ＡＴ＿２が一定間隔離間していることにより、伝送経路Ｐ＿１、Ｐ＿２を有る程度の長さとする場合であっても、伝送経路Ｐ＿１、Ｐ＿２を伝送される送信信号ＳＴの強度低下を回避できる。また、ＶＣＯ８からスイッチＳＷ＿１までの伝送経路においても同様に伝送損失を抑えることができ、送信信号の強度低下が回避される。さらに、ミキサミキサ１６＿ｎに入力するローカル信号ＳＬの強度低下も回避される。よって、レーダ装置として探知精度の低下を防ぐことができる。   Also in this respect, in the first configuration example, the transmission signal ST in the microwave band is transmitted instead of the transmission signal ST in the millimeter wave band, and this is multiplied to the millimeter wave band just before being transmitted, and then transmitted to the transmission antenna AT_1. , Transmitted from AT_2, or multiplied to the millimeter wave band and input to the mixer 16_n, transmission loss in the transmission paths P_1 and P_2 can be suppressed to be smaller than that in the millimeter wave band. Therefore, since the transmission antennas AT_1 and AT_2 are spaced apart from each other by a certain distance, the strength of the transmission signal ST transmitted through the transmission paths P_1 and P_2 is reduced even when the transmission paths P_1 and P_2 are of a certain length. Can be avoided. Similarly, transmission loss can be suppressed in the transmission path from the VCO 8 to the switch SW_1, and a decrease in the strength of the transmission signal is avoided. Furthermore, a decrease in the strength of the local signal SL input to the mixer 16_n is also avoided. Therefore, it is possible to prevent a decrease in detection accuracy as a radar device.

なお、第１の構成例では、送信アンテナＡＴ＿１、ＡＴ＿２が配列された受信アンテナＡＲ＿ｎの両端に配置されることで、図３で示したように倍の数の受信アンテナを備えた場合と等価の構成を実現できる。しかし、送信アンテナＡＴ＿１、ＡＴ＿２の間隔はこれに限定されなくてもよい。送信アンテナＡＴ＿１、ＡＴ＿２から送信されたレーダ信号が物標により反射され受信アンテナＡＲ＿ｎに受信されるまでの往復経路長がそれぞれ異なり、よって、位相と振幅がそれぞれ異なる受信信号が全ての送信アンテナと受信アンテナとの組合せにおいて得られるように送信アンテナＡＴ＿１、ＡＴ＿２が離間していれば、限られた受信アンテナ数でそれ以上の受信アンテナ数を備えた場合と等価な構成を実現できる。いずれにしても、送信アンテナＡＴ＿１、ＡＴ＿２の間隔に応じてある程度の長さの伝送経路Ｐ＿１、Ｐ＿２を必要とする場合に、上記の作用効果を奏することができる。   In the first configuration example, the transmission antennas AT_1 and AT_2 are arranged at both ends of the reception antenna AR_n arranged, which is equivalent to the case where the reception antennas are doubled as shown in FIG. The configuration can be realized. However, the interval between the transmitting antennas AT_1 and AT_2 may not be limited to this. The round-trip path length until the radar signals transmitted from the transmitting antennas AT_1 and AT_2 are reflected by the target and received by the receiving antenna AR_n is different, and therefore, the received signals having different phases and amplitudes are received by all the transmitting antennas. As long as the transmission antennas AT_1 and AT_2 are spaced apart from each other as obtained in combination with the antenna, a configuration equivalent to the case where the number of reception antennas is larger than that with a limited number of reception antennas can be realized. In any case, when the transmission paths P_1 and P_2 having a certain length according to the interval between the transmitting antennas AT_1 and AT_2 are required, the above-described effects can be obtained.

ここで、第１の構成例における各部の動作を説明する。ＶＣＯ８は、制御部６からＦＭ−ＣＷ方式の三角波状の変調信号が入力されると、これに従って、時間に対して周波数が直線的に漸増、漸減するよう変調されたミリ波帯より低い帯域のマイクロ波帯のレ−ダ信号を生成する。例えば、図４で示した従来の構成で、ＶＣＯ８の発振周波数が、レ−ダ信号の中心周波数７６．５ＧＨｚ、周波数変動幅１００ＭＨｚとなるように設定されていたとすると、この第１の構成例では、ＶＣＯ８の発振周波数は、レ−ダ信号の中心周波数が約１９．１（＝７６．５／４）ＧＨｚ、周波数変動幅１００ＭＨｚとなるように設定される。   Here, the operation of each unit in the first configuration example will be described. When the FM-CW triangular modulation signal is input from the control unit 6, the VCO 8 has a lower band than the millimeter wave band that is modulated so that the frequency linearly increases and decreases with time. A microwave band radar signal is generated. For example, in the conventional configuration shown in FIG. 4, if the oscillation frequency of the VCO 8 is set so that the center frequency of the radar signal is 76.5 GHz and the frequency fluctuation range is 100 MHz, the first configuration example The oscillation frequency of the VCO 8 is set so that the center frequency of the radar signal is about 19.1 (= 76.5 / 4) GHz and the frequency fluctuation range is 100 MHz.

このレ−ダ信号は、分配器１０により送信信号ＳＴとロ−カル信号ＳＬとに電力分配されると、送信信号ＳＴは増幅器１２＿１により増幅されてスイッチＳＷ＿１に入力される。   When the power of the radar signal is distributed to the transmission signal ST and the local signal SL by the distributor 10, the transmission signal ST is amplified by the amplifier 12_1 and input to the switch SW_1.

スイッチＳＷ＿１は、制御部６からの制御信号に応答して、送信信号ＳＴの出力先を、伝送経路Ｐ＿１、Ｐ＿２とに交互に時分割で切り替える。こうしてスイッチＳＷ＿１から交互に出力される送信信号ＳＴは、伝送経路Ｐ＿１、Ｐ＿２を介して逓倍器３０＿１、３０＿２に入力される。逓倍器３０＿ｎは、それぞれ入力された送信信号ＳＴの周波数をミリ波帯の所定周波数７６．５ＧＨｚまで逓倍して出力する。すると、周波数が逓倍された送信信号ＳＴは、増幅器２２＿１、２２＿２により増幅され、送信アンテナＡＴ＿１、ＡＴ＿２から送信される。   In response to the control signal from the control unit 6, the switch SW_1 alternately switches the output destination of the transmission signal ST to the transmission paths P_1 and P_2 in a time division manner. Thus, the transmission signal ST output alternately from the switch SW_1 is input to the multipliers 30_1 and 30_2 via the transmission paths P_1 and P_2. The multiplier 30_n multiplies the frequency of the input transmission signal ST to a predetermined frequency of 76.5 GHz in the millimeter wave band and outputs the result. Then, the frequency-multiplied transmission signal ST is amplified by the amplifiers 22_1 and 22_2 and transmitted from the transmission antennas AT_1 and AT_2.

このように、制御部６がスイッチＳＷ＿１の出力先を交互に切り替えることで、ミリ波帯の所定周波数の送信信号ＳＴが送信アンテナＡＴ＿１、ＡＴ＿２から交互に送信される。   As described above, the control unit 6 alternately switches the output destination of the switch SW_1, so that the transmission signal ST having a predetermined frequency in the millimeter wave band is alternately transmitted from the transmission antennas AT_1 and AT_2.

一方、分配器１０により電力分配されたロ−カル信号ＳＬは、伝送経路Ｐ＿４により伝送され、増幅器１２＿２による増幅を経て、逓倍器３０＿３に入力される。そして、逓倍器３０＿３は、入力されたロ−カル信号ＳＴの周波数を、送信信号ＳＴと同じ周波数７６．５ＧＨｚまで逓倍してミキサ１６＿ｎに出力する。   On the other hand, the local signal SL distributed by the distributor 10 is transmitted through the transmission path P_4, is amplified by the amplifier 12_2, and is input to the multiplier 30_3. Then, the multiplier 30_3 multiplies the frequency of the input local signal ST to the same frequency 76.5 GHz as the transmission signal ST, and outputs it to the mixer 16_n.

送信信号として送信されたレ−ダ信号が物標により反射されると、受信信号ＳＲ＿ｎとして受信アンテナＡＲ＿ｎにより受信される。受信信号ＳＲ＿ｎは、増幅器１４＿ｎにより増幅され、ミキサ１６＿ｎに入力される。   When the radar signal transmitted as the transmission signal is reflected by the target, it is received by the reception antenna AR_n as the reception signal SR_n. The received signal SR_n is amplified by the amplifier 14_n and input to the mixer 16_n.

ミキサ１６＿ｎは、送信信号ＳＴと同じ周波数のロ−カル信号ＳＬと受信信号ＳＲ＿ｎとを混合して、送信信号ＳＴと受信信号ＳＲ＿ｎのそれぞれとの周波数差信号（ビ−ト信号）ＳＢ＿ｎを生成する。   The mixer 16_n mixes the local signal SL having the same frequency as the transmission signal ST and the reception signal SR_n to generate a frequency difference signal (bit signal) SB_n between the transmission signal ST and the reception signal SR_n. .

図８は、各信号の時間（横軸）に対する周波数（縦軸）の変化を示す。送信信号ＳＴは上述したように三角波状の変調信号に従い周波数変調される。よって、図８（Ａ）に実線で示すように、送信信号ＳＴの周波数は時間に対し直線的に漸次上昇または下降する。そして、受信信号ＳＲ＿ｎは、図８（Ａ）に点線で示すように、レ−ダ信号が物標との距離を往復する時間ΔＴだけ遅延し、物標との相対速度に応じた周波数ΔＤだけドップラシフトする。   FIG. 8 shows changes in frequency (vertical axis) with respect to time (horizontal axis) of each signal. As described above, the transmission signal ST is frequency-modulated according to a triangular wave-like modulation signal. Therefore, as indicated by a solid line in FIG. 8A, the frequency of the transmission signal ST gradually increases or decreases linearly with respect to time. The received signal SR_n is delayed by a time ΔT in which the radar signal reciprocates the distance from the target as shown by a dotted line in FIG. 8A, and only by a frequency ΔD corresponding to the relative speed with the target. Doppler shift.

このような受信信号ＳＲと送信信号ＳＴとを混合すると、図８（Ｂ）に示すように、両者の周波数差に対応した周波数を有するビ−ト信号ＳＢ＿ｎが生成される。送信信号ＳＴの周波数が上昇する時（以下、周波数上昇期間という）のビ−ト信号の周波数（以下「アップビ−ト周波数」という）ｆｕと、送信信号ＳＴの周波数が下降する時（以下、周波数下降期間という）のビ−ト信号の周波数（以下「ダウンビ−ト周波数」という）ｆｄとから、次の式（１）により物標の相対距離Ｒｄが、式（２）により相対速度Ｒｖが求められる。ここでは、Ｃは電波伝搬速度（光速）、ｆｍは送信信号ＳＴの変調周波数、ΔＦは送信信号ＳＴの周波数変動幅、ｆｏは送信信号ＳＴの中心周波数である。   When such a reception signal SR and transmission signal ST are mixed, a beat signal SB_n having a frequency corresponding to the frequency difference between the two is generated as shown in FIG. 8B. When the frequency of the transmission signal ST increases (hereinafter referred to as a frequency increase period), the frequency of the beat signal (hereinafter referred to as “upbeat frequency”) fu and when the frequency of the transmission signal ST decreases (hereinafter referred to as frequency). The relative distance Rd of the target is obtained from the following equation (1) and the relative velocity Rv is obtained from the equation (2) from the frequency (hereinafter referred to as “downbeat frequency”) fd of the beat signal during the descent period). It is done. Here, C is a radio wave propagation speed (light speed), fm is a modulation frequency of the transmission signal ST, ΔF is a frequency fluctuation range of the transmission signal ST, and fo is a center frequency of the transmission signal ST.

（１） Ｒｄ＝Ｃ・（ｆｕ＋ｆｄ）／（８・ΔＦ・ｆｍ）
（２） Ｒｖ＝Ｃ・（ｆｄ−ｆｕ）／（４・ｆｏ）
このようなビ−ト信号ＳＢ＿ｎは、ミキサ１６＿ｎから、送信信号ＳＴの変調周期毎に、Ａ／Ｄ変換器１８＿ｎを介して制御部６に入力される。 (1) Rd = C · (fu + fd) / (8 · ΔF · fm)
(2) Rv = C · (fd−fu) / (4 · fo)
Such a beat signal SB_n is input from the mixer 16_n to the control unit 6 via the A / D converter 18_n for each modulation period of the transmission signal ST.

制御部６は、送信信号ＳＴの変調周期毎に、ビ−ト信号ＳＢ＿ｎをＦＦＴ処理する。すると、ビ−ト信号ＳＢ＿ｎのそれぞれからは、アップ／ダウンビ−ト周波数ごとに、位相と振幅とが検出される。そして、制御部６は、アップ／ダウンビ−ト周波数ごとに、ビ−ト信号ＳＢ＿ｎを用いてＤＢＦ処理を行う。   The control unit 6 performs FFT processing on the beat signal SB_n for each modulation period of the transmission signal ST. Then, from each of the beat signals SB_n, the phase and amplitude are detected for each up / down beat frequency. Then, the control unit 6 performs DBF processing using the beat signal SB_n for each up / down beat frequency.

ここで、図９を用いて、ＤＢＦ処理について説明する。受信アンテナＡＲ＿ｎの間隔（ｄとする）に比して十分遠方にある物標から到来する受信信号ＳＲ＿ｎは、受信アンテナＡＲ＿ｎに平行に入射するとみなされる。ここで、受信アンテナＡＲ＿ｎが配列された受信面Ｈ１の垂直方向に対し方位角θから受信信号ＳＲ＿ｎが入射したとすると、隣接する受信アンテナＡＲ＿ｎ間での受信信号ＳＲ＿ｎの間には、アンテナ間隔ｄに応じた位相差φ（＝２π・ｄ・ｓｉｎ（θ）／λ、但しλは受信信号ＳＲ＿ｎの波長）が生じる。   Here, the DBF process will be described with reference to FIG. A reception signal SR_n arriving from a target that is sufficiently far away from the interval (d) of the reception antenna AR_n is considered to be incident on the reception antenna AR_n in parallel. Here, assuming that the received signal SR_n is incident from the azimuth angle θ with respect to the vertical direction of the receiving surface H1 on which the receiving antenna AR_n is arranged, the antenna interval d is between the received signals SR_n between the adjacent receiving antennas AR_n. Phase difference φ (= 2π · d · sin (θ) / λ, where λ is the wavelength of the received signal SR_n).

ここで、受信面Ｈ１においては、受信信号ＳＲ＿ｎは位相差φを有するので、受信信号ＳＲ＿ｎの振幅は互いに打ち消しあう部分が生じる。しかし、受信信号ＳＲ＿ｎの到来方向に対する垂直面（受信面Ｈ１に対し角度θの面）Ｈ２においては、すべての受信信号ＳＲ＿ｎは同位相となる（以下、面Ｈ２を同位相面という）。よって、同位相面Ｈ２上では、受信信号ＳＲ＿ｎは互いに振幅を強め合い、その合成振幅は最大となる。このことは、同位相面Ｈ２の角度θが、受信アンテナＡＲ＿ｎの受信利得が最大となる（つまり指向性が一致する）方位角に対応することを意味し、かかる同位相面の角度θを算出することで、受信信号ＳＲ＿ｎの到来方向、つまり物標が位置する方位角が求められる。   Here, on the reception surface H1, since the reception signal SR_n has a phase difference φ, there is a portion where the amplitudes of the reception signals SR_n cancel each other. However, all the received signals SR_n have the same phase (hereinafter, the plane H2 is referred to as the same phase plane) in the plane H2 (plane having an angle θ with respect to the reception plane H1) H2 with respect to the arrival direction of the received signal SR_n. Therefore, on the same phase plane H2, the reception signals SR_n intensify each other and the combined amplitude becomes maximum. This means that the angle θ of the in-phase surface H2 corresponds to the azimuth angle at which the reception gain of the receiving antenna AR_n is maximized (that is, the directivity is the same), and the angle θ of the in-phase surface is calculated. Thus, the arrival direction of the received signal SR_n, that is, the azimuth angle at which the target is located is obtained.

このことを利用し、制御部６は、まず、周波数上昇期間において受信信号ＳＲ＿ｎから得られたアップビ−ト周波数のビ−ト信号ＳＢ＿ｎの合成振幅を求める。ここで、各受信アンテナＡＲ＿ｎでのビート信号ＳＢ＿ｎの振幅をＥ＿ｎとすると、ビート信号ＳＢ＿１を基準としたときのＳＢ＿ｎ（ｎ＝２、３、４）の位相φｎは、φｎ＝２π・ｄ・（ｎ−１）・ｓｉｎ（θ）／λであるので、すべてのビ−ト信号ＳＢ＿ｎ（ｎ＝１、２、３、４）の合成振幅Ｅは式（３）で得られる。   Using this, the control unit 6 first obtains the combined amplitude of the upbeat frequency beat signal SB_n obtained from the reception signal SR_n during the frequency rise period. Here, assuming that the amplitude of the beat signal SB_n at each receiving antenna AR_n is E_n, the phase φn of SB_n (n = 2, 3, 4) with reference to the beat signal SB_1 is φn = 2π · d · ( Since (n-1) · sin (θ) / λ, the combined amplitude E of all the bit signals SB_n (n = 1, 2, 3, 4) is obtained by Expression (3).

Figure 2009162521
Figure 2009162521

ところで、式（３）ではφｉ＝２π・ｄ・（ｉ−１）・ｓｉｎ（θ）／λであることから、合成振幅Ｅはθの関数として表される。よって、制御部６は、かかる関数において合成振幅Ｅの最大値に対応するθを算出する。すると、その算出結果が、周波数上昇期間における、物標が位置する方位角に対応する。   By the way, in Equation (3), φi = 2π · d · (i−1) · sin (θ) / λ, and therefore, the combined amplitude E is expressed as a function of θ. Therefore, the control unit 6 calculates θ corresponding to the maximum value of the combined amplitude E in such a function. Then, the calculation result corresponds to the azimuth angle at which the target is located during the frequency increase period.

制御部６は、同様の処理を周波数下降期間においてダウンビ−ト周波数のビ−ト信号ＳＢ＿ｎに対しても行う。すると、周波数下降期間における、物標が位置する方位角が求められる。   The control unit 6 performs the same processing for the beat signal SB_n having the downbeat frequency in the frequency falling period. Then, the azimuth angle at which the target is located during the frequency drop period is obtained.

このような処理が、周波数上昇期間と周波数下降期間ごとに行われる。すると、周波数上昇期間と周波数下降期間ごとに、方位角方向と周波数方向とにおけるビ−ト信号ＳＢ＿ｎの強度分布が得られる。   Such processing is performed for each frequency increase period and frequency decrease period. Then, the intensity distribution of the beat signal SB_n in the azimuth angle direction and the frequency direction is obtained for each frequency increase period and frequency decrease period.

次に、制御部６は、方位角方向における強度分布がピ−クを形成する周波数を、アップ／ダウンビ−ト周波数として検出する。このとき、対応づけられた両ピ−クの方位角が、物標が位置する方位角として検出される。   Next, the control unit 6 detects the frequency at which the intensity distribution in the azimuth direction forms a peak as the up / down beat frequency. At this time, the azimuth angles of the two peaks associated with each other are detected as the azimuth angle at which the target is located.

また、制御部６は、周波数上昇期間と周波数下降期間の間で、強度または方位角が一致するピ−ク同士を対応付けることにより、アップ／ダウンビ−ト周波数を特定し、上述の式（１）、（２）に従って物標の相対距離と相対速度を算出する。   Further, the control unit 6 specifies the up / down beat frequency by associating peaks having the same intensity or azimuth angle between the frequency increase period and the frequency decrease period, and the above-described equation (1). The relative distance and relative speed of the target are calculated according to (2).

なお、レ−ダ装置２は、図１０に示すように、伝送経路Ｐ＿４により伝送されるロ−カル信号ＳＬを交互に複数の伝送経路Ｔ＿１、Ｔ＿２、Ｔ＿３、Ｔ＿４に出力するスイッチＳＷ＿３と、スイッチＳＷ＿３を通過したロ−カル信号ＳＬの周波数を逓倍してからミキサに入力する逓倍器３２＿１、３２＿２、３２＿３、３２＿４とを設ける構成としてもよい。そして、スイッチＳＷ＿３からの出力タイミングは制御部６により制御され、ロ−カル信号ＳＬは時分割で逓倍器３２＿ｎに入力される。   As shown in FIG. 10, the radar apparatus 2 includes a switch SW_3 that alternately outputs a local signal SL transmitted through the transmission path P_4 to a plurality of transmission paths T_1, T_2, T_3, and T_4, and a switch Multipliers 32_1, 32_2, 32_3, and 32_4 that multiply the frequency of the local signal SL that has passed through SW_3 and then input to the mixer may be provided. The output timing from the switch SW_3 is controlled by the controller 6, and the local signal SL is input to the multiplier 32_n in a time division manner.

ここでは、マイクロ波帯のロ−カル信号ＳＬがスイッチＳＷ＿３を通過した後に、その周波数が逓倍器３２＿ｎによりミリ波帯の所定周波数まで逓倍される。よって、スイッチＳＷ＿３による挿入損失は、ミリ波帯のままでロ−カル信号ＳＬが通過する場合より小さく抑えられる。また、伝送経路Ｐ＿４、Ｔ＿ｎにおける損失も、ミリ波帯のままでロ−カル信号ＳＬが通過する場合より小さく抑えられる。よって、ミキサ１６＿ｎは、十分な強度のロ−カル信号ＳＬを用いて、ビ−ト信号ＳＢ＿ｎを生成することができる。   Here, after the local signal SL in the microwave band passes through the switch SW_3, the frequency is multiplied to a predetermined frequency in the millimeter wave band by the multiplier 32_n. Therefore, the insertion loss due to the switch SW_3 is suppressed to be smaller than when the local signal SL passes in the millimeter wave band. In addition, the loss in the transmission paths P_4 and T_n is also suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Therefore, the mixer 16_n can generate the beat signal SB_n using the local signal SL with sufficient strength.

図１１は、本実施形態におけるレ−ダ装置２の、第２の構成例を説明する図である。このレ−ダ装置２は、次の点において、図６で示した第１の構成例におけるレ−ダ装置２と異なる。   FIG. 11 is a diagram illustrating a second configuration example of the radar apparatus 2 according to the present embodiment. This radar device 2 is different from the radar device 2 in the first configuration example shown in FIG. 6 in the following points.

すなわち、レ−ダ装置２は、複数のＡ／Ｄ変換器１８＿ｎの代わりに１つのＡ／Ｄ変換器１８と、ミキサ１６＿ｎから出力されるビ−ト信号ＳＢ＿ｎを、制御部６からの指示に従って交互にＡ／Ｄ変換器１８に入力するスイッチＳＷ＿２を有する。かかる構成とすることで、複数のＡ／Ｄ変換器１８＿ｎを有する第１の構成例より、回路規模を小さくできる。よって、低コスト化できる。   That is, the radar apparatus 2 receives one A / D converter 18 instead of the plurality of A / D converters 18_n and the beat signal SB_n output from the mixer 16_n according to an instruction from the control unit 6. It has switch SW_2 which inputs into A / D converter 18 by turns. With such a configuration, the circuit scale can be reduced as compared with the first configuration example having the plurality of A / D converters 18_n. Therefore, the cost can be reduced.

なお、ミキサ１６＿ｎが出力するビ−ト信号ＳＢ＿ｎの周波数は、ミリ波帯の送信信号ＳＴと受信信号ＳＲ＿ｎとの周波数差であるので、ある程度低い周波数帯となる。このため、ビ−ト信号ＳＢ＿ｎがスイッチＳＷ＿２を通過する際の挿入損失は小さく抑えられ、ビ−ト信号ＳＢ＿ｎの信号強度は制御部６での信号処理に支障ない程度に確保される。   Note that the frequency of the beat signal SB_n output from the mixer 16_n is a frequency band lower to some extent because it is the frequency difference between the transmission signal ST and the reception signal SR_n in the millimeter wave band. For this reason, the insertion loss when the beat signal SB_n passes through the switch SW_2 is suppressed to be small, and the signal strength of the beat signal SB_n is ensured to such an extent that the signal processing in the control unit 6 is not hindered.

なお、このレ−ダ装置２は、図１２に示すように、ロ−カル信号ＳＬを交互に複数の伝送経路Ｔ＿ｎに出力するスイッチＳＷ＿３と、スイッチＳＷ＿３を通過したロ−カル信号ＳＬの周波数を逓倍してからミキサに入力する逓倍器３２＿ｎとをさらに設ける構成としてもよい。この場合、スイッチＳＷ＿３の入出力タイミングは制御部６により制御され、ロ−カル信号ＳＬは、時分割でミキサに入力される。   As shown in FIG. 12, the radar apparatus 2 uses the switch SW_3 that alternately outputs the local signal SL to a plurality of transmission paths T_n, and the frequency of the local signal SL that has passed through the switch SW_3. A multiplier 32_n that multiplies and then inputs to the mixer may be provided. In this case, the input / output timing of the switch SW_3 is controlled by the control unit 6, and the local signal SL is input to the mixer in a time division manner.

ここでは、図１０に示した第１の構成例の変形例と同様、マイクロ波帯のロ−カル信号ＳＬがスイッチＳＷ＿３を通過した後に、その周波数が逓倍器３２＿ｎによりミリ波帯の所定周波数まで逓倍されるので、スイッチＳＷ＿３による挿入損失は、ミリ波帯のままでロ−カル信号ＳＬが通過する場合より小さく抑えられる。また、伝送経路Ｐ＿４、Ｔ＿ｎによる損失も、ミリ波帯のままでロ−カル信号ＳＬが通過する場合より小さく抑えられる。よって、ミキサ１６＿ｎは、十分な強度のロ−カル信号ＳＬを用いて、ビ−ト信号ＳＢ＿ｎを生成することができる。   Here, as in the modification of the first configuration example shown in FIG. 10, after the local signal SL in the microwave band passes through the switch SW_3, the frequency is increased to a predetermined frequency in the millimeter wave band by the multiplier 32_n. Since the frequency is multiplied, the insertion loss due to the switch SW_3 is suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Further, the loss due to the transmission paths P_4 and T_n is also suppressed to be smaller than when the local signal SL passes in the millimeter wave band. Therefore, the mixer 16_n can generate the beat signal SB_n using the local signal SL with sufficient strength.

図１３は、本実施形態におけるレ−ダ装置が車載用レ−ダ装置として使用される使用状況を説明する図である。レ−ダ装置２は、車両１００の前部フロントグリル内に搭載され、送信信号ＳＴ、受信信号ＳＲ＿ｎを、車両１００のバンパ−やフロントグリルに設けられるレド−ムを透過して送受信することにより、車両１００前方（矢印Ｆ）における先行車両や障害物といった物標が位置する方位角や、物標との相対速度、相対距離を検出する。そして、上述したようにして制御部６が検出した物標が位置する方位角、物標との相対距離、相対速度といった物標情報は、車両１００の車両制御装置（ＥＣＵ：Electronic Control Unit）２００に出力される。   FIG. 13 is a diagram illustrating a usage situation in which the radar device according to the present embodiment is used as a vehicle-mounted radar device. The radar device 2 is mounted in the front front grill of the vehicle 100, and transmits and receives the transmission signal ST and the reception signal SR_n through the radar provided on the bumper and the front grill of the vehicle 100. Then, an azimuth angle at which a target such as a preceding vehicle or an obstacle in front of the vehicle 100 (arrow F) is located, a relative speed and a relative distance with the target are detected. The target information such as the azimuth angle at which the target detected by the control unit 6 is located as described above, the relative distance to the target, and the relative speed is the vehicle control device (ECU: Electronic Control Unit) 200 of the vehicle 100. Is output.

車両ＥＣＵ２００は、これらの検出結果に基づいて、車両１００のスロットル、ブレ−キといったアクチュエ−タを制御して先行車両に追従走行するよう車両の速度を加減し、追従走行制御を行う。また、車両ＥＣＵ２００は、先行車両や障害物との距離が一定以上接近し衝突の蓋然性が大きくなるような場合に、運転者への通知を出力したり、エアバッグなどの安全装置を作動させたりして、衝突対応動作を行う。   Based on these detection results, the vehicle ECU 200 controls the actuators such as the throttle and brake of the vehicle 100 to adjust the vehicle speed so as to follow the preceding vehicle and perform follow-up running control. In addition, the vehicle ECU 200 outputs a notification to the driver or activates a safety device such as an airbag when the distance from the preceding vehicle or an obstacle approaches a certain distance or more and the probability of a collision increases. Then, the collision handling operation is performed.

なお、レーダ装置２は、車両１００の前方監視だけでなく、側方や後方を監視するためのレーダ装置として、車両１００の側面部や後部に設置してもよい。   Note that the radar apparatus 2 may be installed not only on the front side of the vehicle 100 but also on the side surface or the rear part of the vehicle 100 as a radar apparatus for monitoring the side or the rear.

また、上述の構成例では、４つの受信アンテナＡＲ＿１、ＡＲ＿２、ＡＲ＿３、ＡＲ＿４を備えたレ−ダ装置２を例としたが、受信アンテナ数は、これに限られない。   In the above configuration example, the radar apparatus 2 including the four reception antennas AR_1, AR_2, AR_3, and AR_4 is taken as an example, but the number of reception antennas is not limited thereto.

さらに、ＶＣＯ８が出力するレ−ダ信号の周波数帯は上述の例に限られず、必ずしもマイクロ波帯でなくてもよい。また、これに対応して、逓倍器３２＿ｎが逓倍する率も、上述の例に限られず、必ずしもミリ波帯まで逓倍しなくてもよい。スイッチや伝送経路を通過するときの信号の周波数を低くし、その後にこれを逓倍する構成であれば、スイッチの挿入損失による送信信号の強度低下、または、伝送損失による送信信号の強度低下を防ぐという作用効果を奏することができる。   Further, the frequency band of the radar signal output from the VCO 8 is not limited to the above example, and may not necessarily be the microwave band. Correspondingly, the rate of multiplication by the multiplier 32_n is not limited to the above example, and it is not always necessary to multiply to the millimeter wave band. If the configuration is such that the frequency of the signal passing through the switch or transmission path is lowered and then multiplied, it prevents the transmission signal strength from being reduced due to the insertion loss of the switch or the transmission signal from being reduced due to the transmission loss. It is possible to achieve the operational effects.

以上説明したように、本実施形態におけるレーダ装置によれば、所定間隔離間した送信アンテナ対に送信信号を伝送する伝送経路や、伝送経路に送信信号を分配するスイッチを用いても、探知性能の低下を防止できる。   As described above, according to the radar apparatus of the present embodiment, even if a transmission path for transmitting a transmission signal to a pair of transmission antennas separated by a predetermined interval or a switch for distributing the transmission signal to the transmission path is used, the detection performance can be improved. Decline can be prevented.

従来の電子スキャン式レ−ダ装置の構成例を示す図である。It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus. 図１と異なる従来の電子スキャン式レ−ダ装置の構成例を示す図である。It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus different from FIG. 送信アンテナ対と受信アンテナ群の位置関係を説明する図である。It is a figure explaining the positional relationship of a transmitting antenna pair and a receiving antenna group. 図２と異なる従来の電子スキャン式レ−ダ装置の構成例を示す図である。It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus different from FIG. 図４と異なる従来の電子スキャン式レ−ダ装置の構成例を示す図である。It is a figure which shows the structural example of the conventional electronic scan type | mold radar apparatus different from FIG. 本実施形態におけるレ−ダ装置の、第１の構成例を説明する図である。It is a figure explaining the 1st structural example of the radar apparatus in this embodiment. スイッチの挿入損失、伝送経路の伝送損失について説明する図である。It is a figure explaining the insertion loss of a switch, and the transmission loss of a transmission path. 送受信信号の時間に対する周波数の変化を示す図である。It is a figure which shows the change of the frequency with respect to the time of a transmission / reception signal. ＤＢＦ処理について説明する図である。It is a figure explaining DBF processing. 第１の構成例の変形例を説明する図である。It is a figure explaining the modification of the 1st example of composition. 本実施形態におけるレ−ダ装置の、第２の構成例を説明する図である。It is a figure explaining the 2nd structural example of the radar apparatus in this embodiment. 第２の構成例の変形例を説明する図である。It is a figure explaining the modification of the 2nd example of composition. レ−ダ装置が車載用レ−ダ装置として使用される使用状況を説明する図である。It is a figure explaining the use condition in which a radar apparatus is used as a vehicle-mounted radar apparatus.

符号の説明Explanation of symbols

２：レーダ装置、ＡＴ＿ｎ：送信アンテナ、ＡＲ＿ｎ：受信アンテナ、６：制御部、８：ＶＣＯ、ＳＷ＿ｎ：スイッチ、Ｐ＿ｎ、Ｔ＿ｎ：伝送経路、３０＿ｎ、３２＿ｎ：逓倍器 2: radar device, AT_n: transmitting antenna, AR_n: receiving antenna, 6: control unit, 8: VCO, SW_n: switch, P_n, T_n: transmission path, 30_n, 32_n: multiplier

Claims (4)

所定周波数のレ−ダ信号を送信する所定間隔離間した送信アンテナ対と、物標により反射された前記レーダ信号を受信信号として受信する受信アンテナ群とを有し、前記送信アンテナ対から交互に前記レーダ信号を送信して得られる受信信号を用いて前記物標が位置する方位角を検出するレ−ダ装置において、
前記所定周波数より低い周波数のレ−ダ信号を出力する発振器と、
前記発振器が出力するレ−ダ信号を、第１の前記送信アンテナへの第１の伝送経路と、第２の前記送信アンテナへの第２の伝送経路とに分配するスイッチと、
前記第１の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第１の送信アンテナに出力する第１の逓倍器と、前記第２の伝送経路に接続され、前記レ−ダ信号の周波数を前記所定周波数に変換してから前記第２の送信アンテナに出力する第２の逓倍器とを有することを特徴とするレ−ダ装置。 A transmitting antenna pair spaced by a predetermined interval for transmitting a radar signal of a predetermined frequency; and a receiving antenna group for receiving the radar signal reflected by a target as a received signal; and alternately from the transmitting antenna pair In a radar apparatus for detecting an azimuth angle at which the target is located using a received signal obtained by transmitting a radar signal,
An oscillator that outputs a radar signal having a frequency lower than the predetermined frequency;
A switch that distributes the radar signal output from the oscillator to a first transmission path to the first transmission antenna and a second transmission path to the second transmission antenna;
A first multiplier that is connected to the first transmission path, converts the frequency of the radar signal to the predetermined frequency, and then outputs the first signal to the first transmission antenna, and is connected to the second transmission path And a second multiplier for converting the frequency of the radar signal into the predetermined frequency and outputting the second signal to the second transmitting antenna. 請求項１において、
前記スイッチは、これを通過するレ−ダ信号の周波数が第１の周波数のときは第１の挿入損失、前記第１の周波数より大きい第２の周波数のときは前記第の挿入損失より大きい第２の挿入損失を当該レ−ダ信号に与えることを特徴とするレ−ダ装置。 In claim 1,
The switch has a first insertion loss when the frequency of the radar signal passing through the first frequency is a first frequency, and is larger than the first insertion loss when the frequency is a second frequency higher than the first frequency. 2. A radar apparatus, wherein an insertion loss of 2 is given to the radar signal. 請求項１または２において、
前記所定周波数のレーダ信号と前記受信信号との周波数差信号を出力するミキサと、
前記発振器が出力するレ−ダ信号を前記ミキサへの第３の伝送経路に分配する分配器と、
前記第３の伝送経路に接続され、前記レーダ信号を前記所定周波数に変換してから前記ミキサに出力する第３の逓倍器とをさらに有し、
前記周波数差信号を用いて前記物標との相対距離または相対速度をさらに検出することを特徴とするレ−ダ装置。 In claim 1 or 2,
A mixer that outputs a frequency difference signal between the radar signal of the predetermined frequency and the received signal;
A distributor for distributing a radar signal output from the oscillator to a third transmission path to the mixer;
A third multiplier connected to the third transmission path and converting the radar signal to the predetermined frequency and then outputting to the mixer;
A radar apparatus further detecting a relative distance or a relative speed with respect to the target using the frequency difference signal. 請求項１乃至３のいずれかにおいて、
前記所定周波数はミリ波帯であり、前記発振器が出力するレ−ダ信号の周波数はミリ波帯より低いマイクロ波帯であることを特徴とするレ−ダ装置。 In any one of Claims 1 thru | or 3,
The radar apparatus according to claim 1, wherein the predetermined frequency is a millimeter wave band, and a frequency of a radar signal output from the oscillator is a microwave band lower than the millimeter wave band.
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