JP2008523690A - Method and apparatus for performing extended decoding in multiband ultra wideband communication - Google Patents

Abstract

  The data transmission and reception method comprises transmitting a plurality of position data bits and amplitude bits with each pulse. The method also includes decoding position bits and amplitude bits. An ultra-wideband (UWB) system is also described.

Description

  Ultra wideband (UWB) communication involves the transmission of signals that occupy a large band. In UWB systems, the modulated signal is transmitted as a baseband pulse (transmission without a carrier wave) or converted upward (mixed) to a frequency up to a specific carrier frequency. Many UWB applications are limited to radar and military communications. However, due to the possibility of using UWB technology in short-range communications at high data rates, the FCC (Federal Communications Commission) is providing frequency bands from 3.1 GHz to 10.6 GHz for unlicensed equipment.

  UWB communication systems transmit short-term pulses of data in the transmission range described above. As can be seen, the number of frequency components is very large because of the relatively short duration of the pulse in the time domain. This is related to a relatively wide band signal. Thus, a properly designed UWB system provides a significant amount of data transmission in a relatively short time, making the UWB system attractive for high data rate applications.

  Although potentially advantageous, known UWB systems have certain challenges. For example, in one type of UWB system, short duration pulses (usually on the order of psec) are transmitted at the desired rate. This pulse is modulated with the encoded data. When the pulse repetition rate is small, the distance between pulses becomes very large. This is advantageous in a multipath channel because the receiver can resolve different multipath components. However, in order to achieve a high data rate, the repetition rate needs to increase. Due to implementation limitations, this rate cannot increase beyond the limitations.

  Furthermore, in known single-band UWB systems, the signal occupies a large band, so any in-band interference needs to be reduced by using a notch filter. This increases the cost of the device. In order to overcome some of the disadvantages of single band systems, multi-band systems have recently been proposed.

  In a multi-band pulse UWB system, data is modulated into pulses and transmitted in different frequency bands that are interleaved in time. Each of these bands is approximately 500 MHz wide, so up to 14 bands can be used in the allocated spectrum. Compared to single-band systems, multi-band systems provide flexibility in terms of adding / dropping bands based on interference scenarios and data rate requirements.

  The use of a multi-band pulse UWB system is advantageous for the transmission of short duration pulses and provides a relatively wide band over which data can be transmitted. However, as the transmission rate increases, the probability of transmission errors also increases. In order to reduce transmission errors in data transmission, data coding and decoding schemes have been proposed. These coding and decoding schemes (often referred to as forward error correction (FEC) schemes) improve channel capacity and data transmission reliability. The FEC scheme provides redundant information for transmission signals. This is known as channel coding.

  One known channel coding technique is convolutional coding. Convolutional coding operates on serial data and encodes the data for transmission. At the receiver, a method of decoding the received encoded data and recovering the information sequence is used. One such decoding scheme is Viterbi decoding.

  Although known encoding methods are advantageous, there is a need to provide additional reliability with respect to the decoded data.

  The data transmission and reception method includes transmitting a plurality of position data bits and a plurality of amplitude data bits in each of the position data bits, and decoding the position data bits and the amplitude data bits.

  An ultra wideband (UWB) system (400) having a matched filter / correlator (403) passes information about multiple position data bits and information about multiple amplitude data bits to a demapping / convolutional decoder (406). And a demapping / convolutional decoder (406) decodes the position data bits and the amplitude data bits.

  The example embodiments are best understood from the following detailed description when read in conjunction with the accompanying drawings. It is emphasized that the various functions are not necessarily to scale and can be arbitrarily expanded or reduced for clarity of explanation.

  In the following detailed description, for purposes of explanation and not limitation, example embodiments disclosing specific details are set forth in order to provide a thorough understanding of the example embodiments. However, upon review of this disclosure, other embodiments that do not depart from the concept of the described exemplary embodiments will become apparent to those skilled in the art. Moreover, descriptions of well-known devices, elements, methods, and systems are omitted so as not to obscure the description of the exemplary embodiments. Nevertheless, such devices, methods, systems and protocols within the purview of those skilled in the art can be used in accordance with the illustrative embodiments. Finally, where applicable, like reference numerals indicate like features.

  Briefly, exemplary embodiments relate to a method and apparatus for forward error correction in a communication system. In order to improve the reliability of transmitted data by improving the bit error rate (BER), the forward error correction method has the use of amplitude data bits or code bits in the traceback operation. Illustratively, the convolutional coder encodes 1 bit into 2 bits. A coded bit along with one amplitude bit is used to modulate the pulse.

  The receiver demodulates the pulse and inputs information about the position bits and the sign bit to the decoder. As described above, the gain is increased on the order of about 1.5 dB to about 2.0 dB as compared with the coding / decoding method using only the information of position bits, and the reliability of data is improved.

  It should be noted that the exemplary embodiments are described with respect to UWB wireless communication and related elements. It is emphasized that the use of the method and apparatus of the exemplary embodiments is for illustrative purposes only. Obviously, the method and apparatus of the example embodiments can be implemented for other communication applications. Furthermore, the modulation and coding techniques described are also examples of illustrative embodiments. For example, the coding is illustratively Viterbi format coding / modulation. However, other trellis coding modulation (TCM) may be implemented. Of course, other coding modulation techniques will be readily apparent to those skilled in the art. Such techniques can be implemented in harmony with the exemplary embodiments.

FIG. 1 is a conceptual diagram of a time-frequency interleaved multiband UWB system 100 according to an embodiment of the present invention. In this exemplary embodiment, the data modulation pulses are transmitted in frequency band F ₁ 101 during the first time slot. Similarly, in the second time slot, the modulated pulse is transmitted in band F ₃ 102 and during the third time slot, the modulated pulse is transmitted in band F ₅ 103 and during the fourth time slot, The modulated pulse is transmitted in band F ₄ 104 and during the fifth time slot, the modulated pulse is transmitted in band F ₂ . As can be seen, the process is repeated with the transmission of another modulation pulse in band F ₁ 106.

  As mentioned above, it is useful to increase the pulse repetition rate in order to increase the data rate of the system. Further, it may be useful to increase the data rate by increasing the number of bits transmitted per pulse. This can be achieved by using amplitude modulation and / or position modulation techniques well known in the art. A system that combines pulse amplitude modulation (PAM) and pulse position modulation (PPM) in a multi-band UWB system is highly desirable to increase channel throughput. In a PPM-PAM system, some bits are used to modulate the position of the pulse, while the remaining bits are used to modulate the amplitude of the pulse. This can be seen from the examination of FIG. FIG. 2 is a timing diagram of a 4PPM / BPSK (Binary Phase Shit Keying) modulation UWB system.

  In a BPSK modulated pulse, each pulse propagates only one amplitude bit. Thus, the sign of the amplitude provides information about that bit. Therefore, the amplitude bit of the BPSK modulation pulse may be referred to as a sign bit. In contrast, with a 4-PAM modulated pulse, each pulse propagates two amplitude bits. This results in the pulse taking 4 levels. Therefore, at the receiver, the amplitude must be determined in order to determine the two transmitted bits.

The modulation pulse 201 in the first frequency band F ₁ is transmitted during the first time slot 202. The pulse 201 has the illustrated pulse period T _p . Similarly, the modulation pulse 203 of the _third frequency band F3 is transmitted in the second time slot 204, the modulation pulse 205 of the _fifth frequency band F5 is transmitted in the third time slot 206, and the fourth The modulation pulse 207 in the frequency band F ₄ is transmitted in the fourth time slot 204, and the modulation pulse 208 in the _second frequency band F ₂ is transmitted in the fifth time slot 209. The process is repeated and a modulated pulse 211 of the first frequency band F ₁ is transmitted in the first slot of the sequence.

Based on design requirements, the PPM shift T _c may be smaller than the pulse period T _p or larger than the pulse period. PPM shift affects receiver performance. For example, if a correlation-based receiver is used to modulate the data of pulses 201, 203, 205, 207, 209 and 211, the performance of the amplitude and PPM bits depends on the shift. If the shift period is greater than the pulse period, the PPM bit performance is better than the amplitude bit performance. When the shift period is smaller than the pulse period, the amplitude bit performance is better than the PPM bit performance.

  At the receiver, the correlator calculates the correlation between all possible positions and the received pulse. If there is no overlap between different pulse positions (ie if the shift period is greater than the pulse period), the correlation value is clear (assuming a clean channel). The correlation corresponding to the position of the transmitted pulse is substantially maximized, but the remaining correlation values are relatively small.

  However, if there is an overlap between different pulse positions, the correlation at any pulse position will have some contribution from neighboring positions. Therefore, the position information in this case is not as reliable as the position information in other cases.

  For amplitude bits, the integration period for shift periods greater than the pulse width is greater than for other cases. This results in slightly worse performance than the other cases because it adds further noise.

  Normally, the PPM shift period is smaller than the pulse period in order to consider multipath interference. As described above, in this example, the amplitude bits are more robust than the position bits. Channel coding may be used with PPM bits to improve the performance of position bits and make this performance comparable to the performance of amplitude bits.

  As mentioned above, one way to code the channel is to use a convolutional encoder or similar trellis type coding scheme. In the exemplary embodiment described herein, the modulation scheme used with trellis coding / decoding is a 4-PPM / BPSK system. 4-PPM / BPSK is an example of a PPM-PAM modulation system. Again, this is merely an example and it is emphasized that other types and combinations of amplitude modulation and m-PPM modulation schemes may be used.

  The 4-PPM / BPSK system divides the information bits as PPM bits and amplitude / code bits. PPM bits are encoded using a convolutional encoder. As is known, convolutional encoders are usually described in terms of two parameters (code rate and constraint length). The coding length (k / n) is expressed as a ratio of the number of bits to the convolutional encoder with respect to the number of channel signals output by the encoder in a predetermined cycle. For example, a convolutional encoder that receives one bit and outputs two channel signals is called a 1/2 rate convolutional coder.

  The constraint length parameter K indicates the length of the convolutional encoder. That is, the constraint length parameter indicates how many steps are available to supply to the combinational logic that creates the output symbol. Finally, a parameter (m) closely related to K is the number of encoder cycles that are used for encoding with the input bits held after first appearing at the encoder input. This is often referred to as the encoder's memory.

  In this exemplary embodiment, the convolutional encoder is a 1/2 rate convolutional coder. The coder constraint length may be selected based on performance requirements. One way to specify coder performance is by bit error rate (BER) for a given signal to noise ratio (SNR). For example, if one encoder provides a BER = 0.0001 with SNR = 5 dB and the other provides a BER of 0.001 with the same SNR, the first coder is advantageous from a performance standpoint. However, this performance results in additional complexity / hardware costs to name only a few factors that affect costs. Thus, many considerations are made when an encoder is selected.

  The output of the convolutional coder passes through a Gray coder that maps 2 bits to one of four positions, as shown in Table 1 of FIG. This mapping can be understood from a review of FIGS. For example, consider the first time slot 202. The position bit is encoded into two encoded bits ‘11’ and, when Gray encoded, results in ‘2’ in Table 1. Thus, the pulse is transmitted at position 2. In this case, the sign / amplitude bit is 1. Therefore, a pulse having a positive amplitude is transmitted.

The position determines the relative shift in the position of the pulse. The output of the mapper is sent to the PPM modulator, where it is used to shift the position of the pulses. The pulses are multiplied by BPSK modulated code bits. Since the two information bits are encoded into three data bits (two position bits (b ₀ , b ₁ ) and one sign or amplitude bit (b ₂ )), the effective rate of this system is 2/3 is there.

  The output of the PPM modulator is mixed to a specified frequency and transmitted using well known transmission techniques and equipment. The order of the frequency band is determined in advance (the order in FIG. 1) and does not change during the transmission period. The transmission signal can be expressed as:

ただし、p(t)は基本パルスであり、T_bはパルス繰り返し期間であり、T_cは最小シフト期間であり、a_nは符号ビットから導かれ（符号ビット=1の場合にa_n=1、符号ビット=0の場合にa_n=-1）、c_nはシフト量であり符号化ビット（図３の表１）から導かれる。

However, p (t) is the basic pulse, T _b is the pulse repetition period, T _c is the minimum shift time, a _n is derived from the sign bit (a _n = 1 if the sign bit = 1 In the case of sign bit = 0, a _n = −1), and c _n is a shift amount and is derived from the coded bits (Table 1 in FIG. 3).

For 4-PPM modulation, c _n = 0, 1, 2, or 3;

Outside 1

はT_b秒毎に変化する搬送周波数である。搬送周波数はサブセット｛F₁、F₃、F₅、F₂及びF₄｝から選択される。１つのセットの搬送周波数は、f_n=3.5×10⁹+500×10⁶*(n-1)で与えられる。

Is the carrier frequency that changes every _Tb seconds. The carrier frequency is selected from the subset {F ₁ , F ₃ , F ₅ , F ₂ and F ₄ }. One set of carrier frequencies is given by f _n = 3.5 × 10 ⁹ + 500 × 10 ⁶ * (n−1).

The transmission pulses are received at a receiver 400 according to the exemplary embodiment shown in FIG. The receiver 400 includes a mixer 401 coupled to a local oscillator 402 that substantially inputs one of F ₁ -F ₅ depending on the frequency of the pulse. The output of the mixer 401 is input to the matched filter / correlator 403. Correlator 403 outputs correlation value m _k 404, and amplitude / code information a _k 405 is input to demapping / convolutional decoder 406. Matched filter / correlator 402 correlates the shifted version of the pulse template with the received signal and provides the following correlation values:

ただし、ここに記載の例示的な実施例の4-PPM/BPSKシステムではk=0,1,2及び3である。例示的な実施例の残りの説明はnに無関係であるため、m_kがm_k(n)の代わりに使用される。

However, k = 0, 1, 2, and 3 in the exemplary 4-PPM / BPSK system described herein. Since the remaining description of the exemplary embodiment is independent of n, m _k is used instead of m _k (n).

FIG. 5 shows the output 500 of the correlator 403 for an exemplary input pulse with the illustrated coding. For example, a modulation pulse 501 in the first frequency band has a coded bit 10 and amplitude / sign bit 1, and a pulse 502 in the third frequency band has a coded bit 01 and amplitude / sign bit 1. The modulation pulse 503 in the fifth frequency band has a coded bit 00 and an amplitude / sign bit 0. From Equation 2, the correlator 403 calculates m _{0 to} m ₃ for each result shown in FIG.

The position of the pulse and the sign bit of the transmission pulse are derived from m _k shown in equations 3 and 4, respectively.

ブロック406のデマッピング器は、位置信号を２つのデータビットに変換する。ブロック406の畳み込みデコーダは、２ビットの入力データからPPM（位置）ビットをデコードするために、Viterbiアルゴリズム又は同様のトレリスデコード方法を使用する。前述のように、このデコード方式の性能は最適値より小さい。性能は、Viterbiデコーダのソフトメトリックを有する例示的な実施例の方法及び装置により改善される。これらの例示的な実施例についてここで説明する。

The demapping unit at block 406 converts the position signal into two data bits. The convolutional decoder at block 406 uses the Viterbi algorithm or similar trellis decoding method to decode the PPM (position) bits from the two bits of input data. As described above, the performance of this decoding method is smaller than the optimum value. The performance is improved by the method and apparatus of the exemplary embodiment with the Viterbi decoder soft metric. These exemplary embodiments are now described.

  According to an exemplary embodiment, a method and apparatus for improving the performance of PPM bits and amplitude bits incorporates the use of soft metrics and incorporates a sign bit in a trellis traceback sequence. In the above coding scheme, the transmitter provides all coding gains for position bits and essentially no coding gain for sign bits. To improve the performance of the sign bit, the receiver 400 also includes the sign bit in the trellis traceback. This provides an improvement of about 1.5-2.0 dB in system performance compared to a soft decision decode receiver with only PPM bits in the path memory (traceback).

In the receiver 400 of the exemplary embodiment of FIG. 4, no decision is made by the matched filter / correlator 403. Rather, the correlator 403 generates all relevant information for the demapping / decoder 406, including the correlation value and code value. Therefore, for k = 0, 1, 2, and 3, the matched filter calculates the correlation value m _k given by Equation 2. Further, the matched filter calculates the corresponding code information a _k derived from m _k using the following equation:

既知のように、ソフトメトリックは、デコーダのビットについての更なる情報を提供する何らかのメトリックである。表１に示すように定義されたマッピングでは、ビットb₁及びb₀のソフトメトリックは式6を使用して式2のm_kから計算され得る。例示的な実施例では、ソフトメトリックは式6及び7(a)-7(b)により与えられる。

As is known, a soft metric is any metric that provides further information about the bits of the decoder. For the mapping defined as shown in Table 1, the soft metrics for bits b ₁ and b ₀ can be calculated from m _{k in} Equation 2 using Equation 6. In the exemplary embodiment, the soft metric is given by equations 6 and 7 (a) -7 (b).

ただし、mb_xyはビットx=yのメトリックを表す。これから、それぞれビットb₀及びb₁のソフトメトリックsm₀及びsm₁が以下のように計算され得る。

Here, mb _xy represents a metric of bit x = y. From this, the soft metrics sm ₀ and sm ₁ for bits b ₀ and b ₁ respectively can be calculated as follows:

式6又は式7a-7bからのソフトメトリックは、Viterbi計算の分岐メトリックを計算するために使用され得る。

The soft metric from Equation 6 or Equation 7a-7b can be used to calculate the branch metric for Viterbi calculations.

In this illustrative example, the path metric of the nodes of the (i-1) is represented by _{pm i-1 (s')} , the path metric of the nodes of the i is represented by pm _i (s). Here, s ′ and s are general states at the (i−1) -th and i-th nodes, respectively. The path metric is calculated using the following formula:

ただし、bm_i(s’,s)は、ノードi-1の状態s’からノードiの状態sへの分岐の分岐メトリックである。

However, bm _i (s ′, s) is a branch metric of a branch from the state s ′ of the node i−1 to the state s of the node i.

  In the exemplary embodiment, the trellis starts at state 0 as follows:

状態sの（デコーダ406の）パスメモリは、最小のパスメトリックに寄与した状態で更新される。例示的な実施例のデコーダでは、パスメモリは生き残り（survival）符号／振幅ビットで同様に更新される。生き残り符号ビットは、トレースバック長の終了時に振幅／符号ビットを決定する際に役立つ。

The path memory in state s (decoder 406) is updated with the contribution to the minimum path metric. In the decoder of the exemplary embodiment, the path memory is similarly updated with the survival code / amplitude bits. The surviving sign bit helps in determining the amplitude / sign bit at the end of the traceback length.

  In the exemplary embodiment, the surviving code bits are derived as follows:

Outside 2

を状態sの生き残り状態とし、

Is the surviving state of state s,

Outside 3

を

The

Outside 4

からsへの状態遷移の分岐出力ワードとする。表１に記載のマッピングを使用して、

The branch output word for the state transition from to s. Using the mapping described in Table 1,

Outside 5

が

But

Outside 6

から導かれ、生き残り符号ビットが

The surviving sign bit is derived from

Outside 7

で式５のセットa_kから決定される。生き残り符号ビット

_Is determined from the set a _k of Equation 5. Survival sign bit

Outside 8

は生き残り状態

Survived

Outside 9

と共に状態sのパスメモリに格納される。

And stored in the path memory of state s.

  After updating the path memory of all states of node i, the traceback sequence starts from the state with the smallest metric (assuming the decoder is in steady state). Traceback proceeds in a standard way, and the decoded PPM bit 407 is output at the end of the traceback length. Further, a sign / amplitude bit 408 corresponding to the state at the end of the traceback length is output. Thus, using the soft metrics of equations 6, 7 (a) and 7 (b), a traceback can be performed including the sign / amplitude bits 408.

  It should be noted that the exemplary embodiment does not significantly increase the receiver complexity compared to known receivers. The storage of the sign bit in the path memory increases the memory size by (K * L) bits. Where K is the number of states and L is the traceback length.

Outside 10

を計算するロジックも非常に小さくなる。

The logic for calculating is also very small.

  FIG. 6 shows a UWB network 600 according to an exemplary embodiment. The network has an access point (AP) or host 601. Illustratively, UWB network 600 is a wireless network 600 with a centralized MAC layer within AP 601. In particular, the AP 601 operates according to one of the plurality of exemplary protocols described above. The AP 601 provides services to a plurality of wireless stations 602 according to the selected protocol.

  Illustratively, network 600 is a WLAN or wireless personal area network (WPAN), and STAs (devices) 601, 602 operate on computers, mobile phones, personal digital assistants (PDAs) or typically such networks. Similar device. As indicated by the bi-directional arrows, devices 601 and 602 may communicate with each other, and host 601 and device 602 may communicate with each other.

  It should be noted that according to a specific MAC layer protocol, communication from one device of STA 602 to another device of STA 602 is not necessarily direct. Rather, such communication passes through the host 601 and the host 601 transmits the communication (using known scheduling methods) to the correct receiving STA 602.

  It should be further noted that although only a few STAs 602 are shown, this is merely for the sake of brevity. Obviously, many other devices 602 may be used. Finally, it should be noted that the devices 602 need not be the same. Indeed, a large number of different devices that function with the selected protocol may be used in the network 600. In accordance with the exemplary embodiment described above, transmitters that transmit UWB pulses encoded according to the exemplary embodiment may be located at AP 601 and STA 602. Further, the receiver 400 of the exemplary embodiment is included in the AP 601 and STA 602 of the network 600.

  As can be seen from the discussion of the exemplary embodiment above, the advantage of reliability of the decoded data is realized. In known PPM / PAM systems, PAM bits are decoded by a correlator and PPM bits are decoded by a block of a convolutional decoder. This results in unequal error performance between the PAM bit and the PPM bit. In contrast, according to an exemplary embodiment, the performance of the PAM bits is improved by including the PAM bits in the trellis decoder path memory and making a decision based on these bits along with the PPM bits after the traceback. Is done. Illustratively, the receiver of the exemplary embodiment provides a gain of about 1.5 dB to about 2.0 dB compared to a soft decision decoding receiver of PPM bits.

  In view of this disclosure, it should be noted that the various methods and apparatus described with the exemplary embodiment UWB system may be implemented in hardware and software. In addition, various methods, devices, and parameters are included as examples and are not meant to be limiting. In view of this disclosure, while remaining within the scope of the claims, those skilled in the art will implement various exemplary devices and methods in determining their technology and the equipment needed to implement these technologies. Can do.

Conceptual diagram of a time-frequency interleaved 5-band UWB system according to an exemplary embodiment Timing diagram showing pulse transmission, repetition and duration according to an exemplary embodiment Mapping table for use in a coding method according to an exemplary embodiment Schematic block diagram of a multi-band UWB receiver according to an exemplary embodiment Graphical display of received data after demodulation and correlation at a receiver according to an exemplary embodiment Schematic diagram of a UWB system according to an exemplary embodiment

Claims (20)

Transmitting a plurality of position data bits and a plurality of amplitude data bits in each of said position data bits;
Decoding the position data bits and the amplitude data bits;
Data transmission and reception method. The data bits are modulated using pulse position modulation (PPM) / 2 phase shift keying (BPSK) modulation techniques;
The data transmission method according to claim 1, wherein the amplitude data bit is a sign bit.   The method of claim 1, further comprising providing the amplitude data bits to a trellis decoder path memory and tracing back the amplitude data bits.   The method of claim 1, wherein the data transmission is an ultra wideband (UWB) pulse transmission.   4. The method of claim 3, further comprising calculating a soft metric and using a soft metric to determine the position data bits. Metric mb _xy, except for bit x = y, the soft metric sm ₀ and sm ₁ bit b ₀ and b ₁ is
sm ₀ = mb ₀₀ -mb ₀₁
sm ₁ = mb ₁₀ -mb ₁₁
6. The method of claim 5, determined by: For the output m _k of the correlator at multiple pulse positions, mb _xy is
mb ₀₀ = max (abs (m ₀ ), abs (m ₃ ))
mb ₀₁ = max (abs (m ₁ ), abs (m ₂ ))
mb ₁₀ = max (abs (m ₀ ), abs (m ₁ ))
mb ₁₁ = max (abs (m ₂ ), abs (m ₃ ))
7. A method according to claim 6 provided by:   6. The method of claim 5, further comprising calculating a surviving amplitude bit for each of a plurality of nodes and each of a plurality of states and updating a path memory with the surviving amplitude bit.   9. The method of claim 8, wherein the decoding further comprises performing a traceback sequence after the updating and outputting the amplitude bits and the position bits.   The method of claim 9, wherein the amplitude bit and the position bit correspond to a state having a minimum metric at the end of the traceback sequence. The amplitude data bits are derived from the sign a _k of the output from the correlator,
a _k
a _k = sign (m _k )
11. A method according to claim 10 provided by: A matched filter / correlator that provides information about the plurality of position data bits and information about the plurality of amplitude data bits to the demapping / convolutional decoder;
The demapping / convolutional decoder is an ultra wideband (UWB) system that decodes the position data bits and the amplitude data bits. 13. The UWB system according to claim 12, wherein the amplitude data bits are derived from the sign of the correlator output given by a _k = sign (m _k ).   The UWB system according to claim 12, wherein the demapping / convolutional decoder is a trellis decoder having a path memory.   The UWB system of claim 14, wherein the trellis decoder calculates a soft metric.   15. The UWB system according to claim 14, wherein the trellis decoder calculates a surviving amplitude bit for each of a plurality of nodes and each of a plurality of states, and updates the path memory with the surviving amplitude bit.   The UWB system according to claim 14, wherein the trellis decoder updates the amplitude data bits, executes a traceback sequence, and outputs the amplitude data bits and the position data bits. For metric mb _xy of x = y, the soft metric sm ₀ and sm ₁ bit b ₀ and b ₁ is
sm ₀ = mb ₀₀ -mb ₀₁
sm ₁ = mb ₁₀ -mb ₁₁
17. A UWB system according to claim 16 provided by: For the correlator output m _k at different pulse positions, mb _xy is
mb ₀₀ = max (abs (m ₀ ), abs (m ₃ ))
mb ₀₁ = max (abs (m ₁ ), abs (m ₂ ))
mb ₁₀ = max (abs (m ₀ ), abs (m ₁ ))
mb ₁₁ = max (abs (m ₂ ), abs (m ₃ ))
The UWB system of claim 18 provided by: The UWB system is a wireless network having a plurality of wireless stations (STA),
The UWB system according to claim 19, wherein the STA is one or more of a computer, a mobile phone, and a personal digital assistant (PDA).

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