JP2008005633A - Linear motor control system and Stirling refrigerator control system and linear compressor control system using the same - Google Patents

Abstract

A linear motor control system in which occurrence of problems due to inability to detect a stroke of a piston is suppressed is provided.
A stroke control unit outputs a command value of a relatively low carrier frequency after the linear motor is started, and can calculate a piston stroke that is a value determined in advance from an experimental result. Increase the modulation rate to a predetermined value. After that, the stroke control unit increases the modulation factor when the actual stroke value is less than or equal to the stroke command value, while the modulation factor is increased when the actual stroke value is greater than the stroke command value. Decrease. Further, the stroke control unit outputs a command value of a relatively low carrier frequency when the modulation rate is a value between a predetermined value and a specific value larger than the predetermined value, while the modulation rate is a specific value. In the above case, a command value having a relatively high carrier frequency is output.
[Selection] Figure 12

Description

  The present invention relates to a linear motor control system including a linear motor that reciprocates a mover, and to a Stirling refrigerator control system and a linear compressor control system using the linear motor control system.

A linear motor control system including a linear motor that reciprocates a mover is used. In the linear motor control system, control for calculating the stroke of the mover is performed using the value of the voltage applied to the linear motor and the value of the current flowing through the linear motor.
JP-A-9-126147 JP 2003-339188 A JP 2003-314919 A JP 2003-65244 A

  In the conventional linear motor control system, AC power supplied to the linear motor is generated by PWM (Pulse Width Modulation) control. When the pulse width in this PWM control is small, it is difficult to measure an accurate value of the current flowing through the linear motor. For example, when the electric power supplied to the linear motor immediately after the linear motor control system is started is small, the pulse width in the PWM control is small, so that an accurate value of the current cannot be measured. Therefore, the stroke of the mover cannot be calculated. Therefore, when the stroke of the mover is too larger than the target stroke, there is a possibility that problems such as the mover colliding with other parts may occur.

  The present invention has been made in view of the above-described problems, and an object of the present invention is to make it possible to calculate the stroke of the mover as much as possible even in a period when the power supplied to the linear motor is small. It is to provide a linear motor control system in which the occurrence of problems due to the inability to detect the stroke of the motor is suppressed, and a Stirling refrigerator control system and a linear compressor control system using the linear motor control system.

  A linear motor control system according to one aspect of the present invention includes a linear motor that reciprocates a mover, an inverter circuit that converts DC power supplied from a DC power source into AC power, and supplies AC power to the linear motor, And a microcomputer for controlling the inverter circuit by PWM (Pulse Width Modulation). In addition, the system detects the voltage applied to the linear motor and detects the current flowing in the linear motor by detecting a voltage detector that transmits a voltage signal that can specify the voltage value to the microcomputer. And a current detector for transmitting a current signal capable of specifying the value of to the microcomputer.

  The microcomputer calculates the actual stroke value of the mover using the stroke command value output unit that outputs the target stroke command value of the mover, and the voltage value and current value specified by the voltage signal and current signal. A stroke calculation unit that outputs a PWM modulation rate command value and a carrier frequency command value based on a comparison result between a target stroke command value and an actual stroke, and a PWM modulation rate command value And a PWM control unit that controls the inverter circuit in accordance with the command value of the carrier frequency of PWM.

  The stroke control unit outputs a command value having a relatively low carrier frequency after the operation of the linear motor is started, and can calculate a stroke of the mover which is a value determined in advance from an experimental result. Increase the modulation rate to a predetermined value. After that, the stroke control unit increases the modulation factor when the actual stroke value is less than or equal to the stroke command value, while the modulation factor is increased when the actual stroke value is greater than the stroke command value. Decrease. Further, the stroke control unit outputs a command value of a relatively low carrier frequency when the modulation rate is a value between a predetermined value and a specific value larger than the predetermined value, while the modulation rate is a specific value. In the above case, a command value having a relatively high carrier frequency is output.

  According to said structure, when a modulation rate is small, a carrier frequency becomes low. Therefore, even if the modulation rate is the same, the width of the current pulse is increased. As a result, it is possible to properly detect the current pulse even in a period where the modulation rate is small. Therefore, the actual stroke of the mover can be calculated even in a period where the modulation factor is small. As a result, it is possible to suppress the occurrence of problems caused by the fact that the actual stroke of the mover cannot be calculated in a period where the modulation factor is small.

  Further, immediately after the modulation rate becomes equal to or higher than the specific value, the microcomputer may store the modulation rate at that time as the first predetermined modulation rate and stop the supply of power to the linear motor in the inverter circuit. . Thereafter, the microcomputer may cause the inverter circuit to supply power to the linear motor while increasing the modulation rate up to the first predetermined modulation rate in PWM with a relatively high carrier frequency. Further, immediately after the modulation rate becomes equal to or lower than the specific value, the microcomputer may store the modulation rate at that time as the second predetermined modulation rate and stop the inverter circuit from supplying power to the linear motor. . Thereafter, the microcomputer may cause the inverter circuit to supply power to the linear motor while increasing the modulation rate up to the second predetermined modulation rate in PWM with a relatively low carrier frequency.

  According to the above configuration, the operation of the mover is temporarily stopped before switching between the PWM with a relatively high carrier frequency and the PWM with a relatively low carrier frequency. The operation starts again. That is, the aforementioned switching is performed while the operation of the mover is stopped. Therefore, the operation of the mover does not change abruptly due to the aforementioned switching during the operation of the mover.

  In addition, it is desirable that the stroke calculation unit starts calculating the actual stroke after the modulation rate reaches or exceeds a predetermined value.

  In general, when the current value cannot be accurately detected because the modulation rate is small, the error included in the actual stroke value calculated by the stroke calculation unit is large. In this case, if the actual stroke is calculated and the operation of the mover is controlled using the actual stroke value, a malfunction may occur in the operation of the mover. Therefore, during the period in which it is known that the current value cannot be accurately detected in advance, the occurrence of the above-described problems can be prevented by calculating the stroke.

  A linear motor control system according to another aspect of the present invention includes a linear motor that reciprocates a mover, an inverter circuit that converts DC power supplied from a DC power source into AC power, and supplies AC power to the linear motor, And a microcomputer for controlling the inverter circuit by PWM (Pulse Width Modulation). In addition, the system detects the voltage applied to the linear motor and detects the current flowing in the linear motor by detecting a voltage detector that transmits a voltage signal that can specify the voltage value to the microcomputer. Is provided with a current detector that transmits a current signal that can specify the value of the current to the microcomputer, and a DC power generation circuit that generates DC power.

  Further, the microcomputer uses the stroke command value output unit for outputting the target stroke command value of the mover, and the actual stroke value of the mover using the voltage value and the current value specified by the voltage signal and the current signal. A stroke calculation unit that calculates a PWM modulation rate command value based on a comparison result between the stroke command value and the actual stroke, a stroke control unit that controls the DC power control circuit, and a PWM modulation It includes a PWM controller that controls the inverter circuit according to the command value of the rate.

  The stroke control unit controls the DC power generation circuit so that a relatively low DC voltage is applied to the inverter circuit after the operation of the linear motor is started, and is a value determined in advance from experimental results. The modulation factor is increased to a predetermined value at which the stroke of the mover can be calculated. After that, the stroke control unit increases the modulation factor when the actual stroke value is less than or equal to the stroke command value, while the modulation factor is increased when the actual stroke value is greater than the stroke command value. Decrease. Further, the stroke control unit sets the DC power generation circuit so that a relatively low DC voltage is applied to the inverter circuit when the modulation factor is a value between a predetermined value and a specific value larger than the predetermined value. On the other hand, the DC power generation circuit is controlled so that a relatively high DC voltage is applied to the inverter circuit when the modulation rate is equal to or higher than a specific value.

  According to said structure, in the period when a relatively low DC voltage is applied to an inverter circuit, even if a modulation rate becomes large, the maximum value of the alternating voltage applied to a linear motor does not become so large. That is, in this period, the modulation rate can be increased, that is, the width of the current pulse can be increased without significantly increasing the stroke of the mover. As a result, in the period when the AC voltage applied to the linear motor is small, the occurrence of problems due to the inability to calculate the actual stroke of the mover is suppressed.

  Further, immediately after the modulation rate becomes equal to or higher than the specific value, the microcomputer may store the modulation rate at that time as the first predetermined modulation rate and stop the supply of power to the linear motor in the inverter circuit. . Thereafter, the microcomputer may cause the inverter circuit to supply power to the linear motor while controlling the DC power generation circuit so that a relatively high DC voltage is applied to the inverter circuit.

  Further, immediately after the modulation rate becomes equal to or lower than the specific value, the microcomputer may store the modulation rate at that time as the second predetermined modulation rate and stop the inverter circuit from supplying power to the linear motor. . Thereafter, the microcomputer may cause the inverter circuit to supply power to the linear motor while controlling the DC power generation circuit so that a relatively low DC voltage is applied to the inverter circuit.

  According to the above configuration, the operation of the mover is temporarily stopped before switching between the relatively high DC voltage and the relatively low DC voltage, and the operation of the mover is started again after the above switching. Is done. That is, the aforementioned switching is performed while the operation of the mover is stopped. As a result, the operation of the mover does not change abruptly due to the aforementioned switching during the operation of the mover.

  In addition, it is desirable that the stroke calculation unit starts calculating the actual stroke after the modulation rate reaches or exceeds a predetermined value.

  In general, when the current value cannot be accurately detected because the modulation rate is small, the error included in the actual stroke value calculated by the stroke calculation unit is large. In this case, if the actual stroke is calculated and the operation of the mover is controlled using the actual stroke value, a malfunction may occur in the operation of the mover. Therefore, during the period in which it is known that the current value cannot be accurately detected in advance, the occurrence of the above-described problems can be prevented by not calculating the stroke.

  The DC power generation circuit may include a voltage change circuit that can change the DC voltage. In this case, a relatively high DC voltage is generated by turning on the voltage changing circuit, and a relatively low DC voltage is generated by turning off the voltage changing circuit.

  According to the above configuration, switching between a relatively high DC voltage and a relatively low DC voltage can be performed with a simple configuration.

  In the Stirling refrigeration system of the present invention, the above-described linear motor control system is used for controlling the mover of the Stirling refrigeration machine.

  According to the above configuration, it is possible to prevent the movable element from colliding with a peripheral structure such as a displacer and the movable element due to an excessively large stroke of the movable element.

  Moreover, in the linear compression system of this invention, the above-mentioned linear motor control system is used in order to control the needle | mover of a linear compressor.

  According to the above configuration, it is possible to prevent the mover from colliding with the peripheral structure such as a part constituting the compression space due to the stroke of the mover becoming too large.

  According to the present invention, even when the electric power supplied to the linear motor is small, it is possible to calculate the piston stroke as much as possible, thereby causing a problem due to the fact that the piston stroke cannot be detected. It is suppressed.

(Embodiment 1)
First, a linear motor control system according to an embodiment and a Stirling refrigerator control system using the linear motor control system will be described.

  First, the Stirling refrigerator controlled by the linear motor control system of the present embodiment will be described with reference to FIG.

  Drawing 1 is a sectional view showing Stirling refrigerator 40 of an embodiment. In the Stirling refrigerator 40, a cylindrical piston 1 and a displacer 2 are fitted in a cylindrical cylinder 3 composed of two parts. The piston 1 and the displacer 2 are provided via a compression space 9 and have an axis Y as a common drive shaft.

  An expansion space 10 is formed on the tip side of the displacer 2. The compression space 9 and the expansion space 10 communicate with each other via a medium flow passage 11 through which a working medium such as helium flows. A regenerator 12 is provided in the medium flow path 11. The regenerator 12 accumulates the heat of the working medium and supplies the accumulated heat to the working medium. A flange (flange) 3 a is provided in the middle of the cylinder 3. A bounce space (back pressure space) 8 is formed in the collar portion 3a by being sealed with a dome-shaped pressure vessel 4 attached thereto.

  The piston 1 is integrated with the support spring 5 on the rear end side. The displacer 2 is integrated with the support spring 6 through a rod 2 a that passes through the center hole 1 a of the piston 1. The support spring 5 and the support spring 6 are connected by a bolt and a nut 22. As will be described later, when the piston 1 reciprocates, the displacer 2 reciprocates with a predetermined phase difference with respect to the piston 1.

  An inner yoke 18 is fitted on the outer peripheral side of the cylinder 3 in the bounce space 8. The inner yoke 18 faces the outer yoke 17 through a gap 19. A driving coil 16 is fitted inside the outer yoke 17. An annular permanent magnet 15 is movably provided in the gap 19. The permanent magnet 15 is integrated with the piston 1 through a cup-shaped sleeve 14. The inner yoke 18, the outer yoke 17, the driving coil 16, and the permanent magnet 15 constitute a linear motor 13 that moves the piston 1 along the axis Y. Lead wires 20 and 21 are connected to the driving coil 16, and driving power is supplied to the linear motor 13 by the control device 30.

  In the Stirling refrigerator 40 having the above configuration, when the piston 1 reciprocates by the linear motor 13, the displacer 2 reciprocates with a predetermined phase difference with respect to the piston 1. As a result, the working medium moves between the compression space 9 and the expansion space 10. As a result, a reverse Stirling cycle is formed.

  In the Stirling refrigerator 40 of the above-described embodiment, when a drive voltage having a predetermined AC waveform is applied to the linear motor 13, the piston 1 reciprocates at a cycle and a stroke corresponding to the drive voltage having the predetermined AC waveform. Do. Therefore, by controlling the drive voltage applied to the linear motor 13, the period and stroke of the reciprocating motion of the piston 1 can be controlled.

  Next, the operating principle of the free piston type Stirling refrigerator of the above embodiment will be described in more detail. As the piston 1 moves so that the relationship between its position and time draws a sine wave, the working gas in the compression space 9 changes so that the relationship between its pressure and time draws a sine wave, and compression. Heat is released from the space 9 and flows into the expansion space 10 while being cooled by the regenerator 12 provided around the displacer 2.

  The working gas in the expansion space 10 expands due to the movement of the displacer 2, and its temperature decreases. The working gas in the expansion space 10 changes so that the relationship between pressure and time draws a sine wave, and the displacer 2 is reciprocated with a predetermined phase difference with respect to the piston 1.

  Next, an IPM (Intelligent Power Module) 200 and a microcomputer 1000 provided in the control device 30 according to the embodiment will be described with reference to FIGS. 2 to 4A and 4B. As shown in FIG. 2, the IPM 200 is used for controlling the drive voltage of the linear motor 13 of the present embodiment, that is, the linear motor 13 described above. The IPM 200 incorporates an inverter circuit 100. The inverter circuit 100 has four switching elements, and is connected to the linear motor 13 built in the Stirling refrigerator 40 in a manner as shown in FIG. The four switching elements are transistors Gu, Gx, Gv, and Gy, each of which has a flywheel diode connected between the source electrode and the drain electrode.

  As can be seen from FIG. 2, the transistor Gu and the transistor Gx are connected in series, and the transistor Gv and the transistor Gy are connected in series. The linear motor 13 has one terminal connected to a node between the transistor Gu and the transistor Gx, and the other terminal connected to a node between the transistor Gv and the transistor Gy.

  A series circuit of a capacitor C and a capacitor CC is connected in parallel to the inverter circuit 100. The series circuit is connected to the output side of the diode circuit D, the AC power supply G is connected to the input side of the diode circuit D, and a voltage doubler circuit B including a capacitor C and a capacitor CC is configured. Further, a resistor R1 and a resistor R2 are connected between the capacitor C and the capacitor CC and the inverter circuit 100 in parallel to the capacitors C and CC, and a voltage divider for dividing the voltage between the input terminals of the inverter circuit 100. A circuit is configured.

  A capacitor CCC for stabilizing the potential of the node between the resistor R1 and the resistor R2 is connected in parallel to the resistor R2, and the node between the resistor R1 and the resistor R2 is a microcomputer. Connected to 1000 voltage sensor ports. In the microcomputer 1000, a voltage signal that specifies the voltage of the DC power input to the inverter circuit 100 is input to the DC voltage sensor.

  Furthermore, the input terminal of the circuit V functioning as a voltmeter is connected to the two terminals of the linear motor 13 in a one-to-one relationship, and the voltage value obtained by the circuit V is the U-phase voltage sensor port of the microcomputer 1000 and Sent to each of the V-phase voltage sensor ports. Further, a circuit A that functions as an ammeter is provided between the DC power source and the linear motor 13, and a current signal that can specify the current value obtained by the circuit A is transmitted to the current sensor port of the microcomputer 1000. The

  The method for obtaining the voltage value and the current value is as follows. In obtaining the voltage value, the voltage applied to the linear motor (M) is divided by the circuit V, and the divided voltage value is input to the microcomputer 1000. The microcomputer 1000 performs A / D conversion on the voltage value to calculate an actual voltage value. The current value is acquired by amplifying the potential difference between both ends of the shunt resistor S by the circuit A including the operational amplifier, and transmitting a voltage signal that can specify the value of the amplified potential difference to the voltage sensor port of the microcomputer 1000. The The microcomputer 1000 performs A / D conversion on the amplified potential difference value to calculate a current value.

  Further, the microcomputer 100 transmits an ON / OFF switching signal for switching on / off of the voltage doubler switch SW to the voltage doubler switch SW. That is, the microcomputer 1000 can increase or decrease the magnitude of the DC voltage supplied to the inverter circuit 100. However, in the present embodiment, the voltage supplied to the inverter circuit 100 by switching the voltage doubler switch SW is V when the voltage doubler switch SW is off, and the voltage doubler switch SW is on. Sometimes 2V.

FIG. 3 is a block diagram for explaining the configuration of a linear motor control microcomputer 1000 in which one PWM inverter control timer (one channel) is built.
As shown in FIG. 3, the microcomputer 1000 according to the present embodiment includes a clock circuit as an oscillator, a CPU (Central Processing Unit) as arithmetic means, and a RAM (Random Access Memory) as rewritable storage means. And a read-only ROM (Read Only Memory). The ROM stores a program for controlling the transistors as the four switching elements. The RAM is a storage means for temporarily storing the result of the operation performed by the CPU according to the program stored in the ROM, and may include a temporary storage means such as a register. Further, the clock is used to form a clock pulse that is a basis for operating a timer described later, using a signal transmitted from the oscillator.

  Next, the comparators provided in each of the U-phase control circuit and the V-phase control circuit will be described with reference to FIGS. 4A and 4B.

  As shown in FIG. 4A, the comparator has two input terminals. As for the two input terminals, a PWM control signal wave is input to one of them, and a carrier wave is input to the other. The signal wave data is stored in a data table in the ROM. The carrier wave is generated by the up / down timer 1 based on the time signal generated by the clock.

  The relationship between the signal wave input to the comparator and the carrier wave is shown in FIG. 4B (a). The comparator compares the signal wave data with the carrier wave data, and outputs a PWM control signal when the signal wave data is larger than the carrier wave data. In the first half of one cycle of the AC waveform, the transistors Gu and Gx are turned on / off, the transistor Gy is turned on, and Gv is turned off. On the other hand, in the second half of one cycle of the AC waveform, the transistors Gv and Gy are turned on / off, the transistor Gx is turned on, and Gu is turned off. Thereby, a pulse waveform as shown in (b) of FIG. 4B is formed, and an AC voltage having a waveform as shown by a dotted line is applied to the linear motor M. That is, an AC voltage having a sine curve waveform is applied to the linear motor M.

  The frequency of the above-described up / down timer 1 is the carrier frequency for PWM control. The carrier frequency indicates the number of symmetric triangular wave pulses output from the inverter circuit 100 per second, and is the frequency of the carrier wave for PWM control. The effective value of the voltage applied to the linear motor 13 can be changed by increasing or decreasing the carrier frequency. Note that the reciprocal of the carrier frequency is the carrier period. That is, one cycle of the symmetric triangular wave constituting the carrier wave is the carrier cycle.

  In the control of the Stirling refrigerator, the piston 1 and the displacer 2 do not resonate unless they are driven at a predetermined frequency. That is, if the frequency of the reciprocating motion of the piston 1 is significantly different from the resonance frequency of the piston 1 and the displacer 2, the Stirling refrigerator 40 cannot be driven. Therefore, the relationship between the above-described set value data string constituting the PWM signal wave and time must be set so as to draw a sine curve of a predetermined resonance frequency.

  Next, a method for detecting the stroke Xp of the piston 1 of the linear motor (M) will be described. In the Stirling refrigerator 40 of the present embodiment, the stroke Xp of the piston 1 is detected as follows.

First, a steady driving state of the control device 30 will be described with reference to FIGS. 5 and 6.
FIG. 5 shows a voltage V applied to the linear motor (M) in a steady state, a current I flowing through the coil 16 of the linear motor (M), an induced voltage E generated in the coil 16 of the linear motor (M), and a piston. It is the figure which showed the relationship of 1 displacement T. FIG. 6 is an equivalent circuit diagram of the linear motor (M). Also, as shown in FIG. 6, the direction of the current I generated by the induced voltage E is opposite to the direction of the current generated by the applied voltage V.

  As shown in FIG. 5, the phase of the current I is delayed from the applied voltage V by θ due to the influence of the inductance (L shown in FIG. 6) of the linear motor (M). Here, the magnitude of the thrust acting on the linear motor (M) is a value obtained by multiplying the value of the current I by the thrust constant α. Further, as can be seen from the equivalent circuit diagram shown in FIG. 6, the induced voltage E is expressed by the following equation (1).

  Therefore, if the motor winding resistance R and the inductance L are known in advance, the induced voltage E is calculated using the voltage V acquired by the circuit V shown in FIG. 3 and the current I acquired by the circuit A shown in FIG. Can be calculated. The phase difference θ is obtained by calculating the difference between the phase value when the voltage V is peak and the phase value when the current I is peak. Further, the thrust constant α is calculated in advance by experiments, and the motor winding resistance R and the inductance L are values measured in advance.

  The piston stroke Xp is defined by the following equation (2).

  Thus, if the phase difference θ, motor winding resistance R, voltage V, current I, applied frequency f, and thrust constant α are known, the stroke Xp can be calculated. The above-described method for calculating the stroke Xp is disclosed in detail in Japanese Patent Application Laid-Open Nos. 2003-314919 and 2003-65244.

Next, the “current stroke Xp calculation process” will be described with reference to FIG.
In the “calculation process of the current stroke Xp” shown in FIG. 8, first, in S81, a phase difference θ that is the difference between the peak phase of the current waveform and the peak phase of the voltage waveform is calculated. Next, in S82, the microcomputer 1000 calculates the current I flowing through the shunt resistor S using the signal transmitted from the circuit A shown in FIG. Thereafter, in S83, the microcomputer 1000 calculates the voltage V applied to the linear motor (M) using the signal transmitted from the circuit V shown in FIG. Next, in S84, the induced voltage E is calculated using the phase difference θ, the voltage V, the current I, the winding resistance value R, and the above-described equation (A). Thereafter, in S85, the stroke Xp is calculated using the above-described equation (B).

  Specifically, as shown in FIG. 8, the linear motor control system of the present embodiment described above includes a linear motor 13 having a reciprocating mover, an AC power source G and a smoothing circuit P ( D, C, CC) and the inverter circuit 100 that converts the DC power supplied from the aforementioned DC power source into AC power and supplies the AC power to the linear motor 13, and the inverter circuit 100 by PWM (Pulse Width Modulation). And a microcomputer 1000 to be controlled.

  The linear motor control system also includes a motor voltage detector V for detecting an instantaneous value of the motor voltage v (t) applied to the linear motor 13 at a predetermined sampling period, and the linear motor 13 at a predetermined sampling period. Motor current detectors A and S for detecting an instantaneous value i (t) of the motor current flowing in the motor.

  Note that the motor current detectors A and S in this embodiment include a shunt resistor S connected in series to the linear motor 13 between the inverter circuit 100 and the DC power supply and an operational amplifier (Operational) that amplifies the potential difference between both ends. Amplifier) circuit A and the like. The motor voltage detector U (V) has resistors as potential measuring voltage dividers connected to both ends of the linear motor 13. The motor voltage detector V and the motor current detectors A and S are respectively a voltage signal that can specify the instantaneous value of the motor voltage v (t) and a current signal that can specify the instantaneous value of the motor current i (t). Is transmitted to the microcomputer 100.

  The microcomputer 100 outputs a stroke command value output unit 101 that outputs a target stroke of the piston 1 as a mover of the linear motor 13, a stroke control unit 102 that receives stroke command value data, and a transmission from the stroke control unit 102. And a PWM control unit 103 that transmits a PWM pulse signal to the inverter circuit 100 based on the PWM modulation rate command value.

  The stroke command value is determined according to the load condition, and is a value smaller than a limit value at which the piston 1 collides with another part. The modulation rate command is a signal that can specify the PWM modulation rate. Further, the PWM control unit 103 outputs a pulse signal for controlling on / off of the switching elements Gx, Gy, Gu, Gv constituting the inverter circuit 100 based on a predetermined angular velocity ω and a modulation factor command value. . As a result, a predetermined AC voltage is applied to the linear motor 13.

  The microcomputer 1000 has a built-in AD (Analog to Digital) converter, and has a stroke calculation unit 104 that calculates an actual stroke Xp of the mover based on a voltage signal and a current signal. The calculated actual stroke Xp data is transmitted to the stroke control unit 102.

  The stroke control unit 102 transmits a signal that can specify the modulation rate command value and a signal that can specify the command value of the carrier frequency to the PWM control unit 103.

  The motor voltage detector V transmits to the microcomputer 1000 a voltage signal that can specify the instantaneous value of the motor voltage v (t) of the linear motor 13. The microcomputer 1000 uses the AD conversion function to calculate the motor voltage v (t) applied to the linear motor 13 based on a signal that can identify the potential difference between both terminals of the linear motor 13.

  The motor current detectors A and S detect the motor current i (t) flowing through the shunt resistor S connected to the DC (Direct Current) line on the input side of the inverter. The motor current i (t) flows into the shunt resistor S only during the PWM pulse ON period. Therefore, the motor current detectors A and S transmit the potential difference signals at both ends of the shunt resistor S during the above-described ON period to the microcomputer 1000. The microcomputer 1000 converts the potential difference signal into a current signal that can specify the motor current i (t) by an AD conversion function. Therefore, it is desirable that the sampling period of the motor current i (t) coincides with the PWM carrier period.

  The stroke calculation unit 104 receives a voltage signal that can specify the instantaneous value of the motor voltage v (t) and a current signal that can specify the instantaneous value of the motor current i (t), and receives a motor signal for each cycle of the sine wave. The effective value V of the voltage, the effective value I of the motor current, and the phase difference θ are calculated.

  In the PWM control of the present embodiment, one of the PWM control of the standard carrier frequency shown in FIG. 9 and the PWM control of the low carrier frequency (1/2 of the standard carrier number) shown in FIG. 10 is selectively performed. It is.

  In the present embodiment, it is assumed that the minimum pulse width necessary for current detection in inverter circuit 100 and microcomputer 1000 is Ta (sec). This Ta is a value inherent to the inverter circuit 100 and the microcomputer 100. When the control of the standard carrier frequency shown in FIG. 9 is executed, that is, when the carrier period is Tc, the current value can be accurately detected for at least one of the pulses for one period. The minimum modulation rate is Ta / Tc (peak voltage Vpeak = Vdc × Ta / Tc (V)). On the other hand, according to the control of the low carrier frequency shown in FIG. 10, since the carrier period is twice the carrier period Tc (2 × Tc) in the control of the standard carrier frequency, at least one of the pulses for one period is used. The minimum modulation rate for accurately detecting the current value of the pulse is Ta / Tc / 2 (peak voltage Vpeak = Vdc × Ta / Tc / 2 (V)), compared with the control of the standard carrier frequency. The modulation factor is 1/2 and the peak voltage is also 1/2. However, normally, in the calculation of the effective value of the current necessary for detecting the stroke Xp, it is not sufficient to detect the current value of one pulse out of the pulses in one period, so that the above modulation is performed. A value obtained by multiplying the rate by a coefficient k (k> 1) is defined as the lower limit value of the modulation rate at which the effective value of the current can be calculated, that is, the lower limit value of the modulation rate at which the stroke Xp can be detected. The lower limit value of the modulation rate at which the coefficient k and the stroke Xp can be detected is a value inherent to the inverter circuit 100 and the microcomputer 100 and is determined empirically. In addition, according to the determination method, the stroke Xp of the piston 1 is measured in a state where the Stirling refrigerator 40 is driven in advance, and the value of the modulation factor when the stroke Xp is accurately obtained is the lower limit of the modulation factor. Defined as a value. That is, the lower limit value of the modulation factor described above is a value obtained in advance by experiment, and is a substantially minimum value of the modulation factor that can accurately detect the pulse current. Further, if the lower limit value of the modulation rate at which the stroke Xp when the control of the standard carrier frequency in the inverter circuit 100 and the microcomputer 1000 of the present embodiment is executed is detected is M (%), the low carrier The lower limit value of the modulation rate at which the stroke Xp when the frequency control is executed can be detected is half the lower limit value of the modulation rate at which the stroke Xp when the standard carrier frequency control is executed. M / 2 (%). Therefore, if the modulation rate (pulse width Ta / carrier cycle Tc) is the same, the peak voltage Vpeak of the AC voltage applied to the linear motor 13 can be reduced. In other words, if the peak voltage Vpeak is the same, the pulse width can be increased even if the modulation rate is the same. Therefore, according to the control of the low carrier frequency shown in FIG. 10, the pulse width Ta can be increased without increasing the stroke Xp of the piston 1 as compared with the control of the standard carrier frequency shown in FIG. Therefore, even when the voltage applied to the linear motor 13 is small, the stroke Xp can be detected accurately. Therefore, the probability that the collision between the piston 1 and the displacer 2 can be prevented is further increased.

  In the present embodiment, as shown in FIG. 11, the stroke calculation unit 104 calculates the stroke command value determined by the stroke command value output unit 101, that is, whether the load on the Stirling refrigerator 40 is large or small. On the basis of whether or not the actual stroke Xp is equal to or less than the actual stroke Xp, the PWM modulation rate is decreased or increased, that is, the stroke Xp of the piston 1 is decreased or increased, and the piston 1 and the displacer 2 To prevent collisions.

  When the load of the Stirling refrigerator 40 is small, the compression space 9 is in a low temperature and low pressure state, and the stroke Xp tends to be large even when the voltage applied to the linear motor 13 is small. In such a case, immediately after the operation of the Stirling refrigerator 40 is started, the piston 1 and the displacer 2 collide before the modulation factor reaches the lower limit value at which the detection of the stroke Xp can be performed. Is likely to occur.

Hereinafter, the carrier frequency change control according to the present embodiment will be described with reference to FIG.
In the carrier frequency change control of the present embodiment, first, in step S1, it is determined whether or not a control command for starting the Stirling refrigerator 40 is input to the microcomputer 1000. The input of a control command for starting up the Stirling refrigerator 40 is performed by a user.

  In step S1, if the start control command for the Stirling refrigerator 40 is not input, step S1 is repeated. On the other hand, if there is an input of a control command for starting the Stirling refrigerator 40 in step S1, the carrier frequency is set to a half value of the standard value in S2. Next, in step S3, inverter control is started. Further, after the inverter control is started, the modulation rate of the PWM control is gradually increased from zero to M / 2 (%). Thereafter, in step S4, detection of the stroke Xp is started.

  Next, it is determined whether or not the actual stroke Xp of the piston 1 calculated by the stroke calculation unit 104 is equal to or less than the stroke command value output from the stroke command value output unit 101 to the stroke control unit 102. If the actual stroke Xp is equal to or less than the stroke command value in S5, the value of the modulation factor is increased in S6. That is, the microcomputer 1000 determines that there is no problem in increasing the actual stroke Xp of the piston 1 because the actual stroke Xp of the piston 1 is not larger than the target stroke.

  On the other hand, if the actual stroke Xp is larger than the stroke command value in S5, it is determined in step S7 whether or not the voltage modulation rate is M / 2 or less. In S7, if the voltage modulation rate is M / 2 or less, step S5 is executed again. On the other hand, if the voltage modulation rate is not M / 2 or less in step S7, the voltage modulation rate is reduced in step S8. That is, the microcomputer 1000 determines that the stroke Xp of the piston 1 should be reduced because the stroke Xp of the piston 1 is larger than the target stroke.

  Further, after any one of steps S6 and S8 is executed, it is determined in step S9 whether or not the voltage modulation rate is equal to or greater than M + α (%). In step S9, if the voltage modulation rate is not M + α (%) or more, the process of step S5 is executed again.

  On the other hand, if the voltage modulation factor is equal to or higher than M + α (%) in step S9, switching from the low carrier frequency to the standard carrier frequency is performed in step S10.

  This switching may be performed at a time from a low carrier frequency to a standard carrier frequency, but may be performed such that the carrier frequency gradually increases from the low carrier frequency to the standard carrier frequency. Α is a hysteresis width for preventing the above-described switching from occurring frequently.

  Next, in step S11, it is determined whether or not the actual stroke Xp of the piston 1 is equal to or less than the stroke command value. If the stroke Xp is equal to or less than the stroke command value in step S11, the PWM modulation rate is increased in step S12. On the other hand, if the stroke Xp is not less than or equal to the stroke command value in step S11, the PWM modulation rate is lowered in step S13.

  Next, when either step S12 or S13 is completed, it is determined in step S14 whether or not the voltage modulation rate is equal to or less than M (%). If the voltage modulation factor is not less than M (%) in step S14, the process of step 11 is executed again. On the other hand, if the voltage modulation rate is a value equal to or less than M (%) in step S14, the carrier frequency is switched from the standard carrier frequency to the low carrier frequency in S15.

  The carrier frequency may be switched at a stroke from the standard carrier frequency to the low carrier frequency, or may be switched so that the carrier frequency gradually decreases from the standard carrier frequency to the low carrier frequency. .

  Thereafter, the process of S5 is performed again. If a signal for stopping the driving of the Stirling refrigerator 40 is input while the processes from S5 to S15 are repeated, step S1 is performed until a signal for starting the Stirling refrigerator is next input. This process is repeated.

  According to the carrier frequency change control shown in FIG. 11, the relationship between the effective value of the output voltage and the PWM modulation rate as shown in FIG. 12 is obtained.

  As shown in FIG. 12, when the Stirling refrigerator 40 is started, the modulation rate gradually increases from zero to M / 2. At this time, the microcomputer 1000 cannot detect the stroke Xp until the modulation rate becomes M / 2.

Thereafter, when the modulation rate becomes larger than M / 2, detection of the stroke Xp is started.
Next, the modulation rate increases from M / 2 to M. However, when it fluctuates between the modulation factor M / 2 and the modulation factor M, the modulation factor increases or decreases according to the load increase / decrease of the Stirling refrigerator 40 and the command value of the stroke Xp.

  Thereafter, when the modulation rate becomes M + α, processing for switching from PWM control with a low carrier frequency to PWM control with a standard carrier frequency is executed. Thereafter, if the modulation rate is equal to or greater than M, PWM control at the standard carrier frequency is continued.

  However, when the load on the Stirling refrigerator 40 is large, that is, when the pressure in the compression space 9 is high, if the actual stroke Xp obtained by the stroke detection becomes smaller than the stroke command value, the modulation factor Control is performed such that the value gradually decreases. If the modulation rate becomes a value equal to or less than M, processing for switching from PWM control at the standard carrier frequency to PWM control at the low carrier frequency is executed.

  Conventionally, when the load of the Stirling refrigerator 40 is reduced, the modulation factor becomes a value of M or less, and when the actual stroke Xp obtained by the stroke detection is larger than the stroke command value, the stroke Xp Cannot be detected. Therefore, in order to prevent a collision between the piston 1 and the displacer 2, the Stirling refrigerator 40 can only be stopped.

  Immediately after the Stirling refrigerator 40 is started, the modulation factor increases unconditionally from 0 to M. At this time, particularly when the load of the Stirling refrigerator 40 is small, there is a risk that the piston 1 and the displacer 2 collide.

  On the other hand, according to the linear motor control system of the present embodiment, when the control of the standard carrier frequency is being executed, the load on the Stirling refrigerator 40 is reduced, and thereby the modulation factor becomes a value of M or less. First, the modulation rate is lowered to M / 2 by lowering the carrier frequency. Therefore, the incidence rate of the state where the Stirling refrigerator 40 must be stopped decreases. Further, when the Stirling refrigerator 40 is activated, if the carrier frequency is set to be low in advance, the modulation factor that can be increased unconditionally is decreased from M to M / 2. Collision between the piston 1 and the displacer 2 when the load is small can be prevented.

  Further, the carrier frequency change control of the present embodiment may be carrier frequency change control as shown in FIG.

  The carrier frequency change control shown in FIG. 13 is substantially the same as the carrier frequency change control shown in FIG. However, in the carrier frequency change control shown in FIG. 13, the processes in steps S9a to S10a and steps S14a to S15a are different from the processes in the carrier frequency change control shown in FIG.

  In the carrier frequency change control shown in FIG. 13, the current modulation factor (m) value is stored in step 9 a, and the application of voltage to the linear motor 13 is stopped. That is, the PWM modulation factor value at that time is stored, and the operation of the Stirling refrigerator 40 is temporarily stopped. Thereby, the operation of the piston 1 is temporarily stopped. Next, in S10, the control state is changed from the low carrier frequency to the standard carrier frequency. Thereafter, in S10a, the operation of the piston 1 is started. That is, the operation of the Stirling refrigerator 40 is started. At this time, the modulation rate is gradually increased from zero to m.

  In step S14a, the current value of the modulation factor (m) is stored, and the application of voltage to the linear motor 13 is stopped. That is, the current modulation factor value is stored and PWM control is stopped. Thereby, the operation of the piston 1 is stopped. Next, in S15, the operation of the Stirling refrigerator is started at a low carrier frequency. Next, in S15a, the modulation rate is increased from zero to m.

  As described above, according to the carrier frequency modulation control shown in FIG. 13, the Stirling refrigerator 40 is temporarily stopped when the carrier frequency is changed. Therefore, when switching between the PWM control with the standard carrier frequency and the PWM control with the low carrier frequency, occurrence of a situation in which the Stirling refrigerator 40 is damaged due to a malfunction in the operation of the piston 1 is prevented.

  14 and 15 show the relationship between the effective value of the output voltage and the modulation rate in the carrier frequency change control shown in FIG. FIG. 14 shows the relationship between the effective value of the output voltage and the modulation rate when the modulation rate is increasing. FIG. 15 shows the relationship between the effective value of the output voltage and the modulation factor when the modulation factor is decreasing.

  As shown in FIG. 14, when the Stirling refrigerator 40 is started, the modulation rate gradually increases from zero to M / 2. At this time, the microcomputer 1000 cannot detect the stroke Xp until the modulation rate becomes M / 2. Thereafter, when the modulation rate becomes larger than M / 2, detection of the stroke Xp is started.

  Next, the modulation rate increases from M / 2 to M. However, in the values from the modulation factor M / 2 to the modulation factor M, the modulation factor increases or decreases depending on the increase / decrease in the load of the Stirling refrigerator 40. Thereafter, when the modulation rate becomes M + α, the microcomputer 1000 stores the value of the modulation rate m, stops the operation of the Stirling refrigerator 40, and switches from the low carrier frequency PWM control to the standard carrier frequency PWM control. The process for is executed. Thereafter, the operation of the Stirling refrigerator 40 is started, and the modulation factor is gradually increased from zero to m. Thereafter, the modulation rate is increased or decreased based on the comparison result between the actual stroke Xp obtained by the stroke detection and the stroke command value.

  On the other hand, as shown in FIG. 15, the microcomputer 1000 stores the value of the modulation factor m at that time and stops the operation of the Stirling refrigerator 40 when the modulation factor decreases and becomes a value of M or less. Set the modulation rate to zero. Further, the microcomputer 1000 switches from the PWM control with the standard carrier frequency to the PWM control with the low carrier frequency. Thereafter, the microcomputer 1000 starts operation of the Stirling refrigerator 40 and gradually increases the modulation rate from zero to m. Thereafter, the modulation rate is increased or decreased based on the comparison result between the actual stroke Xp obtained by the stroke detection and the stroke command value.

(Embodiment 2)
Next, the linear motor control system and the Stirling refrigerator control system according to the second embodiment will be described with reference to FIGS.

  Since the linear motor control system and the Stirling refrigerator control system of the present embodiment are substantially the same as the linear motor control system and the Stirling refrigerator control system of the first embodiment, the linear motor of the present embodiment will be described below. Differences between the control system and Stirling refrigerator control system and the linear motor control system and Stirling refrigerator control system of the first embodiment will be mainly described.

  As shown in FIG. 16, in the Stirling refrigerator control system of the present embodiment, a voltage doubler changeover switch SW as described in FIG. 2 is used. As shown in FIGS. 2, 17, and 18, the voltage doubler changeover switch SW includes a node between a diode D3 and a diode D4 in the diode circuit D (diodes D1, D2, D3, and D4) and a capacitor C. And a node between the capacitor CC and the capacitor CC. Further, as shown in FIG. 16, the stroke control unit 102 transmits an ON / OFF switching signal to the voltage doubler changeover switch SW and supplies the voltage doubler to the inverter circuit 100 or applies a voltage half the voltage doubler to the inverter. The circuit 100 can be switched.

  17 and 18 show a state where the voltage doubler switch SW is ON and a state where the voltage doubler switch SW is OFF, respectively. 19 and 20 show the PWM pulse waveform and AC voltage waveform when the voltage doubler switch is ON, and the PWM pulse waveform and AC voltage waveform when the voltage doubler switch SW is OFF, respectively. It is shown.

  As can be seen from the comparison between FIG. 19 and FIG. 20, the DC voltage Vdc supplied to the inverter circuit 100 when the voltage doubler switch SW is on is supplied to the inverter circuit 100 when the voltage doubler switch SW is off. It is twice the DC voltage ½ Vdc applied. Therefore, if the modulation rate is the same when the voltage doubler switch SW is off, the voltage applied to the linear motor 13 is applied to the linear motor 13 when the voltage doubler switch SW is on. Half the voltage. In other words, in the state where the voltage doubler selector switch SW is off, if the modulation factor is twice the modulation factor when the voltage doubler selector switch is on, the voltage applied to the linear motor 13 is the voltage doubler selector switch. The operation state in which the piston 1 reciprocates with the same stroke as the stroke when the voltage doubler selector switch is on is maintained substantially the same as the voltage applied to the linear motor 13 when is on.

  In the above embodiment, the carrier frequency does not change and the lower limit value of the modulation rate at which the stroke Xp can be detected is the same regardless of whether the voltage doubler selector switch SW is on or off. It is a constant value (same). Therefore, with respect to the minimum voltage applied to the linear motor 13, the voltage value necessary for detecting the stroke Xp when the voltage doubler switch SW is off is the value of the stroke Xp when the voltage doubler switch is on. It is half the voltage value required for detection. As a result, proper stroke detection can be performed even in a period in which the voltage applied to the linear motor 13 is small.

  Next, voltage doubler switching control in the control method of the Stirling refrigerator of the present embodiment will be described with reference to FIG. Note that the lower limit value of the modulation rate at which the stroke Xp of the present embodiment can be detected is M (%), as in the first embodiment. The method for determining M is the same as described above, in the experiment executed in advance, in the state where the Stirling refrigerator 40 is driven, the stroke Xp of the piston 1 is measured, and the modulation factor when the stroke Xp is obtained accurately Is defined as the value of. In other words, the lower limit value of the modulation rate is a value obtained in advance by experiment, and is a substantially minimum value of the modulation rate that can accurately detect the pulse current.

  As shown in FIG. 21, in the voltage doubler switching control, first, in step S101, it is determined whether or not a control signal for driving the Stirling refrigerator 40 is input to the microcomputer 1000.

  In step S101, if the control signal for starting the Stirling refrigerator 40 is not input to the microcomputer 1000, the process of step S101 is repeated. On the other hand, if a control signal for starting the Stirling refrigerator 40 is input to the microcomputer 1000 in step S101, control for turning off the voltage doubler selector switch SW is executed in step S102.

  Next, in step S103, inverter control is started and the PWM modulation rate is increased to M (%). Next, in step S104, detection of the actual stroke Xp of the piston 1 is started.

  Next, in step S105, it is determined whether or not the stroke Xp is equal to or less than the stroke command value. If the actual stroke Xp is not more than the stroke command value in step S105, the modulation factor is increased in step S106. On the other hand, if the stroke Xp is not less than or equal to the stroke command value in step S105, it is determined in step S107 whether or not the modulation factor is less than or equal to M.

  In step S107, if the modulation factor is a value equal to or less than M, the process of step S105 is repeated. On the other hand, if the PWM modulation rate is a value equal to or greater than M in step S107, the voltage modulation rate is lowered.

  In short, in Step 105 to Step 108, in order to prevent the actual stroke Xp of the piston 1 from becoming larger than the stroke command value and causing the piston 1 and the displacer 2 to collide, the microcomputer 1000 has the stroke command value. From the comparison result between the stroke command value output from the output unit 101 and the actual stroke Xp value, the modulation rate is increased or decreased.

  In addition, after either step S106 or S108 is completed, the process of step S109 is executed. In step S109, it is determined whether or not the modulation rate is 2M + α (%) or more. In step S109, if the modulation rate is not 2M + α (%) or more, the process of step S105 is executed. Note that α is a hysteresis width for preventing frequent switching of control.

  On the other hand, if the modulation rate is 2M + α (%) or more in step S109, the modulation rate is changed to a value half of the current value in step S110, and the voltage doubler switch SW is set to ON. . Thereafter, the process of step S111 is performed.

  In step S111, it is determined whether or not the stroke Xp is equal to or less than the stroke command value. If the stroke Xp is a value equal to or smaller than the stroke command value in step S111, the modulation factor is increased in step S112. On the other hand, if the stroke Xp is not less than or equal to the stroke command value in step S111, the modulation factor is lowered in step S113.

  In short, also in steps 111 to 113, in order to prevent the actual stroke Xp of the piston 1 from becoming larger than the stroke command value and causing the piston 1 and the displacer 2 to collide, the microcomputer 1000 has the stroke command value. From the comparison result between the stroke command value output from the output unit 101 and the actual stroke Xp value, the modulation rate is increased or decreased.

  After any of steps S112 and S113 is completed, it is determined in step S114 whether or not the modulation rate is M (%) or less. If the modulation factor is not M (%) or less in step S114, the process in step S111 is performed again. If the modulation rate is a value equal to or less than M (%) in step S114, the modulation rate is changed to a value twice the current modulation rate in step S115, and the voltage doubler selector switch SW is set to OFF. . Thereafter, the process of S105 is executed. In addition, when the stop command signal of the Stirling refrigerator 40 is input to the microcomputer 1000 while steps S105 to S115 are being executed, the steps until the operation start command signal of the Stirling refrigerator 40 is input. The process of S101 is repeated.

  FIG. 22 shows the relationship between the effective value of the output voltage and the modulation factor when the voltage doubler switching control shown in FIG. 21 is executed. As shown in FIG. 22, in the PWM control in the control method of the linear motor control system and the Stirling refrigerator control system of the present embodiment, first, the modulation factor is changed from zero to M.

  A period between the modulation rate of zero and M is a period during which the stroke Xp cannot be detected. Thereafter, when the modulation rate becomes M, detection of the stroke Xp is started. Next, the modulation rate is increased from M to 2M + α in a state where stroke detection is performed. However, when the modulation factor is between M and 2M + α, the modulation factor gradually increases as the load on the Stirling refrigerator 40 increases, but the modulation factor gradually decreases as the load on the Stirling refrigerator 40 decreases. .

  Thereafter, when the modulation rate becomes 2M + α, the voltage doubler selector switch SW is set to ON. As a result, the relationship between the effective value of the output voltage and the modulation factor can be calculated from the line indicated by the modulation factor-output voltage characteristic when the voltage doubler switch (SW) is OFF to the modulation factor-output when the voltage multiplier switch (SW) is ON. Changes to the line indicated by the voltage characteristics. At this time, the modulation factor is lowered to (2M + α) / 2 when the voltage doubler selector switch SW is turned ON. Thereafter, the modulation rate is gradually increased from (2M + α) / 2.

  In the above-described state, when the load of the Stirling refrigerator 40 is reduced and the modulation rate needs to be lowered, the modulation rate gradually decreases to M. Thereafter, when the modulation factor reaches M, the voltage doubler selector switch SW is set to OFF. At the same time, the modulation rate is increased to 2M. As a result, the relationship between the effective value of the output voltage and the modulation factor can be calculated from the line indicated by the modulation factor-output voltage characteristic when the voltage doubler switch (SW) is ON to the modulation factor-output when the voltage multiplier switch (SW) is OFF. Changes to the line indicated by the voltage characteristics. Thereafter, when the load on the Stirling refrigerator 40 increases, the modulation factor gradually increases, and when the load on the Stirling refrigerator 40 decreases, the modulation factor gradually decreases.

  According to the voltage doubler switching control of the present embodiment, when the load on the Stirling refrigerator 40 is small and the voltage applied to the linear motor 13 is small, the voltage doubler switch SW is set to OFF. At the same time, the modulation factor is set to double. Thereby, the voltage applied to the linear motor 13 is maintained at substantially the same value before and after the switching of the voltage doubler selector switch SW. Therefore, the width of the PWM pulse can be increased while maintaining the stroke of the piston 1. As a result, even when the load on the Stirling refrigerator 40 is further reduced and the modulation rate is reduced, proper stroke detection can be performed.

  Also, if the voltage doubler selector switch SW is previously set to OFF before the modulation factor is unconditionally increased to M when the Stirling refrigerator 40 is started, it is applied to the linear motor 13 that is unconditionally increased. The voltage is reduced to half of the voltage applied to the linear motor 13 which is unconditionally increased when the voltage doubler selector switch SW is off. Therefore, when the load of the Stirling refrigerator 40 is small, it is possible to avoid the occurrence of a problem that the piston 1 and the displacer 2 collide.

  Further, the voltage doubler switching control in the Stirling refrigerator control method of the present embodiment may be as shown in FIG. The voltage doubler switching control shown in FIG. 23 differs from the voltage doubler control shown in FIG. 21 in the processing from steps S100a to S110c and S115a to S115c.

  In the voltage doubler switching control shown in FIG. 23, the current modulation factor (m) value is stored in step 110a, and the application of voltage to the linear motor 13 is stopped. That is, the PWM modulation factor value at that time is stored, and the operation of the Stirling refrigerator 40 is temporarily stopped. Thereby, the operation of the piston 1 is temporarily stopped. Next, in S110b, the voltage doubler switch SW is switched from the off state to the state in which the double voltage switch SW is on. After that, after the operation of the Stirling refrigerator 40 is started, the modulation factor is gradually increased from zero to m / 2.

  In step S115a, the current value of the modulation factor (m) is stored, and the application of voltage to the linear motor 13 is stopped. That is, the current modulation factor value is stored and PWM control is stopped. Thereby, the operation of the piston 1 is stopped. Next, in S115b, the voltage doubler switch SW is switched from the off state to the double voltage switch SW. Next, in S115c, after the operation of the Stirling refrigerator 40 is started, the modulation factor is gradually increased from zero to m / 2.

  According to the voltage doubler switching control shown in FIG. 23, the relationship between the effective value of the output voltage and the modulation rate as shown in FIGS. 24 and 25 is obtained. That is, in the voltage doubler switching control shown in FIG. 23, when the voltage doubler switch SW is switched from ON to OFF and the voltage doubler switch SW is switched from OFF to ON, the Stirling refrigerator 40 is temporarily stopped. Therefore, when the voltage doubler selector switch SW is switched, the modulation factor is once set to zero.

  According to the voltage doubler switching control as shown in FIGS. 23 to 25, the operation of the piston 1 of the Stirling refrigerator 40 is prevented from being discontinuous when the voltage doubler switch SW is switched.

  24 shows the relationship between the effective value of the output voltage and the modulation rate when the modulation rate is increasing, and FIG. 25 shows the effective value and modulation of the output voltage when the modulation rate is reduced. The relationship of rate is shown.

  As shown in FIG. 24, when the Stirling refrigerator 40 is started, the modulation rate gradually increases from zero to M. At this time, the microcomputer 1000 cannot detect the stroke Xp until the modulation rate becomes M. Thereafter, when the modulation rate becomes larger than M, detection of the stroke Xp is started.

  Next, the modulation rate increases from M to 2M + α. However, during the period from the modulation rate M to the modulation rate 2M + α, the modulation rate increases or decreases according to the increase or decrease of the load of the Stirling refrigerator 40. Thereafter, when the modulation rate becomes 2M + α, the microcomputer 1000 stores the value of the modulation rate m, stops the operation of the Stirling refrigerator 40, and switches the voltage double switch SW from the off state to the voltage double switch SW. A process for switching to the ON state is executed. Thereafter, the operation of the Stirling refrigerator 40 is started, and the modulation factor is gradually increased from zero to m / 2. Thereafter, the modulation rate is increased or decreased based on the comparison result between the actual stroke Xp obtained by the stroke detection and the stroke command value.

  On the other hand, as shown in FIG. 25, the microcomputer 1000 stores the value of the modulation factor m / 2 at that time when the modulation factor decreases and becomes a value of M or less, and operates the Stirling refrigerator 40. Stop and set the modulation rate to zero. Further, the microcomputer 1000 switches from the state where the voltage doubler switch SW is on to the state where the voltage doubler switch SW is off. Thereafter, in a period in which the modulation factor is from M to 2M, the microcomputer 1000 starts the operation of the Stirling refrigerator 40 and gradually increases the modulation factor from zero to m. Thereafter, the modulation rate is increased or decreased based on the comparison result between the actual stroke Xp obtained by the stroke detection and the stroke command value.

  In the present embodiment, voltage doubler circuit B is used as a voltage changing circuit that can change the value of the DC voltage applied to inverter circuit 100. For example, the voltage changing circuit may be a PAM (Pulse Amplitude). Any circuit such as a modulation circuit may be used. In short, the voltage changing circuit may be any circuit as long as it can change the value of the DC voltage applied to the inverter circuit 100 according to a command from the microcomputer 1000.

(Embodiment 3)
Next, the linear motor control system and the Stirling refrigerator control system of the third embodiment will be described with reference to FIGS. In the linear motor control system and the Stirling refrigerator control system according to the present embodiment, as shown in FIG. 26, the stroke control unit 102 executes control for switching the voltage doubler selector switch SW and changes the carrier frequency. Perform control for

  That is, the linear motor control system and the Stirling refrigeration system of the present embodiment are characterized by the features of the linear motor control system and the Stirling refrigeration system of the first embodiment and the features of the linear motor control system and the Stirling refrigeration control system of the second embodiment. Are combined.

  In the carrier frequency double voltage switching control in the present embodiment, as shown in FIG. 27, first, in step S301, it is determined whether or not there is an input for starting the Stirling refrigerator 40. If the control signal for starting the Stirling refrigerator 40 is not input to the microcomputer 1000 in S301, the step of step S301 is repeated. On the other hand, if there is an input of a command for starting the Stirling refrigerator 40 in step S301, control for changing the control state from control of the standard carrier frequency to control of the low carrier frequency is executed in step S302. At the same time, control for setting the voltage doubler selector switch to OFF is executed.

  Next, in step S303, inverter control is started and the PWM modulation rate is increased to M / 2 (%). Next, in step S304, detection of the stroke Xp is started. Thereafter, in step S305, it is determined whether or not the stroke Xp is equal to or less than the stroke command value. If the stroke Xp is smaller than the stroke command value in S305, the modulation factor is increased in step S306.

  On the other hand, if the stroke Xp is not less than or equal to the stroke command value in step S305, it is determined in step S307 whether or not the modulation factor is M / 2 or less. In step S307, if the voltage modulation rate is M / 2 or less, the process of step S305 is performed again.

  On the other hand, if the voltage modulation rate is not M / 2 or less in S307, the PWM modulation rate is lowered in step S308. Step S309 is executed even after any of Steps S306 and S308 is completed.

  In step S309, it is determined whether or not the modulation rate is 2M + α (%) or more. In step S309, if the modulation factor is not 2M + α (%) or more, the process of step S305 is executed. On the other hand, if the voltage modulation rate is 2M + α (%) or more in S309, the standard carrier frequency and voltage doubler switch (SW) ON is changed from the low carrier frequency and voltage doubler switch (SW) OFF state in step S310. The control state is switched to the state. At this time, the modulation factor is lowered to (2M + α) / 2 as the voltage doubler selector switch SW is turned on.

  Thereafter, the process of step S311 is performed. In step S311, it is determined whether or not the stroke Xp is equal to or less than the stroke command value. In step S311, if the stroke Xp is equal to or less than the stroke command value, the modulation factor is increased in step S312. On the other hand, if the stroke Xp is not a value equal to or less than the stroke command value in step S311, the modulation factor is lowered in step S313. After either step S312 or step S313 is completed, it is determined in step S314 whether or not the modulation rate is a value equal to or less than M (%).

  In step S314, if the voltage modulation factor is not M (%) or less, the process of step S311 is performed. On the other hand, if the modulation factor is equal to or less than M (%) in step S314, in step S315, the standard carrier frequency and voltage doubler switch (SW) ON state are changed to the low carrier frequency and voltage doubler switch (SW) OFF state. The control state is switched to the state. At this time, the modulation factor is increased to 2M as the voltage doubler switch SW is turned off.

  When the carrier frequency double voltage switching control shown in FIG. 28 is performed, the relationship between the effective value of the output voltage and the modulation rate as shown in FIG. 28 is obtained.

  As shown in FIG. 28, stroke detection cannot be performed properly during the period from the modulation rate of zero to M / 2. Therefore, after the modulation factor is increased from zero to M / 2, detection of the stroke Xp is started. At this time, the microcomputer 1000 is executing control of low carrier frequency and voltage doubler changeover switch (SW) OFF.

  Thereafter, if the modulation rate further increases and exceeds 2M + α, the microcomputer 1000 executes control of the standard carrier frequency and the voltage doubler switch (SW) ON. At this time, the voltage doubler selector switch SW is turned on and the modulation factor is lowered to (2M + α) / 2. As a result, the relationship between the effective value of the output voltage and the modulation factor can be calculated from the line indicated by the modulation factor-output voltage characteristic when the voltage doubler switch (SW) is OFF to the modulation factor-output when the voltage multiplier switch (SW) is ON. Changes to the line indicated by the voltage characteristics.

  On the other hand, when the modulation rate decreases to M in the state where the standard carrier frequency and the voltage doubler selector switch (SW) are ON, the microcomputer 1000 executes control of the low carrier frequency and the voltage doubler selector switch (SW) OFF. Further, the voltage doubler changeover switch SW is turned off and the modulation factor is increased to 2M. As a result, the relationship between the effective value of the output voltage and the modulation factor can be calculated from the line indicated by the modulation factor-output voltage characteristic when the voltage doubler switch (SW) is ON to the modulation factor-output when the voltage multiplier switch (SW) is OFF. Changes to the line indicated by the voltage characteristics.

  Also in the present embodiment, as in the first and second embodiments, during the period in which the stroke detection is performed, based on the comparison result between the actual stroke Xp obtained by the stroke detection and the stroke command value. As a result, the modulation rate is increased or decreased.

  Further, the carrier frequency doubled voltage switching control of the present embodiment may be as shown in FIG. In the carrier frequency double voltage switching control shown in FIG. 29, the processing from step S309a to step S310a and the processing from step S314a to step S315a are different from the carrier frequency double voltage switching control shown in FIG.

  In step S309a, the current modulation factor (m) is recorded, and the application of voltage is stopped, that is, the Stirling refrigerator 40 is stopped. Next, in step S310, switching from the low carrier frequency and voltage doubler switch (SW) OFF state to the standard carrier number and voltage doubler switch (SW) ON state is executed. Thereafter, in step S310a, the operation of the Stirling refrigerator 40 is started and the modulation factor is increased to m / 2. Thereafter, the process of step S311 is executed.

  In step S314a, the current modulation factor m is stored and voltage application is stopped. That is, the Stirling refrigerator 40 is stopped. Thereafter, in step 315, the control state is switched from the standard carrier number and voltage doubler switch (SW) ON state to the low carrier frequency and voltage doubler switch (SW) OFF state. Thereafter, in S315a, after the operation of the Stirling refrigerator 40 is started, the modulation factor is increased to 2 × m. Thereafter, the process of step S305 is executed. When the carrier frequency double voltage switching control shown in FIG. 29 is executed, the relationship between the effective value of the output voltage and the modulation rate is as shown in FIG. When the modulation rate decreases, the relationship between the effective value of the output voltage and the modulation rate as shown in FIG. 31 is obtained.

(Embodiment 4)
Next, the linear motor control system and the Stirling refrigerator control system according to the embodiment of the present invention will be described with reference to FIGS. The linear motor control system and Stirling refrigerator control system of the present embodiment differ from the linear motor control system and Stirling refrigerator control system of Embodiments 1 to 3 only in the stroke detection method described below.

  In the linear motor control system of the present embodiment, all circuit variables (voltage, current, etc.) of the linear motor are regarded as sine waves, and the induced voltage generated in the linear motor is calculated in units of effective values for one cycle. According to this method, the stroke of the mover can be calculated with a simple process and with high accuracy. Hereinafter, the linear motor control system of the present embodiment will be specifically described.

  First, the configuration of the linear motor control system of the present invention will be described with reference to FIGS. FIG. 32 shows the configuration of the linear motor drive circuit. 32, i (t) is a function indicating a motor current flowing through the linear motor, v (t) is a function indicating a motor voltage applied to the linear motor, and e (t) is a reciprocating motion of the mover. Is a function indicating the induced voltage (counterelectromotive force) generated by, R is a constant indicating the winding resistance value of the linear motor, and L is a constant indicating the inductance of the linear motor. Note that t is a variable indicating time.

  When the circuit shown in FIG. 32 is represented by a general expression, the following expression (3) is obtained.

  Further, the induced voltage e (t) is expressed as a function of the induced voltage coefficient α (thrust coefficient) depending on the magnetic characteristics and drive circuit of the linear motor and the velocity vp (t) of the mover (piston). Equation (4) is obtained.

  FIG. 33 shows a motor current i (t), a motor voltage v (t), and an induced voltage e (t when a sinusoidal motor voltage V (t) having an effective value V and an angular velocity ω is applied to the linear motor. ) Shows a change in one cycle of each instantaneous value.

  As shown in FIG. 33, it is assumed that the mover of the linear motor has a motion represented by an ideal sine wave, that is, a simple vibration, and has the same angular velocity as the angular velocity ω of the motor voltage v (t). If the value of the induced voltage coefficient α is constant regardless of the position of the mover, the current i (t) is also an ideal sine wave, and the angular velocity ω of the motor voltage v (t) Have the same angular velocity.

  If the effective value of the motor current is I, the effective value of the motor voltage is V, and the effective value of the induced voltage is E, the motor current i (t), the motor voltage v (t), The instantaneous values of the induced voltage e (t) are expressed by the following equations (5) to (7) with the phase of the current i (t) as a reference.

  In Equation (6), θ is a phase difference between the motor current i (t) and the motor voltage v (t).

In Equation (7), θ ₂ is a phase difference between the motor current i (t) and the induced voltage e (t).

  Further, the term of the induction voltage L · di (t) / dt of the coil in the equation (3) is expressed by the following equation (8).

The above items are illustrated in accordance with the vector notation method of the AC amount as shown in FIG.
Therefore, if the effective value V of the motor voltage, the angular velocity ω, the effective value I of the motor current, and the phase difference θ can be obtained, the effective value E of the induced voltage E, using the following equations (9) and (10): And the phase difference θ ₂ can be calculated.

  The effective value I of the motor current and the effective value V of the motor voltage are expressed by the following equations (11) and (12).

In equation (11), each of i (1), i (2),... I (n-1), i (n) is an instantaneous value of the motor current i (t). n is a natural number. Further, the sampling period of the instantaneous value of the motor current i (t) coincides with the PWM carrier period.

In equation (12), each of v (1), v (2),... V (n−1), v (n) is an instantaneous value of the motor voltage v (t). n is a natural number. The sampling period of the instantaneous value of the motor voltage v (t) is coincident with the PWM carrier period.

  The speed vp (t) of the mover of the linear motor can be summarized by substituting Equation (7) into Equation (4).

It becomes.
When both sides of the equation (13) are integrated over time, the position Xp of the mover is expressed by the following equation (14).

  In Expression (14), Xp0 is a value of the neutral position in one cycle of the mover, that is, a value of the center position of the stroke.

  Therefore, the distance Xpmax between the neutral position Xp0 and the position farthest from the neutral position Xp0 is expressed by the following equation (15).

  Since the distance Xpmax corresponds to the amplitude of the mover, the stroke ST of the mover is a value twice the distance Xpmax, that is, 2Xpmax.

  From the above, if the respective values of the effective value V of the motor voltage, the angular velocity ω, the effective value I of the motor current, and the phase difference θ between the motor current i (t) and the motor voltage v (t) are obtained, the above calculation is performed. The stroke ST of the linear motor can be accurately calculated using the equation.

Next, a method for acquiring each variable of the linear motor driving circuit described above will be described.
Since the angular velocity ω of the motor voltage v (t) and the motor current i (t) corresponds to a control command transmitted from the microcomputer to the inverter circuit, the microcomputer can always grasp the angular velocity ω. is there.

The effective value I of the motor current and the effective value V of the motor voltage are detected as follows. First, the AD (Analog to Digital) of the microcomputer at a predetermined sampling period
Using the conversion function, the instantaneous values of the motor voltage v (t) and the motor current i (t) are measured. Next, the square root of the mean square value of all instantaneous values of the motor current i (t) in one cycle, that is, the effective value I of the motor current is calculated using the above-described equation (11). Further, the square root of the mean square value of all instantaneous values of the motor voltage v (t) in one cycle, that is, the effective value V of the motor voltage is calculated using the above-described equation (12). Note that, under the assumption that each of the motor current i (t) and the motor voltage v (t) is a sine wave, the maximum value and minimum value of the motor current i (t) for one cycle, and one cycle If the maximum value and the minimum value of the motor voltage v (t) are obtained and the effective value I of the motor current and the effective value V of the motor voltage are calculated using these values, the calculation becomes easier.

  In the detection of the phase difference θ described above, first, the motor current i (t) and the motor voltage v (t) are used by using the instantaneous values of the motor current i (t) and the motor voltage v (t). Figure out the phase of each zero cross point. Next, the phase difference between the zero cross points of the motor current i (t) and the motor voltage v (t) is calculated. This phase difference is defined as a phase difference θ between the motor current i (t) and the motor voltage v (t).

  Also in the linear motor of the above-described embodiment, the induced voltage coefficient α decreases due to the influence of the magnetic saturation as in the conventional linear motor. Therefore, in order to solve this problem, in the linear motor of the present embodiment, there is a process for correcting the induced voltage coefficient α using the magnetic strength (effective value Φ of the total number of magnetic fluxes in one cycle). Executed. As a result, the influence of magnetic saturation of the linear motor is removed, and a decrease in the accuracy of calculating the stroke of the mover is prevented.

  In order to perform the above-described correction when the linear motor is operating, the relationship between the magnetic strength (effective value Φ of the number of magnetic fluxes in one cycle) and the induced voltage coefficient α as shown in FIG. It is necessary to prepare an approximate expression or a data table indicating the correlation curve to be shown in advance.

  For this purpose, the induced voltage coefficient α and the effective value Φ of the total number of magnetic fluxes are set under appropriate linear motor operating conditions (the magnitude and frequency of the voltage applied to the linear motor and the load conditions) for creating the correlation curve. taking measurement.

  The method of measuring the induced voltage coefficient α is to measure the stroke ST of the mover with an external sensor (such as a laser displacement meter) under arbitrary operating conditions (the magnitude of the motor voltage, the angular velocity of the motor voltage, and the load condition) The effective value V of the motor voltage, the effective value I of the motor current, and the phase difference θ are measured by an external measuring instrument. The measured stroke ST (= 2Xpmax), the effective value V of the motor voltage, the effective value I of the motor current, and the phase difference θ, the winding resistance value R, the coil inductance L, The induced voltage coefficient α is calculated by substituting the angular velocity ω into the above equation (15).

  Next, how to obtain the effective value Φ of the total number of magnetic fluxes generated in the magnetic circuit of the linear motor drive circuit will be described.

The instantaneous value of the total number of magnetic fluxes φ (t) representing the magnetic strength in the magnetic circuit of the linear motor is equal to the instantaneous value of the number of magnetic fluxes φI (t) generated by the current flowing in the coil and the reciprocation of the magnet of the mover. The sum of the number of magnetic fluxes φmagnet (t) generated by the motion and the instantaneous value is expressed by the following equation (16).

  The instantaneous value of the number of magnetic fluxes φI (t) is expressed by the following equation (17).

  Substituting equation (5) into equation (17) yields the following equation (18).

  The instantaneous value of the number of magnetic fluxes φmagnet (t) is expressed by the following equation (19).

  Assuming that the instantaneous offset value of the number of magnetic fluxes φmagnet (t) is 0 and substituting equation (7) into equation (19), the following equation (20) is obtained.

Furthermore, the vector notation of the AC amount, effective value of the total number of magnetic fluxes [Phi, To illustrate the effective value [Phi _I and the effective value φmagnet the number of magnetic fluxes is as shown in FIG. 36.

  Note that the vector diagram shown in FIG. 36 is drawn based on the phase of the motor current as in the vector diagram shown in FIG.

  In FIG. 36, the effective value Φ of the total number of magnetic fluxes is a vector sum of the effective value ΦI of the number of magnetic fluxes generated by the current flowing through the coil and the effective value Φmagnet of the number of magnetic fluxes generated by the reciprocating motion of the magnet of the mover. Therefore, the effective value Φ of the total number of magnetic fluxes in one cycle is obtained by the following formula (21).

  By the above calculation, data of the induced voltage coefficient α and the effective value Φ of the total number of magnetic fluxes under a certain condition can be obtained. In this way, when plotting data of a large number of induced voltage coefficients α and the effective value Φ of the total magnetic flux number under various conditions, the induced voltage coefficient α and the effective value Φ of the total magnetic flux number as shown in FIG. A correlation curve showing the relationship between

If a data table or an approximate expression for specifying this correlation curve is stored in the memory inside the microcomputer, the effective value of the total number of magnetic fluxes can be obtained if the operating conditions of the linear motor are within a predetermined range. Using Φ, an induced voltage coefficient α suitable for the operating condition can be obtained. Therefore, according to the linear motor control system of the present embodiment, the stroke can be calculated with high accuracy by eliminating the adverse effect of magnetic saturation of the linear motor.

  The correction of the induced voltage coefficient α is not limited to the system that calculates the stroke ST using the effective value E of the induced voltage as described above, but also a system as disclosed in JP-A-2003-65244, That is, the present invention can also be used in a system that calculates a stroke using an instantaneous value (maximum value) of an induced voltage, an induced voltage coefficient, and an angular velocity.

  Specifically, as shown in FIG. 37, the linear motor control system of the present embodiment described above includes a linear motor 13 having a reciprocating mover, an AC power supply G and a smoothing circuit P constituting a DC power supply, The inverter circuit 100 that converts the DC power supplied from the above-described DC power source into AC power and supplies the AC power to the linear motor 13 and the microcomputer 1000 that controls the inverter circuit 100 by PWM (Pulse Width Modulation) are provided. I have.

  In addition, when the difference between the actual stroke Xp and the target stroke command value is large, the stroke control unit 102 changes the PWM modulation rate according to the difference to target the actual stroke Xp. Control to bring it closer to the stroke command value. Therefore, it is possible to increase the accuracy of control of the stroke of the mover.

  The motor voltage detector V transmits to the microcomputer 1000 a voltage signal that can specify the instantaneous value of the motor voltage v (t) of the linear motor 13. The microcomputer 1000 uses the AD conversion function to calculate the motor voltage v (t) applied to the linear motor 13 based on a signal that can identify the potential difference between both terminals of the linear motor 13.

  Motor current detectors A and S detect motor current i (t) flowing through shunt resistor S connected to the DC (Direct Current) line on the input side of inverter circuit 100. The motor current i (t) flows into the shunt resistor S only during the PWM pulse ON period. Therefore, the motor current detectors A and S transmit the potential difference signals at both ends of the shunt resistor S during the above-described ON period to the microcomputer 1000. The microcomputer 1000 converts the potential difference signal into a current signal that can specify the motor current i (t) by the AD function. Therefore, it is desirable that the sampling period of the motor current i (t) coincides with the PWM carrier period.

  The stroke calculation unit 104 receives a voltage signal that can specify the instantaneous value of the motor voltage v (t) and a current signal that can specify the instantaneous value of the motor current i (t), and receives a motor signal for each cycle of the sine wave. The effective value V of the voltage, the effective value I of the motor current, and the phase difference θ are calculated.

  As described above, the effective value V of the motor voltage and the effective value I of the motor current are calculated as follows. The square root of the root mean square value of the total instantaneous values of the motor voltage v (t) in one cycle is a predetermined sampling cycle. And the method using the square root of the root mean square of all instantaneous values of motor current i (t) in one cycle, or the assumption that each of motor voltage v (t) and motor current i (t) is a sine wave Below, there is a method using the maximum value and the minimum value of the motor voltage v (t) and the motor current i (t), respectively.

  In this embodiment, since the instantaneous value of the motor current i (t) is calculated using the potential difference between both ends of the shunt resistor S of the motor current detectors A and S, the ON period of the PWM pulse is short. In this case, that is, at a timing close to the zero cross point of the motor voltage v (t), it is difficult to detect the instantaneous value of the motor current i (t). Therefore, under the assumption that the motor current i (t) is a sine wave, the maximum value and the minimum value during one cycle of the motor current i (t) and the waveform of the motor current stored in advance can be specified. It is desirable to use a method of estimating the total instantaneous value of the motor current i (t) during one cycle using data. In this method, the effective value I of the motor current is calculated using the following calculation formula.

  First, in the memory of the microcomputer 1000, first, as shown in FIG. 38, 1 of the motor current i (t) calculated by using the potential difference signals at both ends of the shunt resistors S of the motor current detectors A and S. A data string of all instantaneous values for a period is stored. Next, the microcomputer 1000 extracts the maximum value Imax and the minimum value Imin of the instantaneous value of the motor current i (t) from the data string, and the maximum value Imax and the minimum value Imin are expressed in the following equation (22). Substituting and calculating the effective value I of the motor current.

  According to the method as described above, the effective value I of the motor current can be accurately calculated even when the instantaneous value of the motor current i (t) at a timing close to the zero cross point of the voltage cannot be accurately detected.

  The phase difference θ between the motor voltage and the motor current is detected by the phase difference between the zero cross point of the motor current i (t) waveform and the zero cross point of the motor voltage v (t) waveform. However, in the current detection method using the shunt resistor described above, when the phase difference θ is small, that is, when the motor voltage zero cross point and the motor current zero cross point are close, the PWM pulse width is small and the motor current It is difficult to detect the zero cross point. In this case, it is desirable to use a method for estimating the phase of the zero cross point of the motor current based on the phase when the motor current becomes the maximum value in one cycle and the phase when the motor current becomes the minimum value.

  Effective value V of motor voltage, effective value I and phase difference θ of motor current calculated by the method as described above, angular velocity ω of sine wave specified by PWM control command by PWM control command, and previously stored in memory If the winding resistance value R and inductance L thus set are substituted into the above-described equation (15), the actual stroke ST of the mover at that time can be calculated.

  Further, in order to eliminate the influence of magnetic saturation of the linear motor 13, the microcomputer 1000 according to the present embodiment has a relationship between the induced voltage coefficient α and the effective value Φ of the total number of magnetic fluxes in the memory as shown in FIG. Is provided with a data table 105 in which approximate expression data or a data string indicating a correlation curve is stored. The correlation curve described above is obtained in advance by experiments.

  When calculating the actual stroke ST of the linear motor 13, first, the effective value Φ of the total number of magnetic fluxes is calculated using the above-described equation (21). Next, the induced voltage coefficient α is corrected using the data table 105 in which the approximate expression or data string of the correlation curve stored in the memory is stored. Thereafter, the stroke ST in which the adverse effect of magnetic saturation of the linear motor 13 is eliminated is calculated using the corrected value of the induced voltage coefficient α.

  Another example of a linear motor control system may be configured as shown in FIG. Next, another example of the linear motor control system shown in FIG. 39 will be briefly described. Since the linear motor control system shown in FIG. 39 can obtain the same effect as the linear motor control system shown in FIG. 37, only different points will be described.

  In another example of the linear motor control system shown in FIG. 39, the motor voltage detector W is provided between the DC power supply and the inverter circuit 100. The DC power voltage signal detected by the motor voltage detector W is transmitted to the motor voltage estimation unit 106 in the microcomputer C. The motor voltage estimation unit 106 estimates the effective value V of the motor voltage using the DC power voltage signal and the modulation rate command value data transmitted from the stroke control unit 102 to the motor voltage estimation unit 106. Then, the effective value V of the estimated motor voltage is transmitted from the motor voltage estimation unit 106 to the stroke calculation unit 104.

  On the other hand, the motor current detector H has a Hall element and is provided between the inverter circuit 100 and the linear motor 13. Further, the motor current detector H detects a change in magnetic flux by the Hall element, and transmits the change in the magnetic flux to the stroke calculation unit 104 in the microcomputer 100 as a current signal of the motor current.

  Next, stroke control processing performed by the microcomputer 1000 will be described with reference to FIGS. 40 and 41. FIG.

  In principle, the stroke control process is performed for each cycle of the linear motor operation. First, in S1, a voltage signal for one cycle, that is, an instantaneous value of the motor voltage v (t) [t = 1 to n] is received. At the same time, in S2, a current signal for one cycle, that is, an instantaneous value of the motor current i (t) [t = 1 to n] is received. That is, in S1 and S2, a voltage signal and a current signal that can specify a voltage waveform and a current waveform for one period are received. Next, in S3, the phase difference θ between the motor current i (t) and the motor voltage v (t) is calculated using the voltage signal and current signal for one cycle. Next, in S4, the angular velocity ω of the motor voltage v (t) or the motor current i (t) is calculated using the voltage signal or current signal for one cycle. However, the angular velocity ω may be a predetermined fixed value and a value stored in the memory.

  Next, in S5, the effective value V of the motor voltage and the effective value I of the motor current are calculated. In the process of S5, v (1), v (2),..., V (n−1), v (n) are sequentially compared in S52 shown in FIG. Next, in S53, the maximum value v (max) and the minimum value v (min) of v (1), v (2), ..., v (n-1), v (n) are determined. Next, in S54, the effective value V = {v (max) −v (min)} / 2√2 of the motor voltage is calculated. Next, in S55, i (1), i (2),..., I (n-1), i (n) are sequentially compared. Next, in S56, the maximum value i (max) and the minimum value i (min) of i (1), i (2), ..., i (n-1), i (n-1) are determined. Next, in S57, the effective value I of the motor current is calculated using Expression (22).

  Thereafter, the process of S6 in FIG. 41 is executed. In S6, the winding resistance value R and the inductance L are read from the memory. Next, in S7, the effective value V of the motor voltage, the effective value I of the motor current, the phase difference θ, the winding resistance value R, and the inductance L are substituted into the above-described equation (1), and the effective value E of the induced voltage is obtained. calculate.

Next, in S8, the effective value V of the motor voltage, the effective value I of the motor current, the effective value E of the induced voltage, the phase difference θ, and the winding resistance value R are substituted into the equation (10), and the phase difference θ ₂ Is calculated. In S9, the effective value E of the induced voltage, the effective value I of the motor current, the angular velocity ω, the inductance L, and the phase difference θ ₂ are substituted into the equation (21) to calculate the effective value Φ of the total number of magnetic fluxes. Next, in S10, the value of the induced voltage coefficient (thrust coefficient) α is corrected using the approximate expression data or the data table (FIG. 35) indicating the effective value Φ of the total number of magnetic fluxes and the correlation curve.

  Next, in S11, the actual stroke ST of the mover is calculated by substituting the induced voltage coefficient α, the effective value E of the induced voltage, and the angular velocity ω into Equation (2). Then, the process of S1-S11 is repeated for every period.

  In the first and second embodiments described above, the actual stroke of the piston 1 is calculated using the voltage signal and the current signal described above. The calculation method has been described in the first and second embodiments. The method is not limited. That is, in the linear motor control system of the present invention, the actual stroke of the piston 1 can be specified using the value of the current applied to the linear motor 13 and the value of the current flowing through the linear motor 13. Any method may be used as long as it is.

(Embodiment 5)
Next, the linear compressor control system according to the embodiment of the present invention will be described with reference to FIG. As the linear motor control system of the linear compressor 540 of the present embodiment, the linear motor control system described in the first to fourth embodiments is used.

  As shown in FIG. 42, the linear compressor 540 includes a cylinder 542 installed in a casing 541, a piston 543 that reciprocates in the cylinder 542, and a linear that is installed on the outer periphery of the cylinder 542 and drives the piston 543. A motor 501, a piston spring (plate spring) 546 that biases the piston 543, and a support mechanism unit that supports the cylinder are provided.

  The linear motor 501 includes an inner yoke 530 disposed on the outer periphery of the cylinder 542, an outer yoke 504 disposed outside the inner yoke 530, and a coil 508 disposed between the inner yoke 530 and the outer yoke 504. And a spacer (not shown) that connects the cup-shaped sleeve 532, the first and second clamp rings 502 and 503 that sandwich the outer yoke 504, and the first and second clamp rings 502 and 503 at a predetermined interval. ) And a support portion 516 that supports the piston spring 546.

  The inner yoke 530 is provided so as to surround the outer periphery of the cylinder 542, and a cylindrical cup-shaped sleeve 532 is disposed so as to surround the inner yoke 530. The cup-shaped sleeve 532 is connected to the piston 543 and has a plurality of magnet pieces constituting the permanent magnet 531 at the tip. Each of the plurality of magnet pieces is disposed between the inner yoke 530 and the outer yoke 504. A cylindrical auxiliary ring 500 is provided on the inner peripheral surface of the cup-shaped sleeve 532 where the permanent magnet 531 is not provided.

  The first clamp ring 502 has a support portion 516 that supports the piston spring 546. A piston spring 546 is connected to the support portion 516 via a support member attached to the support portion 516.

  Further, in the linear compressor 540, a compression space 544 is configured by the cylinder 542, the piston 543, and the facing surface (547). The cylinder 542 is supported by a support mechanism portion in the casing 541. In the example of FIG. 42, the support mechanism portion is mounted on the support plate 549 fixed to the inside of the casing 541 and the support plate 549. And a coil spring 548 that supports the cylinder 542.

  Further, the head cover 545 is fixed to one end side of the cylinder 542 via the plate 547. In the compression space 544, the refrigerant is compressed by the head cover 545 and the head of the piston 543.

  Next, the operation of the linear compressor having the above structure will be described. First, when the coil 508 is energized, a thrust is generated between the cup-shaped sleeve 532 and the permanent magnet 531, and the thrust moves the cup-shaped sleeve 532 along the axial direction of the cylinder 542. Since the cup-shaped sleeve 532 is connected to the piston 543 at this time, the piston 543 moves in the axial direction of the cylinder 542 together with the cup-shaped sleeve 532.

  The refrigerant is introduced into the casing 541 from a suction pipe (not shown), passes through the passages in the head cover 545 and the plate 547, and enters the compression space 544. In the compression space 544, the refrigerant is compressed by the piston 543 and then discharged to the outside through a discharge pipe (not shown).

  If any one of the linear motor control systems of the first to fourth embodiments is used for the linear compressor 540 of the present embodiment, it is possible to prevent the piston 543 from colliding with other parts.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

It is sectional drawing which shows the structure of the Stirling refrigerator of embodiment. It is a figure for demonstrating the structure of the linear motor control system of embodiment. It is a block diagram of the microcomputer of embodiment. It is a figure for demonstrating the comparator in the microcomputer of embodiment. It is a figure which shows the relationship between the PWM signal wave, a carrier wave, a pulse, and an alternating current waveform. It is a reference figure for demonstrating the relationship of the phase of the current of a linear motor, the phase of a voltage, and the phase of a piston. It is an equivalent circuit diagram of a linear motor at a constant time. It is a flowchart for demonstrating the present stroke calculation process of the refrigerator of embodiment. It is a block diagram which shows the structure of a linear motor control system. It is a figure for demonstrating a standard carrier frequency. It is a figure for demonstrating a low carrier frequency. 6 is a diagram for illustrating carrier frequency change control according to Embodiment 1. FIG. 6 is a diagram illustrating a relationship between an effective value of an output voltage and a modulation rate in carrier frequency change control according to Embodiment 1. FIG. FIG. 10 is a diagram for illustrating a modification of the carrier frequency change control according to the first embodiment. FIG. 10 is a diagram illustrating a relationship between an effective value of an output voltage and a modulation rate in a modification of the carrier frequency change control according to the first embodiment. FIG. 10 is a diagram illustrating a relationship between an effective value of an output voltage and a modulation rate in a modification of the carrier frequency change control according to the first embodiment. It is a block diagram which shows schematic structure of the Stirling refrigerator control system of Embodiment 2. FIG. It is a figure for demonstrating the state in which the voltage doubler selector switch is ON. It is a figure for demonstrating the state in which the voltage doubler selector switch is OFF. It is a figure for demonstrating a pulse waveform and an alternating voltage waveform when a voltage doubler selector switch is ON. It is a figure for demonstrating a pulse waveform and an alternating voltage waveform when a voltage doubler selector switch is OFF. 6 is a flowchart for illustrating voltage doubler switching control according to the second embodiment. FIG. 10 is a diagram for explaining a relationship between an effective value of an output voltage and a modulation rate in voltage doubler switching control according to the second embodiment. 10 is a flowchart for illustrating a modification of the voltage doubler switching control of the second embodiment. 6 is a graph showing the relationship between the effective value of the output voltage and the modulation rate in the voltage doubler switching control of the second embodiment. 6 is a graph showing the relationship between the effective value of the output voltage and the modulation rate in the voltage doubler switching control of the second embodiment. FIG. 6 is a block diagram illustrating a schematic configuration of a Stirling refrigeration control system according to a third embodiment. 10 is a flowchart for illustrating carrier frequency double voltage switching control according to a third embodiment. FIG. 10 is a diagram for explaining a relationship between an effective value of an output voltage and a modulation rate in carrier frequency voltage doubler switching control according to the third embodiment. 10 is a flowchart for illustrating a modification of the carrier frequency double voltage switching control of the third embodiment. FIG. 10 is a diagram for explaining a relationship between an effective value of an output voltage and a modulation rate in carrier frequency voltage doubler switching control according to the third embodiment. 10 is a graph showing a relationship between an effective value of an output voltage and a modulation rate in carrier frequency double voltage switching control according to the third embodiment. It is a figure which shows the equivalent circuit structure of a linear motor drive circuit. It is a figure which shows the change of one period of each instantaneous value of a motor voltage, a motor current, and an induced voltage. It is a vector diagram for calculating | requiring the effective value of the induced voltage which arises in a linear motor. It is a graph which shows the correlation of a total magnetic flux number and an induced voltage coefficient. It is a vector diagram for calculating | requiring the total magnetic flux number which arises in the magnetic circuit of a linear motor. FIG. 10 is a block diagram illustrating a configuration of a linear motor control system according to a fourth embodiment. It is a figure for demonstrating the method of calculating the effective value of a motor current using the maximum value and minimum value (peak value) of a motor current. FIG. 10 is a block diagram illustrating a configuration of a linear motor control system according to another example of the fourth embodiment. 10 is a flowchart for illustrating a stroke control process according to the fourth embodiment. 10 is a flowchart for illustrating a stroke control process according to the fourth embodiment. It is a figure which shows the structure of the linear compressor control system in which the linear motor control system of Embodiment 5 was used.

Explanation of symbols

  1 piston, 13 linear motor, 40 Stirling refrigerator, 1000 microcomputer, B voltage doubler circuit.

Claims (9)

A linear motor that reciprocates the mover;
An inverter circuit that converts DC power supplied from a DC power source into AC power and supplies the AC power to the linear motor;
A microcomputer for controlling the inverter circuit by PWM (Pulse Width Modulation);
A voltage detector for detecting a voltage applied to the linear motor and transmitting a voltage signal capable of specifying a value of the voltage to the microcomputer;
A current detector for detecting a current flowing through the linear motor and transmitting a current signal capable of specifying the value of the current to the microcomputer;
The microcomputer is
A stroke command value output unit for outputting a target stroke command value of the mover;
A stroke calculation unit that calculates a value of an actual stroke of the mover using a voltage value and a current value specified by the voltage signal and the current signal;
Based on a comparison result between the target stroke command value and the actual stroke, a stroke control unit that outputs a command value of the PWM modulation rate and a command value of the carrier frequency;
A PWM control unit that controls the inverter circuit according to the command value of the PWM modulation rate and the command value of the carrier frequency of the PWM;
The stroke control unit
After the operation of the linear motor is started, a command value of a relatively low carrier frequency is output, and the value is determined in advance from an experimental result and reaches a predetermined value that can calculate the stroke of the mover. Increase the modulation rate, then
The modulation factor is increased when the actual stroke value is less than or equal to the stroke command value, while the modulation factor is decreased when the actual stroke value is greater than the stroke command value. Let
When the modulation rate is a value between the predetermined value and a specific value larger than the predetermined value, a command value of a relatively low carrier frequency is output, while the modulation rate is equal to or higher than the specific value. Linear motor control system that outputs a command value with a relatively high carrier frequency in some cases. The microcomputer is
Immediately after the modulation rate becomes equal to or greater than a specific value, the modulation rate at that time is stored as a first predetermined modulation rate, and the supply of power to the linear motor is stopped in the inverter circuit. In the PWM with a high carrier frequency, the inverter circuit is supplied with power to the linear motor while increasing the modulation rate up to the first predetermined modulation rate,
Immediately after the modulation rate becomes equal to or lower than a specific value, the modulation rate at that time is stored as a second predetermined modulation rate, and the inverter circuit is stopped from supplying power to the linear motor, and then the relative 2. The linear motor control system according to claim 1, wherein the inverter circuit is configured to supply electric power to the linear motor while increasing a modulation rate up to the second predetermined modulation rate in PWM with a low carrier frequency.   2. The linear motor control system according to claim 1, wherein the stroke calculation unit starts calculating the actual stroke after the modulation rate reaches the predetermined value or more. A linear motor that reciprocates the mover;
An inverter circuit that converts DC power supplied from a DC power source into AC power and supplies the AC power to the linear motor;
A microcomputer for controlling the inverter circuit by PWM (Pulse Width Modulation);
A voltage detector for detecting a voltage applied to the linear motor and transmitting a voltage signal capable of specifying a value of the voltage to the microcomputer;
A current detector for detecting a current flowing through the linear motor and transmitting a current signal capable of specifying the value of the current to the microcomputer;
A DC power generation circuit for generating the DC power,
The microcomputer is
A stroke command value output unit for outputting a target stroke command value of the mover;
A stroke calculating unit that calculates a value of an actual stroke of the mover using a voltage value and a current value specified by the voltage signal and the current signal;
Based on a comparison result between the stroke command value and the actual stroke, a stroke control unit that outputs the PWM modulation rate command value and controls the DC power control circuit;
A PWM control unit for controlling the inverter circuit according to a command value of the PWM modulation rate;
The stroke control unit
After the operation of the linear motor is started, the DC power generation circuit is controlled so that a relatively low DC voltage is applied to the inverter circuit. Increasing the modulation factor to a predetermined value that can be calculated, and then
The modulation factor is increased when the actual stroke value is less than or equal to the stroke command value, while the modulation factor is decreased when the actual stroke value is greater than the stroke command value. Let
The DC power generation circuit is controlled so that the relatively low DC voltage is applied to the inverter circuit when the modulation factor is a value between the predetermined value and a specific value larger than the predetermined value. On the other hand, a linear motor control system that controls the DC power generation circuit so that a relatively high DC voltage is applied to the inverter circuit when the modulation factor is equal to or greater than the specific value. The microcomputer is
Immediately after the modulation rate becomes equal to or greater than a specific value, the modulation rate at that time is stored as a first predetermined modulation rate, and the supply of power to the linear motor is stopped in the inverter circuit. The inverter circuit is supplied with power to the linear motor while controlling the DC power generation circuit so that a high DC voltage is applied to the inverter circuit,
Immediately after the modulation rate becomes equal to or lower than a specific value, the modulation rate at that time is stored as a second predetermined modulation rate, and the inverter circuit is stopped from supplying power to the linear motor, and then the relative The linear motor control system according to claim 1, wherein power is supplied from the inverter circuit to the linear motor while controlling the DC power generation circuit so that a low DC voltage is applied to the inverter circuit.   5. The linear motor control system according to claim 4, wherein the stroke calculation unit starts calculating the actual stroke after the modulation rate reaches the predetermined value or more. The DC power generation circuit includes a voltage changing circuit capable of changing the value of the DC voltage,
The relatively high DC voltage is generated by turning on the voltage changing circuit,
The linear motor control system according to claim 4, wherein the relatively low DC voltage is generated by turning off the voltage circuit.   A Stirling refrigerator control system in which the linear motor control system according to claim 1 is used to control a mover of a Stirling refrigerator.   A linear compressor control system in which the linear motor control system according to claim 1 is used to control a mover of a linear compressor.

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