JP2006295381A - Voltage-current conversion circuit, mixer circuit, and portable device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a voltage-current conversion circuit improved so that the transconductance of a transconductor unit (amplifying unit) does not depend on current and can be controlled in power consumption according to a reception environment. This is the main purpose.
When a current consumption is controlled adaptively and easily in accordance with an environment such as a surrounding interference wave in a voltage-current conversion unit that realizes low distortion, both gain and linearity are deteriorated. The present invention pays attention to a cross-coupled configuration having a current-independent characteristic with respect to a transconductor unit that realizes low distortion, sets Gain by a mirror ratio of a transistor connected in parallel and a negative feedback resistor, and a variable current source As a result, the gain is maintained and the linearity variable control is realized while adaptively controlling the current consumption in accordance with the surrounding environment such as an interference wave.
[Selection] Figure 1

Description

  The present invention generally relates to a voltage-current conversion circuit, and more specifically, the transconductance of a transconductor unit (amplifier unit) does not depend on current, and can control power consumption in accordance with a reception environment. The present invention relates to a voltage-current conversion circuit improved so as to be able to be performed. The present invention also relates to a mixer circuit and a portable device using such a voltage-current conversion circuit.

  In Japanese terrestrial digital television broadcasting (ISDB-T), broadcasting using only one segment (430 kHz) is performed for portable devices. In order to integrate digital terrestrial television broadcasting reception functions into ICs (Integrated Circuits) and incorporate them into battery-powered portable terminals, the reception tuner must have low power consumption, miniaturization, interference wave resistance, and low distortion. This is an important issue.

  FIG. 14 is a block diagram showing a receiving circuit of a conventional TV tuner. The reception circuit includes a low noise amplifier (LNA), a mixer (Mixer: frequency converter), a PPF (Poly Phase Filter), an IFF (IF Filter: a low pass filter for IF band), and an IF VGA (IF A variable gain amplifier for band (Variable Gain Amplifier) and an OFDM demodulator are provided.

  When the IQ orthogonal balance is good, the PPF removes an image band at a mirror image frequency with respect to the LO signal. The OFDM demodulator is an OFDM (Orthogonal Frequency Division Multiplexing) modulated wave demodulator used in digital terrestrial television broadcasting. In the figure, Divider represents a 1/2 frequency divider (IQ generator), and VCO represents a voltage controlled oscillator.

  The operation will be described. The reception circuit transmits a reception signal like LNA-Mixer-PPF-IFF-IFFVGA. An RF (Radio Frequency) signal (470.5-710.5 MHz) received from the antenna is amplified by an LNA, and then mixed by an RF signal and a local signal (LO_I, LO_Q, 470-710 MHz) by a mixer. A frequency signal (IF_I, IF_Q, 500 kHz) is obtained, then the image is suppressed with PPF, the low frequency intermediate frequency is extracted with IFF, and the signal with this intermediate frequency is amplified with IFVGA, and then A / D An analog signal is converted into a digital signal by a converter (not shown), demodulated by an OFDM demodulator, and a digital signal is taken out.

  Note that the portable device according to the present invention uses an LSI in which the LNA to the IFVGA are integrated into a single chip.

  In the image suppression type low frequency intermediate frequency (Low IF) type receiver as described above, the mixer circuit is one of the most important blocks.

  In the mixer circuit, there is a trade-off relationship between lower distortion and lower power consumption of the transconductor section. As one technique for reducing the distortion of a transconductor, a voltage-current conversion circuit having a cross-coupled configuration as shown in FIG. 15 has been proposed (see, for example, Non-Patent Document 1).

  Referring to FIG. 15, the transconductor unit includes a first transistor T1 and a second transistor T2 which connect a collector to a power supply terminal and receive input signals (RF, RFB) from a base. The transconductor portion further includes a third transistor T3 having a base and a collector cross-connected to each other, a collector connected to the emitter end of the first transistor T1, and a collector connected to the emitter end of the second transistor T2. A transistor T4 is provided. The transconductor unit further shares the base / emitter with the third transistor T3, shares the fifth transistor T5 that extracts the current signal from the collector, and shares the base / emitter with the fourth transistor T4, and extracts the current signal from the collector. Transistor T6. The third transistor T3 and the fifth transistor T5 (the fourth transistor T4 and the sixth transistor T6) constitute a current mirror. Here, the current mirror ratio is 1: 1. A negative feedback resistor Rdeg is provided between the emitters of the fifth and sixth transistors T5 and T6 (T3 and T4), and a constant current source is added to each of the emitter terminals 1 and 2.

Now, in the bipolar transistor, the output current is expressed as (1).

Here, V _T is called a thermal potential of electrons and has a value of 26 mV at room temperature. Since the same current flows through the transistors T1 and T3 (T2 and T4), as long as the following necessary conditions (2) and (3) are satisfied,

The input voltage is directly applied to the negative feedback resistor Rdeg as shown in the equation (4).

At this time, the signal current flowing through Rdeg is expressed by Equation (5).

Currents flowing through Iout3 and Iout4 are expressed as (6) and (7).

  These do not include non-linear terms (ln, exp, etc.) and are high frequency inputs using a current mirror configuration, so low distortion operation can be expected.

From equations (6) and (7), the transconductance gm of the differential pair can be expressed as equation (8).

Hidehiko Aoki, “Introduction to Functional Circuit Design of Analog IC”, CQ Publishing, 1992

  Here, the system control of the tuner IC is considered. In the Low-IF tuner IC, referring to FIG. 14, the received signal is transmitted like LNA-Mixer-PPF-IFF-IFFVGA. The linearity is in a trade-off with the current, and generally the distortion component is suppressed by increasing the current. The use of mobile devices is expected in various environments, but when reception sensitivity is good in an environment with few interference waves, each block that is over-spec (excessive current consumption setting) to improve linearity If the current consumption can be adjusted adaptively, the battery duration can be increased.

ＩＩＰ３(ｄＢ)は、式（９）の値を用いて10logIIP3として求めることができる。ここで、Ｇ₁， Ｇ₂，・・・， およびＩＩＰ３₁，ＩＩＰ３₂，・・・はそれぞれ縦列接続順（ＬＮＡ−Mixer−・・・）のＧａｉｎ（利得），ＩＩＰ３をそれぞれ割り当てるものとする。ミキサの線形性に関する式（９）の第２項は、前段のＬＮＡの利得Ｇ１を下げるか、ミキサの線形性ＩＩＰ３₂自体を高く設定することで改善できる。帯域外の妨害波を考慮する時、周波数変換を行うミキサまでの線形性を向上すれば、チューナとしての妨害波耐性が向上する。 The performance of cascaded IIP3 (intermodulation intercept point: linearity index) is given by the following equation (9).

IIP3 (dB) can be obtained as 10logIIP3 using the value of equation (9). Here, G ₁ , G ₂ ,..., And IIP 3 ₁ , IIP 3 ₂ ,... Respectively assign Gain (gain) and IIP 3 in tandem connection order (LNA-Mixer-...). . The second term of the equation (9) regarding the linearity of the mixer can be improved by lowering the gain G1 of the LNA in the previous stage or setting the mixer linearity IIP3 ₂ itself high. When considering out-of-band interference waves, if the linearity up to the mixer that performs frequency conversion is improved, the interference wave resistance as a tuner is improved.

後に制御特性を示す際に参照する。 Similarly, the cascaded noise figure (NF) can be expressed by the following equation (10).

It will be referred to later when showing the control characteristics.

  Next, consider current control of the voltage-current converter. In Non-Patent Document 1, which is a prior art, the transistors do not have parameters such as the number of parallel connections and are configured by one each. Therefore, Iout3: Iout5 = Iout4: Iout6 = 1: 1, and Iin1 (Iin2) Half of the current flows through Iout3 (Iout4). Further, according to the equation (8), Gain is determined only by the value of the negative feedback resistor Rdeg. Therefore, the degree of design freedom is small.

  Furthermore, when the distortion and gain are improved by increasing the currents of the transistors T5 and T6 which are the main components of the differential amplification operation, the circuit configuration requires about twice the current flowing through the optimized transistors T5 and T6. Therefore, there is a problem that an increase in power consumption cannot be avoided.

  The present invention has been made in view of the above problems, and provides an improved voltage-current conversion circuit so that current consumption can be adaptively controlled according to the environment such as ambient interference waves. The purpose is to do.

  Another object of the present invention is to provide an improved frequency converter (mixer) so that current consumption can be adaptively controlled in accordance with an environment such as ambient interference waves.

  Still another object of the present invention is to provide a portable device using such a voltage-current conversion circuit and a mixer.

  The voltage-current conversion circuit according to the present invention includes a transconductor unit that performs voltage-current conversion while maintaining a constant gain, and a variable current source that controls the amount of current flowing through the transconductor unit. The current consumption is controlled according to the reception environment by operating the variable current source.

  According to the present invention, the gain is kept constant and the current consumption is controlled according to the reception environment. Since the gain is constant, readjustment of gain distribution and the like of the entire system is not required, and the battery use efficiency of the portable device is improved by adaptive control according to the reception environment.

  A voltage-current conversion circuit according to another aspect of the present invention includes a transconductor unit that performs voltage-current conversion while maintaining a constant gain, and a variable current source that controls an amount of current flowing through the transconductor unit, and the variable The linearity is controlled according to the reception environment by operating the current source.

  According to the present invention, the gain is kept constant, and the linearity is controlled according to the reception environment by operating the variable current source. Since the gain is constant, the linearity parameter of the mixer can be made independent, and the linearity performance of the entire system can be controlled. Also, control according to the reception environment is possible.

  A voltage-current conversion circuit according to still another aspect of the present invention includes a transconductor unit that performs voltage-current conversion while maintaining a constant gain, and a variable current source that controls an amount of current flowing through the transconductor unit, The noise is controlled according to the reception environment by operating the variable current source.

  According to the present invention, the gain is kept constant, and the noise is controlled according to the reception environment by operating the variable current source. Since the gain is constant, the noise parameter of the mixer can be made independent, and the noise performance of the entire system can be controlled. For example, when the power consumption is reduced, the linearity is degraded, but the noise performance is improved, so that the reception sensitivity is improved somewhat.

  A voltage-current conversion circuit according to still another aspect of the present invention relates to a transconductor unit that performs voltage-current conversion, a first and a second transistor having a collector connected to a power supply terminal and receiving an input signal from the base; A third transistor having a collector connected to the emitter end of the first transistor, a fourth transistor having a collector connected to the emitter end of the second transistor, and a base / emitter connected to the first transistor; A fifth transistor that shares the current signal from the collector with the third transistor, and a sixth transistor that shares the base and emitter with the fourth transistor and extracts the current signal from the collector. Negative feedback resistors are provided between the emitters of the fifth and sixth transistors, and current sources are respectively added to the emitter ends. At least one of the first to sixth transistors is formed by connecting two or more transistors in parallel. By arbitrarily setting the ratio or the size ratio of the number of parallel connections in the first to sixth transistors, the amount of signal current flowing in the fifth and sixth transistors in the total current amount is set. The current source is variable.

  According to the present invention, since it has gm characteristics that do not depend on current, it is possible to control only linearity while suppressing current consumption.

  According to a preferred embodiment of the present invention, the transconductance of the transconductor section is arbitrarily set by the number of parallel connections of the negative feedback resistor and the transistor.

  According to this embodiment, gm can be arbitrarily set by the number of parallel connections of the negative feedback resistor and the transistor while having gm characteristics that do not depend on current.

  According to a further preferred embodiment of the present invention, the negative feedback resistor is variable. According to this embodiment, since gm can be changed by adjusting Rdeg, a desired gm value (Gain value) can be set.

  It is also preferable to realize the negative feedback resistor with a transistor. With this configuration, the resistance value can be controlled using the gate potential of the transistor.

  According to a further preferred embodiment of the present invention, a switch is provided at the collector terminals of the fifth and sixth transistors connected in parallel, and the gain is arbitrarily controlled by operating the switch. With this configuration, it is possible to adjust the mirror ratio (m: n) of the cross-coupled voltage-current conversion circuit and set the gm value (Gain value) by turning the switch ON / OFF.

  The switch is preferably realized by a transistor. With this configuration, it is possible to set the switch ON / OFF using the gate potential of the transistor.

  A mixer according to still another aspect of the present invention includes a first frequency conversion circuit that generates a third signal as a product from a first signal and a second signal, a first signal, and a fourth signal. And a second frequency conversion circuit that generates a fifth signal as a product, and a four-phase mixer having a common transconductor unit for inputting the first signal. The transconductor unit that receives the first signal and performs voltage-current conversion has a collector connected to a power supply terminal, a first and a second transistor that receive an input signal from the base, and a base and a collector that are cross-connected to each other. A third transistor whose collector is connected to the emitter of the first transistor, a fourth transistor whose collector is connected to the emitter end of the second transistor, and a base / emitter shared with the third transistor, And a sixth transistor that shares the base and emitter with the fourth transistor and extracts the current signal from the collector. A resistor is provided between the emitters of the fifth and sixth transistors, and a current source is added to each of the emitter ends. The current source is variable.

  According to the present invention, since the cross-coupled configuration is used, it has a gm characteristic that does not depend on the current, and linearity can control the noise performance according to the reception environment.

  A variable negative feedback resistor is preferably provided between the emitters of the fifth and sixth transistors. By configuring in this way, it is possible to have a gm characteristic that does not depend on current and to reset the gm value.

  More preferably, the negative feedback resistor is realized by a transistor.

  A switch may be provided at the collector terminals of the fifth and sixth transistors connected in parallel, and the gain may be arbitrarily controlled by operating the switch. Even if comprised in this way, it has the gm characteristic which does not depend on an electric current, and can reset a gm value.

  The switch is preferably realized by a transistor. With this configuration, it is possible to set the switch ON / OFF using the gate potential of the transistor.

  A portable device according to still another aspect of the present invention includes a voltage-current conversion circuit or a mixer circuit having the above-described characteristics.

  By using the voltage-current conversion circuit and the mixer in a portable device, the current can be adaptively controlled according to changes in the surrounding environment due to the presence or absence of interference waves. Further, since the gain is kept constant and the current consumption of the portable device is adaptively controlled, an improvement in battery efficiency can be expected.

  As described above, the voltage-current conversion circuit, the mixer circuit, and the portable device using the same according to the present invention utilize the characteristic that the gm value (Gain value) is constant, and adapt the linearity and noise according to the surrounding environment. Control is possible, and the battery efficiency in portable devices and the like is improved.

  The present invention pays attention to a cross-coupled configuration having a current-independent characteristic with respect to a transconductor unit that realizes low distortion, and gain is set by a mirror ratio of a transistor connected in parallel and a negative feedback resistor, and a variable current source By adding, the current consumption can be controlled adaptively according to the environment such as surrounding interference waves. Embodiments of the present invention will be described below with reference to the drawings.

  In this embodiment, while maintaining the circuit topology described in Non-Patent Document 1, the trade-off between linearity and power consumption is first relaxed by using a new parameter called the number of transistors connected in parallel.

  FIG. 1 is a circuit diagram of a voltage-current conversion circuit according to the first embodiment.

  Referring to FIG. 1, a transconductor unit S-100 that performs voltage-current conversion according to the first embodiment has a collector connected to a power supply terminal, and first and second transistors T1, T2 that receive an input signal from a base. With. The base and collector of the third transistor T3 whose collector is connected to the emitter end of the first transistor T1 and the fourth transistor T4 whose collector is connected to the emitter end of the second transistor T2 are cross-connected to each other. The fifth transistor T5 shares the base and emitter with the third transistor T3, and extracts a current signal from the collector. The sixth transistor T6 shares the base and emitter with the fourth transistor T4, and extracts a current signal from the collector. A negative feedback resistor Rdeg is provided between the emitters of the fifth and sixth transistors T5 and T6, and variable current sources are provided at the emitter terminals 1 and 2, respectively. The transistors T1, T3, T2, and T4 are formed by connecting m columns of transistors in parallel. The transistors T5 and T6 are formed by connecting n columns of transistors in parallel.

  Thus, a current mirror is formed between the transistor T3 and the transistor 5 (transistor T4 and transistor T6), and the reference current can be distributed to the ratio of m: n.

  According to the present embodiment, a new parameter based on the number of parallel connections is added while realizing the conventionally proposed low distortion amplification operation, and a degree of freedom in design is created. Furthermore, the current distribution of the fifth and sixth transistors can be optimized (gm can be improved). In general, gm increases in proportion to the operating current. Further, by controlling with the ratio of the number of parallel connections (current mirror ratio), it is possible to effectively use the operating current, and low power consumption can be expected. It should be noted that the operation current can be effectively used by controlling the size ratio of the first to sixth transistors, and low power consumption can be expected.

Further details will be described. The transistors are connected in parallel, and currents represented by the equations (11) and (12) flow through the output nodes 3 and 4 due to the mirror ratio determined by m and n.

  By setting n >> m, it is possible to reduce the currents of the paths located in the m columns on both sides and to reduce power consumption.

From Expressions (11) and (12), gm in the differential pair in the cross-coupled configuration using the parallel connection of the transistors is Expression (13).

  As apparent from the equation (13), gm does not depend on the current value, and is determined by the mirror ratio “n / (m + n)” and the negative feedback resistance (Rdeg). Compared with equation (8), a new parameter of mirror ratio “n / (m + n)” is added in this embodiment.

ミキサの電流を可変電流源によって調整し適応的な制御をする場合、Ｇａｉｎ，ＩＩＰ３，及びノイズの性能が同時劣化すると、両方の影響を考慮する必要があり、制御が複雑になる可能性がある。 On the other hand, in a differential pair using a conventional general negative feedback resistor as shown in FIG. 2, both gain and IIP3 are reduced when the current is reduced. The decrease in Gain is due to the fact that transconductance (gm) depends on the current as shown in Equation (14).

When adaptive control is performed by adjusting the mixer current using a variable current source, if the gain, IIP3, and noise performance deteriorate simultaneously, both effects must be taken into account, which may complicate the control. .

  FIG. 3 shows a graph comparing the formula (13) and the formula (14). As is clear from FIG. 3, in the theoretical calculation, the conventional general gm_Rdeg has a current dependence characteristic, and gm deteriorates as the current decreases. Therefore, the gain that can be obtained by the gm and the output load resistance is lowered. On the other hand, since gm_Cross shown in Expression (13) does not depend on current, gm can be kept constant. FIG. 3 shows Rdeg adjusted so that saturated gm values are equal. In FIG. 1, the mirror ratio of the cross-coupled configuration is m: n = 1: 2.

  In this embodiment, since the gain can be fixed, by providing a variable current source as shown in FIG. 1, the power consumption that has been excessively supplied is ensured in order to ensure linearity in a place where the reception environment is good. Can be suppressed. The current is set so that the required linearity is obtained.

  In the circuit topology shown in FIG. 1, the current distribution ratio can be changed while maintaining the conventional low distortion operation by using the transistors connected in parallel while maintaining the cross-coupled topology of Non-Patent Document 1. Therefore, it is possible to alleviate the trade-off between low distortion and power consumption.

  Next, important characteristics of the voltage-current conversion circuit according to the first embodiment will be described.

  FIG. 4 is a diagram illustrating characteristics of voltage-current conversion. FIG. 4A shows the cross-coupled configuration shown in FIG. 1, where gm is constant, and the slope near the cross point of the differential input is constant without depending on the change in current Iss. When a large amount of current flows, the linear region expands and distortion can be reduced. On the other hand, in the configuration shown in FIG. 2, which is a comparative example, as shown in FIG. 4B, gm depends on the current, so that the slope changes with the current Iss.

  FIG. 5 shows a change in gm due to the mirror ratio expressed by the number of parallel connections of transistors. It was confirmed that the simulation (a) and the theoretical value (b) are in good agreement.

  FIG. 6 is a circuit diagram of a voltage-current conversion circuit according to the second embodiment. Referring to FIG. 6, variable current sources are provided at the emitter ends 1 and 2 of the fifth and sixth transistors T5 and T6, respectively, and the number of parallel connections is equal to the collectors of the fifth and sixth transistors T5 and T6. A number of switches SW1, SW2 and SW3 comprising transistors t21 to t26 are connected. By controlling ON / OFF of SW1, SW2, and SW3, n of the mirror ratio is controlled and a desired gm value is adjusted. Further, by making the negative feedback resistance variable, Rdeg can be set and a desired gm value is adjusted. Also, as shown in the lower right of the figure, replacing the variable negative feedback resistor with a transistor also makes it possible to set the Rdeg of equation (13) and adjust the desired gm value.

  FIG. 7 shows a change in Gain when the negative feedback resistance value Rdeg is changed. The output load resistance was 500Ω and m: n = 1: 2. gm is inversely proportional to Rdeg, and the experimental value and the theoretical value agree with each other.

  FIG. 8 is a circuit diagram of a four-phase mixer circuit according to a third embodiment in which the voltage-current conversion circuit described above is applied to a mixer. The four-phase mixer q100 includes a first switch circuit SW_I that generates a third signal (IF_I, IF_IB) as a product from the first signal (RF, RFB) and the second signal (LO_I, LO_IB); A second switch circuit SW_Q that generates a fifth signal (IF_Q, IF_IQ) as a product from the first signal (RF, RFB) and the fourth signal (LO_Q, LO_QB), and a transconductor unit Gm. A transconductor portion Gm that amplifies and inputs a signal to the first switch circuit SW_I and the second switch circuit SW_Q is shared. The mixer q100 further includes capacitors C3 to C8. Variable current sources are connected to the emitter ends of the fifth and sixth transistors T5 and T6 via transistors T15, T16, and T17, respectively.

  Similar to the amplifier according to the first embodiment, the amount of signal current flowing through the fifth and sixth transistors T5 and T6 in the total amount of current can be set by arbitrarily setting the ratio of the number of transistors connected in parallel (m, n). Can be adjusted. Since the voltage-current conversion circuit has a cross-coupled configuration, Gain is constant without depending on the current, and by making the current flowing through Iin_MIX variable, current control adapted to the surrounding environment can be expected.

  Further, according to the present embodiment, the gm value is determined and the power consumption is controlled by setting the ratio of the number of parallel connections (m, n) or the negative feedback resistance (Rdeg). In addition, the characteristics of the four-phase mixer, such as output phase error compression due to IQ mutual interference, halving of power consumption by sharing the input section, reduction of the number of elements, etc., lower distortion and further reduction of power consumption by the present invention. A synergistic effect can be expected. Further, the insertion of the capacitors C3 to C6 can improve the common mode noise resistance, and the insertion of the capacitors C7 and C8 can attenuate unnecessary harmonics.

  FIG. 9 is a diagram comparing Gain obtained from the mixer according to the third embodiment and Gain obtained from the mixer configured using the comparative example (Rdeg differential pair configuration) illustrated in FIG. 2. In order to reduce power consumption, the mirror ratio was set to m: n = 1: 2. Regarding the characteristics near the lower limit value of the current, the theoretical value shown in FIG. 3 differs from the theoretical value shown in FIG. 3 in that the transistors T1, T3 (T2, T4) located in the side path (m column) with the reduced current are the operation limits. Because.

  The linearity cascade connection is shown in Equation (9), and FIG. 10 shows a plot of the mixer linearity dependence on the LNA-Mixer linearity that determines the jamming tolerance.

  The out of band IIP3 on the vertical axis will be described. Referring again to FIG. 14, in the Low IF architecture, the UHF band (470.5-710.5 MHz) is converted to the IF = 430 kHz band (Center 476 ± 215 kHz) using the LO frequency (470-710 MHz). After that, demodulation is performed. The outside of the UHF band is called “Out of band” and is distinguished from “In band”.

  For example, when the RF frequency is 710.5 MHz and the LO signal is 710 MHz (IF = 500 kHz), there may be a case where an unnecessary strong interference wave (for example, 830 MHz) due to carrying or the like enters. When a strong interference signal outside the band (Out of band) is received together with the UHF band, a distortion component may be generated by the strong interference signal. Therefore, in order to improve the interference wave resistance, the distortion characteristic (IIP3) until the frequency conversion by LNA + Mixer is important. Such out-of-band distortion performance handled by LNA and Mixer is referred to as Out of band IIP3 in this specification. In the IF band, the interference wave 830 MHz itself can be greatly attenuated by the low-pass filter.

  “Default” represents an initial setting value of the design. FIG. 10 is a diagram illustrating a result of an experiment on how much current can be reduced at the sacrifice of distortion characteristics in a good reception environment, with a value designed assuming the worst reception environment with strong interference waves as a default value. It is. The initial setting is IIP3 = −13 dBm. For example, if IIP3 = −24 dBm is acceptable, by setting the initial setting to IIP3 = −24 dBm, it is possible to reduce 2.6 mA while maintaining Gain. . On the contrary, if further linearity is required, it is necessary to flow more current.

  The cascade connection of noise is shown in the equation (10). FIG. 11 shows the dependency of the input conversion noise of the mixer on the NF in the entire tuner. Similar to linearity, if the current is reduced by 2.6 mA, NF will be improved by 0.7 dB. At this time, the linearity is deteriorated, but the reception sensitivity is improved by 0.7 dB by NF improvement. If the current is increased to improve linearity, NF is sacrificed.

  FIG. 12 shows the relationship between Gain and IIP3 with respect to the total current value when a cross-coupled configuration is adopted for the four-phase mixer corresponding to the third embodiment and the current source is made variable. In order to reduce power consumption, the mirror ratio was set to m: n = 1: 2. As long as all transistors operate normally, Gain is constant. Linearity degrades with decreasing current. It can be seen that control is possible with the normal cross-coupled operation in the range of 3.8 mA to 1.2 mA.

  FIG. 13 shows the relationship between the gain and the input referred noise with respect to the total current value when the current source is variable as in FIG. When the current is narrowed against the current setting value of 3.8 mA assuming a poor environment with the largest disturbance wave, the gain is constant, but the current-dependent shot noise etc. decreases, so the input conversion noise decreases (improves). .

  Although the present invention has been described with reference to the embodiments, the present invention is not limited to the above-described embodiments, and various modifications are possible within the scope of the claims, and each is disclosed in a different embodiment. Embodiments obtained by appropriately combining the technical means provided are also included in the technical scope of the present invention.

  The voltage-current conversion circuit and the mixer using the voltage-current conversion circuit of the present invention are suitably used for a transmission / reception device such as a portable terminal that requires low distortion and low power consumption characteristics.

It is a figure which shows the concept of the voltage-current conversion circuit concerning Example 1 of this invention. It is a figure which shows the concept of the conventional voltage-current conversion circuit containing the differential pair with the negative feedback resistance Rdeg shown as a comparative example. It is the graph which compared Formula (13) and Formula (14). FIG. 3 is a diagram illustrating characteristics of the voltage-current conversion circuit according to the first example. It is a figure which shows Gain distribution by a mirror ratio. 6 is a circuit diagram of a voltage-current conversion circuit according to Example 2. FIG. It is a figure which shows the change of Gain at the time of changing Rdeg. It is a circuit diagram of the mixer circuit concerning Example 3 of this invention. It is the figure which compared Gain obtained from the mixer concerning Example 3, and Gain obtained from the mixer comprised using the comparative example shown in FIG. It is a figure which shows the dependence of the theoretical mixer IIP3 with respect to IIP3 normal setting. It is a figure which shows the dependence of the input conversion noise of a mixer with respect to NF of a tuner. It is a figure which shows the current dependence of Gain and IIP3 of the mixer in Example 3. FIG. It is a figure which shows the current dependence of Gain of the mixer in Example 3, and the input conversion noise of a mixer. It is a block diagram which shows the receiving circuit of TV tuner. It is a circuit diagram of the conventional voltage-current conversion circuit which has a cross-coupled configuration.

Explanation of symbols

Gm transconductor T1 1st transistor T2 2nd transistor T3 3rd transistor T4 4th transistor T5 5th transistor T6 6th transistor q100 Four-phase mixer s100, s200 Voltage-current conversion circuit SW_I, SW_Q switch R1, R2, R7 , Rdeg Negative feedback resistance R3, R4, R5, R6 Output load resistance m, n Number of transistors connected in parallel t21-t26 Switch transistor

Claims (15)

A transconductor section that performs voltage-current conversion while keeping the gain constant;
A variable current source for controlling the amount of current flowing through the transconductor unit;
A voltage-current conversion circuit, wherein current consumption is controlled according to a reception environment by operating the variable current source. A transconductor section that performs voltage-current conversion while keeping the gain constant;
A variable current source for controlling the amount of current flowing through the transconductor unit;
A voltage-current conversion circuit, wherein linearity is controlled according to a reception environment by operating the variable current source. A transconductor section that performs voltage-current conversion while keeping the gain constant;
A variable current source for controlling the amount of current flowing through the transconductor unit;
A voltage-current conversion circuit which controls noise according to a reception environment by operating the variable current source. Regarding the transconductor part that performs voltage-current conversion,
First and second transistors having a collector connected to a power supply terminal and receiving an input signal from a base;
A third transistor having a base and a collector cross-connected to each other, a collector connected to the emitter end of the first transistor, a fourth transistor having a collector connected to the emitter end of the second transistor;
A fifth transistor sharing a base emitter with the third transistor and extracting a current signal from the collector;
A sixth transistor that shares a base emitter with the fourth transistor and extracts a current signal from a collector;
A negative feedback resistor is provided between the emitters of the fifth and sixth transistors, and a current source is added to each of the emitter ends;
At least one of the first to sixth transistors is formed by connecting two or more transistors in parallel.
By arbitrarily setting the ratio or the size ratio of the number of parallel connections in the first to sixth transistors, the amount of signal current flowing in the fifth and sixth transistors in the total current amount is set,
A voltage-current conversion circuit characterized in that the current source is variable.   5. The voltage-current conversion circuit according to claim 4, wherein the transconductance of the transconductor unit is arbitrarily set by the number of parallel connections of the negative feedback resistor and the transistor.   6. The voltage-current conversion circuit according to claim 5, wherein the negative feedback resistor is variable.   7. The voltage-current conversion circuit according to claim 5, wherein the negative feedback resistor is realized by a transistor. A switch is provided at the collector terminals of the fifth and sixth transistors connected in parallel;
8. The voltage-current conversion circuit according to claim 4, wherein the gain is arbitrarily controlled by operating the switch.   9. The voltage-current conversion circuit according to claim 8, wherein the switch is realized by a transistor. A first frequency conversion circuit that generates a third signal as a product from the first signal and the second signal, and a second that generates a fifth signal as a product from the first signal and the fourth signal A four-phase mixer having a common transconductor unit for inputting the first signal,
The transconductor unit that receives the first signal and performs voltage-current conversion,
First and second transistors having a collector connected to a power supply terminal and receiving an input signal from a base;
A third transistor having a base and a collector cross-connected to each other, a collector connected to the emitter of the first transistor, a fourth transistor having a collector connected to the emitter end of the second transistor,
A fifth transistor sharing a base emitter with the third transistor and extracting a current signal from the collector;
A sixth transistor that shares a base emitter with the fourth transistor and extracts a current signal from a collector;
A resistor is provided between the emitters of the fifth and sixth transistors, and a current source is added to each of the emitter ends;
A mixer circuit characterized in that the current source is variable.   The mixer circuit according to claim 10, wherein a variable negative feedback resistor is provided between the emitters of the fifth and sixth transistors.   The mixer circuit according to claim 11, wherein the negative feedback resistor is realized by a transistor. A switch is provided at the collector terminals of the fifth and sixth transistors connected in parallel;
The mixer circuit according to claim 10, wherein a gain is arbitrarily controlled by an operation of the switch.   The mixer circuit according to claim 13, wherein the switch is realized by a transistor. A portable device comprising the voltage-current conversion circuit according to any one of claims 1 to 9 or the mixer circuit according to any one of claims 10 to 14.

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