JP2006288083A - Control method of synchronous motor - Google Patents

Abstract

A synchronous motor control method capable of effectively suppressing fluctuations in the rotational speed of a synchronous motor against fluctuations in load torque having periodicity.
A method of controlling the synchronous motor to which a load whose load torque varies during one rotation of the synchronous motor 20 is connected, and (a) calculating a vector corresponding to the variation of the load torque; And (b) controlling the calculated vector to a desired control amount. The calculated vector is a rotation vector, and (b) includes controlling a value obtained by rotating coordinate conversion of the rotation vector to the desired control amount.
[Selection] Figure 1

Description

  The present invention relates to a method for controlling a synchronous motor.

  In International Publication No. 98/08297 (Patent Document 1), an inverter is used to control a load having periodic torque fluctuations (for example, load torque fluctuations corresponding to the rotation angle of a one-cylinder compressor). A technique for performing torque control for suppressing speed fluctuation during one rotation by a synchronous motor is disclosed.

  In Patent Document 1 (FIG. 21), a first neutral point voltage obtained by a resistor connected to each of the three-phase output terminals of the inverter and a second neutral point obtained by a stator winding of each phase of the motor. The difference from the point voltage is integrated to detect the rotor position of the synchronous motor, and in synchronization with the pulsation of the load torque, the fluctuation phase advanced with respect to the phase of the load torque is superimposed on the average value phase command and brushless. A technique is described in which periodic pulsations are suppressed by applying to a DC motor and suppressing fluctuations in the peak value of the integrated signal.

  The technique of the above-mentioned Patent Document 1 has low torque control accuracy in order to calculate the torque fluctuation after estimating the rotor position based on the neutral point voltage, and to suppress periodic pulsation by the superposition method. . Therefore, torque fluctuation cannot be sufficiently suppressed. Moreover, the torque control of the said patent document 1 utilizes the output value obtained from the 1st and 2nd neutral point voltage, and presupposes that the circuit for it is provided. Further, in the torque control of Patent Document 1, the method of estimating the rotor position by integrating the difference between the first and second neutral point voltages has a rough position estimation accuracy.

  On the other hand, it is known that the rotor position is estimated based on the value of the motor current. This position estimation method has high accuracy and can rotate the motor optimally. In a motor control device that estimates the rotor position based on such a motor current value, it is desired to suppress fluctuations in the rotational speed of the motor against fluctuations in load torque having periodicity.

International Publication No. 98/08297 Japanese Patent Laid-Open No. 11-18499

  It is desired to effectively suppress the rotational speed fluctuation of the synchronous motor with respect to the load torque fluctuation having periodicity.

  The objective of this invention is providing the control method of the synchronous motor which can suppress effectively the rotation speed fluctuation | variation of a synchronous motor with respect to the load torque fluctuation | variation which has periodicity.

  The synchronous motor control method of the present invention is a method for controlling the synchronous motor to which a load whose load torque varies during one rotation of the synchronous motor is connected, and (a) a vector corresponding to the load torque variation And (b) controlling the calculated vector to a desired control amount.

  In the synchronous motor control method of the present invention, the calculated vector is a rotation vector, and (b) controls that a value obtained by rotating coordinate conversion of the rotation vector to the desired control amount. It is characterized by including.

  In the synchronous motor control method of the present invention, the desired control amount is a constant value.

  In the synchronous motor control method of the present invention, the vector is obtained based on a current value of the synchronous motor.

  In the synchronous motor control method of the present invention, the vector is obtained for each of the d-axis and q-axis of the synchronous motor.

  In the synchronous motor control method of the present invention, the phase or amplitude of the synchronous motor is controlled based on the calculated vector.

  In the synchronous motor control method of the present invention, the value is a value corresponding to a mechanical angle of the synchronous motor, and the value is obtained using a multiplication value of the electrical angle of the synchronous motor and the number of pole pairs. It is a feature.

  In the synchronous motor control method of the present invention, the synchronous motor is controlled by sensorless control.

In the synchronous motor control method of the present invention, the calculation of the vector is obtained using the following equation [Equation 5] or [Equation 6].

In the synchronous motor control method of the present invention, the calculation of the vector is obtained by using the following equation [Equation 7] or [Equation 8].

  In the synchronous motor control method of the present invention, in (b), the vector is input to the controller, and the output of the controller is feedback controlled, so that the vector is controlled to the desired control amount. It is characterized by including.

  In the synchronous motor control method of the present invention, the desired control amount is zero.

  In the synchronous motor control method according to the present invention, (a) includes calculating the vector using information on an initial phase of the synchronous motor.

  ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to suppress effectively the rotation speed fluctuation | variation of a synchronous motor with respect to the load torque fluctuation | variation which has periodicity.

  Hereinafter, an embodiment of a synchronous motor control method of the present invention will be described in detail with reference to the drawings.

  This embodiment is a synchronous motor connected to a load in which the load torque periodically varies during one rotation of the synchronous motor (for example, a load torque variation corresponding to a rotation angle of a compressor used in an air conditioner). The present invention relates to torque control that suppresses speed fluctuation during one rotation.

As shown in FIG. 9, a load (for example, a compressor) 100 is connected to the synchronous motor (permanent magnet motor, DC brushless motor, IPM motor) 20 of the present embodiment. When the torque of the motor 20 is denoted by T _M and the torque of the load 100 is denoted by T _L , the load torque T _L is constant when the size of the load 100 is constant as shown in FIG. When the load varies periodically (load variation), the load torque T _L varies periodically. Here, since the load 100 and the motor 20 are directly connected by a shaft, it is desirable that the load torque T _L and the motor torque T _M are equal in terms of operational stability, device stability, quietness, and the like.

On the other hand, as shown in FIG. 10, when the motor torque T _M is constant and the load torque T _L changes periodically, the rotational speed of the motor 20 changes transiently accordingly ( Momentarily, the motor speed increases / decreases and returns to the original state).

Against periodic variations in load torque T _L, in order to suppress the fluctuation of the rotational speed of the motor 20 becomes zero to the difference between the load torque T _L and the motor torque T _M is reduced (ideally The motor torque T _M may be controlled as described above. That is, as shown in FIG. 11, by controlling the motor torque T _M in accordance with the periodically varying load torque T _L , the difference between the load torque T _L and the motor torque T _M can be made close to zero. it can.

The motor torque T _M is proportional to the motor current. When the load torque T _L varies, the motor rotation speed varies after the motor current varies. That is, it is the motor current that varies directly with respect to the variation of the load torque _TL . Therefore, in this embodiment, the motor current is estimated or detected, and the motor 20 is controlled based on the motor current. Here, since the motor current can be detected with a sampling period shorter than that of the technique of Patent Document 1, it can be detected with high accuracy.

Periodic variations in load torque T _L with respect to (AC waveform), it is difficult to difference between the load torque T _L controls the motor torque T _M to a value that approaches zero. For the load torque T _L, represented by AC waveform, in terms of a combination of this phase, because it is necessary to control the motor torque T _M. On the other hand, if the influence (load fluctuation component) due to the fluctuation of the load 100 having periodicity is expressed by a DC value, torque control may be performed so that the load fluctuation component becomes zero. Is relatively easy, and stable control is possible.

The basic principle of torque control according to this embodiment will be described below.
As shown in the left graph of FIG. 3, when there is periodic torque fluctuation, only the torque fluctuation component (load fluctuation component) is estimated as a DC value by using a filter. That is, as shown in [Equation 9] below, when an input including a torque fluctuation component is filtered, the output becomes a DC value. As a filter having such characteristics, there is the following [Equation 9].

Here, in the above [Equation 9], the DC value (feature value) obtained as an output corresponds to the magnitude of the load fluctuation (load fluctuation component). In the motor 20 connected to the load 100 in which the load torque T _L periodically varies by controlling the torque using, for example, a controller such as a general PI controller so that the DC value becomes zero, It becomes possible to suppress fluctuations in the rotational speed.

  Here, it is necessary that the input value having the torque fluctuation component, that is, the value of the motor current including the torque fluctuation component can be detected or estimated. Hereinafter, this problem will be described with reference to FIGS.

In FIG. 12, when the motor current including the torque fluctuation component (the fluctuation component of the load torque T _L ) is denoted by ele ′, the value ele ′ is expressed as a combined wave (superimposed wave) of the load torque T _L and the motor current ele. Is done. In this case, when the cycle T _Mech (corresponding to the mechanical angle) of the load torque T _L is equal to the cycle T _ele (corresponding to the electrical angle) of the motor current ele, the load is connected to the motor 20 based on the motor current ele ′. It is difficult to detect whether the size of the applied load 100 is constant or whether the size of the load 100 connected to the motor 20 is fluctuating (presence of torque fluctuation component). By looking at the waveform of the motor current ele ′, the motor current ele ′ is a case where there is a fluctuation in the size of the load 100 connected to the motor 20 (mechanical load fluctuation), and its torque fluctuation component. This is because it is difficult to identify whether the load is connected to the motor 20 or whether the load 100 connected to the motor 20 is constant (the torque fluctuation component is zero).

On the other hand, as shown in FIG. 13, when the cycle T _Mech of the load torque _{TL and} the cycle T ele of the motor current _ele are different, the magnitude of the load 100 connected to the motor 20 is based on the motor current ele ′. It is easy to detect whether the load is constant or whether the size of the load 100 connected to the motor 20 is fluctuating (presence of torque fluctuation component). This is because in this case, the waveform of the motor current ele ′ is clearly different between when the torque fluctuation component is present and when it is absent. Thus, this embodiment is more effective when the period T _Mech of the load torque _{TL and} the period T ele of the motor current _ele are different. From this, this embodiment is suitably used for the motor 20 having two or more pole pairs so that the cycle T _Mech of the load torque _{TL and} the cycle T ele of the motor current _ele are different (number of pole pairs). This is because, in the motor of 1, the cycle T _Mech of the load torque T _L is equal to the cycle T _ele of the motor current ele).

  With reference to FIG. 1, a motor control apparatus according to the synchronous motor control method of the present embodiment will be described.

  In FIG. 1, the motor control device 10 drives a permanent magnet motor 20 having saliency in a sensorless manner. The motor control device 10 includes a PWM inverter 17, a current detector (not shown), coordinate transformation calculation units 27 and 28, a position error estimator 21, a speed estimator 24, an integrator 26, and a speed control. , Current controller 14, voltage generator 15, voltage compensator 16, mechanical angle error estimation (γ) unit 41, mechanical angle error estimation (δ) unit 42, phase control unit 43, And an amplitude control unit 44.

The PWM inverter 17 converts a DC voltage into a three-phase AC voltage.
The current detector (not shown) detects the current i (u, w) of the motor 20.
The coordinate conversion calculation unit 27 converts the detected current i (u, w) into rotational coordinates and outputs a current i (γ, δ).
The mechanical angle error estimation (γ) unit 41 and the mechanical angle error estimation (δ) unit 42 respectively input a current i (γ, δ) and a torque fluctuation component Err () included in the current i (γ, δ). M) (γ, δ) is obtained.

The phase control unit 43 obtains phase compensation θγδ (Err) ^ based on the torque fluctuation component Err (M) γ. The phase compensation θγδ (Err) ^ is input to the current controller 14.
The amplitude controller 44 obtains an amplitude compensation Vm (Err) ^ based on the torque fluctuation component Err (M) δ. The amplitude compensation Vm (Err) ^ is input to the voltage generator 15.

The position error estimator 21 estimates the rotor position error Δθ ^.
The speed estimator 24 estimates the angular speed ωre ^ so that the estimated position error Δθ ^ becomes zero.
The integrator 26 integrates the output ωre ^ of the speed estimator 24 to calculate the rotor position θ (γ, δ) ^.

The speed controller 12 is used to make the error between the speed command value ωre ^* and the output ωre ^ of the speed estimator 24 zero.
The current controller 14 is used to make the error between the γ-axis (d-axis) current command value iγ ^* and the γ-axis current iγ obtained from the actually detected current information zero.
The PWM output value V (u, v, w) ^* is calculated based on the voltage command value V (m) ^* output from the speed controller 12 and the phase command value Vβ ^* output from the current controller 14. .

The voltage generator 15 outputs to the PWM inverter 17 based on the voltage amplitude command V (m) ^* output from the speed controller 12 and the phase V (θ) ^{* of the} voltage command output from the adder 18. Generated voltage command V (u, v, w) ^* .
The voltage compensator 16 inputs the voltage command value V (u, v) ^* to the PWM inverter 17 generated by the voltage generator 15 and corrects the phase and amplitude with respect to the value V (u, v) ^* . The estimated voltage value Vmd (u, v) ^ is output.
The coordinate transformation calculation unit 28 transforms the estimated voltage value Vmd (u, v) ^ output from the voltage compensator 16 into rotational coordinates.

The command values for the motor control device 10 are the angular velocity ωre ^* and the γ-axis current Iγ ^* .
The adder 11 calculates a deviation between the angular velocity command value ωre ^* and the estimated angular velocity value ωre ^. The deviation is input to a speed controller 12 constituted by a PI (proportional integral) controller. The output command output from the speed controller 12 is the voltage amplitude command V (m) ^* . This voltage amplitude command V (m) ^* is an amplitude command of the three-phase command voltage of the motor 20.

The adder 13 calculates a deviation between the γ-axis current command value Iγ ^* and the Iγ detected and calculated from the motor current. The deviation is input to the current controller 14 configured by the PI controller. The output command output from the current controller 14 is a voltage phase command value Vβ ^* . This voltage phase command value Vβ ^* is a phase command of the three-phase command voltage of the motor 20. As shown in FIG. 5, in order to stabilize the operation of the motor 20, the output of the command value Vβ ^* of this voltage phase is passed through a digital LPF, vibration components and the like are reduced, and the command voltage phase is stabilized. May be.

The adder 18 calculates the sum of the voltage phase command Vβ ^* and the estimated rotor position θ (γδ) ^ as the voltage command phase V (θ) ^* . The phase V (θ) ^* of the voltage command is input to the voltage generator 15. In the voltage generator 15, for example, the following command voltage waveform is generated.

Such a voltage command V (u, v, w) ^* is output to the PWM inverter 17. The PWM inverter 17 is configured by an inverter circuit or the like, and generates a PWM waveform. Since this PWM inverter 17 is a general PWM inverter that has been frequently used in the past, the description thereof is omitted.

  When the IPM motor 20 is actually driven by the PWM inverter 17, the phase current of the motor 20 is detected. If this detection circuit uses a CT or the like and an amplifier circuit using an operational amplifier is formed on the secondary side of the CT, a value (waveform) obtained by easily converting the phase current into a voltage signal can be obtained. Since the waveform of the motor phase current is an analog value, it is converted into a digital value by an AD converter or the like and converted into a value that can be calculated.

Further, since the phase currents iu and iw of the motor are currents as seen from the stationary coordinate system, the coordinate transformation calculation unit 27 performs coordinate transformation on the estimated rotational coordinate system. This transformation matrix is the following matrix (similar to [Equation 9] above).

The coordinate transformation performed in the coordinate transformation computation unit 27 using the above [Equation 11] to [Equation 9] outputs the values iγ and iδ as seen in the motor current cycle (T _{ele in} FIG. 13). If the calculated currents i γ and i δ have torque fluctuation components that fluctuate with the rotation period of the load 100 (mechanical period, T _Mech in FIG. 13), the values of the currents i γ and i δ are obtained as shown in FIG. Will fluctuate. That is, the motor current (iγ, iδ) of a motor 20 connected to a load whose load varies periodically, such as a motor in a compressor having a discharge process and a suction process during one rotation, As shown in FIG. 4, the value fluctuates in one rotation cycle.

When the current value (iγ, iδ) is filtered in each of the mechanical angle error estimation (γ) unit 41 and the mechanical angle error estimation (δ) unit 42 as shown in the following equation [Equation 12], Thus, a torque fluctuation component Err (M) (γ, δ) included in the motor current (iγ, iδ) is obtained.

  The torque fluctuation component Err (M) (γ, δ) obtained by the above equation [Equation 12] corresponds to the amplitude of the periodic torque fluctuation and has a constant value. In the above equation [Equation 12], p is the number of pole pairs and represents an integer multiple between the electrical angle and the mechanical angle. θ (γδ) ^ is the estimated rotor position (estimated value of the mechanical angle of the rotor). Using the phase control unit 43 and the amplitude control unit 44, a feedback loop is configured so that each component of Err (M) (γ, δ) disappears (becomes zero) as a torque fluctuation component. As a result, it is possible to suppress fluctuations in the motor rotation speed due to torque fluctuations.

In the above equation [Equation 12], the rotation period of the load 100 is used for the superimposed waveform ele ′ of FIGS. 12 and 13 by using the pole pair number p, that is, the product of the electrical angle θ and the pole pair number p. By _performing rotational coordinate conversion using (T _Mech ), a torque fluctuation component (a fluctuation in the magnitude of the load 100) can be detected as Err (M) (γ, δ).

Note that the above equation [Equation 12] can be variously modified depending on the direction of rotation and the initial phase (see equation [Equation 13] below).

  The configuration of the mechanical angle error estimation (γ) unit 41 will be described with reference to FIG.

  In FIG. 2, a pθγδ (M) creation unit 413 estimates a mechanical position (information on a mechanical angle) pθγδ used in an operation performed in the block denoted by reference numeral 411. Here, the case where the motor 20 is a motor having four pole pairs (four-pole motor) will be described as an example. In this case, when the electrical angle (θγδ input to the mechanical angle error estimation (γ) unit 41) is, for example, 90 °, the mechanical angle is 45 ° or 225 °. That is, there is a possibility that the mechanical angle takes two values for a certain electrical angle. In this case, if it is the first half (0 to 180 °) of the mechanical cycle, the mechanical angle is 90 °, and if it is the second half (180 to 360 °), the mechanical angle is 225 °. The relationship between the electrical angle and two mechanical angles corresponding to the electrical angle is preset in the initial phase estimation unit 415.

  As shown by ele ′ in FIG. 13, the output ωre ^ (how to change the load) of the speed estimator 24 is different between the first half and the second half of the mechanical cycle. Based on the above, it is determined whether the first half or the second half of the mechanical cycle, and the determination result is output to the pθγδ (M) creation unit 413. The pθγδ (M) creation unit 413 specifies the mechanical angle based on the information on the two mechanical angles from the initial phase estimation unit 415 and the determination result, and outputs the information to the calculation block indicated by reference numeral 411. The calculation result (output) of the calculation block is multiplied by a constant K in the calculation block 412 to output Err (M) (γ, δ) as the torque fluctuation component.

  The initial phase estimation unit 415 stores information related to the initial phase every time the motor 20 is started. By using the information of the initial phase, an operation for specifying one of the two mechanical angles corresponding to a certain electrical angle is definitely performed. By using the information of the initial phase, it is possible to effectively stabilize the motor operation and to improve the response when there is an unexpected disturbance or the like.

As shown in FIG. 1, the adder 13 calculates a deviation between the current iγ and the γ-axis current command value iγ ^*, and the deviation is input to the current controller 14. Further, the current iγ is used for calculation in the position error estimator 21 and calculation in the inductance compensator 22. On the other hand, the current iδ is used for calculation in the position error estimator 21 and calculation in the inductance compensator 22.

  The inductance compensation in the inductance compensator 22 is configured to compensate for inductance saturation and modeling error. This inductance compensation is configured to be a function of at least one variable of current (iγ, iδ) and rotational speed. For this compensation, an approximate expression or a table may be used.

The voltage compensator 16 inputs the voltage command value V (u, v) ^* to the PWM inverter 17 generated by the voltage generator 15 and corrects the phase and amplitude with respect to the value V (u, v) ^* . The estimated voltage value Vmd (u, v) ^ is output. The correction performed by the voltage compensator 16 takes into consideration the nonlinearity of input / output in the PWM inverter 17 and is performed so as to correspond to the output from the PWM inverter 17 to the motor 20.

  Hereinafter, the calculation in the position error estimator 21 will be described.

ｋ_Eは、誘起電圧定数である。 The voltage equation based on the motor model in the dq axis real rotation coordinate system is shown in the following formula [Formula 14].

k _E is an induced voltage constant.

The voltage equation based on the motor model in the γ-δ axis estimated rotational coordinate system is expressed by the following equation [Equation 15].

Here, each parameter of the inductance L is expressed by the following equation [Equation 16].

Here, since Δθ is controlled to be zero, Δθ≈0. Therefore, sinΔθ≈0 and cosΔθ≈1. When this approximation is used, the following formula [Equation 17] is obtained for Δθ.

Here, since the induced voltage V is obtained by differentiating the magnetic flux φ, the following equation [Equation 18] is obtained as a relational expression regarding the amount of magnetic flux. That is, by dividing the voltage equation of [Equation 17] by the angular velocity ωre, the magnetic flux equation of [Equation 18] below is obtained.

ここで、電流の過渡項は無視し、ｐ（ｉ）≒０とした。また、Ｋ’は、誘起電圧定数の逆数に相当し、Ｋ’＞０を満たす任意の定数または、関数のように可変にさせてもよい。 Using the approximation of Δθre≈sin Δθre, the following equation [Equation 19] is obtained.

Here, the transient term of the current is ignored and p (i) ≈0. K ′ corresponds to the reciprocal of the induced voltage constant, and may be made variable as an arbitrary constant or function satisfying K ′> 0.

  Since the DC motor rotates, V becomes a value proportional to ω. Therefore, in the equation [Equation 19], Vγ is a value proportional to ω. Therefore, the term of (Vγ−Riγ) / ωre in the above [Formula 19] does not depend on ω. From this, stable control can be performed by obtaining Δθre from the above [Equation 19] and performing control to make Δθre zero.

Ｋ”はＫ”＞０を満たす任意の定数または関数として制御可能である。応答性を重視する場合には、Ｋ”を大きな値とし、応答性よりも安定性を重視したい場合には、Ｋ”＝０とする。この場合には、上記［数１９］と同じ式となる。 Further, when p (i) ≠ 0 and the transient term is taken into consideration, the following equation [Equation 20] is obtained.

K ″ can be controlled as any constant or function that satisfies K ″> 0. In the case where importance is attached to responsiveness, K ″ is set to a large value, and in the case where stability is more important than responsiveness, K ″ = 0 is set. In this case, the equation is the same as the above [Equation 19].

In FIG. 1, R ^* is the winding resistance of the motor. The q-axis inductance Lq ^* is obtained by the inductance compensator 22. The q-axis inductance Lq ^* is a function of iγ, iδ, and ωre obtained in advance by experiments using a motor with a sensor.

  As shown in FIG. 1, the position error Δθre (Δθ ^) obtained by the position error estimator 21 is output to the speed estimator 24.

The position error Δθ ^ calculated by the position error estimator 21 is input to a speed estimator 24 configured by a PI controller. The speed estimator 24 calculates an angular speed estimated value ωre ^ such that Δθ ^ becomes zero. Here, as the speed estimator 24, a general PI controller is usually used. The calculation formula in the speed estimator 24 is as shown in the following formula [Equation 21].

The estimated angular velocity value ωre ^ obtained by the velocity estimator 24 is output to the adder 11 and used for velocity feedback control as described above. A voltage amplitude command V (m) ^* is generated by the speed feedback control. Further, the estimated angular velocity value ωre ^ is integrated by the integrator 26 to calculate the rotor position estimated value θre (γδ) ^. The calculated position estimation value θre (γδ) ^ is input to the voltage phase command adder 18 and the coordinate converters 27 and 28, respectively.

The adder 18 generates a voltage phase command V (θ) ^* input to the voltage generator 15 based on the position estimated value θre (γδ) ^. As described above, the positional deviation of Δθ ^ estimated by the position error estimator 21 is reflected in the voltage phase command V (θ) ^* , so that the positional deviation of Δθ ^ is reflected in the motor 20.

  The angular velocity estimated value ωre ^ is adjusted so that the actual position of the motor 20 matches the estimated value, and the position θre (γδ) ^ on the estimated coordinate system, which is an integral value thereof, is obtained. Feedback control is performed so that the positions of the γ-δ axes coincide.

  As described above, in the motor control device 10 of the present embodiment, the value detected by the sensor is used as the current value, but the command value or the estimated value is used as the voltage value (the voltage sensor is not being used). An inductance estimated value (command value) is obtained by the inductance compensator 22 for the inductance, which is an important factor in the operation of the motor having the saliency (IPM motor). Regarding the angular velocity ω, in the present embodiment, an estimated value is used in consideration of the fact that a transient one is represented, but a command value can also be used.

  In the above embodiment, the synchronous motor control method is applied in the configuration of the motor control device 10 shown in FIG. 1, but the control method is also applied to the load fluctuation control units 410A to 410D in FIGS. Can be applied.

  FIG. 6 shows a general control flow of sensorless control in which a motor current is detected and position estimation (position estimation unit 300B) is performed based on the motor current. The position estimation unit 300B detects the two-phase current of the motor 20B and converts it into the γ-δ coordinate axes. The rotor position and axis deviation of the motor 20B are calculated from the current value after the coordinate conversion, and feedback control is performed on the speed command value ω * so that the axis deviation is zero.

  The torque control unit 400B corresponds to the coordinate transformation calculation units 27 and 28, the mechanical angle error estimation (γ, δ) units 41 and 42, the phase control unit 43, and the amplitude control unit 44 shown in FIG. The load fluctuation control unit 410B corresponds to the mechanical angle error estimation (γ, δ) units 41 and 42, the phase control unit 43, and the amplitude control unit 44 among them. The load fluctuation control unit 410B can detect and estimate only the torque fluctuation component by passing a predetermined filter through the motor current. A control configuration for causing the torque fluctuation to follow a predetermined value is facilitated, and the stability of the control system is improved.

  FIG. 5 corresponds to the configuration example of FIG. 6 described above. In FIG. 5, the internal configuration of the control device 200A and the output destination of the signal of the torque control unit 400A are clearly shown.

  FIG. 7 shows a configuration in which position estimation is performed using a motor current, and is a configuration similar to the technique of Patent Document 2. As shown in FIG. 8, the position detector 500 such as a Hall IC may be provided.

The operation of this embodiment is as follows.
As shown in FIG. 1, compensation for torque control is performed on the speed generation value and the current generation value. For the former, Err (M) δ is calculated by mechanical angle error estimation (δ), and after compensating for amplitude, θδ (M) is compensated. For the latter, Err (M) γ is calculated by mechanical angle error estimation (γ), and after phase compensation, θγ (M) is compensated.

According to this embodiment, the following effects can be achieved.
For example, it becomes possible to suppress torque fluctuations in the low speed range, thereby expanding the operating range. In the compressor of an air conditioner, the operating range in the low speed range is widened, the electricity cost is reduced, and a system capable of high efficiency can be realized.

  Further, since the current corresponding to the torque component is detected or estimated and the torque fluctuation is suppressed, a system that is resistant to disturbances can be constructed. In addition, by applying a filter to the current value, the feedback control can be made to follow a predetermined set value, so that the configuration of the control system is facilitated and the stability of the control is improved.

1 is a block diagram of a motor control device to which an embodiment of a synchronous motor control method of the present invention is applied. It is a block diagram which shows the mechanical angle error estimation ((gamma)) part of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is an image figure for demonstrating the basic principle of the torque control of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is an image figure for demonstrating the basic principle of the torque control of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is a block diagram which shows the structural example of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is a block diagram which shows the other structural example of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is a block diagram which shows the further another structural example of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is a block diagram which shows the further another structural example of the motor control apparatus with which one Embodiment of the control method of the synchronous motor of this invention is applied. It is a figure for demonstrating the relationship between a motor torque and load torque in one Embodiment of the control method of the synchronous motor of this invention. FIG. 6 is another diagram for explaining the relationship between the motor torque and the load torque in the embodiment of the synchronous motor control method of the present invention. FIG. 10 is still another diagram for explaining the relationship between the motor torque and the load torque in the embodiment of the synchronous motor control method of the present invention. It is a figure for demonstrating the relationship between a mechanical period and an electrical period in one Embodiment of the control method of the synchronous motor of this invention. It is another figure for demonstrating the relationship between a mechanical period and an electrical period in one Embodiment of the control method of the synchronous motor of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Motor controller 11 Adder 12 Speed controller 13 Adder 14 Current controller 15 Voltage generation part 16 Voltage compensator 17 PWM inverter 21 Position error estimator 22 Inductance compensator 24 Speed estimator 26 Integrator 27 Coordinate conversion calculation part 28 Coordinate transformation calculation unit 41 Mechanical angle error estimation (γ) unit 42 Mechanical angle error estimation (δ) unit 43 Phase control unit 44 Amplitude control unit

Claims (13)

A method of controlling the synchronous motor to which a load whose load torque varies during one rotation of the synchronous motor is connected,
(A) calculating a vector corresponding to the variation of the load torque;
(B) A step of controlling the calculated vector to a desired control amount. A method for controlling a synchronous motor, comprising: In the synchronous motor control method according to claim 1,
The calculated vector is a rotation vector;
Said (b) includes controlling the value formed by carrying out the rotation coordinate transformation of the said rotation vector to the said desired control amount. The synchronous motor control method characterized by the above-mentioned. In the synchronous motor control method according to claim 1 or 2,
The method for controlling a synchronous motor, wherein the desired control amount is a constant value. In the synchronous motor control method according to any one of claims 1 to 3,
The method for controlling a synchronous motor, wherein the vector is obtained based on a current value of the synchronous motor. In the synchronous motor control method according to any one of claims 1 to 4,
The vector is obtained for each of the d-axis and q-axis of the synchronous motor. In the control method of the synchronous motor according to any one of claims 1 to 5,
A method for controlling a synchronous motor, wherein the phase or amplitude of the synchronous motor is controlled based on the calculated vector. In the synchronous motor control method according to claim 2,
The value is a value corresponding to the mechanical angle of the synchronous motor,
The said value is calculated | required using the multiplication value of the electrical angle of the said synchronous motor, and the number of pole pairs. The synchronous motor control method characterized by the above-mentioned. In the synchronous motor control method according to any one of claims 1 to 7,
The method for controlling a synchronous motor, wherein the control of the synchronous motor is performed by sensorless control. In the synchronous motor control method according to any one of claims 1 to 8,
The calculation of the vector is obtained using the following equation [Equation 1] or [Equation 2].

A method for controlling a synchronous motor. In the synchronous motor control method according to any one of claims 1 to 8,
The calculation of the vector is obtained using the following equation [Equation 3] or [Equation 4].

A method for controlling a synchronous motor. In the synchronous motor control method according to any one of claims 1 to 10,
(B) includes that the vector is input to a controller, and that the output of the controller is feedback controlled to control the vector to the desired control amount. Control method. In the synchronous motor control method according to any one of claims 1 to 11,
The method for controlling a synchronous motor, wherein the desired control amount is zero. In the synchronous motor control method according to any one of claims 1 to 12,
Said (a) includes calculating the said vector using the information regarding the initial phase of the said synchronous motor. The control method of the synchronous motor characterized by the above-mentioned.

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