JP2006149170A - Switching power supply circuit, switching converter device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce circuit components and circuit manufacturing cost as a wide range compatible switching power supply circuit for improving power factor.
Each switching element is configured to perform a switching operation on a rectified output voltage V1 for a commercial AC power supply AC, and then an average value of the DC input voltage Ei is added by adding a phase compensation capacitor Ct2 in a control circuit 1. Try to stabilize. This improves the power factor. Further, by reducing the coupling coefficient k between the primary side and the secondary side in the insulating converter transformer PIT to a predetermined value (for example, k = 0.70 or less), the control range of the switching frequency necessary for stabilization can be reduced. Wide range support is realized. As a result, the conventional active filter can be omitted as a configuration for improving the power factor and supporting a wide range.
[Selection] Figure 1

Description

  The present invention relates to a switching power supply circuit used as a power supply for various electronic devices, for example. The present invention also relates to a switching converter device particularly suitable for a power supply circuit.

JP-A-6-327246 (FIG. 11)

In recent years, with the development of switching elements that can withstand relatively large currents and voltages at high frequencies, most power supply circuits that rectify commercial power supplies to obtain a desired DC voltage are switching power supply circuits. .
The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

By the way, generally, when a commercial power source is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the power source utilization efficiency is impaired.
Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V, for example, to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V AC region such as Europe. A so-called wide-range power supply circuit that is configured so as to be able to be used is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, and fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve each of these problems, a method using a so-called active filter is known as a conventional technique for realizing a power factor improvement and a wide-range configuration (see, for example, Patent Document 1).

The basic configuration of such an active filter is, for example, as shown in FIG.
In FIG. 11, a bridge rectifier circuit Di is connected to a commercial AC power supply AC. An output capacitor Cout is connected in parallel to the positive / negative line of the bridge rectifier circuit Di. By supplying the rectified output of the bridge rectifier circuit Di to the output capacitor Cout, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. This direct-current voltage Vout is supplied as an input voltage to a load 10 such as a subsequent DC-DC converter.

As shown in the figure, the power factor improvement includes an inductor L, a fast recovery diode D, a resistor Ri, a switching element Q, and a multiplier 11.
The inductor L and the diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the output capacitor Cout.
In this case, the switching element Q1 is a MOS-FET, and is inserted between the connection point of the inductor L and the diode D and the primary side ground as shown in the figure.

To the multiplier 11, a current detection line LI and a waveform input line Lw are connected as a feedforward circuit, and a voltage detection line LV is connected as a feedback circuit.
The multiplier 11 detects the level of the rectified current that flows from the current detection line LI and flows to the negative output terminal of the bridge rectifier circuit Di.
Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.
Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV.
The multiplier 11 outputs a drive signal for driving the switching element Q.

  The multiplier 11 first multiplies the rectified current level detected from the current detection line LI as described above by the fluctuation difference of the DC input voltage detected from the voltage detection line LV. Then, a current command value having the same waveform as the AC input voltage VAC is generated based on the multiplication result and the waveform of the AC input voltage detected from the waveform input line Lw.

  Further, the multiplier 11 in this case compares the current command value with the actual AC input current level (detected based on the input from the current detection line L1), and performs PWM control on the PWM signal according to this difference. To generate a drive signal based on the PWM signal. The switching element Q is switched by this drive signal. As a result, the AC input current is controlled to have the same waveform as the AC input voltage, and the power factor is improved so that the power factor approaches one. In this case, since the current command value generated by the multiplier is controlled so that the amplitude changes according to the fluctuation difference of the rectified smoothing voltage, the fluctuation of the rectified smoothing voltage is also suppressed. .

  FIG. 12A shows the input voltage Vin and the input current Iin input to the active filter circuit shown in FIG. The voltage Vin corresponds to the voltage waveform as the rectified output of the bridge rectifier circuit Di, and the current Iin corresponds to the current waveform as the rectified output of the bridge rectifier circuit Di. Here, the waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This is the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectifier circuit Di. Also indicates that the conduction angle is the same as that of the current Iin. That is, a power factor close to 1 is obtained.

FIG. 12B shows a change in energy (power) Pchg input / output to / from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low, thereby maintaining the flow of output power.
FIG. 12C shows a waveform of the charge / discharge current Ichg with respect to the output capacitor Cout. This charge / discharge current Ichg flows corresponding to the accumulation / discharge operation of the energy Pchg in the output capacitor Cout, as can be seen from the fact that it is in phase with the waveform of the input / output energy Pchg in FIG. Current.

  Unlike the input current Vin, the charge / discharge current Ichg has substantially the same waveform as the second harmonic of the AC line voltage (commercial AC power supply AC). In the AC line voltage, a ripple voltage Vd is generated in the second harmonic component as shown in FIG. 12D due to the flow of energy with the output capacitor Cout. The ripple voltage Vd has a phase difference of 90 ° with respect to the charge / discharge current Ichg shown in FIG. The rating of the output capacitor Cout is determined in consideration of processing the second harmonic ripple current and the high frequency ripple current from the boost converter switch that modulates the current.

FIG. 13 shows a configuration example of an active filter having a basic control circuit system based on the circuit configuration of FIG. In addition, about the part made the same as FIG. 11, the same code | symbol is attached | subjected and description is abbreviate | omitted.
A switching pre-regulator 15 is provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. In FIG. 11, the switching pre-regulator 15 is a part formed by the switching element Q, the inductor L, the diode D, and the like.

The control circuit system including the multiplier 11 includes a voltage error amplifier 12, a divider 13, and a squarer 14 in addition.
In the voltage error amplifier 12, the DC voltage Vout of the output capacitor Cout is divided by the voltage dividing resistor Rvo-Rvd and input to the non-inverting input of the operational amplifier 15. A reference voltage Vref is input to the inverting input of the operational amplifier 15. The operational amplifier 15 amplifies a voltage of a level corresponding to the error of the divided DC voltage Vout with respect to the reference voltage Vref by an amplification factor determined by the feedback resistor Rvl and the capacitor Cvl, and outputs the error output voltage Vvea to the divider 13. Output to.

  Further, a so-called feedforward voltage Vff is input to the squarer 14. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input voltage Vin by the averaging circuit 16 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 14 squares the feedforward voltage Vff and outputs it to the divider 13.

The divider 13 divides the error output voltage Vvea from the voltage error amplifier 12 by the square value of the average input voltage output from the squarer 14 and outputs a signal as a result of the division to the multiplier 11.
That is, the voltage loop is made up of a system of a squarer 14, a divider 13, and a multiplier 11. The error output voltage Vvea output from the voltage error amplifier 12 is divided by the square of the average input voltage (Vff) before being multiplied by the rectified input signal Ivac in the multiplier 11. With this circuit, the gain of the voltage loop is kept constant without changing as the square of the average input voltage (Vff). The average input voltage (Vff) has a function of open loop correction that is sent forward in the voltage loop.

An output obtained by dividing the error output voltage Vvea by the divider 11 and a rectified output (Iac) of the positive output terminal (rectified output line) of the bridge rectifier circuit Di via the resistor Rvac are input to the multiplier 11. Here, the rectified output is shown not as voltage but as current (Iac). The multiplier 11 multiplies these inputs to generate and output a current programming signal (multiplier output signal) Imo. This corresponds to the current command value described in FIG. The output voltage Vout is controlled by varying the average amplitude of this current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching drive is performed by a drive signal based on the PWM signal, thereby controlling the level of the output voltage Vout.
Thus, the current programming signal has an average amplitude waveform that controls the input and output voltages. Note that the active filter controls not only the output voltage Vout but also the input current Vin. Since the current loop in the feedforward circuit can be said to be programmed by the rectified line voltage, the input to the subsequent converter (load 10) becomes resistive.

FIG. 14 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG.
In the power supply circuit shown in this figure, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

First, in the power supply circuit shown in FIG. 14, two sets of line filter transformers LFT and LFT and three sets of across capacitors CL are connected to the commercial AC power supply AC in accordance with the connection mode shown in the figure. The bridge rectifier circuit Di is connected through the inrush current limiting circuit 22 shown in the figure.
The inrush current limiting circuit 22 is composed of a parallel connection circuit including an inrush current limiting resistor Ri and a switch SW. For example, the switch SW is turned off by an external signal, so that the commercial AC power supply at the time of starting the power supply circuit Inrush current from the side is restricted.
The rectified output line of the bridge rectifier circuit Di has a normal mode noise filter 4 formed by connecting one set of choke coils LN and two sets of filter capacitors (film capacitors) CN and CN as shown in the figure. It is connected.

The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci via the series connection of the choke coil LN, the inductor Lpc of the power choke coil PCC, and the fast recovery type rectifier diode D10. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIGS. Further, the inductor Lpc and the diode D10 of the power choke coil PCC correspond to the inductor L and the diode D shown in FIG.
In addition, an RC snubber circuit including a capacitor Csn and a resistor Rsn is connected in parallel to the rectifier diode D10 in this figure.

The switching element Q6 corresponds to the switching element Q in FIG. That is, in actually mounting the switching element of the active filter, in this case, the switching element Q6 is connected to the connection point between the power choke coil Lpc and the fast recovery type rectifier diode D10, and the primary side ground (negative rectified output line). It is inserted between.
In this case, a MOS-FET is selected as the switching element Q6. A gate-source resistor R52 is connected between the gate and source of the switching element Q6.

In this case, the active filter control circuit 20 controls the operation of the active filter for improving the power factor so that the power factor approaches 1, and is an integrated circuit (IC) having one stone, for example.
In this case, the active filter control circuit 20 includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, a drive circuit that outputs a drive signal for switching the switching element, and the like. Circuit portions corresponding to the multiplier 11, the error voltage amplifier 12, the divider 13, the squarer 14, and the like shown in FIG. 13 are mounted in the active filter control circuit 20.

  In this case, the feedback circuit is formed so that the voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistors R56 and R57 is input to the terminal T1 of the active filter control circuit 20. .

As the feedforward circuit, first, the rectified output is input to the terminal T3 via the resistor R58. This forms an AC input voltage waveform detection and corresponding feedforward circuit for the averaging circuit.
Further, the rectified current level is input to the terminal T6 via the resistor R60 from the connection point between the negative output terminal of the bridge rectifier circuit Di and the resistor R61 inserted between the primary side grounds. That is, a feedforward circuit is formed as a line corresponding to the current detection line LI in FIG.

Further, the positive rectified output of the bridge rectifier circuit Di via the starting resistor Rs is input to the terminal T4 as the starting voltage. The active filter control circuit 20 is activated by the activation voltage input to the terminal T4 when the power supply is activated.
In the power choke coil PCC, a winding N5 that is transformer-coupled with the inductor Lpc is wound. The alternating voltage excited in the winding N5 is converted into a predetermined low-voltage DC voltage by a half-wave rectifier circuit comprising a diode D11 and a capacitor C11. This low-voltage DC voltage is also input to the terminal T4. Yes. After being activated by the activation voltage, the active filter control circuit 20 is operated by inputting this low-voltage DC voltage as a power source.
The terminal T5 is connected to the primary side ground via a resistor R59.

A drive signal for driving the switching element is output from the terminal T2. The drive signal output from the terminal T2 is output to the gate of the switching element Q6 via the resistor R51.
In the switching element Q6, a gate voltage is generated at both ends of the gate-source resistor R52 in accordance with the applied drive signal. Then, the switching operation is performed so that the gate voltage is turned on when the voltage is equal to or higher than the threshold value, and is turned off when the gate voltage is equal to or lower than the threshold value.

  The switching drive of the switching element Q6 is performed based on PWM control so that the conduction angle of the rectified output current becomes substantially the same as the rectified output voltage waveform as described with reference to FIGS. Done by signal. The conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform. That is, the conduction angle of the AC input current flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC. As a result, the power factor is controlled to be approximately 1. That is, power factor improvement is achieved. In practice, a characteristic of power factor PF = 0.99 to 0.98 is obtained.

Further, depending on the active filter control circuit 20 shown in FIG. 14, the average value of the rectified and smoothed voltage Ei (corresponding to Vout in FIG. 13) = 375 V is constant in the range of AC input voltage VAC = 85 V to 264 V. Also works. That is, a DC input voltage stabilized at 375 V is supplied to the subsequent stage current resonance type converter regardless of the fluctuation range of the AC input voltage VAC = 85 V to 264 V.
The range of the AC input voltage VAC = 85V to 264V continuously covers the commercial AC power supply AC100V system and the 200V system. Therefore, the switching converter in the subsequent stage includes the commercial AC power supply AC100V system and the 200V system. The DC input voltage (Ei) stabilized at the same level is supplied. That is, the power supply circuit shown in FIG. 14 is configured as a wide-range power supply circuit by including an active filter.

  As shown in the figure, the current resonance converter at the latter stage of the active filter includes two switching elements Q1 and Q2. In this case, a half-bridge connection is made such that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and the switching element Q1 is connected in parallel with the rectified and smoothed voltage Ei (DC input voltage). That is, a current resonance type converter by a half bridge coupling method is formed.

The current resonance type converter in this case is a separately excited type, and correspondingly, MOS-FETs are used for the switching elements Q1 and Q2. Clamp diodes DD1 and DD2 are connected in parallel to these switching elements Q1 and Q2, respectively, thereby forming a switching circuit. These clamp diodes DD1 and DD2 form a path through which a reverse current flows when the switching elements Q1 and Q2 are turned off.
The switching elements Q1 and Q2 are driven to be switched by the drive circuit 21 at a required switching frequency at the timing when they are alternately turned on / off. Further, the drive circuit 21 variably controls the switching frequency according to the level of a secondary side DC output voltage Eo described later, thereby stabilizing the secondary side DC output voltage Eo.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 from the primary side to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the switching elements Q1 and Q2, and the other end is connected to the primary side via the series resonance capacitor C1. Connected to ground. Here, the series resonant capacitor C1 forms a series resonant circuit by its own capacitance and the leakage inductance (L1) of the primary winding N1. This series resonance circuit causes a resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied. By this, the operation of the switching circuit composed of the switching elements Q1 and Q2 is made a current resonance type.

Although the description by illustration here is abbreviate | omitted, as a structure of the above-mentioned insulation converter transformer PIT, the EE type | mold core which combined the E type | mold core by a ferrite material is provided, for example. Then, after the winding part is divided between the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the inner magnetic leg of the EE core.
Further, a gap of about 1.0 mm or less is formed with respect to the inner magnetic leg of the EE type core of the insulating converter transformer PIT, and the primary winding N1 and the secondary winding N2 are 0.80 to 0.90. A degree of coupling coefficient is obtained.

  The secondary winding N2 of the insulating converter transformer PIT is provided with a center tap as shown in the figure and connected to the secondary side ground, and then from the rectifier diodes Do1 and Do2 and the smoothing capacitor Co as shown in the figure. A double-wave rectifier circuit is connected. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the drive circuit 21. As described above, the drive circuit 21 switches the switching frequency so that the secondary side DC output voltage Eo is stabilized based on the level of the input secondary side DC output voltage Eo. Elements Q1 and Q2 are driven. That is, stabilization by the switching frequency control method is performed.

  As can be seen from the above description, the power supply circuit shown in FIG. 14 is configured by mounting the conventionally known active filter shown in FIGS. 11 and 13. By adopting such a configuration, the power factor is improved. Moreover, it is so-called wide-range compatible that operates with a commercial AC power supply AC100V system and AC200V system.

However, the power supply circuit having the configuration shown in FIG. 14 has the following problems.
First, since the active filter circuit is a hard switching operation, the level of noise generation is very high, so that a relatively severe noise suppression measure is required.
For this reason, in the circuit shown in FIG. 14, a noise filter including two sets of line filter transformers LFT and three sets of across capacitors is formed for the line of the commercial AC power supply AC. That is, two or more stages of line noise filters are required.
Further, a normal mode noise filter including one set of choke coils LN and two sets of filter capacitors CN is provided for the rectified output line. In addition, an RC snubber circuit is provided for the fast recovery diode D10 for rectification.
Thus, an actual circuit requires countermeasures against noise due to an extremely large number of parts, resulting in a complicated circuit configuration and an increased cost, and an increase in the mounting area of the power circuit board.

In the power supply circuit shown in FIG. 14, the hard switching operation causes a relatively large switching loss in the switching element Q6 and a power loss in the diode D10. In this respect, the power conversion efficiency is effectively improved. There is also a problem of not.
Incidentally, as a result of actually conducting an experiment on the power supply circuit of FIG. 14, the following numerical values were obtained as the power conversion efficiency.
First, the AC → DC power conversion efficiency ηAC → DC in the active filter is about ηAC → DC = 92% when the AC input voltage VAC = 100V, and about ηAC → DC = 95% under the condition of the AC input voltage VAC = 230V. It became.
Here, the power conversion efficiency of the entire power supply circuit shown in FIG. 14 includes the AC → DC power conversion efficiency of the front-stage active filter and the DC → DC power conversion efficiency (ηDC → DC) of the current-resonant converter of the rear stage. It becomes the thing which integrated. In this case, the power conversion efficiency of the current resonance type converter when the average value of the DC input voltage Ei generated by the active filter is the load power Po = 200 W under the condition of Ei = 375 V described above is ηDC → DC is about 94%.
Therefore, the total efficiency of the circuit shown in FIG. 14 is about ηAC → DC = 86.5% when VAC = 100V. The overall efficiency at VAC = 230V is about ηAC → DC = 89.3%.

Further, the power supply circuit of FIG. 14 is provided with an inrush current limiting circuit 22 for suppressing an inrush current when the power supply is started by performing a hard switching operation. For example, in practice, in order to suppress the inrush current, an inrush current limiting resistor Ri as illustrated in FIG. 14 or a power thermistor or the like is inserted into the line of the commercial AC power supply AC. Has been.
The point that such an additional configuration for suppressing the inrush current is required also contributes to an increase in cost and an increase in circuit size.

Therefore, in the present invention, in view of the various problems described above, the switching power supply circuit is configured as follows.
First, the switching power supply circuit according to the present invention includes a current resonance type switching converter at the front stage and a DC / DC converter at the rear stage when viewed from the commercial AC power supply.
And as a switching converter of a front | former stage, it switches by inputting the rectification | straightening output voltage produced | generated by the rectification | straightening voltage which generate | occur | produces the rectification output voltage by inputting commercial alternating current power supply, and performing the rectification operation | movement A switching means formed with a switching element; and a switching driving means for switching the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. Insulating converter transformer formed.
Also, the switching means is formed of at least a leakage inductance component of the primary winding of the insulating converter transformer and a capacitance of a primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type. A primary side series resonance circuit is provided.
Further, a rectifying circuit other than the double-wave rectifying circuit is provided, a rectifying operation is performed on the alternating voltage obtained in the secondary winding, and a DC output voltage is generated by smoothing the rectified output with a smoothing capacitor. Output voltage generation means is provided.
Further, the constant voltage control for the DC output voltage is performed by controlling the switching drive means according to the average value of the DC output voltage and varying the switching frequency of the switching means. Voltage control means, and further, a gap length formed at least at a predetermined position of the core of the insulating converter transformer is set so that a coupling coefficient between the primary side and the secondary side of the insulating converter transformer is not more than a predetermined value. It is what has been.
In addition, a DC / DC converter that generates a secondary DC output voltage by performing DC / DC power conversion by inputting the DC output voltage generated in the switching converter having the above configuration as a DC power supply is provided. It is a thing.

In the present invention, the switching converter device is configured as follows.
That is, first, a rectifying unit that generates a rectified output voltage by performing a rectifying operation by inputting a commercial AC power supply, and a switching element that performs switching by inputting the rectified output voltage generated by the rectifying unit. And a switching driving unit configured to perform switching driving of the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. Insulating converter transformer formed.
Also, the switching means is formed of at least a leakage inductance component of the primary winding of the insulating converter transformer and a capacitance of a primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type. A primary side series resonance circuit is provided.
Further, a rectifying circuit other than the double-wave rectifying circuit is provided, a rectifying operation is performed on the alternating voltage obtained in the secondary winding, and a DC output voltage is generated by smoothing the rectified output with a smoothing capacitor. Output voltage generation means is provided.
Further, the constant voltage control for the DC output voltage is performed by controlling the switching drive means according to the average value of the DC output voltage and varying the switching frequency of the switching means. Voltage control means, and further, a gap length formed at least at a predetermined position of the core of the insulating converter transformer is set so that a coupling coefficient between the primary side and the secondary side of the insulating converter transformer is not more than a predetermined value. It is what has been.

According to the said structure, the structure which performs switching operation | movement about the rectified output voltage of commercial AC power supply is taken in the said switching converter. Further, with respect to the DC output voltage generated by such a switching operation, constant voltage control is performed with respect to the average value.
With such a configuration, the power factor can be improved by the switching converter of the present invention.
In the present invention, the rectifier circuit other than the double-wave rectifier circuit is provided as the secondary-side rectifier circuit. The capacitance can be eliminated, and the parallel resonance operation can be prevented from being performed on the secondary side. In such a configuration, the switching frequency required for stabilizing the secondary side DC output voltage is reduced by reducing the coupling coefficient between the primary side and the secondary side of the insulating converter transformer to a predetermined value or less. It is possible to effectively reduce the variable control range (necessary control range).

As described above, according to the present invention, the variable control range (necessary control range) of the switching frequency necessary for the constant voltage control can be effectively reduced, so that the wide range is supported only by the stabilizing operation by the switching frequency control method. The power supply circuit can be realized.
Further, the power factor can be improved by performing the switching operation on the rectified output voltage of the commercial AC power supply as described above and performing the constant voltage control on the average value of the DC output voltage.
That is, according to the present invention having such a configuration, it is possible to omit an active filter that has been conventionally required for realizing a wide-range power supply circuit that improves power factor.

Since the active filter can be omitted in this way, a reduction in power conversion efficiency can be suppressed as a configuration for achieving the same power factor improvement and wide range compatibility.
In addition, according to the present invention, the current resonance type converter as the above-described switching converter can perform a soft switching operation, and thus switching noise is greatly reduced. No need to strengthen. That is, the number of parts is greatly reduced in this respect, and the power circuit size can be reduced / lightened. In addition, the cost can be reduced accordingly.

Further, according to the configuration of the switching converter in the present invention, unlike the conventional active filter, the primary side and the secondary side can be insulated by providing the insulating converter transformer.
According to this, in the switching power supply circuit of the present invention in which the DC / DC converter is provided in the subsequent stage of the switching converter, the DC / DC converter in the subsequent stage does not need to adopt a configuration for insulation. For example, the converter transformer included in the DC / DC converter can be downsized by eliminating the need to provide an insulation distance between the primary side winding and the secondary side winding for insulation. In addition, the control circuit system that requires feedback from the secondary side to the primary side of the converter transformer can eliminate the need for a photocoupler for insulation, thereby reducing circuit components and circuit manufacturing costs. In addition, the circuit can be reduced in size.

Further, according to the configuration of the switching converter device of the present invention, it is possible to configure a power supply circuit alone without providing a DC / DC converter in the subsequent stage. And if a power supply circuit can be comprised with such a switching converter apparatus single-piece | unit, it will become possible to implement | achieve a power factor improvement and a wide range correspondence by a very simple structure.
For example, in comparison with a conventional current resonance type converter, an additional component necessary for power factor improvement and wide range correspondence can be, for example, only a phase compensation capacitor for obtaining an average value of DC output voltage.

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as the best mode for carrying out the present invention (hereinafter also referred to as an embodiment).
As a basic configuration of the power supply circuit shown in this figure, the portion corresponding to the active filter in the circuit shown in FIG. 14 is configured by a current resonance type converter. In addition, a DC output voltage (DC input voltage Ei) generated by the current resonance converter based on the input of the commercial AC power supply AC is further supplied to the subsequent stage of the current resonance converter (switching converter device). By providing the DC / DC converter 5 that operates as described above, a configuration is adopted in which a secondary side direct-current output voltage (Eo) for a load connected to the power supply circuit is output.

In FIG. 1, as a current resonance type converter (switching converter device) that generates a DC input voltage Ei for the DC / DC converter 5 in the subsequent stage based on the input of the commercial AC power supply AC as described above, a half-bridge coupling method is used. A configuration in which a partial voltage resonance circuit is combined with a separately excited current resonance converter is adopted.
In this current resonance type converter, first, a common mode noise filter is formed for the commercial AC power supply AC by filter capacitors CL and CL and a common mode choke coil CMC.
A full-wave rectifier circuit using a bridge rectifier circuit Di is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter.
The full-wave rectification circuit receives the commercial AC power supply AC and performs full-wave rectification operation, whereby the rectified output voltage V1 shown in the figure is obtained.
In this case, a filter capacitor CN is connected in parallel to the bridge rectifier circuit Di as the full-wave rectifier circuit. By connecting the filter capacitor CN, normal mode noise generated in the line of the commercial AC power supply AC is suppressed.

The current resonance type converter in this case is also provided with a switching circuit in which two switching elements Q1, Q2 by MOS-FETs are connected by half bridge coupling as shown in the figure.
However, unlike the switching circuit provided in the circuit of FIG. 14, the switching circuit in this case is operated by directly inputting the rectified output voltage V1 from the full-wave rectifier circuit. That is, in this case, as can be understood from the above configuration, since the smoothing capacitor for smoothing the rectified output voltage V1 by the bridge rectifier circuit is not provided, the bridge rectifier circuit Di is used as the switching circuit. The rectified output voltage V1 obtained at the rectified output point is switched.

  Even in this case, damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A primary side partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching elements Q1, Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. In this case, for example, a general-purpose IC can be used. The oscillation circuit of the oscillation / drive circuit 2 generates an oscillation signal having a required frequency, and the drive circuit generates a switching drive signal that is a gate voltage for switching the MOS-FET by using the oscillation signal. The voltage is applied to the gates of the switching elements Q1, Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be continuously turned on / off at alternate timings according to the switching frequency according to the cycle of the switching drive signal.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series connection of the primary side series resonant capacitor C1. As a result, the switching output is transmitted. The other end of the primary winding N1 is connected to the primary side ground.

Here, the insulating converter transformer PIT has a structure as shown in the sectional view of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with resin etc. is provided by the shape divided | segmented so that it might mutually become independent about the primary side and the secondary side winding part. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding (N1) and the secondary side winding (N2) are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side Due to the different winding regions, the windings are wound around the inner magnetic legs of the EE type core. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G is formed as shown in the figure for the inner magnetic leg of the EE type core. In this case, as the gap G, for example, a gap length of about 2.0 mm or more is set, and as the coupling coefficient k between the primary side and the secondary side, for example, a loosely coupled state with k = 0.70 or less is obtained. ing. As an actual coupling coefficient k, a gap length of about 2.8 mm was set and k = 0.65 was set. The gap G can be formed by making the inner magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

Incidentally, in the power supply circuit having the conventional current resonance type converter including the power supply circuit shown in FIG. 14, the gap formed in the core of the insulating converter transformer PIT is, for example, 1.0 mm as described above. By setting the degree, k = 0.8 to 0.9 was obtained as the coupling coefficient k.
In other words, in the present embodiment, the degree of coupling between the primary side and the secondary side of the insulating converter transformer PIT is set lower than that of the conventional configuration.

Returning to FIG.
The insulating converter transformer PIT generates a predetermined leakage inductance L1 in the primary winding N1 with the structure described with reference to FIG. Further, as described above, the primary winding N1 and the primary side series resonance capacitor C1 are connected in series. Accordingly, a series resonance circuit (primary side series resonance circuit) is formed by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1.
In addition, the primary side series resonance circuit is connected to the switching output points of the switching elements Q1 and Q2, and therefore the switching outputs of the switching elements Q1 and Q2 are transmitted to the primary side series resonance circuit. Become. In the primary side series resonance circuit, the resonance operation is performed by the transmitted switching output, so that the operation of the primary side switching converter is a current resonance type.

By the way, according to the description so far, the primary side switching converter shown in this figure has an operation as a current resonance type by the primary side series resonance circuit (L1-C1) and the above-described primary side partial voltage resonance circuit (Cp / / L1) and partial voltage resonance operation.
That is, on the primary side of the power supply circuit shown in this figure, a configuration in which the resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit is adopted. Here, the switching converter in which two resonance circuits are combined in this way is also referred to as a “composite resonance converter”.

An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited (induced) in the secondary winding N2 of the insulating converter transformer PIT.
In this case, a full-wave rectifying / smoothing circuit including a bridge rectifier circuit formed by connecting rectifier diodes D1 to D4 as illustrated and a set of smoothing capacitors Ci is provided for the secondary winding N2.
In this full-wave rectifying / smoothing circuit, in one half cycle of the alternating voltage excited by the secondary winding N2, the pair of rectifier diodes [D1, D4] of the bridge rectifier circuit is turned on, and is connected to the smoothing capacitor Ci. The operation of charging the rectified current is obtained. Further, in the other half cycle of the alternating voltage excited in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Ci by conducting the pair of rectifier diodes [D2, D3] is obtained.
As a result, as the voltage across the smoothing capacitor Ci, a level corresponding to the same multiple of the alternating voltage level excited in the secondary winding N2 is obtained.

The both-end voltage obtained in the smoothing capacitor Ci in this way is supplied as a DC input power source for the DC / DC converter 5 at the subsequent stage as a DC input voltage Ei shown in the figure.
The DC / DC converter 5 is configured to perform a switching operation on the DC input voltage Ei and generate a secondary DC output voltage Eo to be supplied to the load. In this case, the DC / DC converter 5 is also configured as a current resonance type converter, for example. Further, as the secondary side DC output voltage Eo, it is assumed that, for example, three systems of Eo1, Eo2, and Eo3 illustrated are output.

The DC input voltage Ei is also branched and input as a detection voltage for the control circuit 1 as shown.
The control circuit 1 is provided to stabilize the DC input voltage Ei by a switching frequency control method, and as shown in the figure, a shunt regulator Q5, detection resistors R10 and R11, a resistor R9, a phase compensation capacitor Ct1, a phase compensation capacitor. Ct2, resistor R12, and photocoupler PC are provided at least.
In the control circuit 1, the level of the DC input voltage Ei is detected at the connection point of the detection resistors R10-R11 inserted in series between the positive terminal of the smoothing capacitor Ci and the secondary side ground. A control signal corresponding to the detected voltage is supplied to the control terminal of the shunt regulator Q5.
The shunt regulator Q5 has an anode connected to the secondary side ground and a cathode connected to the cathode of the photodiode in the photocoupler PC. Further, in this case, the anode of the photodiode is connected to the connection point between the positive terminal of the smoothing capacitor Ci and the resistor R10 through a series connection of the resistor R9.

  According to the configuration described so far, a control current corresponding to the level of the DC input voltage Ei detected and input to the control terminal of the shunt regulator Q5 flows through the photodiode connected to the shunt regulator Q5. Then, this control current is transmitted to the primary side phototransistor and supplied to the oscillation / drive circuit 2 as shown.

In the control circuit 1 in this case, the phase compensation capacitor Ct2 is connected in parallel between the control terminal and the cathode of the shunt regulator Q5. For convenience of explanation, this phase compensation capacitor Ct2 is provided. The operations performed will be described later.
In this case, a series connection circuit including a phase compensation capacitor Ct1 and a resistance R12 is provided in parallel to the detection resistor R10. Depending on the series connection circuit, the phase delay of the error amplifier as the control circuit 1 is provided. A compensation circuit is formed.

In the oscillation / drive circuit 2, the switching elements Q1 and Q2 are driven and controlled such that the switching frequency is varied according to the level of the control current supplied from the control circuit 1 side as described above. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied.
By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit changes, and the amount of electric power transmitted from the primary winding N1 to the secondary winding N2 side of the insulating converter transformer PIT is reduced. Although it changes, this operates so as to stabilize the level of the DC input voltage Ei.

Although details will be described later, as a switching frequency control method in the power supply circuit of the present embodiment, a frequency range higher than the resonance frequency fo of the primary side series resonance circuit is set as a variable range of the switching frequency. That is, a so-called upper side control method is adopted.
As a general matter, a series resonance circuit has the lowest resonance impedance at the resonance frequency. Therefore, when the upper side control method based on the resonance frequency of the series resonance circuit is employed as in the present embodiment, the resonance impedance is increased as the switching frequency fs increases. Become.
Therefore, for example, when the DC input voltage Ei decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. This lowers the resonance impedance and increases the amount of power transmission from the primary side to the secondary side, and thus the DC input voltage Ei increases.
On the other hand, when the DC input voltage Ei increases due to a light load tendency, the switching frequency is controlled to be higher. As a result, the resonance impedance is increased and the power transmission amount is reduced, so that the DC input voltage Ei is lowered. In this way, the DC input voltage Ei is stabilized by changing the switching frequency.

FIG. 3 shows the operation waveform of the main part of the current resonance converter in the previous stage in the power supply circuit shown in FIG.
Here, the power supply circuit shown in FIG. 1 adopts a configuration corresponding to a wide range as will be described later. Correspondingly, FIG. 3 shows operation waveforms when the AC input voltage VAC = 100 V and the AC input voltage VAC = 230 V, respectively.

In order to obtain the experimental results shown in this figure, each part of the circuit of FIG. 1 was selected as follows.
Insulating converter transformer PIT: Gap G = 2.8 mm, primary winding N1 = 25T, secondary winding N2 = 25T, coupling coefficient k = 0.65,
・ Primary side series resonance capacitor C1 = 0.056μF
・ Filter capacitor CN = 1μF
・ Smoothing capacitor Ci = 1000μF
・ Phase compensation capacitor Ct1 = 0.047μF, resistance R12 = 3.3kΩ
・ Phase compensation capacitor Ct2 = 3.3μF
In addition, the induced voltage level per turn of the secondary winding N2 was set to 5 V / T or more by the selection of each part.
Further, the capacitance of the above-described phase compensation capacitor Ct1 is increased compared to about Ct1 = 1000 pF in the conventional configuration (configuration without the phase compensation capacitor Ct2). This is a result of adjusting the phase delay compensation of the control circuit 1 corresponding to the case where the phase compensation capacitor Ct2 is provided.

First, as shown in FIG. 3, as the rectified output voltage V1 of the bridge rectifier circuit Di obtained at both ends of the filter capacitor CN shown in FIG. Note that, when the AC input voltage VAC = 230 V, a DC component of 100 V is superimposed on the rectified output voltage V1 as shown in the figure, which is a charge charge of the filter capacitor CN.
The peak level of the rectified output voltage V1 is 140Vp when the AC input voltage VAC = 100V, and 325Vp when the AC input voltage VAC = 230V.

Further, the primary side series resonance current Io flowing through the primary side series resonance circuit has a waveform according to the switching period of the switching elements Q1 and Q2, and the envelope corresponds to the peak level of the rectified output voltage V1. Thus, a waveform having a peak in both positive and negative polarities is obtained.
In this case, the peak level (absolute value) of the primary side series resonance current Io is 15 Ap for both the AC input voltage VAC = 100V and 230V.

Further, the alternating voltage V2 excited by the secondary winding N2 of the insulating converter transformer PIT is also a waveform with a switching cycle, and its envelope changes to both positive and negative polarities according to the peak level of the rectified output voltage V1. It becomes a peak.
The peak level (absolute value) of the alternating voltage V2 is clamped at the level of the DC input voltage Ei.

Then, the DC input voltage Ei has a waveform in which a ripple voltage ΔEi having a cycle approximately twice that of the commercial AC power supply AC is superimposed as shown in the figure.
As can be understood from the description of FIG. 1, in the current resonance type converter according to the embodiment, the switching elements Q1 and Q2 perform the switching operation with respect to the rectified output voltage V1 of the bridge rectifier circuit Di that rectifies the commercial AC power supply AC. By being configured to do.
According to the actual measurement result, ΔEi when the AC input voltage VAC = 100 V is about 5 Vp-p, and ΔEi when the AC input voltage VAC = 230 V is about 10 Vp-p.

Here, in the circuit of FIG. 1, the switching elements Q1 and Q2 perform the switching operation with respect to the rectified output voltage V1 as described above. As shown in FIG. 3, as the primary side series resonance current Io, the conduction period of the average waveform (envelope) is obtained over almost a half cycle of the commercial AC power supply AC.
According to the configuration of FIG. 1, the conduction period of the AC input current IAC is equivalent to the conduction period of such a primary side series resonance current Io.
That is, according to this, since the conduction period of the primary series resonance current Io extends over a half cycle of the commercial AC power supply AC as described above, the commercial AC power supply can be used as the conduction period of the AC input current IAC. It will be obtained over approximately half the AC period.
In this way, the conduction period of the AC input current IAC is made closer to the half cycle of the commercial AC power supply AC (AC input voltage VAC), which means that the conduction angle of the AC input current IAC is expanded. It is intended.

Further, as shown in FIG. 3, the DC input voltage Ei in this case actually has a ripple corresponding to the commercial AC power supply cycle.
According to this, depending on the operation of the constant voltage control system by the control circuit 1 and the oscillation / drive circuit 2 as described with reference to FIG. The switching frequency control that follows this is also performed. That is, the level of the primary side series resonance current Io is controlled so as to cancel such a component of the ripple ΔEi.

If the level of the primary side series resonance current Io is controlled so as to cancel the ripple voltage ΔEi in this way, the envelope of the primary side series resonance current Io, which is sinusoidal in FIG. A waveform whose absolute value level increases or decreases in a corresponding cycle is obtained.
Depending on this, the envelope of the primary side series resonance current Io does not have a sinusoidal shape unlike the one shown in FIG. 3, and a higher frequency component (a component with a period twice that of the commercial AC power supply AC corresponding to ΔEi). ) Is superimposed, and even if the conduction angle of the AC input current IAC is increased as described above, the power factor itself is improved by the amount of harmonics superimposed. Will not be able to be planned.

Therefore, in the embodiment, in the control circuit 1, the phase compensation capacitor Ct2 is added according to the connection form described above.
By adding the phase compensation capacitor Ct2, the control circuit 1 forms a low-pass filter for the detection input corresponding to the DC input voltage Ei, thereby extracting a low-frequency component of the DC input voltage Ei and generating a ripple component. Removed. That is, as a result, a control signal having a level corresponding to the average value Ei-avr obtained by averaging the DC input voltage Ei is supplied to the control terminal of the shunt regulator Q5.
The control signal corresponding to the average value Ei-avr of the DC input voltage Ei is supplied to the control terminal of the shunt regulator Q5 in this way, and this constant is controlled in the constant voltage control system including the control circuit 1 in this case. An operation of stabilizing the value Ei-avr is performed.

If the average value Ei-avr of the DC input voltage Ei is stabilized in this way, the envelope of the waveform of the primary side series resonance current Io is simply the rectified output voltage V1 as shown in FIG. A waveform in which the positive / negative absolute value level increases in accordance with the increase is obtained. That is, in this case, the switching frequency control that follows the component of the ripple voltage ΔEi is not performed, so that the waveform envelope of the primary side series resonance current Io has a sine wave shape corresponding to the waveform of the rectified output voltage V1. It can be a waveform.
Then, the envelope of the waveform of the primary side series resonance current Io is made sinusoidal in this way, so that the waveform of the AC input current IAC can be made sinusoidal as well.

Thus, according to the configuration of FIG. 1, the switching elements Q1 and Q2 perform the switching operation with respect to the rectified output voltage V1, so that the conduction angle of the AC input current IAC can be increased, and Since the DC input voltage Ei is stabilized at the average value Ei-avr, the waveform of the AC input current IAC can be made closer to a sine wave.
That is, the conduction angle of the AC input current IAC can be increased in this way, and the power factor can be improved by making the waveform close to a sine wave.

  As understood from the above-described operation, the ripple voltage ΔEi of the DC input voltage Ei detected and input is removed as a low-pass filter by the phase compensation capacitor Ct2 added in the control circuit 1 in this case. Thus, it is sufficient that the filter coefficient is set.

By the way, the current resonance type converter in the power supply circuit of FIG. 1 adopts a configuration corresponding to a wide range capable of operating in accordance with both input of the AC100V system and AC200V system in addition to the power factor improving operation described above. It is.
Hereinafter, the operation corresponding to the wide range by the circuit configuration of FIG. 1 will be described.

First, as understood from the above description, the current resonance type converter shown in FIG. 1 can stabilize the output voltage on the secondary side (in this case, the DC input voltage Ei) by the switching frequency control method. A configuration as a resonance type converter is realized.
Here, when the secondary side DC output voltage is stabilized by the switching frequency control method in this way, conventionally, there is generally a variable control range of the switching frequency for stabilization. It tends to be a relatively wide range. Since the variable control range required for stabilization of the secondary side DC output voltage is relatively wide as described above, the AC 100 V system and the AC 200 V system can be conventionally operated only by the stabilization operation by the switching frequency control method. It has been very difficult to realize a wide-range configuration that supports input.

FIG. 15 shows the constant voltage control characteristic in the conventional current resonance type converter by the relationship between the switching frequency fs and the level of the secondary side DC output voltage Eo.
As a conventional current resonance type converter, the power supply circuit shown in FIG. 14 may be considered as a configuration in which the previous active filter is omitted. That is, it is a current resonance type converter in which the gap length of the insulating converter transformer PIT is set to about 1.0 mm, for example, and the coupling coefficient k is set to about 0.8 to 0.9.

In FIG. 15, first, as a general matter, the series resonance circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs in the upper side control, the level of the secondary side DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo. As it gets away from fo, it goes down.
Therefore, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs under the condition that the load power Po is constant, the switching frequency fs is equal to the resonance frequency fo of the primary side series resonance circuit as shown in FIG. It becomes a peak at the same time, and shows a change in a quadratic curve that decreases as the frequency moves away from the resonance frequency fo.

  Further, a characteristic is obtained that the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs is shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  Under such characteristics, when the secondary side DC output voltage Eo is stabilized by the upper side control so that Eo = tg, the switching frequency required in the conventional power supply circuit is reduced. The variable range (necessary control range) is a range indicated as Δfs.

Here, assuming that it corresponds to a single range of AC 100 V system, for example, the fluctuation range of AC input voltage VAC = 85 V to 120 V, and further, from the maximum load power Pomax of the secondary side DC output voltage Eo which is the main DC power supply, to the minimum load power Assume that the load fluctuation range up to Pomin is, for example, Po = 200 W to 0 W.
In this case, the variable range of the switching frequency fs that can be varied for constant voltage control of the secondary side DC output voltage Eo in the conventional current resonance type converter is fs = 80 kHz to 200 kHz or more, and Δfs is correspondingly 120 kHz or more. It will be a wide range.

In consideration of this, when considering the realization of the wide range only by the switching frequency control, first, the wide range corresponds to the AC input voltage range of, for example, AC85V to 288V. Accordingly, the level fluctuation range of the secondary side DC output voltage Eo in the case of corresponding to the same load fluctuation range as compared with the case of corresponding to the single AC100V system only or the AC200V only single range as described above is naturally. It tends to expand.
In order to perform constant voltage control on the secondary side DC output voltage Eo whose level fluctuation range tends to expand corresponding to such an AC input voltage range, a wider switching frequency control range is required. For example, when the required control range Δfs (about fs = 80 kHz to 200 kHz) in the AC100V single range described above is set, the control range of the switching frequency fs for wide range correspondence is expanded to about 80 kHz to 500 kHz. Necessity comes out.

However, as described above, the upper limit of the drive frequency that can be handled as an IC (oscillation / drive circuit 2) for driving a current switching element is limited to about 200 kHz. Even if the switching drive IC capable of driving at a high frequency as described above is configured and mounted, the power conversion efficiency is significantly reduced when the switching element is driven at such a high frequency. Therefore, it is not practical as an actual power supply circuit.
Under such circumstances, it has been very difficult to achieve a wide range only by a stabilizing operation by the switching frequency control method.

Therefore, in the power supply circuit shown in FIG. 1, the coupling coefficient k in the insulating converter transformer PIT is set to a value lower than that of the conventional converter as described in FIG. .
With such a configuration, a wide-range configuration that can operate in response to the input of the AC 100 V system and the AC 200 V system is realized. This will be described with reference to FIG.

FIG. 4 shows the constant voltage control characteristics in the previous stage current resonance type converter in the power supply circuit of FIG. 1 by the relationship between the switching frequency fs and the average value Ei-avr of the DC input voltage Ei.
In this figure, at the same time, the constant voltage control characteristic for the secondary side DC output voltage Eo in the power supply circuit when the coupling coefficient k in the insulating converter transformer PIT is set to the conventional setting is shown by a one-dot chain line.
For confirmation, it is assumed that so-called upper side control is adopted as the switching frequency control method in the current resonance converter in the previous stage in the power supply circuit of FIG.

In FIG. 4, even in this case, since the resonance impedance becomes the smallest at the resonance frequency fo in the series resonance circuit, the relationship between the DC input voltage Ei and the switching frequency fs in the upper side control is that the switching frequency fs is The average value Ei-avr of the DC input voltage Ei increases as it approaches the resonance frequency fo, and the average value Ei-avr decreases as the distance from the resonance frequency fo increases.
Accordingly, in this case as well, the average value Ei-avr of the DC input voltage Ei with respect to the switching frequency fs under the condition that the load power Po is constant is, as shown in the figure, the switching frequency fs is the resonance frequency fo of the primary side series resonance circuit. It becomes a peak at the same time, and shows a quadratic curve-like change that decreases as the distance from the resonance frequency fo increases.

  In this case, the average value Ei-avr corresponding to the same switching frequency fs also has a characteristic of shifting so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. Therefore, even in this case, if the switching frequency fs is fixed, the average value Ei-avr tends to decrease as the load becomes heavy.

  Here, the characteristic curve (characteristic curve 1) at the time of the maximum load power Pomax (Po = 200 W) in the case of the circuit of FIG. 1 shown in this figure and the characteristic curve (characteristic curve 2) at the time of Pomax in the conventional circuit. As can be seen from comparison with (2), approximately the same relatively steep quadratic curve is obtained as both characteristic curves at the maximum load power. On the other hand, when the minimum load power Pomin (Po = 0W) is compared, the curve of the circuit of FIG. 1 shown as the characteristic curve 3 and the conventional curve shown as the characteristic curve 4 are very gentle in the prior art. It can be understood that a steep characteristic is obtained in the circuit of FIG.

Then, when such characteristics are obtained, when trying to stabilize the average value Ei-avr by upper side control so that Ei-avr = tg, stabilization is achieved in the conventional power supply circuit. The required switching frequency variable range (necessary control range Δfs) is a range indicated by a relatively wide ΔfsB as shown in the figure because the characteristic curve at the minimum load power Pomin is gentle as described above. It becomes.
On the other hand, the necessary control range Δfs for stabilizing Ei-avr in the power supply circuit shown in FIG. 1 is more steep than the above ΔfsB because the characteristic curve at the minimum load power Pomin is steeper than before. This is the range indicated as the reduced ΔfsA.

In this way, in the embodiment in which the coupling coefficient k is set to a predetermined value lower than the conventional value, the increase in the switching frequency especially at the minimum load power Pomin is suppressed, so that it is necessary for stabilization. The control range Δfs is greatly reduced.
Here, in the characteristic diagram of FIG. 4, the required control range Δfs in the case of one single range of the AC 100V system or the 200V system is shown, but in the other range, the required control range Δfs is also significantly increased. Reduction will be achieved. That is, similarly, the increase of the switching frequency at the time of the minimum load power Pomin is particularly suppressed, so that the required control range Δfs can be greatly reduced in comparison with the conventional one in both the AC100V system and the AC200V system. It is.
The reduction of the required control range Δfs in each single range in this way means that the required control range Δfs is also larger than the conventional control range Δfs even when it corresponds to the input from the AC 100V system to the AC 200V system. It can be greatly reduced.

  In this way, the required control range Δfs when the input from the AC 100V system to the AC 200V system is correspondingly reduced, so that the current resonance converter in the previous stage in the power supply circuit of FIG. A configuration corresponding to a wide range can be realized only by the stabilization operation by the frequency control method.

  By the way, as a current resonance type converter that stabilizes the DC output voltage on the secondary side by such a switching frequency control method, for example, in the current resonance type converter in the circuit of FIG. However, when the secondary-side rectifier circuit is a double-wave rectifier, the switching frequency control range is particularly wide.

First, in the case of a double-wave rectifier circuit, the secondary winding N2 is center-tapped to form two secondary winding portions. In these two secondary winding sections, in one half cycle of the alternating voltage excited by the secondary winding N2, the rectified current is [one secondary winding section → rectifier diode Do1 → smoothing capacitor Co. ] Flow. Further, in the other half cycle of the alternating voltage, the rectified current flows through [the other secondary winding portion → rectifier diode Do2 → smoothing capacitor Co].
That is, in the two-wave rectification, the two secondary winding portions are in a state in which current flows only in one half cycle and does not flow in the other half cycle.

According to such a double-wave rectification operation, a required capacitance exists between the two secondary winding portions wound around the bobbin of the insulating converter transformer PIT.
Since the inter-line capacitance is present in this way, the secondary side of the insulating converter transformer PIT in this case is equivalently the secondary as shown in FIG. The capacitor C2 is connected in parallel to the winding N2.

Since the capacitor C2 is connected in parallel to the secondary winding N2, in this case also on the secondary side, the parallel is caused by the leakage inductance (L2) of the secondary winding N2 and the capacitance (C2) of the capacitor C2. Thus, a resonance circuit (partial resonance circuit) is formed, and the resonance operation similar to that of the primary side partial resonance circuit (L1 // Cp) can be obtained on the secondary side.
Incidentally, the capacitance of the capacitor C2 is determined by the number of litz wire bundles used as the secondary winding N2 and the window area of the bobbin around which the secondary winding N2 is wound. As a result of experiments on the circuit, it is very small, about 100 pF to 500 pF.

Thus, in the current resonance type converter in which the secondary side rectifier circuit is a double-wave rectifier circuit and a parallel resonant circuit is formed also on the secondary side, the secondary side DC output as shown in FIG. As the constant voltage characteristic of the voltage Eo, the characteristic shown in FIG.
In FIG. 16, first, a parallel resonant circuit is also formed on the secondary side as described above. When the resonant frequency of the primary side series resonant circuit is fo1, the secondary side parallel resonant circuit is A resonance frequency fo2 exists.
In addition, since there are two different resonance points in this way, the characteristic curve particularly at Pomin corresponds to the peak and secondary resonance frequency fo2 according to the primary resonance frequency fo1. A bimodal curve having two peaks and a peak as shown in the figure is obtained.

  In this case, when the capacitance of the capacitor C2 is relatively small as described above, the level of the secondary side DC output voltage Eo tends to be relatively low under heavy load conditions. The resonance point on the secondary side does not become apparent (characteristic curve at Pomax). However, if the secondary-side DC output voltage Eo tends to rise sharply by approaching a no-load state due to a light load trend, does the secondary-side resonance point become obvious? As shown, a bimodal characteristic curve such as a characteristic curve at Po = 0 in the figure can be obtained.

When comparing the characteristic curve of this bimodal with the characteristic curve at the same Po = 0 W in FIG. 15, the bimodal curve shown in FIG. 16 has no load compared to the case where it is a single-peaked curve. It can be seen that the switching frequency at the time tends to be higher.
And, according to this, as can be seen by comparing Δfs in each figure, the required control range Δfs of the switching frequency becomes wider in the bimodal in FIG.

FIG. 17 is a diagram showing an example of fluctuation characteristics of the switching frequency fs with respect to load fluctuations in a conventional current resonance type converter when the secondary side rectifier circuit is a double-wave rectifier circuit.
From this characteristic diagram, as described above, when the dual-wave rectifier circuit is used, when the load power Po is around 0 W, the secondary resonance point becomes obvious and the switching frequency tends to increase rapidly. Become.
According to this characteristic diagram, when Pomax = 200 W, the switching frequency fs is about 75 to 80 kHz, whereas in the vicinity of Pomin = 0 W, the switching frequency fs rapidly increases to about 170 kHz.

  In this way, when a double-wave rectifier circuit is configured on the secondary side as a configuration of a conventional current resonance type converter, the necessary control due to the presence of two resonance points by the resonance circuit on the primary side and the secondary side In addition to the expansion of the range Δfs, the necessary control range Δfs tends to expand.

Therefore, as shown in FIG. 1, as an embodiment, the secondary side rectifier circuit includes a bridge rectifier circuit, and is a rectifier circuit other than the double-wave rectifier circuit.
Here, for example, under the setting of the coupling coefficient k of the insulating converter transformer PIT described with reference to FIG. 2, a double-wave rectifier circuit is provided as a secondary side rectifier circuit as in the case of the circuit of FIG. Let's consider the case.
First, since it is a double-wave rectifier circuit, even in this case, there is no change in the capacitance between the secondary windings divided by the center tap. A parallel resonant circuit is equivalently formed on the side, and as a result, even in this case, the characteristic curve for the constant voltage control, particularly when the load power Po = 0 W, is the bimodal curve shown in FIG. Will be obtained.
In this way, the characteristic curve itself is not a single peak but a double peak characteristic. Therefore, in the case of a double-wave rectifier circuit, the required control range of the switching frequency is also set by the setting of the coupling coefficient k of the embodiment. Δfs cannot be reduced. This is because the characteristic curve is a double peak, and as a switching frequency in the vicinity of no load, the characteristic of rapidly increasing as shown in FIG. 17 is maintained. As a result, it is almost impossible to effectively reduce the necessary control range Δfs, and as a result, the necessary control range Δfs does not change even if the coupling coefficient k is set to a predetermined value or less.

On the other hand, if a parallel resonant circuit is not formed on the secondary side and the characteristic curve of constant voltage control is a single peak, as can be understood from the characteristic diagram of FIG. 4, by setting the coupling coefficient k, The switching frequency at the time of no load (at the time of Pomin) can be set to a lower value than when the conventional coupling coefficient is set, so that the rapid increase characteristic of the switching frequency near the no load is effectively improved. It is done.
For this reason, according to the embodiment in which the rectifier circuit on the secondary side is other than the double-wave rectifier circuit and the coupling coefficient k is reduced to a predetermined value or less, the necessary control range Δfs can be effectively reduced. Thus, it is possible to realize a configuration corresponding to a wide range only by a stabilizing operation by the switching frequency control method.

FIGS. 5 and 6 show the results of actual experiments on the power supply circuit shown in FIG. 1, including power factor PF, AC → DC power conversion efficiency ηAC → DC, and average value Ei-avr of DC input voltage Ei. Each characteristic is shown.
The AC → DC power conversion efficiency shown in each figure is the total power conversion based on the AC → DC power conversion efficiency of the current-resonance converter in the previous stage and the DC → DC power conversion efficiency of the DC / DC converter 5 in the subsequent stage. It shows the efficiency.
Further, FIG. 5 shows each characteristic with respect to fluctuations in the load power Po = 200 W to 0 W, and FIG. 6 shows each characteristic with respect to fluctuations in the AC input voltage VAC = 85 V to 288 V.
Further, in FIG. 5, the characteristics at the AC input voltage VAC = 100V are indicated by a solid line, the characteristics at the AC input voltage VAC = 230V are indicated by a broken line, and FIG. 6 shows the experimental results when the load power Po = 200 W is constant. Show.
In obtaining the experimental results shown in these figures, each part of the power supply circuit in FIG. 1 was set to the values described in FIG.

First, in FIG. 5, the AC → DC power conversion efficiency has a characteristic that increases as the load power Po increases at both the AC input voltage VAC = 100V and 230V. When the AC input voltage VAC = 100 V, the maximum value was obtained when the load power Po = 200 W, and about ηAC → DC = 90.2% was obtained.
Further, even when the AC input voltage VAC = 230 V, the maximum value was obtained when the load power Po = 200 W, and the result was about ηAC → DC = 89.5%.
Note that even when the overall power conversion efficiency ηAC → DC shown in this figure is used, the power conversion efficiency ηDC → DC in the subsequent DC / DC converter 5 is changed to ηDC → DC = as in the case of the circuit of FIG. The result in the case of 94% is shown.

Here, according to the configuration of the power supply circuit shown in FIG. 1, by setting the level of the DC input voltage Ei (average value Ei-avr) generated by the current resonance converter in the previous stage high, the DC / The DC → DC power conversion efficiency in the DC converter 5 is improved.
In this case, the average value Ei-avr of the DC input voltage Ei is set to about Ei-avr = 420 V, so that the DC → DC power conversion efficiency (ηDC → DC = 94) in the DC / DC converter 5 as described above. %)It was set. The generation level of the DC input voltage Ei can be set by the winding ratio of the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT according to the circuit configuration of FIG. That is, in this case, the above-described Ei-avr = 420V is set by setting the primary winding N1 = secondary winding N2 = 25T shown in FIG.
According to the experiment, the AC → DC power conversion efficiency in the previous stage current resonance type converter shown in FIG. 1 in the case where such Ei-avr = 420 V is set is as follows: load power Po = 200 W The results were as follows: when AC input voltage VAC = 100 V, ηAC → DC = about 96.0%, and when AC input voltage VAC = 230 V, ηAC → DC = about 95.2%.
This is the AC → DC power conversion efficiency in the active filter shown in FIG. 14, compared with ηAC → DC = 92% at AC input voltage VAC = 100V and ηAC → DC = 95% at 230V. It is an improvement.

Further, as shown in FIG. 5, the power factor PF tends to increase at both the AC input voltage VAC = 100V and 230V as the load power Po increases. According to the experiment, when the load power Po = 200 W, the power factor PF = 0.99 is obtained at the AC input voltage VAC = 100 V, and the power factor PF = 0.83 is obtained at the AC input voltage VAC = 230 V.
Further, as shown in FIG. 6, when the load power Po = 200 W, the power factor PF with respect to the fluctuation of the AC input voltage VAC = 85V to 288V is about 0.8 or more, and this result also shows that FIG. It can be understood that a practically sufficient power factor is obtained in the power supply circuit.

  Further, the average value Ei-avr is controlled to be constant with respect to fluctuations in the load power Po as shown in FIG. And with respect to the fluctuation | variation of alternating current input voltage VAC, as shown in FIG. 6, the alternating current input voltage VAC becomes substantially constant in the range of 85V to 288V.

From the description so far, according to the configuration of the current resonance type converter of the embodiment, it is possible to realize both improvement of the power factor and compatibility with a wide range.
In realizing the power factor improvement and wide-range compatibility, the active source filter shown in FIG. 14 is not mounted in the source circuit shown in FIG. In other words, depending on the configuration shown in FIG. 1, it is possible to improve the power factor and support a wide range without mounting an active filter.

As described above, since the active filter is not required as a configuration for improving the same power factor and supporting a wide range, the circuit of FIG. 1 can improve the overall power conversion efficiency. .
In other words, the active filter has a hard switching operation, has a large switching loss, and cannot improve the power conversion efficiency effectively. However, the current resonance converter according to the present embodiment can perform a soft switching operation. This is because the switching loss can be reduced and the power conversion efficiency can be effectively improved.
As described above, this is because the AC → DC power conversion efficiency in the current resonance type converter of the embodiment is VAC = 100V and 230V, as compared with the AC → DC power conversion efficiency in the conventional active filter. It is clear from the improvement.

In addition, since the active filter can be eliminated as described above, the number of circuit components can be reduced.
That is, in the power supply circuit shown in FIG. 1, since the operation of the current resonance converter in the previous stage is a soft switching operation as described above, the level of switching noise is greatly reduced as compared with the active filter shown in FIG. Is done. For this reason, as shown in FIG. 1, if a single-stage noise filter composed of a pair of common mode choke coils CMC and an across capacitor CL is provided, it is possible to sufficiently satisfy the power disturbance standard. The Further, as shown in FIG. 1, the normal mode noise of the rectified output line is taken by only one set of filter capacitors CN.
By reducing the number of parts as a noise filter in this way, the cost of the power supply circuit is reduced and the circuit board is reduced in size and weight.

Further, when the configuration of the power supply circuit of FIG. 1 is adopted, since the isolated converter transformer PIT is provided in the current resonance type converter provided in the preceding stage, the DC / DC converter 5 in the subsequent stage is already insulated from the primary side. can get. In other words, this eliminates the need for the DC / DC converter 5 in this case to adopt a configuration for insulating the primary side and the secondary side.
For example, the converter transformer provided in the DC / DC converter 5 in this case can take a non-insulated configuration. According to this, it is not necessary to take an insulation distance when winding the primary side winding and the secondary side winding, and the converter transformer can be downsized accordingly.
The DC / DC converter 5 is also configured to stabilize the DC output voltage Eo. For example, the DC / DC converter 5 is configured to feed back the output voltage level in the same manner as that provided in the current resonance converter shown in FIG. When the voltage control system is provided, a photocoupler for insulating the primary side and the secondary side can be eliminated.
Furthermore, even when an overvoltage protection circuit or the like is provided, a photocoupler for insulating the primary side and the secondary side can be eliminated.
In this way, since the configuration for insulation can be omitted, circuit components and manufacturing costs can be reduced as the DC / DC converter 5 in this case.

  Further, according to the configuration of FIG. 1, when the power supply circuit is started, the inrush current from the commercial AC power supply AC can be prevented from flowing by the soft start function by the oscillation / drive circuit 2. This eliminates the need for the inrush current suppression configuration (inrush current limiting circuit 22) as provided in the circuit of FIG. 14, and also in this respect, circuit components and manufacturing costs can be reduced. .

  From these comparisons, according to the power supply circuit of the present embodiment shown in FIG. 1, after solving various problems of the circuit of FIG. 14 having the active filter, it is the same as the case of having the active filter. In addition, a practically sufficient power factor can be obtained, and furthermore, higher power conversion efficiency can be obtained.

Next, FIG. 7 shows a configuration of a modification on the secondary side of the current resonance converter in the previous stage shown in FIG.
In FIG. 7, the configuration on the primary side is the same as that shown in FIG.
In the modification shown in FIG. 7, a voltage doubler half-wave rectifier circuit is configured as the secondary-side rectifier circuit.

First, the voltage doubler half-wave rectifier circuit includes two sets of smoothing capacitors Ci including a rectifier diode D1 and a rectifier diode D2, and a smoothing capacitor Ci1 and a smoothing capacitor Ci2, as shown in the figure.
In this voltage doubler half-wave rectifier circuit, first, the anode of the rectifier diode D1 is connected to one end portion (winding end portion) of the secondary winding N2. The cathode of the rectifier diode D1 is connected to the positive terminal of the smoothing capacitor Ci1.
The negative terminal of the smoothing capacitor Ci1 is connected to the positive terminal of the other smoothing capacitor Ci2, as shown, and the negative terminal of the smoothing capacitor Ci2 is connected to the secondary side ground. In addition, the other end (winding start end) of the secondary winding N2 is connected to a connection point between the negative terminal of the smoothing capacitor Ci1 and the positive terminal of the smoothing capacitor Ci2.
Further, as shown in the figure, a rectifier diode D2 is inserted between a connection point between one end of the secondary winding N2 and the anode of the rectifier diode D1 and the secondary side ground. The rectifier diode D2 is inserted so that the anode side is connected to the secondary side ground.

In the voltage doubler half-wave rectifier circuit having the above configuration, first, the rectifier diode D1 conducts in one half cycle of the alternating voltage excited in the secondary winding N2, and charges the smoothing capacitor Ci1 with the rectified current. As a result, a voltage is generated at both ends of the smoothing capacitor Ci1 at a level corresponding to an equal multiple of the alternating voltage level excited by the secondary winding N2. In the other half cycle, the rectifier diode D2 conducts to charge the smoothing capacitor Ci2 with a rectified current, and both ends of the smoothing capacitor Ci2 correspond to an equal voltage level of the alternating voltage excited by the secondary winding N2. Generate a voltage according to the specified level.
As a result, in one cycle of the alternating voltage excited in the secondary winding N2, there is a level corresponding to twice the alternating voltage level excited in the secondary winding N2 at both ends of the series connection of the smoothing capacitors Ci1 to Ci2. Thus, the DC input voltage Ei can be obtained.

Thus, depending on the configuration shown in FIG. 7, each of the smoothing capacitors Ci is charged only in one half cycle of the alternating voltage excited in the secondary winding N2, and the smoothing capacitor Ci (Ci1- As the voltage level obtained in Ci2), a double voltage half-wave rectification operation is obtained, which is a level corresponding to twice the alternating voltage level.
In this case, the DC input voltage Ei is also branched and supplied as a detection input of the control circuit 1 similar to that shown in FIG.

Here, for confirmation, in this case as well, since the secondary side rectifier circuit has a configuration other than the double-wave rectifier circuit, the same coupling coefficient k (gap length) as in the circuit of FIG. ), A configuration compatible with a wide range is realized.
Also in this case, the primary side configuration shown in FIG. 1 is adopted, and stabilization by the control circuit 1 is performed with respect to the average value Ei-avr of the DC input voltage Ei, thereby improving the power factor.

In addition, when the secondary side rectifier circuit is a voltage doubler half-wave rectifier circuit as shown in FIG. 7, in order to obtain the same DC input voltage Ei level, the number of turns of the secondary winding N2 is The bridge rectifier circuit of FIG. 1 can be further reduced by half.
That is, according to this, compared with the conventional circuit of FIG. 14, the number of turns of the secondary winding N2 is reduced to ¼. Therefore, when the configuration on the secondary side of FIG. The insulation converter transformer PIT can be downsized.

FIG. 8 shows another modification of the secondary side configuration of the current resonance type converter according to the embodiment.
In this figure as well, the configuration on the primary side is the same as that shown in FIG.
The modification shown in FIG. 8 includes a voltage doubler full wave rectifier circuit as a secondary side rectifier circuit.

First, in this case, the secondary winding N2 is divided into secondary winding portions N2A / N2B by applying a center tap. Here, a bridge rectifier circuit having the same connection form as that of the circuit of FIG. 1 is connected to the entire secondary winding N2.
Specifically, the end of the secondary winding N2 which is the winding end of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode D1 and the cathode of the rectifier diode D2. The Further, the end on the secondary winding portion N2B side, which is the winding start end of the secondary winding N2, is connected to a connection point between the anode of the rectifier diode D3 and the cathode of the rectifier diode D4.
The connection point between the cathode of the rectifier diode D1 and the cathode of the rectifier diode D3 is connected to the positive terminal of the smoothing capacitor Ci. Also in this case, the negative terminal of the smoothing capacitor Ci is connected to the secondary side ground.

In this case, the positive terminal of the capacitor Cc is connected to the center tap of the secondary winding N2. Further, the negative terminal of the capacitor Cc is connected to the connection point between the anodes of the rectifier diodes D2 and D4.
As shown in the figure, the connection point between the anodes of the rectifier diodes D2 and D4 is connected to the secondary side ground.

  In the voltage doubler full-wave rectifier circuit configured as described above, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding portion N2A → smoothing capacitor Cc → rectifier diode D2 → secondary. It flows through the circulation path of the winding portion N2A]. Further, in the other half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is caused by a circulation path of [secondary winding portion N2B → smoothing capacitor Cc → rectifier diode D4 → secondary winding portion N2B]. Flowing. In other words, in this case, for the smoothing capacitor Cc, a DC voltage is obtained at a level corresponding to the same multiple of the alternating voltage level excited in the secondary winding portion N2A and the secondary winding portion N2B in each half cycle. It is like that.

In addition, in the above half cycle of the alternating voltage excited in the secondary winding N2, the rectified current branches off from the circulation path [secondary winding portion N2B → rectifier diode D3 → smoothing capacitor Ci → It also flows through the path of the rectifier diode D2].
As a result, in the half cycle, the smoothing capacitor Ci is charged with a level in which the alternating voltage of the secondary winding N2B and the voltage across the smoothing capacitor Cc as described above are superimposed. That is, as the voltage across the smoothing capacitor Ci, a level corresponding to twice the alternating voltage level obtained in the secondary winding is obtained.

  Further, also as the other half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is branched from the above circulation path [secondary winding portion N2A → rectifier diode D1 → smoothing capacitor Ci → rectifier diode. D4] also flows, and in this case as well, the voltage across the smoothing capacitor Ci is the alternating voltage level obtained in the secondary winding portion by the alternating voltage of the secondary winding portion N2A and the charge of the capacitor Cc. A level corresponding to twice this is obtained.

From such a rectification operation, in the rectification circuit in this case, the smoothing capacitor Ci is charged in each half cycle of the alternating voltage obtained in the secondary winding N2. As the charging potential, a level corresponding to twice the alternating voltage induced in the secondary winding portion as described above is obtained.
From this, it can be understood that a double voltage full wave rectification operation is obtained by the secondary side rectifier circuit shown in FIG.

  Even when the voltage doubler full wave rectifier circuit is provided as the secondary side rectifier circuit in this way, in order to obtain the same DC input voltage Ei level, the total number of turns of the secondary winding N2 is shown in FIG. It can be equivalent to the circuit. Therefore, even when the configuration of FIG. 8 is adopted, the size of the insulating converter transformer PIT similar to that of the circuit of FIG. 1 can be reduced.

For confirmation, the secondary side configuration shown in FIG. 8 is provided with a center tap on the secondary winding N2 as in the case of the circuit shown in FIG. However, in the configuration of FIG. 8, the parallel resonance operation (partial voltage resonance operation) does not occur on the secondary side as in the case of the conventional double-wave rectifier circuit. In the configuration of FIG. 8, as can be understood from the above description, a rectification current flows in each secondary winding part in each half cycle of the alternating voltage, and the winding is different from the case of the conventional double-wave rectification. This is because there is no line-to-line capacitance between the secondary winding portions because there is no period during which the rectified current does not flow in the portion.
Since no parallel resonance operation occurs on the secondary side as described above, the required control range Δfs as in the case of FIG. 1 is reduced by setting the coupling coefficient k described in FIG. 2 even in the configuration of FIG. It is something that can be done.

FIG. 9 shows still another modified example of the configuration on the secondary side of the current resonance type converter according to the embodiment.
Also in FIG. 9, the configuration on the primary side is the same as in the case of FIG.
The modification of FIG. 9 includes a quadruple voltage rectifier circuit as a secondary side rectifier circuit.

This quadruple voltage rectifier circuit is formed by including four rectifier diodes by rectifier diodes D1 to D4, capacitors Cc1, Cc2, and smoothing capacitors Ci1, Ci2 as shown in the figure.
In this case, a series connection of a capacitor Cc1 (negative terminal → positive terminal) → rectifier diode D1 (anode → cathode) is connected to one end (end of winding) of the secondary winding N2 as shown in the figure. And the positive terminal of the smoothing capacitor Ci1 is connected. The negative terminal of the smoothing capacitor Ci1 is connected to the other end (winding end) of the secondary winding N2.
The positive terminal of the smoothing capacitor Ci2 is connected to the connection point between the negative terminal of the smoothing capacitor Ci1 and the other end of the secondary winding N2, and the negative terminal of the smoothing capacitor Ci2 is connected to the secondary side ground. It is connected.

Further, a series connection circuit of a capacitor Cc2 (positive terminal → negative terminal) → rectifier diode D4 (cathode → anode) is inserted between the one end of the secondary winding N2 and the secondary side ground. ing.
The anode of the rectifier diode D3 is connected to the connection point between the capacitor Cc2 and the rectifier diode D4 as shown. The cathode of the rectifier diode D3 is connected to the connection point of the smoothing capacitors Ci1 and Ci2 and the other end of the secondary winding N2.
Further, the anode of the rectifier diode D2 is connected to the connection point between the cathode of the rectifier diode D3 and the other end of the secondary winding N2. The cathode of the rectifier diode D2 is connected to the connection point between the capacitor Cc1 and the rectifier diode D1.

In the quadruple voltage rectifier circuit configured as described above, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding N2 → rectifier diode D2 → capacitor Cc1 → secondary winding N2. ] Through the circulation path. Similarly, in the other half cycle of the alternating voltage, the rectified current flows through [secondary winding N2 → capacitor Cc2 → rectifier diode D3 → secondary winding N2] through the circulation path.
In other words, even in this case, a DC voltage having a level corresponding to the same multiple of the alternating voltage level excited in the secondary winding N2 is obtained at both ends of the capacitors Cc1 and Cc2 in the corresponding half cycle. Become.

Even in this case, the rectified current branches from the circulation path and flows through the following paths in each half cycle.
First, in one half cycle of the alternating voltage described above, the rectified current branches and flows through the path [smoothing capacitor Ci 2 → rectifier diode D 4 → capacitor Cc 2 → secondary winding N 2]. At this time, charged charges are obtained at both ends of the capacitor Cc2 during this period by the previous circulation path. For this reason, depending on the rectification current path as described above, the smoothing capacitor Ci2 is charged with a potential due to the alternating voltage obtained in the secondary winding N2 and the superimposed charge of the capacitor Cc2. Become.
That is, a smoothing capacitor Ci2 generates a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2.

Further, in the other half cycle of the above alternating voltage, the rectified current branches and flows also through the path [capacitor Cc 1 → rectifier diode D 1 → smoothing capacitor Ci 1 → secondary winding N 2]. The smoothing capacitor Ci1 is charged with respect to the alternating voltage of the secondary winding N2 that has received the superimposed charge charge obtained by the capacitor Cc1.
In other words, even with the smoothing capacitor Ci1, a DC voltage having a level corresponding to twice the alternating voltage level obtained at the secondary winding N2 is obtained as the voltage across the smoothing capacitor Ci1.

  In this way, a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is generated at both ends of the smoothing capacitor Ci1 and the smoothing capacitor Ci2. As a result, a DC input voltage Ei having a level corresponding to four times the alternating voltage level excited by the secondary winding N2 is obtained at both ends of the series connection of the smoothing capacitor Ci1 and the smoothing capacitor Ci2. .

  When such a quadruple voltage rectifier circuit is employed, the number of turns of the secondary winding N2 can be reduced to about ¼ that in the case of FIG. 1, thereby further reducing the size of the insulating converter transformer PIT. It becomes possible.

Further, the circuit diagram of FIG. 10 shows a configuration of a modified example of the primary side of the current resonance type converter according to the embodiment.
As a modification of the primary side, the configuration of the switching converter is changed from the half bridge coupling method to the full bridge coupling method as shown in the figure.
In FIG. 10, the same parts as those already described in FIG.

In FIG. 10, as a full bridge coupling system, as shown in the figure, the half bridge connection of the switching elements Q3 and Q4 is connected in parallel to the half bridge connection of the switching elements Q1 and Q2.
As for the switching elements Q3 and Q4, similarly to the switching elements Q1 and Q2, a damper diode DD3 and a damper diode DD4, which are body diodes, are connected in parallel between the drain and the source, respectively.

In addition, in this case, a primary side series resonant circuit comprising a series connection of the primary winding N1 of the insulating converter transformer PIT and the primary side series resonant capacitor C1 is connected as follows.
First, one end (winding end) of the primary winding N1, which is one end of the primary side series resonance circuit, is connected to a connection point between the source of the switching element Q3 and the drain of the switching element Q4. The connection point between the source of the switching element Q3 and the drain of the switching element Q4 is one switching output point in the full-bridge coupling switching circuit system.
Further, with respect to the other end portion side of the primary side series resonance circuit, the other end (end of winding end) of the primary winding N1 is connected to the other switching output point via the series connection of the primary side series resonance capacitor C1. The connection is made to the connection point between the source of a certain switching element Q1 and the drain of the switching element Q2.

  In this case, the primary-side partial resonance capacitor Cp2 is connected in parallel with the source and drain of the switching element Q4. The primary side partial resonance capacitor Cp2 also forms a parallel resonance circuit (partial voltage resonance circuit) by its own capacitance and the leakage inductance L1 of the primary winding N1, and voltage resonates only when the switching elements Q3 and Q4 are turned off. Get voltage resonant operation.

  The oscillation / drive circuit 2 in this case is configured to drive four switching elements Q1 to Q4. Depending on the oscillation / drive circuit 2, switching driving is performed such that the group of switching elements [Q1, Q4] and the group of switching elements [Q2, Q3] are alternately turned on / off.

  Here, for example, as the load condition becomes a heavy load trend, the current flowing through the switching converter increases, the load on the circuit components increases, and the power loss also increases. Therefore, if the full-bridge coupling is used as described above, the necessary load current is provided by four switching elements. For example, each component is more than the case of the half-bridge coupling system including two switching elements. This reduces the burden on the power source and reduces power loss, which can be advantageous for heavy load conditions.

  Even in the case of full bridge coupling as described above, each switching element Q1 to Q4 performs a switching operation on the rectified output voltage V1 of the bridge rectifier circuit Di, and the control circuit 1 and the oscillation / drive circuit shown in the figure. By performing the operation of stabilizing the average value Ei-avr of the DC input voltage Ei by the constant voltage control system according to 2, the power factor can be improved as in the case of the circuit of FIG.

The present invention is not necessarily limited to the configuration as the above-described embodiment.
For example, as a switching element, an element other than a MOS-FET may be employed as long as it is an element that can be used in an separately excited type, such as an IGBT (Insulated Gate Bipolar Transistor). Also, the constants of the component elements described above may be changed according to actual conditions. Further, for example, the circuit configuration for generating a DC voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Further, the wide range compatible and power factor improving configuration according to the present invention can be applied to a self-excited current resonance type converter.

  Further, the configuration of the secondary side rectifier circuit is not limited to that exemplified in the embodiment, and is a configuration other than the double-wave rectifier circuit and does not cause a line-to-line capacitance in the secondary winding. If so, other rectifier circuit configurations may be employed.

In addition, as the current resonance type converter of the embodiment, the DC / DC converter 5 is provided in the subsequent stage to configure the power supply circuit. However, the DC / DC converter 5 is omitted as the power supply circuit, and the DC input voltage Ei. Can be configured to be supplied to the load side.
However, as the DC input voltage Ei generated by the current resonance type converter according to the embodiment, a ripple of several V is generated as is apparent from the waveform diagram of FIG. Therefore, in practice, a configuration may be adopted in which the DC input voltage Ei is supplied to the load side through a configuration for suppressing such ripples.

It is a circuit diagram which shows the structure of the switching power supply circuit as embodiment in this invention. It is sectional drawing which shows the structural example of the insulation converter transformer with which the current resonance converter (switching converter apparatus) of a front | former stage is provided in the switching power supply circuit of embodiment. It is a wave form diagram which shows the operation | movement waveform of the principal part in the power supply circuit of embodiment. It is a figure which shows the switching frequency control range (necessary control range) according to load fluctuation | variation as a constant voltage control characteristic of the power supply circuit of embodiment. It is a figure which shows each characteristic of the power factor with respect to the load variation of the power supply circuit of embodiment, AC-> DC power conversion efficiency, and the average value of DC input voltage. It is a figure which shows each characteristic of the power factor with respect to the alternating current input voltage fluctuation | variation of embodiment, AC-> DC power conversion efficiency, and the average value of direct current input voltage of embodiment. It is the circuit diagram which showed the modification about the structure of the secondary side of the current resonance type | mold converter of the front | former stage in the power supply circuit of embodiment. FIG. 6 is a circuit diagram showing another modification of the configuration on the secondary side of the current resonance converter in the previous stage in the power supply circuit according to the embodiment. FIG. 10 is a circuit diagram showing still another modification example of the configuration on the secondary side of the current-resonance converter in the previous stage in the power supply circuit according to the embodiment. It is the circuit diagram which showed the modification about the structure of the primary side of the current resonance converter of the front | former stage in the power supply circuit of embodiment. It is a circuit diagram which shows the basic circuit structure of an active filter. It is a wave form diagram which shows the operation | movement in the active filter shown in FIG. It is a circuit diagram which shows the structure of the control circuit system of an active filter. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art. It is the figure which showed the constant voltage control characteristic in the case of setting the coupling coefficient of a primary side and a secondary side as the conventional setting. It is the figure which showed the constant voltage control characteristic of the conventional power supply circuit which made the secondary side rectifier circuit the double wave rectifier circuit. It is the figure which showed about an example of the change characteristic of the switching frequency with respect to the load fluctuation | variation about the conventional power supply circuit which made the secondary side rectifier circuit the double wave rectifier circuit.

Explanation of symbols

  1 control circuit, Q5 shunt regulator, Ct2 phase compensation capacitor, 2 oscillation / drive circuit, Di bridge rectifier circuit, CN filter capacitor, Q1, Q2, Q3, Q4 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, Cp, Cp1, Cp2 Primary side partial resonance capacitor, N1 primary winding, N2 secondary winding, N2A, N2B secondary winding part, D1-D4 (secondary side) rectifier diode, Ci (secondary side) smoothing capacitor , Cc, Cc1, Cc2 capacitors

Claims (9)

Rectifying means for generating a rectified output voltage by inputting a commercial AC power supply and performing a rectifying operation;
Switching means formed with a switching element that performs switching by inputting the rectified output voltage generated by the rectifying means; and
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A DC output voltage that is provided with a rectifier circuit other than the double-wave rectifier circuit, performs a rectification operation on the alternating voltage obtained in the secondary winding, and smoothes the rectified output with a smoothing capacitor to generate a DC output voltage. Generating means;
Constant voltage control configured to perform constant voltage control on the DC output voltage by controlling the switching drive means according to the average value of the DC output voltage and varying the switching frequency of the switching means. Means, and
A switching converter in which a gap length formed at least at a predetermined position of the core of the insulating converter transformer is set so that a coupling coefficient between the primary side and the secondary side of the insulating converter transformer is not more than a predetermined value; ,
And a DC / DC converter that generates a secondary DC output voltage by inputting the DC output voltage generated in the switching converter as a DC power source and performing DC / DC power conversion.
A switching power supply circuit. The constant voltage control means is
By providing a low pass filter for extracting the low frequency component of the DC output voltage, it is configured to obtain an average value of the DC output voltage,
The switching power supply circuit according to claim 1. The DC output voltage generating means is
As the rectification operation, it is configured to perform full-wave rectification operation by a bridge rectification circuit,
The switching power supply circuit according to claim 1. The DC output voltage generating means is
As the rectifying operation,
The smoothing capacitor is charged only in one half cycle of the alternating voltage excited in the secondary winding, and the DC output voltage is generated at a level corresponding to twice the alternating voltage level. Configured to perform voltage half-wave rectification operation,
The switching power supply circuit according to claim 1. In the insulating converter transformer, a center tap is applied to the secondary winding to form a first secondary winding portion and a second secondary winding portion,
The DC output voltage generating means is the rectifying operation,
The smoothing capacitor is charged in each half cycle of the alternating voltage excited by the secondary winding, and the first secondary winding portion and the second secondary winding are provided at both ends of the smoothing capacitor. The voltage doubler full wave rectification operation is performed to generate the DC output voltage at a level corresponding to twice the alternating voltage level obtained with each of the units,
The switching power supply circuit according to claim 1. The DC output voltage generating means is
As the rectifying operation,
Configured to perform a quadruple voltage rectification operation configured to generate the DC output voltage at a level corresponding to four times the alternating voltage level excited in the secondary winding;
The switching power supply circuit according to claim 1.   2. The switching power supply circuit according to claim 1, wherein the switching means has two switching elements connected by a half-bridge coupling method.   The switching power supply circuit according to claim 1, wherein the switching means includes four switching elements connected by a full bridge coupling method. Rectifying means for generating a rectified output voltage by inputting a commercial AC power supply and performing a rectifying operation;
Switching means formed with a switching element that performs switching by inputting the rectified output voltage generated by the rectifying means; and
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A DC output voltage that is provided with a rectifier circuit other than the double-wave rectifier circuit, performs a rectification operation on the alternating voltage obtained in the secondary winding, and smoothes the rectified output with a smoothing capacitor to generate a DC output voltage. Generating means;
Constant voltage control configured to perform constant voltage control on the DC output voltage by controlling the switching drive means according to the average value of the DC output voltage and varying the switching frequency of the switching means. Means, and
The gap length formed at least at a predetermined position of the core of the insulating converter transformer is set so that the coupling coefficient between the primary side and the secondary side of the insulating converter transformer is not more than a predetermined value.
A switching converter device.

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