JP2005318555A - Data transfer system and control program for base station - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a data transfer system and a control program for a base station by which a decoding error caused by tag coding cycle deviation (modulation frequency deviation) can be improved by widening an allowable range for variations of resonant frequencies of a transponder, improving the detection rate of the transponder, and improving the yield of an IC (transponder) by a simple configuration. <P>SOLUTION: The present invention relates to a data transfer apparatus comprised of a passive transponder and a base station, wherein the passive transponder transmits information by modulating the load of a carrier wave and the base station includes a quadrature detector for outputting I and Q signals by performing quadrature detection on the load-modulated carrier wave, an A-D converter for sampling the I and Q signals at a predetermined frequency and converting them into digital data, an adder for adding the respective digital data of the I and Q signals and outputting a result as a digital waveform, and an extraction section for extracting information from the digital waveform. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a data transfer system and a base station control program for transmitting information between a passive transponder and a base station via a coil on each side without being directly connected.

  In the data transfer system, as shown in FIG. 5, a passive transponder (for example, an RF-ID (wireless IC tag including an antenna and an IC)) 100 and a base station (for example, an RF-ID reader / writer). In the data transmission with the gate 101, a control unit (not shown) controls on / off of the switch 102 at a predetermined modulation frequency based on specific data stored in advance, so that a load that can be connected (capacitor Cload) ) Is connected / disconnected to the coil 102 of the transponder 100 and is subjected to load modulation (that is, amplitude modulation). This RF-ID is affixed to a book or the like as a tag, and even if a plurality of books are stacked, the stored information is different, so that individual discrimination is possible.

  That is, by connecting the capacitor Cload to the coil of the transponder 100, the resonance frequency is shifted, and the coil 104 provided in the base station 101 acting as a transmitting antenna and a receiving antenna has an amplitude in the signal of the coil 104. Inducing variation. At this time, the coil 103 of the transponder 100 forms part of a resonant circuit tuned to the frequency transmitted from the base station 101, and maximum energy transmission occurs from the base station 101 to the transponder 100. The data is demodulated by analyzing the amplitude modulation of the signal in the coil 104.

  Here, as shown in FIG. 4, in the conventional data transfer apparatus, in order to analyze the amplitude modulation in the coil 104, the coupler 201 is coupled to the coil 104, the AC signal component is extracted, and the linear detector 202 linearly analyzes it. Detection is performed, and a predetermined frequency region is extracted by a BPF (band pass filter) 203. Then, the signal level is amplified to a predetermined intensity by an AGC (auto gain control amplifier) 204, and the presence or absence of the signal, that is, whether the level is 1 (H level) / 0 (L level) is determined. The decoding processing unit 206 decodes the data transmitted from the transponder 100 to the base station 101 based on the determination result.

  Here, as can be seen from the equation shown in FIG. 5, the resonance frequency (f) changes due to variations in the inductance of the coil 103 and the capacitor Cload. For this reason, an amplitude difference between when the resonance frequency of the switch is changed and when it is not changed cannot be obtained sufficiently, and the signal cannot be compared by the comparator 205. For this reason, as shown in FIG. 6, the capacitance of the externally attached capacitor Cload is adjusted to trimming.

However, a low-cost RF-ID or the like has a problem that the manufacturing cost cannot be reduced because a process of creating a capacitance component (capacitor Cload) outside and adjusting it by trimming increases.
For this reason, the base station receiving apparatus having the configuration shown in FIG. 7 uses the circuit shown in FIG. 7 to detect not only the intensity but also the phase change to improve the decoding error due to the tag encoding period deviation. Since the modulation analysis of the signal is facilitated, the capacitor Cload can be built in the RF-ID, and the price of the RF-ID can be reduced (see Patent Document 1).
JP 09-247227 A

  However, in the above conventional example, although the phase matching process is performed, since the baseband signal (encoded data to be transmitted) is reproduced substantially by the in-phase component, the resonance frequency of the transponder and the base station is reduced. When the deviation (that is, the fluctuation range) is large, the reach distance is short, and therefore the amplitude displacement is small, the S / N ratio is deteriorated, and the detection rate is lowered.

The variation in the resonance frequency is based on many factors such as variations in the capacitor C2, temperature characteristics of the capacitance, and variations in the dielectric constant of a detection object (such as a book) to be attached. For this reason, in the above conventional example, when a transponder is used as a tag, the one that satisfies the detection rate is selected, so that the yield deteriorates and it is difficult to reduce the manufacturing cost.
Furthermore, in Patent Document 1, since the signal is detected by directly logically combining the amplitudes of the obtained I signal and Q signal after orthogonal transformation, the S / N ratio is not improved. The detection rate cannot be improved sufficiently.

  The present invention has been made in view of such circumstances, and with a simple configuration, the allowable range of variation in the resonance frequency of the transponder is increased, the detection rate of the transponder is improved, and the yield of the IC (transponder) is improved. Another object of the present invention is to provide a data transfer system and a base station control program capable of improving a decoding error due to a tag encoding cycle deviation (modulation frequency deviation).

  The present invention has been made to solve the above problems, and the invention according to claim 1 is a data transfer system including a passive transponder and a base station, and the passive transponder transmits information. The base station performs quadrature detection of the load-modulated carrier wave and outputs an I signal and a Q signal, and each of the I signal and the Q signal is sampled at a predetermined frequency. The information is extracted from the digital waveform, an A / D converter that converts the data into digital data, an adder that adds and synthesizes the digital data of each of the I and Q signals, and outputs the resulting digital waveform. And a data transfer system.

  According to a second aspect of the present invention, an adaptive control unit that obtains a weighting coefficient to be multiplied to each of the digital data so as to maximize the S / N ratio of the combined digital waveform signal, and each of the digital data, The data transfer system according to claim 1, further comprising a multiplier for multiplying the weighting factor.

  The invention according to claim 3 is the data transfer system according to claim 1 or 2, wherein the load modulation is capacitive load modulation.

  The invention according to claim 4 is characterized in that the A / D converter generates digital data at a sampling frequency that is twice or more the modulation frequency. The data transfer system according to the item.

  Further, the invention according to claim 5 is characterized in that the extraction unit determines a signal level of a digital waveform in synchronization with the sampling frequency, and a preset encoding in synchronization with the sampling frequency. 5. The data transfer system according to claim 1, further comprising a matched filter that obtains an inner product of a pattern and the digital waveform and decodes the information as a peak value. 6. is there.

  The invention described in claim 6 further includes a synchronization circuit that obtains a deviation of the modulation frequency, extracts a decoding timing based on the deviation, and controls the decoding timing in the matched filter. 5. The data transfer system according to 5.

  According to a seventh aspect of the present invention, there is provided a plurality of offset estimation circuits in which the synchronization circuit gives delays corresponding to different deviations to the digital waveform and outputs a correlation value by autocorrelation, The selection circuit that extracts the deviation of the modulation frequency having a high correlation and outputs the estimated frequency signal of the modulation frequency having this deviation is compared with the estimated frequency signal from the selection circuit and the digital waveform delayed by the delay circuit. The data transfer system according to claim 6, further comprising a timing demodulation circuit that finely adjusts the frequency of the estimated frequency signal.

  According to an eighth aspect of the present invention, in the data according to any one of the first to seventh aspects, the transponder includes a sheet-like planar coil and is affixed to a detection object. It is a transfer system.

  According to a ninth aspect of the present invention, there is provided a control program for a base station that performs wireless communication with a passive transponder. And a first step of outputting a Q signal, a second step of sampling each of the I signal and the Q signal output in the first step at a predetermined frequency and converting each of the I signal and the Q signal into digital data, The third step of adding and synthesizing the digital data of each of the I signal and Q signal converted in step 2 and outputting as a digital waveform, and the digital waveform output in the third step, A base station control program for causing a computer to execute a fourth step of extracting information. A gram.

  According to a tenth aspect of the present invention, there is provided a fifth step of calculating a weighting factor by which each digital data is multiplied so as to maximize the S / N ratio of the combined digital waveform signal, and the second step. The base station control program according to claim 9, further causing a computer to execute a sixth step of multiplying each digital data converted in the step by the weighting factor calculated in the fifth step.

  According to an eleventh aspect of the invention, in the first step, information transmitted by capacitive load modulation of a carrier wave is received from the transponder, and an I signal and a Q signal are output by performing quadrature detection. 11. The base station control program according to claim 9 or 10, wherein the base station control program is characterized by the following.

  According to a twelfth aspect of the present invention, in the second step, each of the I signal and the Q signal output in the first step is sampled at a sampling frequency that is twice or more of a modulation frequency, 12. The base station control program according to claim 9, wherein the base station control program converts the data into data.

  The invention according to claim 13 is a seventh step of determining a signal level of a digital waveform in synchronization with the sampling frequency, a coding pattern set in advance in synchronization with the sampling frequency, The base station control according to any one of claims 9 to 12, for causing the computer to further execute an eighth step of obtaining an inner product with the digital waveform and decoding the information as a peak value. It is a program.

  According to a fourteenth aspect of the present invention, the computer further includes a ninth step of calculating a deviation of the modulation frequency, extracting a decoding timing based on the deviation, and controlling the decoding timing in the eighth step. 14. A base station control program according to claim 13 for execution.

  The invention according to claim 15 provides a tenth step of giving each of the delays corresponding to different deviations to the digital waveform and outputting a plurality of correlation values by autocorrelation, and a plurality of outputs outputted in the tenth step. An eleventh step of extracting a deviation of the modulation frequency having the highest correlation from the correlation value of the output, and outputting an estimated frequency signal of the modulation frequency having the deviation; an estimated frequency signal output in the eleventh step; and a delay 15. The base station control program according to claim 14, further comprising: causing the computer to execute a twelfth step of comparing the digital waveform thus obtained and performing fine adjustment of the frequency of the estimated frequency signal.

  As described above, in the data transfer system according to the embodiment of the present invention, the determiner corrects the S / N ratio of the synthesized digital waveform signal to the maximum in the amplitude-modulated (and phase-modulated) waveform. Since the amplitude of the Q component (Q signal) is also added to the amplitude of the I component (I signal) as a real component of amplitude modulation, and the input waveform is extracted as a composite value of the real number, the resonance frequency of the transponder It is possible to widen the tolerance for variation, improve the S / N ratio of the input waveform, and reproduce the baseband signal even if the transmitted energy is weak, Fine adjustment of the capacity for setting the resonance frequency is not required, and this capacity can be built in the IC, and the manufacturing cost can be greatly reduced.

  In addition, the data transfer system according to the embodiment of the present invention obtains the inner product (correlation) between the input baseband signal and the coding pattern by the matched filter, and detects the signal transmitted by the correlation value. In addition, the coding period is estimated from the peak period of the correlation value in the inner product result to compensate for the transponder frequency offset (modulation frequency deviation). For this reason, the data transfer system according to the embodiment of the present invention prevents the detection rate of the baseband signal (data to be transmitted) from being lowered due to the phase shift of decoding, and prevents the reception quality from being deteriorated as in the conventional example, that is, Decoding error due to tag coding period deviation (modulation frequency deviation) is improved.

  A data transfer system according to an embodiment of the present invention is a data transfer apparatus including a passive transponder and a base station. The passive transponder transmits information by load modulation of a carrier wave, and the base station receives an input load. A quadrature detector that quadrature-detects the modulated carrier wave and outputs an I signal and a Q signal; an A / D converter that samples each of the I signal and the Q signal at a predetermined frequency and converts them into digital data; , An adder that adds and synthesizes the digital data of each of the I signal and the Q signal and outputs the resultant as a digital waveform, and an extraction unit that extracts information from the digital waveform.

Hereinafter, a data transfer apparatus according to a first embodiment of the present invention will be described with reference to the drawings.
FIG. 1 is a diagram illustrating a configuration example of a receiving unit in a base station of a data transfer apparatus according to a first embodiment. The configuration of the transponder is the same as the configuration of the transponder 101 in FIG. The coils 103 and 104 are sheet-like coils (planar coils).
The coupler 2 is coupled to a junction point between the capacitor 105 and the resistor 106 (parasitic resistance of the coil 104), for example. The quadrature detector 3 performs quadrature detection of the modulated carrier wave input from the coupler 2 and outputs an I component and a Q component to BPFs (band pass filters) 4 and 5, respectively. BPFs 4 and 5 are band-pass filters that pass the modulation frequency band. Here, the data frequency-modulated in the transponder 100 is, for example, data encoded by Manchester code.

  The ADCs (A / D converters) 6 and 7 sample the analog alternating current signal input from the BPFs 4 and 5 at a predetermined sampling frequency, and output the result as digital amplitude data. The multipliers 8 and 9 multiply the weighted coefficients α and β output from the adaptive control units 10 and 11, respectively, by the digitally converted amplitude data. The adder 12 adds the amplitude data of the I signal and the Q signal. The determiner 13 determines whether the added data is greater than “0”. The matched filter 14 determines a correlation between a pre-stored encoding pattern (for example, Manchester code pattern) and determination data input in time series from the adder 12. Based on the correlation value from the matched filter 14, the decoding processing unit 15 performs a process of decoding baseband signal data (decoding from Manchester code).

Next, an example of the operation of the data transfer apparatus according to the first embodiment of the present invention will be described with reference to FIG.
The coil 104 couples with the coil 103 of the transponder 100 to transmit energy. At this time, when the resonant frequencies of the transponder 100 and the base station 101 are the same, the transmitted energy is maximized.

  For example, the coupler 2 is connected to a connection point between the capacitor 105 and the resistor 106 in FIG. 4 and extracts AC data (received signal) of the coil 104. The quadrature detector 3 performs quadrature detection of the AC data input from the coupler 2 and outputs an I signal and a Q signal. The BPF 4 passes a frequency component of a predetermined frequency band of the I signal as an I-B signal. Similarly, the BPF 5 passes a frequency component in a predetermined frequency band as a Q-B signal. Here, the predetermined frequency band is in the vicinity of the modulation frequency at which the switch 102 in the transponder 100 is turned on / off.

  The ADC (Analog / Digital Converter) 6 receives an I-B signal input from the BPF 4 at a predetermined sampling frequency, for example, a frequency eight times the modulation frequency (for example, 70 kHz) (for example, sampling at 560 kHz). Frequency), and the sampled zero-cross AC signal is A / D converted and output as an I-D signal (digital value amplitude data). Similarly, the ADC 7 samples the Q-B signal input from the BPF 5 at the predetermined sampling frequency, performs A / D conversion of the input zero-cross AC signal, and generates a Q-D signal. Output.

  The multiplier 8 multiplies the ID signal by a predetermined weight coefficient α and outputs a multiplication result. Similarly, the multiplier 9 multiplies the Q-D signal by a predetermined weight coefficient β and outputs a multiplication result. The adder 12 adds and synthesizes the I-D signal and the Q-D signal multiplied by the weighting factors α and β, respectively, and reproduces the waveform as digital data (hereinafter, digital waveform). Then, the determiner 13 compares the digital waveform with a reference reference waveform, determines whether it is “H (+1)” level data or “L (−1)” level data, and performs time series. The matched filter 14 outputs a determination result, that is, “1” as bit data if “H”, and “0” as bit data if “L”.

  Further, the determiner 13 obtains the average amplitude value of the digital waveform of the training signal sent from the transponder 100 at the start of communication, and outputs the amplitude difference between the average amplitude value and “0 V” to the adaptive control unit 10. To do. Furthermore, the determiner 13 similarly outputs the amplitude difference between the amplitude average value of the training signal and “0 V” to the adaptive control unit 11. Each time the amplitude difference is input, the adaptive control unit 10 sets a new weighting factor so as to minimize the amplitude difference (to maximize the S / N ratio) based on the previously output weighting factor α. α is generated and output to the multiplier 8. Similarly, each time the amplitude difference is input, the adaptive control unit 11 generates a new weighting factor β based on the weighting factor β output last time so as to minimize this amplitude difference, and the multiplier 9 Output to. That is, the S / N ratio is maximized by the weighting coefficients α and β obtained by the adaptive control units 10 and 11, the influence of noise in the synthesized signal is eliminated, and both are virtually in-phase components (real components) as digital Additive synthesis is possible as a waveform.

  Here, in the above description, the adaptive control units 10 and 11 perform the above-described processing using an LMS (Least Mean Squares) adaptive control algorithm, but the present invention is not limited to this, and other types such as RLS (Recursive Least Squares). An adaptive control algorithm can also be used. The training signal described above is a zero-cross signal in which the “H (+1)” level and the “L (−1)” level are arranged in time series and periodically. Ideally, when the average value is calculated, This is a data signal having a multi-bit configuration that is “0”.

  The input and analysis of the training signal described here is performed only when communication between the transponder 100 and the base station 101 is started. As described above, when the weighting factors α and β are determined in the adaptive control units 10 and 11 by analyzing the training signal, the weighting factors α and β are used until the end of communication. Each does not calculate a new weighting factor. That is, in the actual analysis of the baseband signal, the adder 12 adds and combines the ID signal and the QD signal obtained by multiplying the weighting coefficients α and β obtained from the training signal by the multipliers 8 and 9. Do.

  Next, after the predetermined weighting factors α and β are set by the training signal, the determiner 13 adds and synthesizes the I-D signal and the Q-D signal multiplied by the weighting factors α and β. Data, that is, waveform data that is actually transmitted information is input from the adder 12. Then, the determiner 13 determines the signal level of the waveform data, that is, whether it exceeds the reference value “0”, that is, the “H (+1)” level, or is “0” or less, that is, Whether the level is “L (−1)” is determined in synchronization with the sampling frequency, and the determination results are output in time series.

  Further, the adder 12 can add and synthesize the amplitude value when the in-phase component (I signal) and the quadrature component (Q signal) are close to the maximum SN ratio (signal-to-noise ratio). By the load modulation at 100, the modulated real and imaginary two impedance components can be reproduced as a one-dimensional signal (amplitude of the digital waveform) on the time axis. That is, the determiner 13 corrects the amplitude-modulated (and phase-modulated) waveform so that the S / N ratio between the I signal and the Q signal is maximized, and the amplitude of the Q component is also used as a real component of the amplitude modulation. , The input waveform is extracted as a real composite value.

  Next, the matched filter 14 obtains an inner product (comparison is obtained by comparison) between the digital waveform input in time series and a preset coding pattern in synchronization with the sampling frequency. That is, the matched filter 14 obtains the correlation of the waveform shape of the coding pattern and the digital waveform. If the digital waveform has the same shape as the coding pattern, the matched filter 14 has a positive correlation and a positive correlation signal ( For example, “H” level) is output. Here, the matched filter has a FIFO (first in first out) type register that sequentially shifts each data of the digital waveform to be compared in synchronization with the sampling frequency, and in synchronization with the sampling frequency, The inner product of each data of the digital waveform in the register and the data at the corresponding position of the coding pattern is calculated (correlation is obtained), and the respective shapes are compared. Here, for example, the encoding pattern is set as a pattern that changes from the “H” level of the Manchester code to the “L” level.

  On the other hand, the matched filter 14 compares the encoded pattern with the digital waveform, and if the correspondence between “H” and “L” between the encoded pattern and the digital waveform is reversed, it is determined that the correlation is negative. A negative correlation signal (for example, “L” level) is output as the peak of. Here, the matched filter 14 detects the encoding period from the period of the peak of the result of the inner product, that is, the positive correlation signal and the negative correlation signal, and compensates the frequency offset (deviation) of the digital waveform.

In the matched filter 14, by setting the peak level of the correlation value to a predetermined value, if a certain degree of correlation between the digital waveform and the coding pattern is obtained, data can be reproduced, and variations in modulation frequency can be prevented. The shared range can be expanded.
Then, the decoding processing unit 15 decodes the Manchester code based on the correlation peak value (H or L) from the matched filter 14 to obtain a bit data string, and determines the information transmitted from the transponder 100.

  With the configuration of the receiving unit described above, the S / N ratio of the input waveform can be improved as shown in FIG. In this figure, the S / N ratio of the I signal (data used in the conventional determination) is indicated by a broken line (■), the S / N ratio of the Q signal is indicated by a one-dot chain line (♦), and the digital waveform (I The S / N ratio of the combined waveform of the signal and the Q signal is indicated by a solid line (●). Compared to the conventional case where only the in-phase amplitude is used, when the frequency modulation by the amplitude modulation occurs, when the 1 MHz modulation occurs at the resonance frequency of 13.56 MHz, the present invention is the conventional example, that is, the I signal. Compared to the case of using only S, the S / N ratio is improved by 15 dB.

  As shown in FIG. 3, it is assumed that a transponder 100, for example, an RF-ID, is passed between the coils 104 of the base station 101, that is, between the gates (Master, Slave). At this time, the coil surface of the RF-ID coil 103 is perpendicular to the magnetic field between the gates (x-axis direction) as in the case of “A (parallel to the y-axis-x-axis surface)”, that is, the coil of the gate If the coil surfaces of the coil 104 and the RF-ID coil 103 are parallel to each other, a sufficient magnetic flux density is obtained on the coil surface, energy is efficiently transmitted, and a sufficient S / N ratio can be obtained. Here, the coil surface of the gate coil 104 is parallel to the x-axis-z-axis surface.

However, as in the case of “B (parallel to x-axis and z-axis)” and “C (parallel to x-axis and y-axis)”, it is parallel to the magnetic field (x-axis direction), that is, the coil of the gate. If the coil surface between the coil 104 and the RF-ID coil 103 becomes nearly vertical, the magnetic flux density on the coil surface cannot be obtained sufficiently, the energy transmission becomes inefficient, and the S / N ratio is greatly deteriorated. Depending on the situation, an area where the signal detection itself is impossible occurs, and the passage of the RF-ID cannot be recognized.
Since the transponder (RF-ID) is affixed to an object to be detected, such as a book, the angle of the coil surface of the coil 103 changes at any time with respect to the magnetic field depending on how the person holds it.

  In the data transfer system according to the embodiment of the present invention, even when the magnetic field and the coil surface are not perpendicular to each other as in “B” and “C” described above and only an insufficient amplitude change is obtained, As described above, the amplitude of each component of the I signal and the Q signal is added to substantially increase the real number component and the baseband signal is reproduced. Compared to the conventional example using only the I signal component. Then, it becomes possible to widen the allowable range with respect to the variation of the resonance frequency of the transponder, and to greatly improve the S / N ratio. -Recognize the passage of ID, that is, improve the detection rate.

  In the data transfer system according to the embodiment of the present invention, the encoding pattern of a signal transmitted by frequency modulation is a Manchester code, and bit data is encoded by a combination of H level and L level. Even if there is a slight frequency deviation, there is an effect that a positive / negative correlation can be easily obtained by the matched filter.

Next, in addition to the configuration of the first embodiment, the data transfer device according to the second embodiment gives each of the delays corresponding to the different deviations shown in FIGS. 8 and 9 to the digital waveform from the adder 12, Correlators 23 ₀ to 23 _m in a plurality of offset estimation circuits 21 that output correlation values by autocorrelation, and the deviation of the modulation frequency having the highest correlation is extracted from the estimated correlation values, and the modulation frequency having this deviation is extracted. A selection circuit 24 that outputs an estimated frequency signal; a timing demodulation circuit 22 that performs fine adjustment of the frequency of the estimated frequency signal by comparing the estimated frequency signal from the selection circuit 24 with the digital waveform delayed by the delay circuit; have.

Next, a configuration as the second embodiment will be described below.
In order to remove the surrounding interference signals (electromagnetic waves generated by nearby fluorescent lamps, personal computers, etc.) and improve the SN ratio of the received signal, BPFs 4 and 5 in FIG. 1 are centered on the modulation frequency in the transponder 100. It is conceivable to narrow the band.
As a result, the SN ratio is improved, but higher-order components are removed from the waveform of the received signal. Therefore, when the received signal is decoded, the resonance frequency (timing for sampling and decoding) has a frequency deviation. It is difficult to achieve synchronization in decoding in accordance with the modulation frequency in the transponder 100.

Therefore, the base station according to the second embodiment has a configuration in which the synchronization circuit shown in FIG. 8 is interposed between the adder 12 and the determiner 13 shown in FIG.
As a result, the synchronization circuit can give the matched filter 14 decoding timing corresponding to the modulation frequency (including deviation) of the received signal.
Therefore, by narrowing the BPFs 4 and 5 to improve the S / N ratio, the defect that the synchronization at the time of decoding is difficult to achieve with respect to the deviation of the modulation frequency in the transponder 100 at the resonance frequency of the received signal and the base station is improved. It becomes possible to do.

Hereinafter, the synchronous circuit inserted between the adder 12 and the determiner 13 will be described with reference to FIG. FIG. 8 is a block diagram showing a configuration example of the synchronization circuit.
The synchronization circuit includes a delay sample circuit 20 that delays the synthesized signal (digital waveform) D input from the adder 12 in units of frames of the received signal including the training signal, and a modulation frequency (that is, a resonance frequency) of the received signal. An offset estimation circuit 21 that estimates the deviation and a timing demodulation circuit 22 that matches the phase of the received signal and the decoding timing are provided.
In the base station 101 according to the second embodiment, the configuration other than the synchronization circuit is inserted between the adder 12 and the determiner 13 is the same as that of the first embodiment. Will not be described except for different operations.

Next, the operation of the synchronization circuit will be described with reference to FIGS.
FIG. 9 is a block diagram showing a configuration example of the offset estimation circuit 21 in FIG. 8, and FIG. 10 is a block diagram showing a configuration example of the timing demodulation circuit 22 in FIG.
The delay sample circuit 20 corresponds to the modulation frequency of the received signal, obtains the decoding timing in the matched filter 14, and then uses the combined signal input from the adder 12 at least for the frame ( The delay is made by the interval of the length of the training signal added to the actual data of the received signal. For example, if the training signal is 8 bits and each bit is detected by 14 sampling pulses corresponding to the resonance frequency, the sampling signal is delayed by a time interval of 112 cycles and output as a delayed signal DS. become.

On the other hand, the combined signal D is input from the adder 12 to the offset estimation circuit 21 without being delayed.
Here, the offset estimation circuit 21 includes a plurality of correlators (23 ₀ to 23 _m ) including a delay unit (25 _{0 to} 25 _m ), a multiplier (26), and an integration unit (27).
Each of the delay units delays the input composite signal D by a time proportional to a predetermined multiple of the sampling pulse period.
For example, when 0 ≦ m ≦ 4 based on the length of the sampling pulse of 14 cycles, five correlators 23 ₀ , 23 ₁ , 23 ₂ , 23 ₃ , and 23 ₄ are included in the offset estimation circuit. 21.
Here, the training signal is a signal of 8 bits, that is, 4 pulses of “H” and “L” data continuously.

Here, 12 cycles for correlating values virtually to correspond to the frequency deviation of the modulation frequency, in the correlator _{23 0,} 23 _1, 23 _2, 23 3, 23 _4, respectively, the delay unit 25 ₀ sampling pulses min shift, the delay unit 25 ₁ is shifted 13 cycles of the sampling pulse, shifted 14 periods of the delay unit 25 ₂ sampling pulse, the delay unit 25 ₃ is shifted 15 cycles of the sampling pulse, a delay unit 25 ₄ of the sampling pulse Shift 16 cycles.
Each multiplier 26 multiplies the input composite signal D by the composite signal D delayed by each of the delay units 25 _{0 to} 25 ₄ to obtain the area of the overlapping portion, and multiplies this area. The training pulse is added by the accumulating unit 27.

Thus, the correlator 23 ₀ as long in pulse width is 12 cycles and outputs the maximum correlation value, the correlator 23 ₁ outputs the maximum correlation value as long in pulse width is 13 cycles, correlation vessel 23 ₂ as long a pulse width of 14 cycles and outputs the maximum correlation value, the correlator 23 ₃ pulse width to output the maximum correlation value as long at 15 cycles, the correlator 23 ₄ If the pulse width is 16 cycles, the maximum correlation value is output.

Here, each of the correlators 23 ₀ to 23 ₄ obtains an integrated value as an autocorrelation according to a delayed state in 8 bits of the training signal, so that it is compared with 7 bits (for comparison with the delayed one). (Corresponding to the first one bit does not overlap) is accumulated (the overlapping time of each bit), and this is output as a correlation value.
At this time, when the correlators 23 ₀ to 23 ₄ complete the integration process of the overlapping portion for 7 bits, the correlator 23 ₀ to 23 ₄ outputs an end signal to the selection circuit 24 assuming that the correlation value calculation process is completed.
Therefore, among the correlation values output from each correlator, the deviation of the correlator that outputs the largest correlation value has a period width closest to the frequency deviation of the received signal.

Therefore, the deviation of the modulation frequency of the received signal at (+/−) 10% (13 cycles / 15 cycles) and (+/−) 20% (12 cycles / 16 cycles) with 14 sampling pulses as a standard value. Can be detected.
When the end signal is input from each of the correlators 23 ₀ to 23 ₄ , the selection circuit 24 selects the correlator having the highest numerical value from the correlation values input from these correlators, and this correlation An estimated timing signal SG having a period of the number of sampling pulses corresponding to the counter is output to the timing demodulation circuit 22.
At this time, the selection circuit 24 detects the end of the training signal by inputting an end signal indicating the end of the correlation calculation process from each correlator, and starts the fine adjustment process for the timing demodulation circuit 22. A fine adjustment start signal for requesting is output.

The timing demodulation circuit 22 includes a multiplier 32, an LMS unit 31, and an MLSE (maximum likelihood sequence estimation) unit 30, optimizes an input estimated timing signal SG, and outputs a delay output from the delay sample circuit 20. The signal is synchronized, and the synchronized timing is output to the matched filter 14 as a timing signal TG.
Here, the MLSE unit 30 calculates the difference between the estimated timing signal SG generated from the training signal (a replica of the training signal in the received signal) and the delayed signal DS (the delayed training signal) input from the delay sample circuit 20. The frequency value to be multiplied by the estimated timing signal SG to be given to the multiplier 32 (transversal filter) is estimated by the LMS unit 31 and the LMS algorithm so that the square value which is the absolute value of the difference is minimized. The frequency value is output to the multiplier 32 as the adjusted frequency value γ.

Then, the multiplier 32 multiplies the estimated timing signal SG by the adjusted frequency value γ input from the LMS unit 31, performs fine adjustment of the frequency deviation, and supplies the finely adjusted estimated timing signal SG to the MLSE unit 30. Output.
At this time, the fine adjustment processing of the frequency deviation of the LMS unit 31 and the MLSE unit 30 described above is performed over 8 bits of the training signal, and corresponds to the modulation frequency in the transponder 100, that is, the timing corresponding to the bit period of the received signal. The signal TG is output.
Then, when the fine adjustment process using the training signal is completed, the MLSE unit 30 fixes the adjustment frequency value γ to the LMS unit 31, ends the fine adjustment process, and converts the actual data after the training signal into a digital waveform. The result is output to the determiner 13 as DD.

As a result, the matched filter 14 synchronizes with the input timing signal TG, and as previously described, the digital filter DD (that is, the delay signal DS) output from the timing demodulation circuit 22 is preset and internally set in advance. An inner product with the encoded pattern is obtained, and a correlation value as decoded data is output.
As described above, the data transfer system according to the second embodiment of the present invention obtains a rough modulation frequency deviation by the offset estimation circuit 21 in addition to the effects of the first embodiment, and sets the decoding timing to this modulation frequency. Therefore, the timing demodulation circuit 22 finely adjusts the adjusted decoding timing to obtain the timing signal TG, so that the decoding timing of the received signal is obtained with high accuracy by using a small number of training signals. It is possible to narrow the BPF frequency band, improve the S / N ratio of the received signal, and perform decoding processing in accordance with the timing of the received signal.

  Note that a program for realizing the functions of the base station 101 (first and second embodiments) in FIG. 1 is recorded on a computer-readable recording medium, and the program recorded on the recording medium is stored in the computer system. The received signal from the transponder 100 may be decoded by reading and executing. The “computer system” here includes an OS and hardware such as peripheral devices. The “computer system” includes a WWW system provided with a homepage providing environment (or display environment). The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Further, the “computer-readable recording medium” refers to a volatile memory (RAM) in a computer system that becomes a server or a client when a program is transmitted via a network such as the Internet or a communication line such as a telephone line. In addition, those holding programs for a certain period of time are also included.

  The program may be transmitted from a computer system storing the program in a storage device or the like to another computer system via a transmission medium or by a transmission wave in the transmission medium. Here, the “transmission medium” for transmitting the program refers to a medium having a function of transmitting information, such as a network (communication network) such as the Internet or a communication line (communication line) such as a telephone line. The program may be for realizing a part of the functions described above. Furthermore, what can implement | achieve the function mentioned above in combination with the program already recorded on the computer system, and what is called a difference file (difference program) may be sufficient.

It is a block diagram which shows the structural example of the receiving part in the base station of the data transfer apparatus by one Embodiment of this invention. It is a graph which shows the improvement of S / N ratio by this invention (a horizontal axis | shaft: Resonance frequency (MHz), a vertical axis | shaft: S / N ratio (dB)). It is a conceptual diagram for demonstrating the relationship of the coil surface of a gate and RF-ID. It is a block diagram which shows the structural example of the receiving part in the base station of the conventional data transfer apparatus. It is a conceptual diagram for demonstrating the structural example of a passive transponder (for example, RF-ID) and a base station. It is a conceptual diagram explaining the trimming which adjusts the dispersion | variation in the capacity | capacitance at the time of manufacture in the conventional transponder. It is a block diagram which shows the other structural example of the receiving part in the base station of the conventional data transfer apparatus. It is a block diagram which shows one structural example of the synchronous circuit by the 2nd Embodiment of this invention. FIG. 9 is a block diagram illustrating a configuration example of an offset estimation circuit 21 in FIG. 8. FIG. 9 is a block diagram illustrating a configuration example of a timing demodulation circuit 22 in FIG. 8.

Explanation of symbols

2 ... coupler, 3 ... quadrature detector, 4,5 ... BPF (bandpass filter), 6,7 ... ADC (A / D converter), 8,9 ... multiplier DESCRIPTION OF SYMBOLS 10,11 ... Adaptive control part, 12 ... Adder, 13 ... Determinator, 14 ... Matched filter, 15 ... Decoding processing part, 20 ... Delayed sample signal, 21 * ..Offset estimation circuit, 22... Timing demodulation circuit, 23 ₀ , 23 _m ... Correlator, 24... Selection circuit, 25 ₀ , 25 _m . 27 ... Integrating unit, 30 ... MLSE unit, 31 ... LMS unit, 100 ... Transponder, 101 ... Base station, 103,104 ... Coil

Claims (15)

A data transfer system composed of a passive transponder and a base station,
The passive transponder performs transmission of information by carrier load modulation,
The base station is
A quadrature detector that quadrature-detects the load-modulated carrier wave and outputs an I signal and a Q signal;
An A / D converter that samples each of the I signal and the Q signal at a predetermined frequency and converts them into digital data;
An adder that adds and synthesizes the digital data of each of the I signal and the Q signal and outputs a digital waveform;
An extraction unit for extracting the information from the digital waveform;
A data transfer system comprising: An adaptive control unit for obtaining a weighting coefficient to be multiplied to each of the digital data so as to maximize the S / N ratio of the combined digital waveform signal;
A multiplier for multiplying each of the digital data by the weighting factor;
The data transfer system according to claim 1, further comprising:   The data transfer system according to claim 1, wherein the load modulation is capacitive load modulation.   4. The data transfer system according to claim 1, wherein the A / D converter generates digital data at a sampling frequency that is twice or more of a modulation frequency. 5. The extraction unit is
A determiner for determining a signal level of the digital waveform in synchronization with the sampling frequency;
In synchronization with the sampling frequency, a matched filter that obtains an inner product of a preset coding pattern and the digital waveform, and decodes the information as a peak value;
5. The data transfer system according to claim 1, further comprising:   6. The data transfer system according to claim 5, further comprising a synchronization circuit that obtains a deviation of the modulation frequency, extracts a decoding timing based on the deviation, and controls the decoding timing in the matched filter. The synchronization circuit comprises:
A plurality of offset estimation circuits that apply delays corresponding to different deviations to the digital waveform and output correlation values by autocorrelation;
A selection circuit that extracts a deviation of the modulation frequency having the highest correlation from the correlation value and outputs an estimated frequency signal of the modulation frequency having the deviation;
A timing demodulation circuit that performs fine adjustment of the frequency of the estimated frequency signal by comparing the estimated frequency signal from the selection circuit with the digital waveform delayed by the delay circuit;
The data transfer system according to claim 6, further comprising:   The data transfer system according to any one of claims 1 to 7, wherein the transponder includes a sheet-like planar coil and is attached to an object to be detected. A control program for a base station that performs wireless communication with a passive transponder,
A first step of receiving information transmitted by load modulation of a carrier wave from the transponder and outputting an I signal and a Q signal by performing quadrature detection;
A second step of sampling each of the I signal and the Q signal output in the first step at a predetermined frequency and converting them to digital data, respectively;
A third step of adding and synthesizing the digital data of each of the I signal and Q signal converted in the second step, and outputting them as a digital waveform;
A fourth step of extracting the information from the digital waveform output in the third step;
A base station control program that causes a computer to execute. A fifth step of calculating a weighting factor by which each digital data is multiplied so as to maximize the S / N ratio of the combined digital waveform signal;
A sixth step of multiplying each of the digital data converted in the second step by the weighting factor calculated in the fifth step;
The base station control program according to claim 9, further comprising:   11. The method according to claim 9, wherein in the first step, information transmitted by capacitive load modulation of a carrier wave is received from the transponder, and an I signal and a Q signal are output by performing quadrature detection. The base station control program described.   In the second step, each of the I signal and Q signal output in the first step is sampled at a sampling frequency that is twice or more of a modulation frequency, and converted into digital data, respectively. The base station control program according to any one of claims 9 to 11. A seventh step of determining the signal level of the digital waveform in synchronization with the sampling frequency;
An eighth step of obtaining an inner product of a preset coding pattern and the digital waveform in synchronization with the sampling frequency, and decoding the information as a peak value;
The base station control program according to any one of claims 9 to 12, for causing the computer to further execute.   The base according to claim 13, further comprising: calculating a deviation of the modulation frequency; extracting a decoding timing based on the deviation; and causing the computer to further execute a ninth step of controlling the decoding timing in the eighth step. Station control program. A tenth step of applying delays corresponding to different deviations to the digital waveform and outputting a plurality of correlation values by autocorrelation;
An eleventh step of extracting a deviation of the modulation frequency having the highest correlation from the plurality of correlation values output in the tenth step, and outputting an estimated frequency signal of the modulation frequency having the deviation;
A twelfth step in which the estimated frequency signal output in the eleventh step is compared with the delayed digital waveform to finely adjust the frequency of the estimated frequency signal;
15. The base station control program according to claim 14, further comprising:

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