JP2004364410A - Switching power supply circuit - Google Patents

Abstract

For example, in a switching power supply circuit corresponding to a load power of 250 W or more, an improvement in power factor and a reduction in power loss are achieved.
A quadruple voltage rectifier circuit for obtaining a DC input voltage at a level approximately four times the level of a commercial AC power supply suppresses a peak level of a current flowing through a switching element to reduce switching loss.
At this time, of the four smoothing capacitors constituting the quadruple voltage rectifier circuit, the capacitance of two smoothing capacitors that can obtain a rectified smoothed voltage at a level approximately equal to that of the commercial AC power supply is determined by the capacitance of these smoothing capacitors. It is set so as to be larger than the capacitances of the other two smoothing capacitors that can obtain a rectified smoothed voltage at a level approximately twice that of the commercial AC power supply in which the charged electric charge and the commercial AC power supply are superimposed. As a result, a greater difference can be given to the capacitances of the respective rectification current paths formed during the period when the commercial AC power supply has the positive / negative polarity, and the rectification currents to flow simultaneously in the respective rectification current paths can be temporally reduced. And the conduction angle of the rectified current is increased to improve the power factor.
[Selection diagram] Fig. 1

Description

[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.
[0002]
[Prior art]
In recent years, with the development of switching elements capable of withstanding relatively large currents and voltages at high frequencies, most power supply circuits that rectify commercial power and obtain a desired DC voltage are switching-type power supply circuits. .
The switching power supply circuit is used as a power source for various electronic devices as a high-power DC-DC converter while increasing the switching frequency to reduce the size of a transformer and other devices.
[0003]
FIG. 10 shows an example of a switching power supply circuit configured based on the invention previously filed by the present applicant. The power supply circuit shown in this drawing has a configuration corresponding to the conditions of a load power Po = 250 W or more and an AC input voltage VAC = 100 V system.
[0004]
In the power supply circuit shown in FIG. 10, first, a common mode choke coil CMC and a filter capacitor CL are connected to the commercial AC power supply AC as shown in the drawing to remove harmonics superimposed on the commercial AC power supply AC. Filter to be formed.
[0005]
A rectifier circuit system for rectifying the commercial AC power supply AC to obtain a rectified smoothed voltage Ei (DC input voltage) includes two rectifier diodes D1 and D2 and two smoothing capacitors Ci1 and Ci2. By connecting these parts as shown in the figure, a voltage of 1 Ev corresponding to the same-level level as that of the commercial AC power supply AC is obtained as the voltage between both ends of the smoothing capacitors Ci1 and Ci2. As the rectified smoothed voltage Ei (DC input voltage) obtained at both ends of the smoothing capacitors Ci1-Ci2 connected in series, a level of 2Ev corresponding to twice the AC input voltage VAC is obtained.
That is, in this case, a voltage doubler rectifier circuit is formed.
For example, when the AC input voltage VAC is a 100 V system and the AC input voltage is at a level equal to the AC input voltage VAC under a relatively heavy load condition of load power Po = 250 W or more, the peak current flowing to the subsequent switching element It has been found that the power loss increases accordingly. Therefore, if a DC input voltage corresponding to a level twice as high as the AC input voltage VAC is set by the voltage doubler rectifier circuit as described above, the level of the peak current flowing through the switching element can be suppressed.
[0006]
Then, the rectified and smoothed voltage Ei obtained by rectifying and smoothing the commercial AC power supply AC is input as a DC input voltage to a subsequent primary-side switching converter.
In this case, a separately-excited current resonance type converter is provided as a switching converter that performs a switching operation by inputting the DC input voltage Ei. This current resonance type converter includes two switching elements Q1 and Q2 as shown in the figure.
[0007]
In this case, MOS-FETs are selected as the switching elements Q1 and Q2, and the switching elements Q1 and Q2 are connected as shown to form a switching circuit based on a so-called half-bridge coupling system. .
Clamp diodes DD1 and DD2 are connected in parallel to the switching elements Q1 and Q2 in the direction shown in the figure.
Further, among the switching elements Q1 and Q2, a partial resonance capacitor Cp for partial voltage resonance is connected in parallel to the switching element Q2.
[0008]
These switching elements Q1 and Q2 are driven by a drive circuit 6 shown. The drive circuit 6 is activated by electric power input via an activation resistor RS shown in the figure.
[0009]
The insulating converter transformer PIT is provided for transmitting the switching output of the primary side switching converter to the secondary side.
Although illustration is omitted here, the insulating converter transformer PIT includes, for example, an EE-type core, and a primary winding N1 and a secondary winding N are formed on a center magnetic leg of the EE-type core by using a bobbin or the like. N2 is wound so as to ensure an insulating state.
In addition, a gap having a width of, for example, about 1.5 mm to 2.0 mm is formed in the center magnetic leg of the EE type core, for example, so that the coupling coefficient k between the primary winding N1 and the secondary winding N2 is increased. A state of loose coupling of about 0.8 is obtained. This prevents occurrence of abnormal oscillation at the time of intermediate load.
[0010]
One end of the primary winding N1 of the insulating converter transformer PIT is grounded to the primary side ground. The other end is connected between the source and the drain of the switching elements Q1 and Q2 via the series resonance capacitor C1. By being connected in this manner, the switching outputs of the switching elements Q1 and Q2 are transmitted to the primary winding N1.
Further, depending on such a connection form, the primary winding N1 and the series resonance capacitor C1 are connected in series. However, the primary side is determined by the leakage inductance of the primary winding N1 and the capacitance of the series resonance capacitor C1. A series resonance circuit will be formed. With this primary-side series resonance circuit, the switching operation of the switching elements Q1 and Q2 is made a current resonance type.
[0011]
For the secondary winding N2 of the insulating converter transformer PIT, a center tap is provided as shown in the drawing to connect to the secondary side ground, and then a double-wave rectification composed of rectifier diodes DO1, DO2 and a smoothing capacitor CO. Circuit is connected. As a result, the secondary DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load (not shown), and is also branched and input as a detection voltage for the control circuit 5 described below.
[0012]
The control circuit 5 obtains, as a control output, a current or voltage whose level is varied according to the level of the secondary DC output voltage EO, for example. This control output is supplied to the drive circuit 6.
The drive circuit 6 alternately turns on / off a high-side drive signal and a low-side drive signal to be supplied to the switching elements Q1 and Q2 according to the input control output level. While maintaining the timing, the operation is performed so that the frequency of each drive signal is changed in a synchronized state.
Thus, the switching frequency of the switching elements Q1 and Q2 is variably controlled according to the control output level (that is, the secondary DC output voltage EO level) supplied from the control circuit 5.
By changing the switching frequency, the resonance impedance of the primary side series resonance circuit changes. By changing the resonance impedance in this way, the amount of current supplied to the primary winding N1 of the primary-side series resonance circuit changes, and the power transmitted to the secondary side also changes. As a result, the level of the secondary side DC output voltage EO changes, and constant voltage control is performed.
[0013]
FIG. 11 is an operation waveform diagram of the switching converter in the power supply device shown in FIG. Note that FIG. 11 shows experimental results when the AC input voltage VAC = 100 V and the load power Po = 350 W.
First, in FIG. 11, a period in which the drain-source voltage VQ2 of the switching element Q2 has the positive peak level Vp and a period in which the positive element has the zero level alternately appear. / Off timing. That is, it is turned off during the period of the peak level Vp, and turned on during the period of the 0 level.
Along with this, the drain current IQ2 of the switching element Q2 flows according to the illustrated waveform during a period when the drain-source voltage VQ2 is at the peak level Vp, and becomes zero level during a period when the drain-source voltage VQ2 is at the zero level. ing.
Also, the drain current IQ1 of the switching element Q1 is 180 ° out of phase with the drain current IQ2 of the switching element Q2. That is, switching is performed such that the switching elements Q1 and Q2 are alternately turned on / off.
[0014]
In this figure, the peak level Vp of the voltage VQ2 is set to 260V. In this case, the period in which the drain-source voltage VQ2 of the switching element Q2 is Vp and the period in which it is at the 0 level have a period of 5 μs. This cycle corresponds to the switching frequency obtained under the conditions of AC input voltage VAC = 100 V and load power Po = 350 W.
The peak levels (Ap1, Ap2 in the figure) of the drain currents IQ1, IQ2 of the switching elements Q1, Q2 are both 5.5 A.
[0015]
The primary side series resonance current I1 is obtained by combining the drain currents IQ1 and IQ2 of the switching elements Q1 and Q2. From this, the peak level (Ap-p in the figure) of the primary-side series resonance current I1 is 5.5 Ap + 5.5 Ap = 11 Ap-p.
Further, the secondary side DC output voltage EO is made to be a constant voltage as described above, and becomes constant at, for example, 200 V as shown in the figure.
[0016]
Here, as shown in FIG. 10, when the input stage is constituted by the voltage doubler / half-wave rectifier circuit, it is known that the power factor becomes low.
FIG. 12 shows the waveforms of the AC input voltage VAC, the AC input current IAC, and the DC input voltage Ei in the circuit of FIG. 10 when the AC input voltage VAC = 100 V and the load power Po = 350 W. When the stage is constituted by the voltage doubler / partial rectification circuit, the waveform of the AC input current IAC shown in the figure becomes a pointed shape compared to the waveform of the AC input voltage VAC. That is, at this time, there is a large difference in the conduction angle between the AC input current IAC and the AC input voltage VAC, and the power factor is thereby reduced.
The power factor at this time is about 0.58.
The peak level of the AC input current IAC at this time is 15 Ap as illustrated. The peak level of the AC input voltage VAC is 141 V, and the ripple component ΔEi of the DC input voltage Ei is ΔEi = 23 V with respect to the average level 260 V of the DC input voltage Ei as shown in the figure.
The power conversion efficiency ηAC → DC at this time is about ηAC → DC = 92%.
[0017]
Depending on the decrease of the power factor, the reactive power increases, and there is a possibility that electronic devices connected to the commercial AC power supply may be adversely affected.
Therefore, conventionally, a power choke coil is added to improve the power factor.
The circuit shown in FIG. 13 is obtained by adding a power choke coil for improving the power factor as the configuration of FIG. 10 corresponding to the condition of the system of the load power Po = 250 W or more and the AC input voltage VAC = 100 V.
As can be seen from the comparison with FIG. 10, in the power supply circuit shown in FIG. 13, the power choke coil PCH is inserted in series with the line of the commercial AC power supply AC based on the configuration of FIG. ing. With this power choke coil PCH, the power in the frequency band of the commercial AC power supply is smoothed, and the conduction angle of the AC input current IAC is enlarged. This improves the power factor to about PF = 0.75.
[0018]
FIG. 14 shows waveforms of the AC input voltage VAC, the AC input current IAC, and the DC input voltage Ei in the power supply circuit shown in FIG. 13 when the AC input voltage VAC = 100 V and the load power Po = 350 W.
As can be seen by comparing FIG. 14 with the previous FIG. 12, in the circuit of FIG. 10 described above, the peak level of the AC input current IAC was 20 A with respect to the peak level Vp = 141 V of the AC input voltage VAC. On the other hand, in the circuit shown in FIG. 13, the peak level of the AC input current IAC is reduced to 15A.
That is, in this case, in the circuit shown in FIG. 13, the conduction angle of the AC input current VAC is larger than that of the circuit of FIG. 10, and approaches the conduction angle of the AC input voltage VAC. The power factor is improved in the circuit shown in FIG. 13 accordingly.
[0019]
The operation waveforms of the switching converter of the power supply circuit shown in FIG. 13 are almost the same as those shown in FIG.
However, at this time, 240 V is obtained as the peak level Ap of the drain-source voltage VQ2 of the switching element Q2. The peak level Ap1 of the drain current of the switching element Q1 is 6 Ap, and the peak level Ap2 of the drain current of the switching element Q2 is also 6 Ap.
The current level Ap-p between the positive and negative peaks of the primary side series resonance current I1 in this case is 6Ap + 6Ap = 12Ap-p.
The AC → DC power conversion efficiency at this time is ηAC → DC = 90.3%.
[0020]
Here, in obtaining the above-described experimental results, each part of the power supply circuit shown in FIG. 10 was selected as follows.
・ Smoothing capacitor Ci = 1000μF
・ Insulated converter transformer PIT: Ferrite core of EER42, gap G = 1mm
Primary winding N1 = 30T (turn), Secondary winding N2 = 50T (25T + 25T at the center tap)
-Series resonance capacitor C1 = 0.033 µF
[0021]
In the circuit shown in FIG. 13, each part was selected as follows.
・ Smoothing capacitor Ci = 1000μF
・ Insulated converter transformer PIT: Ferrite core of EER42, gap G = 1mm
Primary winding N1 = 28T (turn), Secondary winding N2 = 50T (25T + 25T at the center tap)
-Series resonance capacitor C1 = 0.039 µF
-Power choke coil PCH (inductance L) = 5 mH
[0022]
In addition, as another conventional technique related to the present invention, for example, the following patent documents can be cited.
[Patent Document]
JP-A-9-117137
[Problems to be solved by the invention]
As described above, in the conventional power supply circuit shown in FIG. 13, the power factor is improved by inserting the power choke coil PCH in series with the AC line.
However, by providing the power choke coil PCH in this way, the inductance component becomes a DC resistance, and power loss occurs.
For example, in the circuit of FIG. 13, the inductance L is set to about 5 mH as described above in order to obtain a power factor PF = 0.75 on the assumption that the power supply harmonic distortion regulation is cleared. A DC resistance of about 2Ω is generated.
In addition, the provision of the power choke coil PCH causes iron loss and copper loss here.
As a result, in the circuit shown in FIG. 13, the AC → DC power conversion efficiency decreases by 1.7% from ηAC → DC = 92.0% in FIG. DC is reduced to 90.3%.
[0023]
In support of this, the peak levels (Ap1, Ap2) of the drain currents of the switching elements Q1, Q2 in the circuit of FIG. 13 shown in FIG. 11 are 5.5 Ap in the case of the previously described circuit shown in FIG. On the other hand, it has increased to 6 Ap.
Further, as can be seen from a comparison between FIGS. 12 and 14, the average value of the DC input voltage Ei is reduced to about 240 V from 260 V in the circuit of FIG. Further, the ripple component ΔEi of the DC input voltage Ei at this time has also increased from ΔEi = 23V in the circuit of FIG. 10 to ΔEi = 24V.
[0024]
In addition, such a power choke coil PCH is a relatively large component among components constituting a power supply circuit, and providing such a component hinders miniaturization and weight reduction of the circuit.
For example, as described above, in the circuit shown in FIG. 13, the inductance L is set to about 5 mH, but the power choke coil PCH in this case weighs about 300 g, thereby increasing the circuit weight and the circuit area. Had to be forced.
[0025]
[Means for Solving the Problems]
In view of the above problems, the present invention is configured as a switching power supply circuit as follows.
That is, first, a rectifying / smoothing circuit that receives a commercial AC power supply and generates a rectified / smoothed voltage, and a switching converter unit that receives the rectified / smoothed voltage as a DC input voltage and causes a switching element to perform a switching operation. .
And a first rectifier diode element for rectifying the commercial AC power and a rectified output of the first rectifier diode element being charged during a period in which the commercial AC power is negative. A first rectified current path including a first smoothing capacitor that generates a voltage between both ends of a level approximately equal to that of the commercial AC power supply;
A second rectifier diode for rectifying the commercial AC power on which the voltage across the first smoothing capacitor is superimposed during a period in which the commercial AC power has a positive polarity; and a rectified output of the second rectifier diode. Is charged, and a second rectifying current path including a second smoothing capacitor that generates a voltage between both ends of the commercial AC power supply at a level approximately twice as high as that of the commercial AC power supply;
During the period in which the commercial AC power supply has a positive polarity, a third rectifier diode element for rectifying the commercial AC power supply and a rectified output of the third rectifier diode element are charged, so that the commercial AC power supply is approximately A third rectifying current path comprising a third smoothing capacitor for generating a voltage at both ends of the same level;
A fourth rectifying diode element for rectifying the commercial AC power on which the voltage across the third smoothing capacitor is superimposed during a period in which the commercial AC power has a negative polarity, and a rectified output of the fourth rectifying diode element And a fourth rectifying current path including a fourth smoothing capacitor that generates a voltage between both ends of the commercial AC power supply at a level approximately twice that of the commercial AC power supply.
Then, by forming a capacitor series circuit in which the second smoothing capacitor and the fourth smoothing capacitor are connected in series, the voltage across the capacitor series circuit is approximately four times as high as that of the commercial AC power supply. And the first smoothing capacitor has a larger capacitance than the fourth smoothing capacitor, and the third smoothing capacitor has a larger capacitance than the second smoothing capacitor. Also had a large capacitance.
[0026]
According to the above configuration, the switching power supply circuit of the present invention includes a rectification / smoothing circuit that receives a commercial AC power supply and generates a rectified / smoothed voltage, and a switching converter that receives and inputs the rectified / smoothed voltage and operates. You.
In addition, the rectifying and smoothing circuit is configured to generate a rectified and smoothed voltage having a level corresponding to approximately four times the commercial AC power supply level. That is, it is configured as a quadruple voltage rectifier circuit. In the quadruple voltage rectifier circuit according to the present invention, two systems of rectified current paths of the first and fourth rectified current paths are formed during the period when the AC input voltage is negative, and similarly, during the period when the AC input voltage is positive. , The second and third rectified current paths are formed.
In the rectifying and smoothing circuit, the first smoothing capacitor has a larger capacitance than the fourth smoothing capacitor, and the third smoothing capacitor has a larger capacitance than the second smoothing capacitor. Have been. Due to this capacitance difference, it is possible to cause a shift in the phase of the rectified current flowing through the two rectified current paths in each of the positive and negative periods of the AC input voltage.
[0027]
BEST MODE FOR CARRYING OUT THE INVENTION
FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention. For example, the power supply circuit shown in this figure is, like the power supply circuits shown in FIGS. 10 and 13 described above, a load power Po = 250 W or more and corresponds to an AC 100 V system as an AC input voltage.
First, in the power supply circuit shown in FIG. 1, a noise filter including two filter capacitors CL and a common mode choke coil CMC is connected to the line of the commercial AC power supply AC. As this noise filter, a filter capacitor CL, a common mode choke coil CMC, and a filter capacitor CL are connected in parallel in this order to a commercial AC power supply AC.
This noise filter removes harmonics superimposed on the commercial AC power supply AC.
[0028]
A rectifying / smoothing circuit including rectifying diodes D1, D2, D3, and D4 and smoothing capacitors Ci1a, Ci1b, Ci2a, and Ci2b is provided downstream of the noise filter.
In this rectifying / smoothing circuit, the smoothing capacitor Ci1a has its negative terminal connected to the positive line of the commercial AC power supply AC through the winding of the common mode choke coil CMC, so that the positive line of the commercial AC power supply AC is connected. Has been inserted against. The anode of the rectifier diode D1 and the cathode of the rectifier diode D2 are connected to the positive terminal of the smoothing capacitor Ci1a.
The anode of the rectifier diode D2 is connected to the negative terminal of the commercial AC power supply AC, thereby being connected to the negative line of the commercial AC power supply AC.
[0029]
The smoothing capacitor Ci2a has a positive terminal connected to the negative terminal of the smoothing capacitor Ci1a, and a negative terminal connected to the cathode of the rectifier diode D4 and the anode of the rectifier diode D3. The anode of the rectifier diode D4 is connected to the primary side ground as shown, and the cathode of the rectifier diode D3 is connected to the anode of the rectifier diode D2 to be connected to the negative line of the commercial AC power supply AC. You.
[0030]
The smoothing capacitors Ci1b and Ci2b are connected in series and inserted between the cathode of the rectifier diode D1 and the primary side ground as shown in the figure. The connection point of the smoothing capacitors Ci1b and Ci2b is connected to the connection point of the rectifier diodes D2 and D3, thereby being connected to the negative line of the commercial AC power supply AC.
Then, the voltage between both ends of the smoothing capacitors Ci1b and Ci2b connected in series is set as the rectified smoothed voltage of this power supply circuit, and is supplied to the subsequent switching converter as the operating power.
In this case, a low-speed recovery type diode is selected as the rectifier diodes D1 to D4.
[0031]
The rectifying and smoothing circuit having the above configuration is configured as a quadruple voltage rectifying circuit that generates a rectified and smoothed voltage Ei (V4) having a level substantially equal to four times the peak level of the AC input voltage VAC. The operation will be described below.
[0032]
FIG. 2 is a diagram for explaining an operation obtained in the rectifying / smoothing circuit shown in FIG. 1. FIG. 2A shows a rectifying current path during a period when the AC input voltage VAC has a negative polarity. (B) shows the rectified current paths during the period when the AC input voltage VAC has a positive polarity.
First, during the period in which the AC input voltage VAC is negative, the rectified current is [commercial AC power supply AC → smoothing capacitor Ci2b → rectifier diode D4 → smoothing capacitor Ci2a → commercial AC power supply, as indicated by the arrow in FIG. AC] (fourth rectified current path).
That is, as described later, during this period, the rectified output of the rectifier diode D4 inserted in the rectified current path is charged to the smoothing capacitor Ci2b.
[0033]
As shown in the figure, the rectified current is branched and also flows through a path (first rectified current path) of [commercial AC power supply AC → rectifier diode D2 → smoothing capacitor Ci1a → commercial AC power supply AC]. When a rectified current flows through the path, the rectified output of the rectifying diode D2 is charged in the smoothing capacitor Ci1a. As a result, a voltage at both ends corresponding to the level (1 Ev) of the AC input voltage VAC is generated in the smoothing capacitor Ci1a.
[0034]
Subsequently, during the period in which the AC input voltage is positive, as shown in FIG. 2B, the rectified current is [commercial AC power supply AC → smoothing capacitor Ci1a → rectifier diode D1 → smoothing capacitor Ci1b → commercial AC power supply AC). ] (Second rectified current path), and charges the rectified output of the rectifier diode D1 to the smoothing capacitor Ci1b.
Here, due to the rectification operation in the half cycle in which the AC input voltage VAC has the positive polarity shown in FIG. 2A, the voltage (1Ev) across the smoothing capacitor Ci1a corresponding to the equal-level level of the AC input voltage VAC. It has been obtained. Therefore, at this time, depending on the formation of the current path as described above, a rectified current flows such that the voltage across the smoothing capacitor Ci1a is superimposed on the AC input voltage VAC. As a result, the smoothing capacitor Ci1b can obtain a voltage (2Ev) at both ends corresponding to twice the level of the AC input voltage VAC.
[0035]
Also, during the period in which the AC input voltage VAC has a positive polarity, the rectified current also flows through a path of [commercial AC power supply AC → smoothing capacitor Ci2a → rectifier diode D3 → commercial AC power supply AC] (third rectified current path). The smoothing capacitor Ci2a is charged with the rectified output of the rectifier diode D3 inserted in this current path. That is, during this period, a rectified smoothed voltage having a level of 1 Ev corresponding to the same-level level of the AC input voltage VAC is obtained at both ends of the smoothing capacitor Ci2a.
Then, when the AC input voltage reaches the next half cycle in which the polarity becomes negative, the rectified current becomes [commercial AC power supply AC → smoothing capacitor Ci2b → rectifier diode D4 → smoothing capacitor Ci2a, as shown in FIG. → Commercial AC power supply AC] flows through the fourth rectified current path. That is, by forming such a current path, a rectified current flows by superimposing the voltage of 1Ev obtained on the smoothing capacitor Ci2a on the AC input voltage VAC as described above. In this period, as described above, the rectified output of the rectifier diode D4 is charged to the smoothing capacitor Ci2b. Therefore, both ends of the smoothing capacitor Ci2b correspond to the level twice the AC input voltage VAC. A rectified smoothed voltage based on the level of 2Ev can be obtained.
[0036]
By repeating the operation corresponding to each of the positive / negative periods of the AC input voltage VAC, a rectified smoothed voltage of 2Ev can be obtained at each end of the smoothing capacitor Ci1b and the smoothing capacitor Ci2b. As a result, the level of the rectified smoothed voltage Ei obtained at both ends of these smoothing capacitors Ci1b-Ci2b connected in series is as follows.
2Ev + 2Ev = 4Ev
Thus, a quadruple voltage rectifier circuit that can obtain a rectified smoothed voltage Ei represented by 4Ev is formed. In this case, since the AC input voltage is set to the AC 100 V system, a DC voltage of the 400 V system is obtained as the rectified smoothed voltage Ei, and is supplied to the subsequent switching converter.
[0037]
Here, in the present embodiment, as described above, the smoothing capacitors Ci1a and Ci2a that can obtain the voltage across the level corresponding to 1Ev, and the voltage across the 1Ev and the AC input voltage VAC (corresponding to 1Ev) are superimposed. The relationship between the capacitances of the smoothing capacitors Ci1b and Ci2b that can obtain the voltage across the level of 2Ev is set as follows.
First, as for the smoothing capacitors Ci1a and Ci2b in which the AC input voltage VAC is on the rectified current path during the period of the positive polarity, the capacitance of the smoothing capacitor Ci1a is set to be larger than that of the smoothing capacitor Ci2b. I do.
Similarly, as for the smoothing capacitors Ci2a and Ci1b in which the AC input voltage VAC is in the rectified current path during the period of the negative polarity, the capacitance of the smoothing capacitor Ci2a is set to be larger than that of the smoothing capacitor Ci1b. It shall be.
Then, in the present embodiment, the capacitance of Ci1a = Ci2a is obtained by a set of smoothing capacitors that can obtain a voltage between both ends of a level corresponding to 1Ev and a set of smoothing capacitors that can obtain a voltage between both ends of a level corresponding to 2Ev. , Ci1b = Ci2b.
For example, in this case, as the smoothing capacitors Ci1a and Ci2a, a capacitance that is 10 times or more that of the smoothing capacitors Ci1b and Ci2b is selected.
In this case, the withstand voltage of the smoothing capacitor Ci1a and the smoothing capacitor Ci2a is set to 200V, and the withstand voltage of the smoothing capacitor Ci1b and the smoothing capacitor Ci2b is set to 400V.
[0038]
As a switching converter that operates by inputting the DC input voltage Ei generated by the rectifying and smoothing circuit, a separately excited current resonance configured by half-bridge coupling of two switching elements Q1 and Q2 as shown in FIG. With a shape converter.
Here, as shown in the figure, two switching elements Q1 (high side) and Q2 (low side) of a MOS-FET are connected by half-bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the respective drains and sources of the switching elements Q1 and Q2 in the direction shown.
In this case, the withstand voltage of the switching elements Q1 and Q2 is 800V.
[0039]
A partial resonance capacitor Cp is connected in parallel between the drain and the source of the switching element Q2. A partial voltage resonance circuit is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. In other words, a partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1 and Q2 are turned off is obtained.
[0040]
A drive signal is applied to the gates of the switching elements Q1 and Q2 by the control circuit 1 including an oscillation circuit, a control circuit, a protection circuit, and the like for driving the switching elements Q1 and Q2 in a separately excited manner.
In this control circuit 1, an oscillation signal of a required frequency is generated by an internal oscillation circuit. The oscillation circuit varies the frequency of the oscillation signal according to the level of the control signal obtained according to the level of the secondary-side DC output voltage EO as described later.
The control circuit 1 uses the oscillation signal generated by the oscillation circuit to generate a high-side drive signal and a low-side drive signal using rectangular waves. Then, the high-side drive signal generated in this manner is supplied to the switching element Q1, and the low-side drive signal is supplied to the switching element Q2.
[0041]
According to the above description, the high-side drive signal is applied to the switching element Q1. As a result, a waveform corresponding to the high-side drive signal is obtained as the gate-source voltage of the switching element Q1. In other words, within one switching cycle, a period in which a rectangular pulse having a positive polarity is generated and a period in which 0 V is obtained are obtained.
Then, by the gate-source voltage, the switching element Q1 is first turned on at a timing at which a positive-polarity rectangular wave pulse is obtained within one switching cycle. That is, in order for the switching element Q1 to be turned on, it is necessary to apply a voltage of an appropriate level equal to or higher than the gate threshold voltage (≒ 5 V), for example. Since the voltage between the gate and the source as the positive pulse is set to 10 V, a state in which the pulse is turned on corresponding to the period in which the positive pulse is applied is obtained. When the voltage between the gate and the source becomes 0 V and becomes equal to or lower than the gate threshold voltage, the state is switched to the off state. With such timing, the switching element Q1 performs a switching operation so as to be turned on / off.
[0042]
On the other hand, a low-side drive signal is applied to switching element Q2. The gate-source voltage of the switching element Q2 obtained in response to the drive signal has the same waveform as the gate-source voltage of the switching element Q1, and the timing is the same as the gate-source voltage. As a result, a waveform having a phase difference of 180 ° can be obtained.
From this, the switching element Q2 is switched and driven by the timing of turning on / off alternately with the switching element Q1.
That is, in this way, depending on the control IC2, the switching elements Q1 and Q2 are controlled to be turned on / off alternately.
[0043]
The insulating converter transformer PIT transmits the switching output of the switching elements Q1 and Q2 to the secondary side, and is wound with a primary winding N1 and a secondary winding N2.
One end of the primary winding N1 of the insulating transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the illustrated primary parallel resonance capacitor C1. The other end is connected to the primary side ground.
Here, a primary side series resonance circuit is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1. Then, as described above, since the primary-side series resonance circuit is connected to the switching output point, the switching output of the switching elements Q1 and Q2 is transmitted to the primary-side series resonance circuit. In the primary side series resonance circuit, resonance operation is performed in accordance with the transmitted switching output, whereby the operation of the primary side switching converter is of a current resonance type.
[0044]
According to the above description, as the primary-side switching converter shown in this figure, the operation as a current resonance type by the primary-side series resonance circuit (L1-C1) and the partial voltage resonance circuit (Cp // L1) described above A voltage resonance operation is obtained.
In other words, the power supply circuit shown in this figure adopts a form in which a resonance circuit for making the primary-side switching converter a resonance type is combined with another resonance circuit. In this specification, such a switching converter is referred to as a composite resonance converter.
[0045]
Although not described here, the structure of the insulating converter transformer PIT includes, for example, an EE-type core obtained by combining an E-type core made of a ferrite material. Then, after the winding site is divided on the primary side and the secondary side, the primary winding N1 and the secondary winding N2 described below are wound around the center magnetic leg of the EE type core. .
[0046]
A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage according to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2.
For the secondary winding N2, a center tap is provided and connected to the secondary side ground, and then a double-wave rectification circuit including rectifier diodes DO1 and DO2 and a smoothing capacitor CO is connected as shown in the figure. I have. As a result, the secondary DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary DC output voltage EO is supplied to a load (not shown), and is also branched and input as a detection voltage for the control circuit 1 described above.
[0047]
The control circuit 1 obtains, as a control signal, a current or voltage whose level is varied according to the level of the secondary DC output voltage EO, for example.
Then, in accordance with the level of the control signal, the timing for alternately turning on / off the high side drive signal to be output to the switching element Q1 and the low side drive signal to be output to the switching element Q2 is determined. After being kept, it operates so that the frequency of each drive signal can be varied in a synchronized state.
Accordingly, the switching frequency of the switching elements Q1 and Q2 is variably controlled according to the level of the control signal (that is, the level of the secondary DC output voltage EO).
By changing the switching frequency, the resonance impedance of the primary side series resonance circuit changes. By changing the resonance impedance in this way, the amount of current supplied to the primary winding N1 of the primary-side series resonance circuit changes, and the power transmitted to the secondary side also changes. As a result, the level of the secondary side DC output voltage EO changes, and the stabilization is achieved.
[0048]
FIG. 3 is a waveform chart showing the operation of the power supply circuit according to the present embodiment having the above configuration. This figure shows the experimental results when the AC input voltage VAC = 100 V and the load power Po = 350 W.
In order to obtain the experimental results shown in FIG. 3, each part is selected as follows.
・ Insulated converter transformer PIT: Ferrite core of EER-42, gap G = 1mm
Primary winding N1 = 40T (turn), Secondary winding N2 = 50T (25T + 25T at the center tap)
-Series resonance capacitor C1 = 0.018 µF
Smoothing capacitor Ci1a = Smoothing capacitor Ci2a = 3900 μF
・ Smoothing capacitor Ci1b = smoothing capacitor Ci2b = 390 μF
In this figure, the illustrated time point t1 indicates zero cross timing when the level of the AC input voltage VAC changes from a positive polarity to a negative polarity. In addition, a time point t4 indicates zero cross timing when the polarity changes from negative polarity to positive polarity. That is, in this case, the time point t1 to the time point t4 indicate a half cycle in which the AC input voltage VAC has a negative polarity, and a period from this time point t4 to the next time point t1 indicates a half cycle in which the AC input voltage VAC has a positive polarity.
[0049]
The illustrated rectified currents ID1, ID2, ID3, and ID4 indicate currents flowing through the rectified diodes D1 to D4 shown in FIG. 1, respectively.
The potential V1 shown in FIG. 1 indicates a potential between the connection between the smoothing capacitor Ci1a and the rectifier diode D1 and the primary side ground, as shown in FIG.
The potential V2 is a potential between the connection point between the rectifier diode D4 and the smoothing capacitor Ci2a and the primary side ground. In this case, the potential V2 is regarded as a sum of the potential of the smoothing capacitor Ci2a and the potential of the AC input voltage VAC. be able to.
The potential V3 is a potential between the connection point of the smoothing capacitors Ci1b and Ci2b and the primary side ground, and indicates a voltage across the smoothing capacitor Ci2b.
The potential V4 indicates the voltage (rectified smoothed voltage Ei) across the series connection circuit of the smoothing capacitors Ci1b-Ci2b.
[0050]
First, in the half cycle from time t1 to time t4 when the AC input voltage VAC has a negative polarity, the rectifier diodes D2 and D4 conduct during a period in which the level of the potential V2 is higher than the level of the potential V3.
In this case, the potential V2 is obtained as the sum of the potential of the smoothing capacitor Ci2a and the potential of the AC input voltage VAC as described above. As described above with reference to FIG. 2, the smoothing capacitor Ci1a is charged during the period in which the AC input voltage VAC corresponding to the periods t1 to t4 has a positive polarity, so that the commercial AC power AC A voltage at both ends is generated at a level of 1 Ev corresponding to the equal-level of.
From this, in the period t1 to t4, the potential V2 rises to the level of 2Ev as the level of the AC input voltage VAC becomes the negative peak level as shown in FIG. As a result, a waveform falling to the level of 2Ev is obtained.
On the other hand, since the potential V3 is a voltage between both ends of the smoothing capacitor Ci2b, a waveform in which a ripple component is superimposed on the basis of the vicinity of the illustrated 2Ev as a rectified smoothed voltage is obtained.
During the half period from the time point t1 to the time point t4, the potential V3 falls below the level of 2Ev, and the potential V2 rises to near the level of 2Ev according to the level transition of the AC input voltage VAC as described above, and V3 <V2 During this period, the rectifier diodes D2 and D4 become conductive (periods t2 to t3 in the figure).
[0051]
In this way, the rectifier diodes D2 and D4 conduct, so that the rectifier currents ID2 and ID4 flow during the period from the time point t2 to the time point t3.
In the case of the present embodiment, as the rectified currents ID2 and ID4, the rectified current ID4 flows earlier in time and the rectified current ID2 flows later in time as shown in the figure.
[0052]
Here, in the present embodiment, the reason why the rectified current ID2 flows with a time delay with respect to the rectified current ID4 is as follows.
First, as described in FIG. 1, the capacitance of the smoothing capacitors Ci1a and Ci2a is set to be larger than the capacitance of the smoothing capacitors Ci1b and Ci2b.
In this case, since the smoothing capacitor Ci2a and the smoothing capacitor Ci2b are inserted in the fourth rectifying current path where the rectifier diode D4 performs rectification, the capacitance in this path is the sum of the smoothing capacitor Ci2b and the smoothing capacitor Ci2a. The capacitance is obtained as a series connection.
Therefore, as the capacitance in the fourth rectified current path,
Ci2b × Ci2a / Ci2b + Ci2a
Will be represented by
On the other hand, the capacitance in the first rectification current path where the rectification diode D2 performs rectification is based on the capacitance of only the smoothing capacitor Ci1a (= Ci2a) because only the smoothing capacitor Ci1a is inserted here. It becomes.
Therefore, the capacitance of the fourth rectified current path and the capacitance of the first rectified current path are as follows.
(Ci2b × Ci2a / Ci2b + Ci2a) <Ci1a
Will be represented by
Further, in the present embodiment, the capacitance of the smoothing capacitors Ci1a and Ci2a is set to be about 10 times the capacitance of the smoothing capacitors Ci1b and Ci2b. The magnitude of the capacitance of the current path is based on this capacitance difference, and is further increased by a factor of ten.
[0053]
Here, as the capacitance increases, the phase of the current flowing therethrough is delayed. That is, in this case, the phase of the current flowing through the first rectified current path is relatively delayed as compared with the phase of the fourth rectified current path.
Therefore, as a relative relationship, the rectified current flows earlier in the fourth rectified current path, and the rectified current flows later in the first rectified current path.
[0054]
Thus, in the period from t2 to t3 in which the rectified current flows during the period when the AC input voltage VAC is negative, the rectified current ID4 flows earlier in time and the rectified current ID2 flows later in time as shown in FIG. Become like
As the rectified currents ID4 and ID2 flow at such timings, these rectified currents are combined. The AC input current IAC flowing during the period when the commercial AC power supply AC has the negative polarity is as shown in the figure. Thus, a substantially M-shaped waveform having two peaks corresponding to the waveforms of the currents ID4 and ID2 is obtained.
[0055]
Another reason is as follows.
For example, as described with reference to FIG. 2A, during the half period in which the AC input voltage VAC has a positive polarity, the smoothing capacitor Ci2a is charged. Therefore, when the next AC input voltage shifts to a period in which the AC input voltage has a negative polarity, the charge is sufficiently accumulated in the smoothing capacitor Ci2a. In this state, the voltage based on the polarity that discharges the smoothing capacitor Ci2a The application will be applied.
On the other hand, since one of the smoothing capacitors Ci1a has been discharged in the positive half cycle, when the AC input voltage VAC shifts to the negative polarity, the charge is reduced. Here, a voltage is applied according to the polarity for charging.
Thus, when the AC input voltage VAC shifts to a negative polarity, the fourth rectified current path in which the smoothing capacitor Ci2a is inserted is more rectified than the first rectified current path in which the smoothing capacitor Ci1a is inserted. Current flows more easily.
Thus, in the period t2 to t3 in which the rectified current flows during the period when the AC input voltage VAC has the negative polarity, the rectified current flows faster in the fourth rectified current path and the other first rectified current path is better. , A slow flowing operation is obtained.
[0056]
Next, a half cycle from the time point t4 to the next time point t1 when the AC input voltage VAC has a positive polarity will be described.
During the period from the time point t4 to the time point t1, the rectifier diodes D1 and D3 conduct while the illustrated level of the potential V1 exceeds the level of the potential V4.
At this time, the potential V1 can be viewed as a composite value of the voltage between both ends of the smoothing capacitors Ci1a and Ci2b and the level of the AC input voltage VAC.
As described above, the smoothing capacitor Ci1a is charged with the rectified current in the half cycle (period t1 to t4) immediately before the half cycle of the period t4 to t1. That is, at the time point t4 when the AC input voltage VAC shifts to the region of positive polarity, a potential of 1Ev level is obtained at both ends of the smoothing capacitor Ci1a.
Further, as the voltage across the smoothing capacitor Ci2b, a waveform that changes near the level of 2Ev as a rectified smoothed voltage is obtained as described above.
Therefore, as the potential V1, at the time point t4 when the AC input voltage VAC becomes the 0 level, the level of 3Ev is obtained by the voltage across the smoothing capacitors Ci1a and Ci2b. Then, as the AC input voltage VAC rises to the peak level in the positive polarity region, a level of 4Ev is obtained as shown in the figure.
On the other hand, since the potential V4 is a voltage between both ends of the series connection circuit including the smoothing capacitors Ci1b and Ci2b, a waveform in which a ripple is superimposed on the basis of 4Ev is obtained as a smoothed output of the quadruple voltage rectifier circuit. ing.
In the period from the time point t4 to the time point t1, the level of the potential V1 becomes a level corresponding to 4Ev as the level of the AC input voltage VAC increases, and during the period in which the level of the potential V4 becomes lower than 4Ev, The rectifier diodes D1 and D3 become conductive (period t5 to t6).
[0057]
Thus, depending on the conduction of the rectifier diodes D1 and D3, the rectifier currents ID1 and ID3 flow during the period from the time point t5 to the time point t6.
At this time, as can be seen with reference to FIG. 2B, the capacitance of the second rectified current path is based on the series connection of the capacitances of the smoothing capacitors Ci1a and Ci1b, and is set as the third rectified current path. Is based on only the smoothing capacitor Ci2a.
Also in this case, the rectification current is charged to one smoothing capacitor Ci1a in the immediately preceding half cycle, so that the rectification current flows through the second rectification current path including the smoothing capacitor Ci1a. This is a state in which it is easy to flow (see FIG. 2A). Further, since the other smoothing capacitor Ci2a discharges, the path on the side of the rectifier diode D3 is in a state where the rectified current hardly flows.
For this reason, even in the period from t5 to t6 in which the rectified current flows during the period in which the AC input voltage VAC has the positive polarity, the rectified current ID1 flows earlier in time and the rectified current ID3 flows later. It becomes.
As a result, a substantially M-shaped waveform having two peaks corresponding to the waveforms of these currents ID1 and ID3 is obtained as the AC input current IAC flowing during the period when the AC input voltage VAC is positive. It will be something that can be done.
[0058]
As can be understood from the above description, in the present embodiment, the input stage in the switching power supply circuit is constituted by the quadruple voltage rectifier circuit, so that two rectified current paths are provided for each half cycle of the AC input voltage VAC. Is formed.
Then, the capacitance of the smoothing capacitors Ci1a and Ci2a is made larger than the capacitance of the smoothing capacitors Ci1b and Ci2b, so that the capacitance of one of these two paths is larger than that of the other. By increasing the size, rectified currents that should flow simultaneously in these two rectified current paths are caused to flow with a time difference.
Thus, the conduction period of the rectified current flowing during each half cycle of the AC input voltage VAC can be extended as compared with the case where the rectified current flows simultaneously in the two rectified current paths. Accordingly, the peak level of the AC input current IAC is suppressed, and the conduction angle of the AC input current IAC is increased, so that the power factor can be improved.
In the conventional circuit shown in FIG. 10, the peak level of the AC input current IAC is 20 Ap, but in this example, it is reduced to 10 Ap as shown in FIG.
[0059]
By the way, according to the above description of FIG. 3, as the difference between the capacitances of the smoothing capacitors Ci1a and Ci2b with respect to the smoothing capacitors Ci1b and Ci2b increases, the time difference between the periods in which the rectification current flows through the two rectification current paths relatively increases. Can be expanded. That is, a larger power factor can be obtained. However, as a practical problem, as the capacitance difference increases, components having a large capacitance must be selected as the smoothing capacitors Ci1a and Ci2a, resulting in an increase in size and cost.
Therefore, in the present embodiment, in consideration of such a realistic situation, the magnitude relationship between the capacitances of the smoothing capacitors Ci1a and Ci2a with respect to the smoothing capacitors Ci1b and Ci2b is set to 10 times.
In this embodiment, as described above, the capacitance of the smoothing capacitors Ci1a and Ci2a is set to 3900 μF, and the capacitance of the smoothing capacitors Ci1b and Ci2b is set to 390 μF, whereby an appropriate power factor of about 0.7 is obtained. Is to be obtained. Naturally, the magnitude relationship between the capacitances of the smoothing capacitors Ci1a and Ci2a with respect to the smoothing capacitors Ci1b and Ci2b is not limited to 10 times, but includes the value of the power factor PF actually required. It may be changed according to various conditions described above.
[0060]
Further, in the present embodiment, since the input stage is a quadruple voltage rectifier circuit, the drain current flowing through the switching elements Q1 and Q2 is larger than in the case of the conventional voltage doubler rectifier circuit shown in FIGS. Can be reduced.
According to an experiment, a result of 5.5 Ap in the case of the circuit of FIG. 10 including the conventional voltage doubler rectifier circuit was obtained to be 2.8 Ap in the present example. Further, this is greatly reduced as compared with 6.0 Ap in the case of the circuit of FIG.
[0061]
Further, since the drain current flowing through each switching element is reduced and the switching loss is reduced, the primary side series resonance current is 11 Ap-p in the circuit of FIG. 10 and 12 Ap-p in the circuit of FIG. However, in this example, a result of 5.6 Ap was obtained.
[0062]
In addition, due to the reduction of the switching loss, the ripple component ΔEi of the DC input voltage Ei becomes ΔEi = 23 V in the circuit of FIG. 10, ΔEi = 24 V in FIG. 13 and ΔEi = 22 V in the present example. Reduced results were obtained.
[0063]
Further, as the AC → DC power conversion efficiency, ηAC → DC = 92.0% in the conventional circuit shown in FIG. 10, whereas ηAC → DC = 93.8% in this example, which is 1.8. % Can be obtained. At this time, an experimental result that the AC input power can be reduced by 7.3 W as compared with the circuit of FIG. 10 was obtained.
In addition, in comparison with the circuit of FIG. 13, the result is improved from ηAC → DC = 90.3% to 3.5%.
[0064]
According to this example as described above, when compared with the circuit of FIG. 10, it is possible to significantly reduce the power loss and improve the power conversion efficiency while greatly improving the power factor.
Further, when compared with the circuit of FIG. 13, the power factor can be reduced to about the same level without providing the power choke coil PCH, and the power conversion efficiency can be greatly improved.
[0065]
Here, as described above, depending on the power supply circuit of the first embodiment, the power factor is improved to about 0.70, and a power factor similar to that of the circuit including the power choke coil PCH shown in FIG. 13 is obtained. However, at this time, as shown in FIG. 3, the waveform of the AC input current IAC is substantially M-shaped. For this reason, for example, the distortion of the power supply harmonic component of the seventh or higher order becomes relatively large.
In order to suppress the distortion of the power supply harmonic component of the seventh or higher order, a power choke coil PCH is provided in the configuration of FIG. 1 as shown in FIG. It is conceivable to respond to this.
[0066]
FIG. 4 is a circuit diagram showing a configuration example of a switching power supply circuit according to a second embodiment of the present invention.
In this figure, the parts already described with reference to FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted. Further, in this case, the following description will be continued assuming that each unit is selected in the same manner as in FIG.
In the switching power supply circuit according to the second embodiment, one end of the power choke coil PCH is connected to the negative line of the commercial AC power supply AC via a common mode choke coil CMC. The other end is connected to a connection point between the negative terminal of the smoothing capacitor Ci1a and the positive terminal of the smoothing capacitor Ci2a.
That is, the power choke coil PCH is inserted in series with the line of the commercial AC power supply AC.
[0067]
The operation of the power supply circuit according to the second embodiment having the above configuration will be described with reference to FIG.
FIG. 5 is a diagram showing a rectified current path in the rectifying / smoothing circuit of the power supply circuit shown in FIG. 4, and a solid line in the figure indicates a rectified current path formed during a period when the AC input voltage VAC has a negative polarity. The dashed line indicates a rectified current path formed during a period when the AC input voltage VAC has a positive polarity.
As can be seen by comparing the rectified current path shown in this figure with the rectified current path shown in FIG. 2 above, the rectifying and smoothing circuit of the second embodiment is similar to that of the first embodiment. It can be seen that the rectified current is flowing through the path.
In this case, since the power choke coil PCH is inserted in the negative line of the commercial AC power supply AC as described above, the rectification is performed during the period when the AC input voltage VAC is positive / negative. The current flows through the power choke coil PCH.
[0068]
FIG. 6 is a diagram illustrating waveforms of the AC input voltage VAC and the AC input current IAC in the power supply circuit according to the second embodiment. The operation waveforms of the other units shown in FIG. 3 are almost the same as those in the circuit of FIG.
As can be seen by comparing the waveform of the AC input current IAC shown in FIG. 6 with the waveform of the AC input current IAC in the power supply circuit of the first embodiment shown in FIG. Also in the power supply circuit, the peak level of the AC input current IAC is suppressed to about 10 Ap as in the case of the first embodiment.
That is, since the rectifying current flows through the same route after having the same rectifying and smoothing circuit configuration as the circuit of FIG. 1, the peak level of the AC input current IAC is reduced according to the same principle as that of FIG. Thus, the conduction angle is increased accordingly.
[0069]
In addition, in the power supply circuit according to the second embodiment, since the power choke coil PCH is inserted into the negative line of the commercial AC power supply AC as described above, the power The distortion generated in the waveform of the input current IAC is reduced.
That is, depending on the provision of the power choke coil PCH in this manner, the AC input current IAC is generated by the resonance operation of the inductance L of the power choke coil PCH and the capacitance of the four smoothing capacitors Ci provided in the rectifying and smoothing circuit. Will be smoothed. Thus, in the first embodiment, the M-shaped distortion occurs near the peak level of the AC input current IAC, but in the present embodiment, the distortion can be reduced. is there.
[0070]
For confirmation, since the power supply circuit according to the second embodiment has the basic configuration of FIG. 1, the power supply circuit does not include the power choke coil PCH but has a higher power than the circuit of FIG. 10. The rate can be greatly improved.
The power choke coil PCH in this case is inserted for the purpose of reducing the M-shaped distortion generated near the peak level of the AC input current IAC. Therefore, the power choke coil PCH here is It is sufficient if the inductor has a relatively small inductance L that is sufficient to reduce such distortion in the M-shaped portion.
For this reason, as for the power choke coil PCH to be provided in the circuit of the second embodiment, the inductance L can be made smaller as compared with the circuit of FIG. 13 which employs a power factor improving configuration using a power choke coil. it can.
For example, in this case, in order to obtain the waveform of the AC input current IAC shown in FIG. 6, the inductance L may be set to about 1 mH, whereby the weight of the power choke coil PCH in FIG. It is possible to reduce the size and size to about 100 g from 300 g (inductance L = 5 mH) according to the conventional configuration.
[0071]
As described above, according to the second embodiment, the power choke coil PCH can reduce the M-shaped distortion of the AC input current IAC. Can be. Thereby, for example, it is possible to cope with the case where the power supply harmonic distortion regulation value is specified for the seventh or higher order.
At this time, for example, when the inductance L of the power choke coil PCH is set to 1 mH as described above, an experimental result that the power factor PF is about 0.80 has been obtained.
[0072]
Also in this case, since the high DC input voltage Ei is obtained by the quadruple voltage rectifier circuit as in the case of the circuit of FIG. 1, the peak level of the drain current flowing through the switching elements Q1 and Q2, and The ripple component ΔEi generated in the DC input voltage Ei can be reduced, and the power conversion efficiency can be improved.
[0073]
In addition, as described above, in this case, a power factor improving configuration using a quadruple voltage rectifier circuit is employed, and a power choke coil PCH is provided. , The power choke coil PCH can be made smaller and lighter than the circuit of FIG.
Since the power choke coil PCH can be reduced in size and weight in this way, iron loss and copper loss in the power choke coil PCH can be reduced, and power loss can be further reduced as compared with the circuit of FIG. .
[0074]
That is, according to the power supply circuit of the second embodiment, the power supply harmonic distortion regulation can be cleared similarly to FIG. 13 while the power loss is significantly reduced as compared with the case of FIG. is there.
According to the experiment, the AC → DC power conversion efficiency ηAC → DC in the circuit of this example is ηAC → DC = 93.1% when the load power Po = 350 W, which is 90.3% in the case of the circuit of FIG. % Is improved by 2.8%. In addition, the result that the AC input power was reduced by 11.7 W as compared with the circuit of FIG. 13 was obtained.
[0075]
Next, FIG. 7 shows a configuration example of a switching power supply circuit according to a third embodiment.
Note that FIG. 7 shows only the configuration of the input stage in the switching power supply circuit, and the other portions have the same configuration as those in FIGS. I do. Also in FIG. 7, the same reference numerals are given to the parts already described in FIGS. 1 and 4, and description thereof will be omitted.
The power supply circuit according to the third embodiment has a configuration corresponding to the load power Po = 250 W or more as in the circuits of FIGS. 1 and 4, and has a so-called AC input voltage VAC corresponding to a 100 V system and a 200 V system. This realizes a configuration compatible with a wide range.
[0076]
First, in FIG. 3, the power choke coil PCH in this case is divided into an inductor L1 and an inductor L2 as shown by providing a tap output. Then, the terminal t2 of the relay switch S1 is connected to the tap output as shown in the figure.
A terminal t3 of the relay switch S1 is connected to an end of the power choke coil PCH opposite to a connection point with the filter capacitor CL.
In this case, the power choke coil PCH has an inductance L of, for example, 5 mH. At this time, it is assumed that the tap output is applied to a position where the inductance relationship between the inductor L1 and the inductor L2 formed thereby is, for example, inductor L1 = 1 mH and inductor L2 = 4 mH.
[0077]
The relay switch S1 employs a two-contact switch that switches by connecting the terminal t2 and the terminal t3 to the terminal t1 alternatively.
The terminal t1 of the relay switch S2 is connected to a connection point between the rectifier diode D2 and the rectifier diode D3.
[0078]
In this case, the connection point between the rectifier diode D2 and the rectifier diode D3 is connected to the terminal t1 of the relay switch S2.
The relay switch S2 is also switched such that the terminal t2 and the terminal t3 are alternatively connected to the terminal t1. The terminal t2 of the relay switch S2 is connected to a connection point between the smoothing capacitor Ci1b and the smoothing capacitor Ci2b as shown in the figure. The terminal t3 is open.
[0079]
Further, a rectifier circuit switching module 10 is provided for such a configuration.
The rectifying circuit switching module 10 is provided with a rectifying and smoothing circuit including a diode D10 and a smoothing capacitor C10 as shown in the figure. In this rectifying / smoothing circuit, the anode of the diode D10 is connected to one end of the common mode choke coil CMC, so that it is connected to the negative line of the commercial AC power supply AC. The cathode is connected to the positive terminal of the smoothing capacitor C10. The negative terminal of the smoothing capacitor C10 is connected to the primary side ground.
The rectifier circuit switching module 10 has a detection terminal connected to a connection point of the diode D10 and the smoothing capacitor C10, and thereby converts a DC component of the AC input voltage VAC obtained on the negative line of the commercial AC power supply AC. It is possible to detect and input the level of the AC input voltage VAC.
[0080]
The rectifier circuit switching module 10 includes a relay RL as shown in the figure, and controls the conduction of the relay RL to perform the terminal switching control of the relay switch S1 and the relay switch S2.
Such switching of the relay switches S1 and S2 is performed based on the level of the AC input voltage VAC detected as described above.
That is, in this case, in the rectifier circuit switching module 10, the relay RL is turned on, for example, in response to the detected level of the AC input voltage VAC being, for example, 150 V or less, and the terminal t2 is connected to the relay switches S1, S2. To select.
Further, in response to the detected level of the AC input voltage VAC being, for example, 150 V or more, the switching control is performed by turning off the relay RL, for example, and selecting the terminal t2 in the relay switches S1, S2. .
Accordingly, the terminal t2 is selected by the relay switches S1 and S2 in response to the case where 150 V or less is detected as the level of the AC input voltage VAC, and it is determined that the AC input voltage VAC of the 100 V system is obtained. be able to.
Then, when 150 V or more is detected as the level of the AC input voltage VAC, and it is determined that the AC input voltage VAC by the AC 200 V system is obtained, the terminal t3 can be selected by the relay switches S1 and S2. .
[0081]
Here, when the terminal t2 is selected in both the relay switches S1 and S2 corresponding to the AC 100 V system as described above, first, on the relay switch S1 side, only the inductor L1 in the power choke coil PCH is enabled.
In this case, since the inductance of the inductor L1 is 1 mH as described above, the power choke coil PCH in this case has the same inductance as that of the circuit shown in FIG. It can be seen that it is provided.
At this time, on the relay switch S2 side, by selecting the terminal t2, a state in which the connection point of the smoothing capacitors Ci1b and Ci2b is connected to the connection of the rectifier diodes D2 and D3 is obtained. That is, in this case, a quadruple voltage rectifier circuit smoothing circuit having a connection configuration equivalent to that of the circuit of FIG. 4 is formed.
From these facts, in the circuit shown in FIG. 7, a configuration almost equivalent to that of the circuit of FIG. 4 as the second embodiment can be obtained corresponding to the AC 100 V system. In other words, with the circuit shown in FIG. 7, an operation equivalent to that of the second embodiment can be obtained in the case of the AC 100 V system, and as a result, the second embodiment The same effect as in the case of is obtained.
[0082]
On the other hand, when the terminal t3 is selected in the relay switches S1 and S2 as described above corresponding to the AC 200 V system, the series connection of the inductor L1 and the inductor L2 in the power choke coil PCH becomes effective. Thus, in the power choke coil PCH in this case, an inductance L of 5 mH can be obtained.
[0083]
In this case, on the relay switch S2 side, the connection point of the smoothing capacitors Ci1b and Ci2b and the connection point of the rectifier diodes D2 and D3 are cut off (disconnected) by selecting the terminal t3 as described above. You.
In this way, the connection between the connection point of the smoothing capacitors Ci1b and Ci2b and the connection point of the rectifier diodes D2 and D3 is cut off, so that 2Ev is applied to both ends of the series connection circuit including the smoothing capacitors Ci1b and Ci2b. Is obtained, and a voltage doubler rectifying / smoothing circuit capable of obtaining a level corresponding to the above is formed.
[0084]
Here, FIG. 8 shows a rectification current path formed in the circuit of FIG. 7 when the switching is performed by the rectification circuit switching module 10 corresponding to the AC 200 V system.
Also in this figure, the rectified current path during the period when the AC input voltage VAC is negative is indicated by a solid line, and the rectified current path during the period when the AC input voltage VAC is positive is indicated by a broken line.
First, during a period in which the AC input voltage VAC has a negative polarity, the rectified current is as shown by a solid line arrow [commercial AC power supply AC → power choke coil PCH → rectifier diode D2 → smoothing capacitor Ci1a → commercial AC power supply AC]. Flows by route.
Also, during the period in which the AC input voltage VAC is negative, the rectified current is [commercial AC power supply AC → power choke coil PCH → rectifier diode D2 → rectifier diode D1 → smoothing capacitor Ci1b → smoothing capacitor Ci2b → rectifier diode as shown in the figure. D4 → smoothing capacitor Ci2a → commercial AC power supply AC].
[0085]
During the period in which the AC input voltage is positive, as indicated by the broken line arrow in the figure, [commercial AC power supply AC → smoothing capacitor Ci1a → rectifying diode D1 → smoothing capacitor Ci1b → smoothing capacitor Ci2b → rectifying diode D4 → rectifying diode A rectified current flows through a path of D3 → power choke coil PCH → commercial AC power supply AC].
Also in this period, the rectified current branches and flows through the path of [commercial AC power supply AC → smoothing capacitor Ci2a → rectifier diode D3 → power choke coil PCH → commercial AC power supply AC].
[0086]
Here, for the smoothing capacitor Ci1a in this case, as described above, during the period when the AC input voltage VAC is negative, [commercial AC power supply AC → power choke coil PCH → rectifier diode D2 → smoothing capacitor Ci1a → commercial AC Power supply AC]. Then, the charge of the smoothing capacitor Ci1a is transferred during the period of [AC input voltage VAC → smoothing capacitor Ci1a → rectifying diode D1 → smoothing capacitor Ci1b → smoothing capacitor Ci2b →...] During the period when the AC input voltage VAC is positive. Is discharged by Thus, both ends of the series connection circuit formed by the smoothing capacitors Ci1b and Ci2b in this case have both ends based on the 2Ev level obtained by superimposing the 1Ev level based on the charge charged in the smoothing capacitor Ci1a on the 1Ev level based on the AC input voltage VAC. A voltage is obtained.
[0087]
In addition, the smoothing capacitor Ci2a is charged by the above-described route of [commercial AC power supply AC → smoothing capacitor Ci2a → rectifier diode D3 → power choke coil PCH → commercial AC power supply AC] during a period when the AC input voltage VAC has a positive polarity. Will be done. The charge of the smoothing capacitor Ci2a is calculated during the period when the AC input voltage VAC is negative [commercial AC power supply AC → power choke coil PCH → rectifying diode D2 → rectifying diode D1 → smoothing capacitor Ci1b → smoothing capacitor Ci2b →. ..] is discharged. That is, at both ends of the series connection circuit formed by the smoothing capacitors Ci1b and Ci2b in this case, the voltage across the 2Ev level obtained by superimposing the 1Ev level by the charge of the smoothing capacitor Ci2a on the 1Ev level by the AC input voltage VAC. Can be obtained.
[0088]
In this manner, in the circuit of FIG. 7, when the AC input voltage VAC is 200 V, the rectification by the level of 2Ev as the voltage between both ends of the smoothing capacitors Ci1b-Ci2b during the period when the AC input voltage VAC is positive / negative. A smooth voltage is obtained.
Thus, in the rectifying and smoothing circuit in this case, a voltage doubler rectifying operation is performed in which a level corresponding to twice the commercial AC power supply level is obtained as the DC input voltage Ei.
[0089]
By the way, in order to cope with the wide range as described above, by performing the voltage doubler rectification operation corresponding to the AC 200 V system, a power factor improving effect as obtained by the quadruple voltage rectification operation performed in the AC 100 V system is obtained. Can not be done. That is, in this case, the rectified current does not flow with a time lag in each path depending on the rectification operation by the voltage doubler rectifier circuit itself.
[0090]
Therefore, as described above, when the power choke coil PCH is inserted so that the inductance is switched with respect to the commercial AC power supply line, and the voltage doubler rectifier circuit is used, the inductance is increased from 1 mH. It can be switched to 5 mH. Thus, when the voltage doubler rectifier circuit operates, the power factor can be improved to the extent that the inductance of the power choke coil PCH at this time is sufficient. If it is not necessary to particularly improve the power factor when switching to the voltage doubler rectifier circuit, it is not necessary to adopt a configuration for switching the inductance of the power choke coil PCH as shown in the figure. As in the case of the second embodiment, a fixed power choke coil PCH at 1 mH may be provided.
[0091]
FIG. 9 is a diagram showing waveforms of the AC input voltage VAC and the AC input current IAC in the circuit of FIG. This figure shows the experimental results when the AC input voltage VAC = 230 V and the load power Po = 350 W. Also in this case, it is assumed that the same elements as those in the circuit of FIG.
In this case, as the AC input voltage VAC, a peak level of 325 Vp is obtained as shown in the figure. As shown in the figure, a sinusoidal waveform is obtained as the AC input current IAC with respect to such a waveform of the AC input voltage VAC, and its peak level is suppressed to about 4 Ap.
FIG. 9 also shows that the conduction angle of the AC input current IAC is enlarged and the power factor is improved.
[0092]
In this way, depending on the configuration of the third embodiment, a power factor improving configuration with low power loss by quadruple voltage rectification is realized in the AC 100 V system, and the power factor is improved even in the case of the AC 200 V system. Is realized.
According to the experiment, the result that the power factor PF = 0.79 was obtained under the conditions of the AC input voltage VAC = 230 V and the load power Po = 350 W as the power factor PF in the circuit of FIG.
Further, the AC → DC power conversion efficiency ηAC → DC under the same conditions is ηAC → DC = 95.0%, which is 1.5% higher than the conventional configuration corresponding to the 200 V system and having the voltage doubler rectifier circuit. It will improve. At this time, the input power is reduced by 5.9 W. At this time, the DC input voltage Ei was 605 Vp.
[0093]
Note that the present invention is not limited to the configuration of the power supply circuit described above.
In the above embodiment, the switching converter that operates by inputting the rectified smoothed voltage Ei is a separately excited half-bridge current resonance type converter, but may be a self-excited type. Further, instead of the half-bridge method, a full-bridge coupling method including four switching elements may be used. In addition, as the switching element, an element other than the MOS-FET such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor may be employed. Further, the constants of the respective component elements described above may be changed according to actual conditions and the like. Also, the switching converter may be of a type other than the current resonance type, such as a voltage resonance type.
Further, for example, a circuit configuration for generating a secondary-side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
[0094]
【The invention's effect】
As described above, in the present invention, the switching power supply circuit includes the quadruple voltage rectification circuit as the rectification and smoothing circuit for generating the DC input voltage (rectified and smoothed voltage), so that the switching power supply circuit has approximately four times the level of the commercial AC power supply. A DC input voltage is obtained.
In this way, by obtaining a DC input voltage that is approximately four times that of the commercial AC power supply, the current level flowing through the switching element can be reduced by that much compared to the conventional configuration in which the input stage is a voltage doubler rectifier circuit. Switching loss can be reduced. That is, the power loss can be significantly reduced as compared with the conventional circuit.
[0095]
At this time, as the capacitance of the four smoothing capacitors constituting the quadruple voltage rectifier circuit, the capacitance of the two smoothing capacitors for generating a rectified smoothed voltage equal to the commercial AC power supply level is used. By making the capacitance larger than the capacitance of two smoothing capacitors capable of obtaining a rectified smoothed voltage twice as high as the level, the capacitance existing in each rectified current path during the period when the commercial AC power supply has positive / negative polarity Can be set so as to enlarge the difference. As a result, each of the rectification currents flowing through the two rectification current paths, which are formed in each of the periods in which the commercial AC power supply has the positive / negative polarity, can be flowed so as to be shifted in time. The conduction angle of the AC input current flowing to the AC power supply can be increased.
That is, the power factor can be improved without providing a power choke coil as in the related art.
By improving the power factor without using a power choke coil, power loss due to iron loss and copper loss when a power choke coil is provided can be eliminated. The power conversion efficiency is improved.
[Brief description of the drawings]
FIG. 1 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a first embodiment of the present invention;
FIG. 2 is a diagram illustrating a rectification current path in a rectification smoothing circuit provided in the switching power supply circuit according to the first embodiment.
FIG. 3 is a waveform chart showing operation waveforms of respective units in the switching power supply circuit according to the first embodiment.
FIG. 4 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a second embodiment;
FIG. 5 is a diagram illustrating a rectified current path in a rectifying / smoothing circuit provided in a switching power supply circuit according to a second embodiment.
FIG. 6 is a waveform diagram showing waveforms of an AC input voltage and an AC input current in the switching power supply circuit according to the second embodiment.
FIG. 7 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a third embodiment;
FIG. 8 is a diagram illustrating a rectified current path in a rectifying / smoothing circuit provided in a switching power supply circuit according to a third embodiment.
FIG. 9 is a waveform chart showing waveforms of an AC input voltage and an AC input current in the switching power supply circuit according to the third embodiment.
FIG. 10 is a circuit diagram showing a configuration of a conventional switching power supply circuit.
FIG. 11 is a waveform chart showing operation waveforms of various parts in the circuit shown in FIG.
FIG. 12 is a waveform chart showing operation waveforms of respective units in the circuit shown in FIG.
FIG. 13 is a circuit diagram showing a configuration of a switching power supply circuit having another conventional configuration.
FIG. 14 is a waveform chart showing operation waveforms of various parts in the circuit shown in FIG.
[Explanation of symbols]
Reference Signs List 1 control circuit, 10 rectifier circuit switching module, D1-D4 rectifier diode, Ci1a, Ci1b, Ci2a, Ci2b smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, Cp partial resonance capacitor, N1 Primary winding, RL relay, S1, S2 relay switch, CL filter capacitor, CMC common mode choke coil

Claims (4)

A rectifying and smoothing circuit that receives a commercial AC power supply and generates a rectified and smoothed voltage;
Switching converter means for inputting the rectified smoothed voltage as a DC input voltage and performing a switching operation by a switching element,
The rectifying and smoothing circuit includes:
During the period in which the commercial AC power supply has a negative polarity, a first rectifier diode element that rectifies the commercial AC power supply and a rectified output of the first rectifier diode element are charged, so that the commercial AC power supply is approximately A first rectified current path including a first smoothing capacitor that generates a voltage at both ends of a unity level;
A second rectifier diode for rectifying the commercial AC power on which the voltage across the first smoothing capacitor is superimposed during a period in which the commercial AC power has a positive polarity; and a rectified output of the second rectifier diode. Is charged, and a second rectifying current path including a second smoothing capacitor that generates a voltage between both ends of the commercial AC power supply at a level approximately twice as high as that of the commercial AC power supply;
During the period in which the commercial AC power supply has a positive polarity, a third rectifier diode element for rectifying the commercial AC power supply and a rectified output of the third rectifier diode element are charged, so that the commercial AC power supply is approximately A third rectifying current path comprising a third smoothing capacitor for generating a voltage at both ends of the same level;
A fourth rectifying diode element for rectifying the commercial AC power on which the voltage across the third smoothing capacitor is superimposed during a period in which the commercial AC power has a negative polarity, and a rectified output of the fourth rectifying diode element And a fourth smoothing capacitor that generates a voltage between both ends of the commercial AC power supply at a level approximately twice that of the commercial AC power supply.
By forming a capacitor series circuit in which the second smoothing capacitor and the fourth smoothing capacitor are connected in series, a rectified smoothing voltage having a level approximately four times as high as that of the commercial AC power supply as a voltage across the capacitor series circuit. And the first smoothing capacitor has a larger capacitance than the fourth smoothing capacitor, and the third smoothing capacitor has a larger static capacitance than the second smoothing capacitor. A switching power supply circuit having a capacitance. The switching power supply circuit according to claim 1, wherein an inductor having an inductance equal to or less than a predetermined value is inserted in series with the line of the commercial AC power supply. A connection point between the second smoothing capacitor and the fourth smoothing capacitor in the rectifying and smoothing circuit and a negative line of the commercial AC power supply are connected / disconnected in accordance with the level of the input commercial AC power supply. By connecting, a quadruple voltage rectification operation for obtaining a rectified and smoothed voltage of a level approximately four times that of the commercial AC power supply as a voltage across the capacitor series circuit, and a rectification of a level approximately twice that of the commercial AC power supply 2. The switching power supply circuit according to claim 1, further comprising a rectifying operation switching unit that switches between a voltage doubler rectifying operation for obtaining a smoothed voltage. An inductor inserted in series with the line of the commercial AC power supply;
4. The switching power supply circuit according to claim 3, further comprising an inductor switching unit configured to switch the inductance of the inductor to a predetermined value in accordance with a level of the input commercial AC power supply. 5.

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