JP2004080985A - Power supply device, hard disk device and IC using the same - Google Patents

Abstract

Provided is a power supply device that achieves both control loop stabilization and responsiveness.
A power supply device according to the present invention feeds back to an output voltage signal error amplifier by connecting a filter provided separately from a power system LC smoothing filter and a differential amplifier inside the error amplifier. In addition, the power supply device of the present invention may be configured such that the output of the power-system LC smoothing filter is added to a control circuit having upper and lower limit detection, and the transient control is performed separately from the duty control of the power MOSFET which is the upper / lower semiconductor switching element in a steady state. When the load changes, the duty is forcibly set to 0% or 100%.
[Selection diagram] Fig. 1

Description

[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a power supply device having a stabilized control loop and a device using the same.
[0002]
[Prior art]
In a power supply device provided with a conventional loop stabilization method, a signal is fed back from an LC smoothing filter of a power system to an error amplifier, and a phase is compensated by the error amplifier to stabilize a control loop. In this prior art, an aluminum electrolytic capacitor is used for the power LC smoothing filter (for example, Non-Patent Document 1).
[0003]
On the other hand, in order to cope with downsizing of the power supply device, it is necessary to use a ceramic capacitor of a chip component instead of the aluminum electrolytic capacitor of the power LC smoothing filter. However, chip ceramic capacitors have equivalent series resistance
(ESR) is as small as several mΩ, and under actual use conditions, the total ESR is as small as 1 mΩ or less because ceramic capacitors are connected in parallel. For this reason, damping of the ESR as in the case of using an aluminum electrolytic capacitor cannot be expected, and it is difficult to stabilize the control loop.
[0004]
When a ceramic capacitor having a small ESR is used for the power-system LC smoothing filter in the above-mentioned conventional technology, a signal is oscillated because the damping effect of the ESR cannot be expected, and phase compensation becomes difficult. Further, even if the phase compensation can be performed by narrowing the operation band of the error amplifier in the related art, the response of the power supply is significantly slowed down. further,
When changing the constant of the LC smoothing filter, there is a troublesome adjustment of the phase compensation condition of the error amplifier each time.
[0005]
Further, a method is described in which an output of a CR smoothing filter connected to both ends of an inductor of an output LC smoothing filter is returned to an error amplifier having a low input impedance (for example, Patent Document 1). In this technique, since the voltage and current signals of the power supply output are extracted using the CR smoothing filter, the circuit configuration of the error amplifier needs to have low input impedance. Therefore, it is necessary to reduce the R value of the CR smoothing filter, and 0.47 μF, 100Ω is used as the constant of the CR smoothing filter. Therefore, since the CR smoothing filter having this constant cannot be mounted on the power supply IC on a chip, there is a problem that the power supply device cannot be miniaturized because it remains as an external component.
[0006]
[Non-patent document 1]
ZHANG et al, "Low-voltage on-board DC / DC modules for next generations of data processing circuits," IEEE Tran. on Power Elect. vol. 11, no. 2Mar. 1996 P328-337
[Patent Document 1]
USP5,877,611
[0007]
[Problems to be solved by the invention]
If the power supply device is downsized, it is difficult to stabilize the control loop. An object of the present invention is to provide a power supply device that is easy to stabilize a control loop while being downsized.
[0008]
[Means for Solving the Problems]
The present invention can solve the above problem by using a power supply device that feeds back a signal through a high-speed filter separately from a power filter for stabilizing a control loop.
[0009]
BEST MODE FOR CARRYING OUT THE INVENTION
An embodiment of the present invention will be briefly described.
[0010]
The power supply device of the present invention includes a power supply device for a step-down DC-DC converter.
The signal is fed back to the error amplifier through a CR smoothing filter provided separately from the LC smoothing filter.
[0011]
In addition, the power supply device of the present invention may be configured such that the output of the power-system LC smoothing filter is added to a control circuit having upper and lower limit detection, and the transient control is performed separately from the duty control of the power MOSFET which is the upper / lower semiconductor switching element in a steady state. Means are provided for forcibly setting the duty to 0% or 100% when the load changes.
[0012]
Further, the power supply device of the present invention is provided with an oscillator and a phase shift circuit, which are provided in common for the plurality of power supply units, in order to prepare a plurality of power supply units and operate them in parallel. Unit upper / lower power
The drive pulse phase of the MOSFET is shifted to a phase obtained by dividing 360 ° by the number of parallel circuits, and all the parallel power supply units are operated with the same phase drive pulse during a transient load change.
[0013]
Hereinafter, embodiments will be described with reference to the drawings.
[0014]
(Example 1)
FIG. 1 shows a power supply device according to the present embodiment. In FIG. 1, Vi is an input terminal, and Vo is an output terminal. An upper power MOSFET Q1 is connected to the input terminal Vi, and a lower power MOSFET Q2 is connected to the ground potential side. An LC smoothing filter, which is a power system output filter including an inductor L and a capacitor Co, and a CR smoothing filter, including a resistor R and a capacitor C, are connected in parallel between the power MOSFETs Q1 and Q2. An output terminal Vo is provided at the center of the filter.
One input (−) of the error amplifier EA is connected to the middle point of the CR smoothing filter. Here, the capacitor Co of the LC smoothing filter is a chip ceramic capacitor.
[0015]
The reference voltage Vref is connected to the other input (+) of the error amplifier EA. The gates of the power MOSFETs Q1 and Q2 are connected to the output of the error amplifier EA via a pulse width modulation (PWM) oscillator PWM and a driver DRV. The power MOSFETs Q1, Q2 are driven in opposite phases and alternately conduct. In this embodiment, the output voltage Vout is lower than the input voltage Vin.
[0016]
Next, the circuit operation of FIG. 1 will be described. The input voltage applied to the input terminal Vi is converted into a voltage through a CR smoothing filter by ON / OFF control of the upper power MOSFET Q1 and the lower power MOSFET Q2. The converted voltage VFB is compared with the reference voltage Vref and the error amplifier EA, and the error voltage is amplified and generated at the output of the error amplifier EA. This error voltage is converted into a PWM pulse by the pulse width modulation oscillator PWM. This PWM pulse is converted by a driver DRV into an on / off time ratio (duty: α) for driving the upper power MOSFET Q1 and the lower power MOSFET Q2, and is subjected to negative feedback control so that the error voltage becomes zero. VFB becomes equal to the reference voltage Vref. In this case, the converted voltage VFB obtained through the CR smoothing filter in the steady state is proportional to the duty α of the input voltage Vin. Therefore, the relational expression of VFB = Vref = α · Vin holds. Here, since the duty α is defined by ON time / (sum of ON time and OFF time), it takes a value between 0 and 1.
[0017]
In the case of a normal step-down converter, it is known that the voltage conversion rate in a steady state is equal to the ratio between the output voltage and the input voltage, that is, the duty, so the output of the LC smoothing filter, that is, the output obtained at the output terminal Vo The voltage Vout is obtained by a relational expression of Vout = α · Vin, where Vin is an input voltage and α is a duty.
[0018]
From the above two equations, the relationship of Vout = VFB = α · Vin is established. Therefore, the output voltage is equivalent to directly controlling the output voltage Vout of the output terminal Vo if the duty α can be indirectly controlled by another method without directly controlling the output of the LC smoothing filter by feedback. A voltage proportional to the duty α of the input voltage Vin is obtained at the output terminal Vo. In other words, by driving the power MOSFETs Q1 and Q2 and performing negative feedback control on the output of the CR smoothing filter, a desired voltage proportional to the duty α of the input voltage Vin also changes the output voltage Vout of the output of the LC smoothing filter. Obtainable.
[0019]
This embodiment is a first-order lag control method using a CR smoothing filter in the control loop as voltage conversion means by duty control of the upper power MOSFET Q1 and the lower power MOSFET Q2. Since there is no next delay and the control loop does not become a vibration system, no vibration waveform is generated in the output, and the loop becomes stable. Therefore, according to the present embodiment, the control loop can be stabilized even if a chip ceramic capacitor having a small ESR is used as the capacitor of the LC smoothing filter.
[0020]
Next, the magnitude relationship between the corner frequency and the switching frequency of the two smoothing filters will be described. If the corner frequency of the CR smoothing filter is fCR, the corner frequency of the LC smoothing filter is fLC, and the switching frequency is fSW, these are set to fLC <fCR <fSW, and the loop stability can be secured. From this relationship, the feedback from the CR smoothing filter has a higher operating frequency than the feedback from the LC smoothing filter, so that a high-speed response is possible. In addition, if fLC and fCR are set to frequencies that are somewhat apart from each other, it is not necessary to change the CR smoothing filter constant even if the LC smoothing filter constant is changed, and the degree of freedom in design can be increased. For high-speed operation at a switching frequency of 1 to 6 MHz, the constants of the LC smoothing filter and the CR smoothing filter can be, for example, 0.2 μH, 220 μF, 20 pF, and 200 kΩ, respectively. If the values of the capacitor and the resistor are in this order, the CR smoothing filter can be mounted on a semiconductor integrated circuit chip (on-chip), and external components are not required. This is because if the power supply device shown in FIG. 1 is an IC having the same terminal arrangement (pin compatible) as the conventional power supply control IC,
A conventional printed circuit board can be used as it is simply by replacing the IC. Since the constants of the CR smoothing filter are 20 pF and 200 kΩ, the error amplifier EA requires an amplifier configuration having a high input impedance. Therefore, the error amplifier EA is preferably used in a configuration in which the input of the differential amplifier inside the error amplifier and the output of the CR smoothing filter are directly connected.
[0021]
FIG. 2 is an explanatory diagram of a chip layout when a CR smoothing filter is incorporated in a semiconductor chip in the power supply device of FIG. In FIG. 2, C is a built-in capacitor and R is a built-in resistor, which are mounted on the same semiconductor substrate as the error amplifier EA, the pulse width modulation oscillator PWM, the driver DRV, and the power MOSFETs Q1, Q2.
[0022]
In the above, the filter that feeds back to the error amplifier of the control loop has been described as an example of the CR smoothing filter. However, a similar effect can be obtained by using another filter circuit having good response instead. Further, although the power MOSFET has been described as an example of the semiconductor switching element, an IGBT may be used instead.
[0023]
(Example 2)
FIG. 3 shows this embodiment. 3, the same components as those in FIG. 1 are denoted by the same reference numerals. FIG. 3 differs from FIG. 1 in that a CR smoothing filter is connected to both ends of an inductor L of the LC smoothing filter. In this embodiment, since the capacitance of the capacitor Co of the output LC smoothing filter is large, the inductor connection end side of the capacitor Co can be regarded as the ground potential. Also in this embodiment, the same effect as that of FIG. 1 can be obtained, and a small capacitance change due to a temperature change of the capacitor Co of the LC smoothing filter can be negatively fed back. Even if a chip ceramic capacitor having a small ESR is used, the control loop can be controlled. Stability can be improved. Also in this case, the constant of the embodiment of FIG. 1 can be used as the constant of the CR smoothing filter. FIG. 4 is an explanatory diagram of a chip layout when a CR smoothing filter is incorporated in a semiconductor chip in the power supply device of FIG.
[0024]
(Example 3)
FIG. 5 shows a power supply device in which the transient fluctuation detection circuit TVD is further provided in the first embodiment. The transient fluctuation detection circuit TVD detects a transient load fluctuation between the output voltage Vout of the output terminal Vo and a voltage obtained by adding upper and lower limit voltage widths ± Δ to the reference voltage Vref, and controls the duty of the pulse width modulation oscillator PWM. . FIG. 6 shows a specific example of the pulse width modulation oscillator PWM and the transient fluctuation detection circuit TVD.
[0025]
6, the pulse width modulation oscillator PWM is a variable oscillator including a voltage / current conversion circuit V / I, current source MOSs 110 and 120, inverters INV11 and INV12, a capacitor 105, and a flip-flop FF. Further, the transient fluctuation detection circuit TVD includes comparators CMP1 and CMP2, and switches MOS SW1 to SW1.
SW4, constant current sources I1 to I4, and inverters INV1 to INV8.
[0026]
The transient change detection circuit TVD includes a window comparator including two comparators CMP1 and CMP2, compares the output voltage Vout with a voltage obtained by adding upper and lower limit voltage widths ± Δ to a reference voltage Vref, and operates the output voltage Vout. To determine the pulse duty α of the PWM oscillator PWM shown in FIG. This means that the transient fluctuation detecting circuit TVD switches the control method between the steady state and the transient load fluctuation to a control mode suitable for the operating state.
[0027]
From the outputs of the two comparators CMP1 and CMP2, three types of information are obtained: (a) when the load current decreases, (b) a steady state, and (c) when the load current increases. These cases will be described with reference to FIG.
[0028]
(A) is a case of the condition of Vo ≧ (Vref + Δ). At this time, the output duty of the pulse width modulation oscillator PWM is forcibly set to 0%. Therefore, the switches MOS SW1 and SW4 are turned on, the switches MOS SW3 and SW2 are turned off, and the current of the constant current source I1 flows into the inverter INV11 in addition to the current of the current source MOS110, and the current of the constant current source I4 is Since the current of the current source MOS 120 is extracted, the current flowing through the inverter INV12 becomes zero. Therefore, the upper power MOSFET Q1 is turned off, the lower power MOSFET Q2 is turned on, and the duty becomes 0%. Also in this case, in order to completely set the duty α to 0%, it is preferable to set the current values of the constant current sources I1 to I4 to the total current of the differential pair operating current of the voltage / current conversion circuit V / I. .
[0029]
(B) is a case of the condition of (Vref + Δ)>Vo> (Vref−Δ). In this case, all the switch MOS switches SW1 to SW4 are off, and operate at the current ratio determined by the control command from the error amplifier EA. Since this current ratio is equal to the duty ratio, a voltage proportional to the duty α of the input voltage Vin is obtained as the output voltage Vout.
[0030]
(C) is a condition of Vo ≦ (Vref−Δ), and the duty is forcibly set to 100%. In this case, the switches MOS SW3 and SW2 are turned on, the switches MOS SW1 and SW4 are turned off, the current of the constant current source I3 is added to the current of the current source MOS 120 and flows to the inverter INV12, and the current of the constant current source I2 is Since the current of the MOS 110 is extracted, the current flowing through the inverter INV11 becomes zero. Therefore, the upper power MOSFET Q1 is turned on and the lower power MOSFET Q2 is turned off, and the duty becomes 100%. Also in this case, in order to completely set the duty α to 100%, it is preferable to set the current values of the constant current sources I1 to I4 to the total current of the differential pair operating current of the voltage / current conversion circuit V / I. .
[0031]
In the present embodiment, the voltage generated at the output terminal Vo during the transient load change is
The duty α of the pulse width modulation oscillator PWM is forcibly switched to 0% or 100% so as to be within the upper and lower limit voltage width ± Δ added to Vref, and the output voltage Vout is rapidly suppressed to within Vref ± Δ. I do. Then, when the stationary state is entered, the output voltage is finally settled to a voltage proportional to the duty α of the input voltage.
[0032]
As described above, in the present embodiment, the control mode is automatically switched depending on the transient load change and the steady state, so that, for example, a transient load change of about 10 A having a high current change rate (di / dt) of 500 A / μs is performed. However, both high-speed response and stabilization of the output voltage in a steady state can be achieved.
[0033]
Next, another embodiment of the pulse width modulation oscillator PWM will be described with reference to FIG. 20 can be achieved by a combination of the oscillator OSC, the one-shot multivibrator OSM, and the V / I converter VI. The constant-period pulse generated by the oscillator OSC can be set by the MOS 130 and the constant current source I5 as a constant current required for determining a desired period of a current flowing through the current sources MOS 110 and 120 of the pulse width modulation oscillator PWM of FIG. When the pulse of this fixed cycle is applied to the clock terminal CLK of the one-shot multivibrator OSM, the terminal voltage of the capacitor CT temporarily becomes zero, but the error voltage of the error amplifier EA is converted by the V / I converter VI. The converted current charges the capacitor CT. Then, a time until the charging voltage reaches a predetermined threshold value is obtained as a PWM pulse. Thus, a series of pulse width modulation oscillation operations can be repeated. That is, a PWM pulse proportional to the error voltage of the error amplifier EA can be obtained.
[0034]
This pulse width modulation oscillator PWM is used as an effective means in the multi-phase control of FIGS. 11 and 12 described later. In this case, it is necessary to insert a phase shift circuit after the oscillator OSC for the multi-phase operation.
[0035]
(Example 4)
This embodiment is shown in FIGS. FIG. 8 shows the embodiment of FIG. 3 in which a transient fluctuation detection circuit TVD is provided, and the same effects as in FIG. 5 can be obtained. FIGS. 9 and 10 show the circuit diagram of FIGS. 1 and 3 in which the input of the transient fluctuation detection circuit TVD is taken from the midpoint of the series circuit composed of the capacitor C3 and the resistor R3 provided at both ends of the inductor L of the LC smoothing filter. It was made. As a result, the phase of the inductor L current, which can be detected by the series circuit of the capacitor C3 and the resistor R3, and the charge / discharge phase of the output capacitor Co can be matched, so that the excess / excess charge due to the charge / discharge of the output capacitor Co from the inductor L current. Can be minimized. For this reason, in addition to high-speed response and high stability, it is possible to reduce the fluctuation (ripple) of the output voltage at the time of a transient load fluctuation.
[0036]
(Example 5)
This embodiment is a multi-phase embodiment in which a plurality of power supply units of the first to fourth embodiments are operated in parallel. In this embodiment, two or more power supplies of the same type shown in the first to fourth embodiments are combined. Hereinafter, a two-phase process will be described as an example.
[0037]
FIG. 11 shows an example in which the power supply unit of FIG. 8 is multi-phased. In FIG. 11, a new oscillator OSC and a phase shift circuit PSFT are newly provided for multi-phase operation, and a two-phase pulse whose phase is shifted by 180 ° is generated by these oscillators. And PWM2 to realize multi-phase control.
[0038]
FIG. 12 shows an example of the power supply device of FIG. 11 in detail. In FIG. 12, a pulse width modulation oscillator PWM1 includes a voltage / current conversion circuit V / I1 and a one-shot multivibrator OSM1, and operates in a steady state by receiving a pulse signal from a phase shift circuit PSFT.
[0039]
The operation of FIG. 12 will be described using the operation mode of FIG. The operation state mode will be described similarly to the third embodiment. Hereinafter, the description will be made using the power supply of Phase 1 shown in the upper half of FIG. (A) When Vout ≧ (Vref + Δ), the output duty of the pulse width modulation oscillator PWM is forcibly set to 0%. Therefore, the reset RST of the one-shot multivibrator OSM1 is turned on, and the duty becomes 0%. (B) In the case of (Vref + Δ)>Vout> (Vref−Δ), the pulse from the phase shift circuit PSFT is received as the clock CLK by the operation of the normal one-shot multivibrator, and the current value and the timing of the current source MOS 210 An on-pulse width determined by the capacitance value of the capacitor CT1 is generated. This on-pulse width is a control mode that operates at a current ratio determined by control from the error amplifier EA. That is, since this current ratio is equal to the duty, the output voltage Vout becomes a voltage proportional to the duty α of the input voltage Vin. In the case of (c) Vout ≦ (Vref-Δ), the duty is forcibly set to 100%. For this reason, both ends of the capacitor CT1 which is a timing capacitor are short-circuited by the MOS switch M21 to keep the ON state, and the duty is set to 100%. In addition, the detection result of the overcurrent detection circuit OC1 is added to the reset RST to prevent the element breakdown due to the overcurrent of the upper power MOSFET Q1. The operation is the same with the power supply of Phase 2 in the lower half of FIG.
[0040]
In the above operation, in the steady state, the inductor currents of the two power supplies operate in opposite phases with a 180 ° phase shift, and during transient load changes, the inductor currents of the two power supplies have the same phase, which corresponds to a sudden load change. In the present embodiment, not only the output current is increased by using a plurality of power supply devices, but also the ripple of the output voltage is reduced.
[0041]
In the case where more than two power supply units are provided, an oscillator and a phase shift circuit which are shared by a plurality of power supply units are provided. In a steady state, the drive pulse phase of the upper / lower power MOSFET of each power supply unit is set. Phase shifts by 360 ° divided by the number of power supply units arranged in parallel, and in the event of a transient load change, all parallel power supply units are driven by drive pulses of the same phase as in (a) and (c) above. I do. For example, in the case of four power supply units, the phases may be shifted to 0 ° (reference), 90 °, 180 °, and 270 °.
[0042]
(Example 6)
An example of an IC chip configuration of the power supply control device of the present invention will be described.
[0043]
FIG. 14 is an example of a one-chip configuration of the circuit configuration of FIG. In FIG. 14, the circuit is the same as that of the LC smoothing filter, except that a CR circuit including a capacitor C3 and a resistor R3 for detecting a current phase of a transient fluctuation detection circuit TVD and a boost circuit including a diode DBT and a capacitor CBT are externally provided. The functions are integrated on a single semiconductor substrate on a chip.
[0044]
On-chip circuits and functions include a CR smoothing filter including a capacitor C and a resistor R, an error amplifier EA, a reference voltage Vref, a pulse width modulation oscillator PWM, a dead band circuit DBU, a dead band circuit DBL, a level shift circuit LS, Driver DRVU, driver DRVL, upper / lower power MOSFETs Q1, Q2, overcurrent detection circuit OC, transient fluctuation detection circuit TVD, upper / lower limit voltage generation circuit VΔ, soft start circuit SS, under voltage lockout circuit UVLO, power There is a good circuit PWRGD. Instead of obtaining the reference voltage Vref from the bandgap reference circuit, the reference voltage Vref may be obtained by receiving a digital signal corresponding to a VID (Voltage Identification) code and using an on-chip D / A converter shown in FIG. Although not shown, the one-chip power supply control IC of the present embodiment has a function based on VRM 9.1 proposed by Intel.
[0045]
Although the case where the upper power MOSFET Q1 is an NMOS has been described with reference to FIG. 14, it may be a PMOS. In this case, an external boost circuit is not required, but since it is necessary to drive the gate of the PMOS with the potential from the input terminal Vi, the voltage generator for this purpose is provided on-chip.
[0046]
The voltage supplied to the input terminal Vi and the power supply terminal Vcc is made the same, for example, 5V or
The voltage may be 12 V, or 12 V for the input terminal Vi and 5 V for the power supply terminal Vcc, and different voltages may be used. When the voltage to be supplied to the input terminal Vi and the power supply terminal Vcc is different, the power supply terminal Vcc of 5 V may be supplied from the outside, or 5 V may be generated and supplied from the 12 V of the input terminal Vi by an on-chip circuit. When supplying 12 V to the input terminal Vi, the boost circuit of FIG. 14 is connected to a zener diode of about 7 V in series with the diode DBT so that the gate voltage of the upper power MOSFET does not become excessive.
[0047]
In the operation of the soft start circuit, the output signal of the transient fluctuation detection circuit for high-speed response at the time of turning on the power may be masked.
[0048]
(Example 7)
FIG. 16 shows a multi-phase IC chip configuration of this embodiment. FIG. 16 illustrates a multi-phase circuit configuration of an IC chip, and differs from FIG. 14 of the sixth embodiment in that an oscillator OSC and a phase shift circuit PSFT are added to the IC chip. The IC pins required by the multi-phase operation are provided with terminals for providing phase pulses φ1 to φ4 corresponding to the number of multi-phases to the own / other IC chip, a reference voltage Vref and a transient fluctuation detection circuit TVD. There is a terminal for supplying the output of the lower limit voltage generation circuit VΔ.
[0049]
In the multi-phase configuration, first, IC chips are prepared by the number of desired multi-phases, and one of the IC chips is used as a master. Specifically, the oscillator OSC and the switch SWr are activated by the selection signal SEL0 of the master IC chip, and a desired number of phases is designated by two bits of the selection signals SEL1 and SEL2. Next, when the phase pulses φ2 to φ4, the reference voltage Vref, and the outputs V + Δ, V−Δ of the upper / lower limit voltage generation circuit VΔ are supplied from the master IC chip, φ, Vref, Vref + Δ, Vref−Δ are respectively supplied to the other IC chips. Multiphase is achieved by adding.
[0050]
In the present embodiment, the number of multi-phases is illustrated as 4, but the number of phases is not limited, and the number of selection signals for setting the number of phases can be changed, and the phase shift circuit PSFT has a circuit configuration corresponding to the number of phases. By changing them and incorporating them in an IC chip, the number of multiphases can be increased or decreased as appropriate.
[0051]
(Example 8)
FIG. 17 shows an embodiment in which the power supply control IC chip of the present invention is mounted on a printed wiring board. FIG. 17 shows that the power supply control IC is a BGA (Ball Grid Array), and the inductor L and the capacitor Co are mounted on the printed wiring board PB as chip components, thereby enabling small-sized and high-density mounting. Here, the capacitor Co is a chip ceramic capacitor. Although not shown, in addition to this, in this embodiment, a CR circuit of the capacitor C3 and the resistor R3, a boost circuit, and an input capacitor are mounted as chip components on the printed wiring board PB. Further, in addition to the chip mounting by the BGA, a chip size package (CSP) may be mounted.
[0052]
Furthermore, in the case of multi-phase support, in addition to the above-described chip mounting of a plurality of power control ICs, an MCM (Multi Chip Module) may be mounted. In addition, a control unit including an error amplifier, a PWM oscillator, and the like, and a driver unit having a built-in power MOSFET, in which two IC chips are separated may be similarly mounted on a printed wiring board.
[0053]
As described above, according to the present embodiment, the elimination of the pin neck, the improvement of the heat radiation, and the miniaturization of the power supply device printed wiring board can be realized.
[0054]
(Example 9)
This embodiment is shown in FIG. FIG. 18 shows an embodiment applied to a hard disk drive (HDD) device. The HDD device includes a magnetic recording disk, a magnetic head, a magnetic disk rotation driving device, a magnetic head driving device, a magnetic head position control device, and an input / output signal control device. Power is supplied to the HDDn from the DC-DC converters DC-DC1 to DC-DCn, which are the power supply devices described in the first to eighth embodiments. The DC-DC converters DC-DC1 to DC-DCn, which are the power supply devices shown in FIG. 18, use a single-phase power supply device or a multi-phase power supply device according to the current capacity of the HDD device to which power is supplied.
[0055]
(Example 10)
Next, an embodiment in which the control method of the present invention is applied to an insulation type DC-DC converter will be described. FIG. 19 shows an example of application to a forward converter. In FIG. 19, CR smoothing filters C and R are provided at both ends of the inductor L of the forward type converter as shown in FIG. 3, and the error amplifier EA is used by using the relationship between the voltage VFB at the middle point of the CR smoothing filter and the reference voltage Vref. Generates an error amplification, and further converts the voltage into a PWM pulse using a pulse width modulation oscillator PWM. This PWM pulse is applied to the gate of the power MOSFET QD that drives the transformer T1 through the transformer T2, and is subjected to negative feedback control. Thereby, the output terminal V _O , A desired output voltage is constantly obtained. This method does not feed back from the power system LC smoothing filter, so that a power supply system with good loop stability can be constructed. Therefore, it is effective when a ceramic capacitor is used for C of the LC smoothing filter.
[0056]
In the above description, the CR smoothing filter of FIG. 3 has been described, but the method of FIG. 1 is also possible. Further, it can also be realized by using a photocoupler instead of the transformer T2. In FIG. 19, a single type of forward converter is described. However, the present invention can also be applied to a two-type isolated DC-DC converter such as a forward type, a push-pull type, a half-bridge type, and a full-bridge type.
[0057]
(Example 11)
Next, an embodiment in which the control method of the present invention is applied to a commercially available power supply IC will be described. FIG. 21 shows a case where, for example, a PWM control IC HIP6311A manufactured by Intersil and a power MOSFET IC ILS6571 with a built-in driver are used as power supply ICs that are generally sold. The middle point of one of the CR smoothing filters C and R provided at both ends of the inductor L is connected to a feedback terminal FB of a PWM control IC HIP6311A via a buffer amplifier BA and a resistor RIN for achieving high input impedance, and the other end. The middle point of the CR smoothing filters C3 and R3 is connected to a transient fluctuation detection circuit TVD including a reference power supply LT1790A-2.5 and a comparator LT1715 manufactured by Linear Technology. The three operation state modes of the PWM pulse signal PWM1 (desired duty α), duty 0% α0, and duty 100% α100 output from the PWM control IC correspond to two signals a obtained by the transient fluctuation detection circuit TVD. , B are selectively switched by the selector HD74HC153 as shown in FIG. 22, and the selection signal Y is output to the PWM terminal of the power MOSFET IC with a built-in driver. This indicates that the control method of the present invention can be easily applied to a power supply device using a commercially available power supply IC. The application of the present invention is not limited to the products described in the above embodiments. When the transient fluctuation detection circuit TVD is not used, the present invention can be achieved by directly connecting the PWM pulse signal PWM1 output from the PWM control IC to the PWM terminal of the power MOSFET IC with a built-in driver.
[0058]
Although not shown, the power supply devices according to the first to eighth embodiments can be applied to VRMs, DC-DC converters for portable devices, general-purpose DC-DC converters, and the like.
[0059]
As a result, the power supply device of the present invention does not include the second-order lag of the power-system LC smoothing filter in the control loop, thereby improving the stability of the control loop. As a result, a chip ceramic capacitor having a small ESR can be used for the LC smoothing filter, so that the power supply device can be downsized.
[0060]
Further, since the control circuit having the upper and lower limit detection performs high-speed response control at the time of a transient load change, it is possible to provide a power supply device that can respond to a high current change rate (di / dt).
[0061]
Furthermore, in the power supply device according to the present embodiment, multiphase can be easily achieved, and both a large output current and a reduction in ripple voltage can be achieved.
[0062]
【The invention's effect】
According to the present invention, the stability of the control loop can be improved, and the degree of freedom in design can be increased, so that the power supply device can be downsized.
[Brief description of the drawings]
FIG. 1 is a circuit block diagram of a power supply device according to a first embodiment.
FIG. 2 is an explanatory diagram of an IC in which a CR filter is built in a semiconductor chip in the power supply device of FIG. 1;
FIG. 3 is a circuit block diagram of a power supply device according to a second embodiment.
FIG. 4 is an explanatory diagram of an IC in which a CR filter is built in a semiconductor chip in the power supply device of FIG. 3;
FIG. 5 is a circuit block diagram of a power supply device according to a third embodiment.
FIG. 6 is a circuit diagram showing details of FIG. 5;
FIG. 7 is a diagram illustrating an operation state mode of FIG. 6;
FIG. 8 is a circuit block diagram of a power supply device according to a fourth embodiment.
FIG. 9 is a circuit block diagram of another power supply device according to the fourth embodiment.
FIG. 10 is a circuit block diagram of still another power supply device according to the fourth embodiment.
FIG. 11 is a circuit block diagram of a multiphase power supply device according to a fifth embodiment.
FIG. 12 is a circuit diagram showing details of FIG. 11;
FIG. 13 is a diagram showing an operation state mode of FIG.
FIG. 14 is a circuit block diagram illustrating a chip configuration of a power supply device according to a sixth embodiment.
15 is an explanatory diagram of a VID code input D / A converter applied to FIG.
FIG. 16 is a circuit block diagram of a multiphase-compatible chip according to a seventh embodiment.
FIG. 17 is an explanatory view of mounting a power supply control IC according to an eighth embodiment on a printed wiring board.
FIG. 18 is an explanatory diagram of an HDD device according to a ninth embodiment.
FIG. 19 is an explanatory diagram of the tenth embodiment.
FIG. 20 is an explanatory diagram showing another embodiment of the pulse width modulation oscillator PWM.
FIG. 21 is an explanatory diagram of Embodiment 11 applied to a general power supply IC.
FIG. 22 is a diagram illustrating an operation state mode of FIG. 21;
[Explanation of symbols]
105, Co, C, C1 to C4, CT1, CT2, CBT ... capacitors, 110, 110 ', 120 ... current source MOS, BGA ... pin grid array chip, CMP1 to CMP4 ... comparators, D / A ... D / A converter, DBT: diode, DBU, DBL: dead band circuit, DC-DC1 to DC-DCn: DC-DC converter, DRV, DRV1, DRV2, DRVU, DRVL: driver, EA, EA1, EA2: error amplifier FF: flip-flop, GND: ground, HDD1 to HDDn: HDD device, I1 to I4: constant current source, INV1 to INV12: inverter, L, L1, L2: inductor, LINE: power supply line, LS: level shift circuit, OC, OC1, OC2 ... overcurrent detection circuit, OR1, OR2 ... OR circuit, SC: oscillator, OSM1, OSM2: one-shot multivibrator, PB: printed wiring board, PG: power ground, PSFT: phase shift circuit, PWM, PWM1, PWM2: pulse width modulation oscillator, PWRGD: power good circuit, Q1 , Q3: upper power MOSFET, Q2, Q4: lower power MOSFET, R, R1 to R4: resistor, SS: soft start circuit, SW1 to SW4, M21, M21 ': switch MOS, TVD, TVD1, TVD2: transient fluctuation Detection circuit, UVLO: Under voltage lock-out circuit, Vcc: Power supply terminal, Vi: Input terminal, V / I, V / I1, V / I2: Voltage / current conversion circuit, Vo: Output terminal, Vref: Reference Voltage, VΔ: Upper and lower limit voltage generation circuit, Δ: Upper and lower limit voltage width.

Claims (16)

Step-down DC-DC comprising a power semiconductor switching element, a driving unit for the power semiconductor switching element, a pulse width modulation oscillator for supplying a driving signal to the driving unit, and an error amplifier for supplying an error signal to the oscillator In the converter power supply,
A power system output filter for passing the output power, and a filter provided separately from the power system output filter, wherein the output of the separately provided filter and the differential amplifier inside the error amplifier are directly connected and output to the error amplifier. A power supply device for returning a signal. The power supply device according to claim 1, wherein the power output filter is an LC filter including an inductor and a capacitor, and the separately provided filter is a CR filter including a capacitor and a resistor. A power supply device provided in parallel with an LC filter and feeding back an output signal to an error amplifier through the CR filter. 2. The power supply device according to claim 1, wherein the power system output filter is an LC filter including an inductor and a capacitor, and the separately provided filter is a CR filter including a capacitor and a resistor. A CR filter is provided at both ends of the inductor, and an output signal is fed back to the error amplifier through the CR filter. The power supply device according to claim 2 or 3,
When the frequency of the CR filter is fCR and the frequency of the LC filter is fLC, a relationship of fLC <fCR is satisfied. 5. The power supply device according to claim 4, wherein the power supply device further includes a transient fluctuation detection circuit, wherein the transient fluctuation detection circuit detects an output voltage from an output terminal of the power system output filter, and the output voltage is determined in advance. A signal that sets the duty of the pulse width modulation oscillator to 0% is output when the output voltage exceeds a predetermined upper limit, and the duty of the pulse width modulation oscillator is set to 100% when the output voltage is equal to or lower than a predetermined lower limit. A power supply device for outputting a signal to perform 5. The power supply device according to claim 4, wherein the power supply device further includes a transient fluctuation detecting circuit, and the transient fluctuation detecting circuit detects a transition from an output terminal of a CR circuit newly provided at both ends of an inductor of a power output filter. Detecting an output voltage, outputting a signal that sets the duty of the pulse width modulation oscillator to 0% when the output voltage exceeds a predetermined upper limit, and outputting the signal when the output voltage is equal to or lower than a predetermined lower limit. A power supply device for outputting a signal for setting the duty of the pulse width modulation oscillator to 100%. Step-down DC-DC comprising a power semiconductor switching element, a driving unit for the power semiconductor switching element, a pulse width modulation oscillator for supplying a driving signal to the driving unit, and an error amplifier for supplying an error signal to the oscillator In a power supply device having a plurality of power supply units for a converter,
Each of the plurality of power supply units includes a power output filter that passes output power, and a filter provided separately from the power output filter, and outputs an output of the separately provided filter and a differential amplifier in the error amplifier. A power supply device, which is directly connected and feeds back an output signal to the error amplifier. The power supply device according to claim 7, wherein the plurality of power supply units are operated in parallel, the plurality of power supply units have the pulse width modulation oscillator in common, and the output of the pulse width modulation oscillator is phase-shifted. A power supply device, wherein the power supply device shifts and supplies the phase-shifted signal to the plurality of power supply units. 9. The power supply device according to claim 8, wherein each of the plurality of power supply units includes a transient fluctuation detection circuit, and the transient fluctuation detection circuit detects an output voltage from an output terminal of the power system output filter, and outputs the output voltage. When the voltage exceeds a predetermined upper limit, a signal for setting the duty of the pulse width modulation oscillator to 0% is output. When the output voltage is equal to or less than a predetermined lower limit, the duty of the pulse width modulation oscillator is changed. A power supply device, which outputs a signal that makes it 100%. The power supply device according to claim 1,
An output voltage is detected from the output terminals of the power semiconductor switching element, the driving means of the power semiconductor switching element, the pulse width modulation oscillator, the error amplifier, and the power system output filter, and the output voltage is determined in advance. When the output voltage exceeds the upper limit, a signal for setting the duty of the pulse width modulation oscillator to 0% is output. When the output voltage is equal to or lower than a predetermined lower limit, the duty of the pulse width modulation oscillator is set to 100%. A power supply device, wherein a transient detection circuit for outputting a signal is formed on the same semiconductor substrate. A magnetic recording disk, a magnetic head, a magnetic disk rotation driving device, a magnetic head driving device, a magnetic head position control device, an input / output signal control device, and a hard disk device including a power supply device for supplying power;
The power supply device includes: a power semiconductor switching element; driving means for the power semiconductor switching element; a pulse width modulation oscillator for supplying a driving signal to the driving means; and an error amplifier for supplying an error signal to the oscillator. A step-down DC-DC converter power supply device,
A power system output filter for passing the output power, and a filter provided separately from the power system output filter, wherein the output of the separately provided filter and the differential amplifier in the error amplifier are directly connected and output to the error amplifier. Return the signal,
The power system output filter is an LC filter including an inductor and a capacitor, the separately provided filter is a CR filter including a capacitor and a resistor, and the CR filter is provided in parallel with the LC filter. Where fCR is the frequency of the LC filter and fLC is the frequency of the LC filter, the relationship of fLC <fCR is established, and a power supply unit that feeds back an output signal to the error amplifier through the CR filter is provided. A power supply device, wherein the power supply device according to claim 4 is applied to an insulation type DC-DC converter. 5. A power supply device according to claim 4, wherein a combination of an oscillator, a one-shot multivibrator, and a V / I converter is used as the pulse width modulation oscillator of the power supply device. 14. The power supply according to claim 13, wherein a phase shift circuit is inserted after the oscillator, and a one-shot multivibrator is provided for each phase. apparatus. 2. The power supply device according to claim 1, wherein the error amplifier has a low input impedance, and is connected to an output of the separately provided filter via a buffer amplifier having a high input impedance. . An IC, wherein the power supply device according to any one of claims 1 to 10, 13, and 14 is incorporated in a semiconductor chip.

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